Note: Descriptions are shown in the official language in which they were submitted.
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LASER RANGE FINDER
BACKGROUND OF THE INVENTION
Field of the Invention
The present invention relates to distance or range
measuring equipment. More particularly, the present
invention relates to a laser based range finder.
Description of the Prior Art
Laser based distance and range measuring equipment has
been used for a number of years to provide extremely
accurate distance measurements to a remote target or
object. A representative instrument including a range
finder is shown in U.S. Pat. No. 5,359,404. Although a
highly accurate and reliable device, its great distance
ranging capability and inherent complexity translates to a
cost and form factor most suitable only for certain
specific applications.
SUMMARY OF THE INVENTION
The present invention is embodied in a laser range
finder including an internal power supply unit for
providing electrical power thereto which includes a laser
transmit section for generating a number of laser pulses
for transmission to a target. A laser receive section
receives reflected laser pulses from the target and a
precision timing section is coupled to the laser transmit
section and the laser receive section for determining a
flight time of the laser pulses and the reflected laser
pulses. A central processor section is coupled to the
precision timing section for determining a range to the
target derived from the flight time and a display is
coupled to the central processor section for visually
displaying the range to said target. In a particular
embodiment, the display may comprise an LCD display within
the field of view of an optical sight for aiming the laser
range finder.
In a more specific embodiment, the laser range finder
includes a crystal reference oscillator coupled to the
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central processor and the precision timing section for
providing a reference clock signal. Also provided is a
manually actuatable trigger switch coupled to the power
supply unit for causing the laser transmit section to
transmit the laser pulses toward the target and a manually
actuatable mode switch coupled to the central processor
unit for selecting a desired target acquisition mode based
upon the reflectivity of the target or the possible
presence of intervening partial obstructions. Successive
actuations of the mode switch display a plurality of target
acquisition modes for the laser range finder on the
display.
An embodiment of the range finder also includes an
automatic noise threshold section coupled to the laser
receive section which is responsive to the central
processor section for determining a desired signal-to-noise
ratio for the reflected laser pulses and provide a series
of possible return pulse values to the central processor.
The central processor is operative to place a preselected
number of the possible return pulse values in a stack until
a predetermined number of the return pulse values coincide
within a specified precision, at which time an average of
the predetermined number of the return pulse values are
utilized to determine the range to the target. The desired
signal-to-noise ratio of the automatic noise threshold
circuit may be determined by the central processor unit in
response to manual selection of a target reflectivity type
through the mode switch.
A precision timing section of the laser range finder
comprises means for determining a zero time value for the
laser pulses in addition to means for determining a
calibration time value as well. Means are also be provided
for determining a laser flight time value for the laser
pulse wherein the range to the target may be computed by
the central processor section as directly related to the
quantity (laser flight time minus zero time) divided by the
quantity (calibration time minus zero time).
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In one embodiment, the precision timing section
provides a start timer signal to the central processor
prior to transmission of one of the laser pulses from the
laser transmit section and a stop timer signal in response
to receipt of a corresponding one of the reflected laser
pulses by the laser receive section.
The range finder includes a self-calibrating,
precision timing circuit and method for determining a range
to a target based upon a flight time of a pulse toward the
target. The circuit comprises means for initially
establishing first and second reference voltage levels
together with means for unclamping the second reference
voltage level and means for allowing the second reference
voltage level to then diminish at a first rate to the first
reference voltage level. Further provided are means for
storing a first reference time extending from the step of
unclamping until the first and second reference voltage
levels are determined to be equal. Means are also provided
for then re-establishing the first and second reference
voltage levels together with means for again unclamping the
second reference voltage level. Additional means are
provided for increasing the second reference voltage level
at a second higher rate than the first rate for a
predetermined period of time to establish a third reference
voltage level together with means for then allowing the
third reference voltage level to diminish at the first rate
to the first reference voltage level at which time, a
second reference time extending from the step of again
unclamping until the first and third reference voltage
levels are equal is additionally stored. The first and
second reference voltage levels are again re-established
and the second reference voltage level is further
unclamped. Means are provided for again increasing the
second reference voltage level at the second higher rate
for a period of time related to the flight time of the
pulse to the target to establish a fourth reference voltage
level, together with means for then allowing the fourth
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reference voltage level to diminish at the first rate to
the first reference voltage level. A third reference time
extending from the unclamping of the second reference
voltage level until the first and fourth reference voltage
levels are equal is then stored and the range to the target
may be computed as proportional to the quantity of the
(third reference time minus the first reference time)
divided by the quantity of the (second reference time minus
the first reference time).
In a particular embodiment the establishing means
comprises a transistor switch for coupling a capacitor to
a source of the second voltage while the unclamping means
may comprise a second transistor switch for decoupling the
capacitor from the second voltage source. The allowing
means comprises a third transistor switch coupling a
resistor to the capacitor to bleed off the charge
therefrom .
The means for increasing the second reference voltage
level comprises means for applying a charge to the
capacitor at the second rate and the predetermined time
period specified may be determined by reference to a
crystal oscillator. In a particular embodiment, the second
charging rate is substantially 1000 times the first
discharging rate.
A further embodiment includes an automatic noise
threshold system for operative association with central
processing and signal receiving sections of a signal
transmitting device for discriminating between an actual
return signal and associated noise. The system includes
means responsive to the central processing section for
determining a desired signal-to-noise ratio for a series of
possible signal pulses, including both noise and actual
signal pulses received through the signal receiving
section. The possible signal pulses each have a
representative pulse value with respect to a pulse
previously transmitted from the signal transmitting device.
Further included are means responsive to the central
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processing section for placing up to a preselected number
of the possible signal pulse values in a stack until a
predetermined number of them coincide within a specified
precision. The value of one or more of the predetermined
5 number of the possible signal values is then considered to
be representative of the actual return signal. The
predetermined number of the possible signal pulse values
may further be averaged to represent the actual return
signal to a greater degree of precision.
In a more particular embodiment, the signal
transmitting device is a laser range finder and the pulse
value of the possible signal pulses corresponds to a
possible flight time of the pulses transmitted from the
laser range finder to a target.
The means for determining the desired signal-to-noise
ratio comprises a detector coupled to an output of the
signal receiving section for producing a substantially
constant noise pulse firing rate output and an operational
amplifier coupled to an output of the detector for
providing a threshold signal to the receiving section in
response thereto.
The threshold signal may be supplied to a summing node with
at least one noise level setting signal from the central
processor section for further determining the actual
threshold signal depending upon the target characteristics.
The invention is further embodied in a method for
discriminating between an actual return signal and
associated noise in a signal receiving section of a signal
transmitting device. The method comprises the steps of
transmitting a series of signal pulses to a target and
receiving a number of possible reflected signal pulses
therefrom with the possible reflected signal pulses
including both noise and actual signal pulses. A
representative pulse value is assigned for each of the
possible reflected signal pulses with respect to the series
of signal pulses transmitted to the target and each of the
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representative pulse values is compared with other ones of
the representative pulse values.
Each of the representative pulse values are compared until
any predetermined number of the representative pulse values
coincide within a specified precision and the actual return
signal is determined to be represented by the predetermined
number of the representative pulse values or they are
averaged to produce a greater precision value.
The present invention is a precise, yet accurate and
reliable laser range finder which may be economically
produced and is adapted to individual portable use in a
unit potentially weighing less than a pound with an on
board battery based power supply. Moreover, the compact
instrument herein provided has a number of user selectable
target acquisition operational modes which may be invoked
depending on the distance,,type and reflectivity of the
target being sighted.
This range finder provides highly accurate precision
range measurements of up to 1000 yards or more with a
resolution of less than 1 yard, and has a number of user
selectable target acquisition and enhanced precision -
measurement modes which may be viewed on an in-sight
display during aiming and operation of the instrument.
Extremely efficient self-calibrating precision timing and
automatic noise threshold circuits incorporated in the
design provide a compact, low-cost, highly accurate and
reliable ranging instrument for a~multitude of uses.
Through the use of an in-sight display, distance or
range information can be shown while the user may also view
and select the instrument's mode of operation through
successive actuations of a push button mode switch while
simultaneously sighting the target object. A precision
mode of operation may also be invoked in which an even more
precise measurement to an object may be achieved following
an initial measurement together with the visual indication
of a "precision flag" on the in-sight display.
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A highly precise range measurement is made possible
through the use of a novel and efficient timing circuit
which makes use of the instrument's internal central
processing unit crystal oscillator. A likewise unique
automatic noise threshold determining circuit allows for
instrument operation with a low signal-to-noise ratio to
optimize sensitivity and performance in conjunction with a
processor based pulse discrimination procedure which,
nevertheless assures accurate range measurements.
The unit herein disclosed can be utilized in a
multitude of endeavors including such recreational
activities as golf where it can be utilized to very
accurately determine the distance to a flag or pin as well
as to trees and other natural objects. The principles of
the invention are further applicable to the design of a
laser based "tape measure" where ranges can be precisely
measured with resolutions of on the order of an inch or
less.
The laser range finder is operative to initially
determine the range to the target to a first degree of
precision and then may be operated to continue to determine
the range to the target to a second higher degree of
precision, which range to the second higher degree of
precision may be displayed in the display accompanied by an
indication of the displayed range having been determined to
the second higher degree of precision.
DETAILED DESCRIPTION OF THE DRAWINGS
The foregoing and other features and objects of the
present invention and the manner of attaining them will
become more apparent and the invention itself will be best
understood by reference to the following description of a
preferred embodiment taken in conjunction with the
accompanying drawings, wherein:
Fig. 1 is a simplified logic block diagram of a laser
range finder in accordance with the present invention
illustrating the significant functional aspects thereof,
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inclusive of a laser signal transmitting and receiving
section, central processing unit and the precision timing
and automatic noise threshold sections thereof;
Fig. 2 is a detailed schematic diagram of the laser
transmit section of Fig. 1 illustrating, inter alia, the
laser signal producing diode and the associated driving and
reference signal producing circuitry;
Fig. 3 is an additional detailed schematic diagram of
the laser receive section of Fig. 1 illustrating, inter
l0 alia, the laser signal receiving diode, transimpedance
amplifier and the precision comparator for establishing the
Vthreshold and RX(Out+) signals for the precision timing
and automatic noise threshold circuits;
Figs. 4 and 5 are further detailed schematic diagrams
of the precision timing section of the laser range finder
of Fig. 1 illustrating the circuit nodes for establishing
the voltages V1 and V2 during the zero, calibration ("CAL")
and laser firing phases of operation;
Fig. 6 is an additional detailed schematic diagram of
the central processing unit ("CPU") portion of the laser
range finder of Fig. 1 illustrating the CPU, associated
oscillator and the in-sight liquid crystal display ("LCD")
for displaying measured distances to an operator of the
laser range finder in addition to the various signals for
operative association with the precision timing and
automatic noise threshold sections thereof;
Figs. 7A, 7B and 7C are individual graphic
representations of the voltages V1 and V2 of certain of the
precision timing section circuit nodes during the zero,
calibration and laser firing phases of operation from which
the values ZeroTIME~ CaITIME and LaserTIME are derived to
enable rapid and accurate calculation of the distance to an
object from the laser range finder; and
Fig. 8 is a final detailed schematic diagram of the
automatic noise threshold section of the laser range finder
of Fig. 1 illustrating the various components thereof as
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The precision timing section 34 provides a number of
signals to the CPU section 28 including a TIMER and /RX
DETECT signals as shown and receives a RUN/CLAMP signal
back therefrom. The CPU section 28 provides a number of
5 signals to the precision timing section 34 including a HOLD
OFF, NORM/CAL, /RESET, and a CAL DITHER signal. The
automatic noise threshold section 36 also receives a number
of inputs from the CPU section 28 including a number of
noise set ("NSET") signals and a REFLECTION MODE signal to
l0 operatively control its function.
With reference additionally now to Fig. 2, the laser
transmit section 18 is shown in more detail. The laser
transmit section 18 receives a transmit ("TX") BIAS signal
on supply line 50 of approximately 110 to 140 volts for
application through resistor 52 to the emitter of
transistor 54. The emitter of transistor 54 is coupled to
its base by means of a resistor 58 which also couples the
collector of transistor 56 to resistor 52. The emitter of
transistor 56 is connected to circuit ground on ground line
60. A capacitor 62 couples the emitter of transistor 54 to
the cathode of the laser emitting diode 20 which has its
anode also connected to circuit ground 60. An additional
diode 64 is coupled in parallel with the laser emitting
diode 20 having its anode connected to the cathode of the
laser emitting diode 20 and its cathode connected to
circuit ground 60. A resistor 66 is placed in parallel
with the laser emitting diode 20 and the diode 64.
A source of +5 volts is also received by the laser
transmit section 18 on supply line 68 through resistor 70.
Resistor 70 is coupled to the emitter of transistor 72 as
well as to circuit ground 60 through a capacitor 74. A
resistor 76 couples the emitter of transistor 72 to its
base which is coupled through resistor 78 to line 80 for
supplying a /FIRE signal to the CPU section 28 (shown in
Fig. 1) .
An additional diode 82 has its anode connected to the
collector of transistor 72 and its cathode coupled to
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well as the signals coupling the same to the laser receive
section and CPU. '
DESCRIPTION OF A PREFERRED EMBODIMENT-
With reference now to Fig. 1, a logic block diagram of
a laser range finder 10 in accordance with the present
invention is shown. The laser range finder 10 includes, in
pertinent part, a main power supply unit ("PSU") 12 as
operatively controlled by a trigger switch 14. The main
power supply unit 12 is coupled to a high voltage ( "HV" )
l0 power supply unit 16 for supplying operating power in
conjunction with the main power supply unit 12 to a laser
transmit section 18.
The laser transmit section 18 activates a laser
emitting diode 20 for directing a laser signal toward an
object in the operation of the laser range finder 10. The
laser transmit section 18 also supplies a /FIRE signal to
the central processing unit ("CPU") section 28 as will be
more fully described hereinafter.
The main power supply unit 12 also supplies operating
- _20 power to a laser receive section 22 which further has as an
input a signal generated by a laser receiving diode 24 as
the laser signal emitted from the laser emitting diode 20
is reflected from an object back thereto. The laser
receive section 22 supplies a Vthreshold signal and
RX(OUT+) signal, respectively, to an automatic noise
threshold section 36 and a precision timing section 34 both
of which will be described in more detail hereinafter._,
The CPU section 28 receives as one input a signal from
a mode switch 26 by means of which an operator can change
the operating mode and functional operation of the laser
range finder 10. An oscillator 30 supplies a clocking
signal to the CPU section 28 as well as to the precision
timing section 34. The CPU section 28 provides an output
indicative of the distance from the laser range finder 10
to an object as sighted through a viewing scope 11 thereof
on an in-sight liquid crystal display ("LCD") 32.
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voltage source and circuit ground 60 through the node
intermediate resistor 142 and resistor 144.
The precision comparator 134 which may, in a preferred
embodiment, comprise a MAX 913 low power precision
transistor-transistor logic ("TTL") comparator available
from Maxim Integrated Products, Inc., Sunnyvale, CA, has
its "V-", "LE" and ground ("GND") inputs connected to
circuit ground 60 as shown. A capacitor 146 couples the "-
" output of the precision comparator 134 to circuit ground
60 as shown. The "O+" output of the precision comparator
134 is supplied through a resistor 148 to line 100 to
provide the RX(OUT+) signal while the "-" output of the
precision comparator 134 is supplied through resistor 150
to line 102 to provide the Vthreshold signal.
With reference additionally now to Fig. 4, a portion
of the precision timing section 34 (shown in Fig. 1) is
illustrated. A CPU clock ( "CLK" ) signal is input to the
precision timing section 34 on line 152 to the CLK input of
a serial in/parallel out shift register 160 from the
oscillator 30 as previously shown in Fig. 1. An additional
input to the shift register 160 is received on line 154
comprising a NORM/CAL signal from the CPU section 28 to the
data set B ("DSB") input thereof. The active low clear
("CLR") input and DSA input are held high as shown.
An additional input to the precision timing section 34
is received from the CPU section 28 (shown in Fig. 1) on
line 156 comprising a /RESET signal for input to the reset
("R") inputs of D type flip-flop 158 and flip-flop 162.
The Q output of flip-flop 158 is supplied as one input to
an invertor comprising a portion of a NAND Schmitt trigger
168 through a low pass filter comprising resistor 164 and
capacitor 166 as shown. The remaining input to the
invertor 168 is connected to a source of +5 volts.
A resistor 172 couples a source of +5 volts to the
collector of transistor 174 having its emitter coupled to
circuit ground. The collector terminal of transistor 174
is coupled through capacitor 170 to the input of the
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circuit ground 60 through resistor 86. A capacitor 84
couples the cathode of diode'82 to the common connected
collector of transistor 54 and base of transistor 56. The
common connected collector of transistor 54 and base of
transistor 56 are coupled through a voltage divider network
comprising resistor 88 and resistor 90 to circuit ground.
A resistor 92 coupled between resistor 88 and resistor 90
provides a REF signal on line 94 for application to the
precision timing section 34 (shown in Fig. 1).
With reference additionally now to Fig. 3, the laser
receive section 22 is shown in more detail. The output
signals of the laser receive section 22 are the signals
RX(OUT+) and Uthreshold provided on lines 100 (Figs. 4 and
8) and 102 (Fig. 8) respectively for application to the
precision timing section 34 and automatic noise threshold
section 36 as previously shown in Fig. 1. A source of +50
volts providing a receive ("RX") BIAS signal is input to
the laser receive section 22 from the HV power supply unit
16 on supply line 104. A low pass filter network 106
comprising resistors 108 and 112 in conjunction with
capacitors 110 and 114 couples the supply line 104 to
circuit ground 60 to provide a bias signal to the cathode
of the laser receiving diode 24. The laser receiving diode
24 has its anode connected to the base of transistor 118
which, in conjunction with transistors 120, 122, and 124
comprises a transimpedance amplifier 116 providing an
output on node 126 which is capacitively coupled to the "+"
input of a precision comparator 134., A source of +5 volts
is input to the laser receive section 22 from the main
power supply unit 12 (shown in Fig. l) for input to the
transimpedance amplifier 116 through a low pass filter
comprising resistor 130 and capacitor 132. The +5 volt RX
supply voltage is also coupled to the V+ input of the
precision comparator 134 through resistor 136 and is
coupled to circuit ground through capacitor 138. The "+"
input of the precision comparator 134 is connected between
the plus 5 volt RX
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invertor 168 coupled to the Q output of flip-flop 158.
Transistor 174 has its based coupled to circuit ground
through resistor 176 and receives a HOLD OFF signal on node
178 received from the CPU section 28.
The flip-flop 158 receives an input to its CLK
terminal on line 94 comprising the REF output signal from
the laser transmit section 18 (shown in Fig. 1). Its data
("D") input is coupled to a source of +5 volts and the Q1
output of the shift register 160 is provided to the active
low set ("S") input as shown. The Q output of flip-flop
158 is supplied as one input to a transmit gate 204 having
its other input coupled to the output of an invertor
comprising an additional NAND Schmitt trigger 202.
Invertor 202 has one input connected to a source of +5
volts and another input connected to the Q output of flip-
flop 162. Flip-flop 162 has its S input coupled to the Q7
output of shift register 160 and its D input connected to
the output of invertor 168. The Q output of flip-flop 162
is supplied on line 184 to comprise a /RX DETECT signal for
input to the CPU section 28 (shown in Fig. 1). The flip-
flop 162 has its CLK input connected to line 100 for
receiving the RX(OUT+) signal from the laser receive
section 22 (shown in Fig. i) which is also supplied as one
input to NAND Schmitt trigger 180. The other input of NAND
Schmitt trigger 180 is connected to line 184 through
resistor 182 and coupled to circuit ground through
capacitor 186. The output of Schmitt trigger 180 is
supplied to the base electrode of transistor 200 which has
its collector terminal coupled to circuit ground. Line
196, comprising an analog-to-digital ("A/D") POWER
CORRECTION signal is supplied to the emitter terminal of
transistor 200 through resistor 198 as well as to the
collector terminal of transistor 190 which is coupled to
circuit ground through capacitor 194. The /RESET signal on
line 156 is supplied to the base terminal of transistor 190
through resistor 188. A source of +5 volts is connected to
the emitter of transistor 190 as well as through resistor
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192 to the base of transistor 190 to provide an operating
bias. '
Referring additionally now to Fig. 5, the remaining
portion of the precision timing section 34 (shown in block
form in Fig. 1) is illustrated. The HOLD OFF signal output
from CPU section 28 to the precision timing section 34 is
supplied on line 258 through resistor 256 to node 178 for
input to the base of transistor 174 (shown in Fig. 4).
The output of transmit gate 204 appearing on node 206
is supplied through resistor 208 to the base terminal of
transistor 210. A source of +5 volts is supplied to the
emitter terminal of transistor 210 through the series
connection of resistor 216 and resistor 222. The node
intermediate resistors 216 and 222 is coupled to circuit
ground through the parallel combination of capacitors 218
and 222 as well as to the output of comparator 236 through
resistor 246 to provide aTIMER signal on line 250 for
input to the CPU section 28 as will be more fully described
hereinafter. The source of +5 volts is also connected to
the base terminal of transistor 210 through the series
connection of resistors 216 and 224. A V1 node 228 at the
common connected base of transistor 212 and emitter of
transistor 214 is coupled through a source of +5 volts
through resistor 216 and resistor 226. Node 228 is
connected through resistor 230 to V2 node 232 which, in
turn, is connected to circuit ground through resistor 240.
A capacitor 238 couples V1 node 228 to circuit ground. V2
node 232 is connected to the "-" input of comparator 236.
V1 node 228 is connected to line 254 from the CPU section
28 (shown in Fig. 1) to receive the CAL DITHER signal
through resistor 252.
The collector terminal of transistor 210 is coupled to
the collector terminals of transistors 212 and 214 as well
as to the "+" terminal of comparator 236 which, in turn, is
coupled to circuit ground through capacitor 244. A
RUN/CLAMP signal output from the CPU section 28 (shown in
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Fig. 1) is furnished on line 260 through resistor 248 for
input to the base terminal of transistor 214.
With reference additionally now to Fig. 6, the CPU
section 28 is shown in greater detail. The CPU section 28
5 comprises, in pertinent part, a microcomputer 270 which
may, in a preferred embodiment, comprise a ST6240 device.
An 8 megaHertz ("MHz") crystal 274 forms a portion of the
oscillator 30 for providing an oscillator ("OSCIN") and
oscillator out ("OSCOUT") signal to the microcomputer 270
l0 as well as supplying a CPU CLK signal on line 152 for input
to the precision timing section 34 as previously described.
The VDD input of microcomputer 270 is coupled to a source
of +5 volts and the /RESET input thereof is held high
through pull up resistor 276 which is coupled to circuit
15 ground through capacitor 278. Output from the
microcomputer 270 is taken on a display bus 280 comprising
the communication ("COM") lines COM 1-COM 4 and S16-S28
lines for input to the LCD display 32.
An A/D LOW BATTERY signal, a TRIGGER signal, and a
POWER CONTROL signal are input to the microcomputer 270 on
lines 284, 286, and 288 respectively. The A/D LOW BATTERY
signal on line 284 is also supplied to the "-" input of
comparator 296 which is coupled to circuit ground through
capacitor 304. The "+" input of comparator 296 is coupled
to a source of +5 volts through resistor 298 which is also
coupled to circuit ground through the parallel combination
of resistor 300 and capacitor 302. The output of
comparator 296 appearing on line 306 provides a SHUTDOWN
signal for the laser range finder 10 in the event the
onboard battery voltage drops below a predetermined limit.
The microcomputer 270 supplies the HOLD OFF signal on
line 258, the RUN/CLAMP signal on line 260, the CAL DITHER
signal on line 254, the /RESET signal on line 156 and the
NORM/CAL signal on line 154 for input to the precision
timing section 34 as has been previously described. The
microcomputer 270 receives as outputs from the precision
timing section 34 the /RX DETECT signal on line 184 and the
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TIMER signal on line 250. Additional inputs to the
microcomputer 270 are the /FIRE signal on line 80 from the
laser transmit section 18 (shown in Fig. 1) as well_as the
A/D POWER CORRECTION signal on line 196 from the-precision
timing section 34 (as shown in Fig. 4). A MODE input
signal on line 294 is received from the mode switch 126
which is otherwise held to a +5 volts through resistor 292.
Microcomputer 270 supplies an NSET1 and NSET2 signal on
lines 308 and 310 respectively as well a REFLECTION MODE
signal on line 312 for input to the automatic noise
threshold section 36 (as shown in Fig. 1).
In overall operation, a reference signal (REF) on line
94 is generated by the laser transritit section 18 (shown in
Fig. 2) when the laser range finder 10 is fired by placing
a current pulse through the laser emitting diode 20 in
response to manual actuation of the trigger switch 14.
The REF signal on line 94 is derived from the current
placed through the laser emitting diode 20 and not from the
light pulse itself and is sufficiently precise for
accurately indicating the time of the laser firing. The
REF signal is ultimately input to the CLK input terminal of
flip-flop 158, which has its Q output coupled to the
transmit gate 204, which then turns on the current-switch
comprising transistor 210, and starts charging the
capacitor 244. When the receive pulse RX(OUT+) on line 100
comes back from the laser receive section 22 (shown in Fig.
3), it triggers the flip-flop 162 at its CLK input. Flip-
_ flop 162 has its Q output coupled to the input of invertor
202 which then shuts the transmit gate 204 off, stopping
the current pulse. At this point, a constant current sink
discharges capacitor 244.
In this manner, capacitor 244 is charged up with a
relatively large current, on the order of 10 milliamps, and
later discharged with a small current, on the order of 10
microamps, applied over the entire flight time of the laser
pulse from its firing from the laser emitting diode 20 to
its reflection from a target back to the laser receiving
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diode 24. Because the laser range finder 10 is intended
for a shorter maximum range than other laser based range
finding instruments, the use of this technique doss not
require a separate counting oscillator followed by an
interpolation operation and the entire flight time is
essentially stretched by a factor of 1000 and then the
stretched result is counted. By charging capacitor 244 at
a fast rate and then discharging it and then monitoring the
time it takes to discharge, the flight time is expanded so
that the slower clock in the CPU section 28 can then count
it accurately. The microcomputer 270 utilized in the CPU
section 28 has a 1.5 microsecond resolution and, because
the incoming flight time has been expanded by a factor of
1,000 on the input side to the precision timing section 34,
it is the equivalent of a 1.5 nanosecond resolution, which
corresponds to a measurement resolution for the laser range
finder 10 of on the order of nine inches. Therefore, given
that the laser range finder 10 is intended to be a one-yard
instrument with a nine-inch resolution, sufficient
resolution is provided to be able to measure distances up
to a thousand yards to a one-yard accuracy.
The precision timing section 34 of the laser range
finder 10 has three distinct modes of operation including
a zero calibration, fixed pulse width calibration and laser
measurement function as will be more fully described
hereinafter. The portion of the precision timing section
34 comprising transistors 210, 214, and 212 (shown in Fig.
5) is the essence of the integrating flight time expander.
Transistor 210 functions as a current switch which is
turned on for the duration of the laser flight time in the
laser mode of operation and is also turned on for the
duration of whatever calibration pulse is placed into it
during the calibrate mode. In the latter instance, a
calibration pulse is supplied by the shift register 160 via
flip-flop 158 and the start and end of the calibration
pulse is gated via transmit gate 204 to actually turn the
transistor 210 on and off in order to function as a current
CA 02203278 1997-04-21
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:~1 s AU G 1996
18
source, typically sourcing 10 milliamps of current. It
should be noted that prior td turning transistor 210 on,
transistor 214 must first be turned off and, when the
system is in the reset state ready to start 'the whole
measurement sequence, transistor 210 is off. Transistor
212, which is the current sink in the system, is always on,
and typically sinks on the order of 10 microamps of
current. In the reset condition, transistor 214 is on, and
that clamps the voltage at the top plate of capacitor 244
to a voltage level designated as V1 at node 228. A voltage
V2 is defined as the voltage at node 232 at the "-" input
of comparator 236. It should also be noted that a metal
oxide semiconductor field effect transistor ("MOSFET") may
be utilized for transistor 210 and would exhibit a much
lower offset than the bipolar device shown. However, due
to the lower cost of bipolar transistors and the fact that
any offset cancels during the processing of the signal, a
bipolar transistor is entirely adequate for this purpose.
When transistor 214 is on, the voltage on the positive
plate of capacitor 244 is clamped to voltage V1, plus a
fixed offset due to the transistor 210, which is small and
typically on the order of 50 millivolts. During the zero
calibration function, transistor 214 is turned on by
holding the RUN/CLAMP signal on line 260 high, thereby
applying a positive current to its base through resistor
248. To initiate the zero calibration, the TIMER signal on
line 250 is asserted and supplied to the microcomputer 270
of the CPU section 28. Utilizing the ST6240 unit shown in
Fig. 6, when the microcomputer TIMER pin is held high, the
device is counting. Conversely, the microcomputer stops
counting when the pin is allowed to go low. In operation,
the output comparator 236, determines whether or not the
voltage at the top plate of capacitor 244 is greater or
less than V2, and its output determines whether the TIMER
pin on the microcomputer 270 is high or low. In the normal
reset condition, the output of the comparator 236 is high,
which means the timer is active. In sequence, the
CA 02203278 1997-04-21
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19
microcomputer 270 initiates the TIMER function and then
turns off transistor 214 by 'lowering the control signal
RUN/CLAMP on line 260, to unclamp capacitor 244. Capacitor
244 then starts discharging towards zero due to the current
being drained out of it via transistor 212 at a rate of
about ten microamps. When it has discharged such that the
charge removed drops the voltage V1 at node 228 to the
level of V2, the output of the comparator 236 changes state
to stop the TIMER function. (In the particular embodiment
shown, V1 is typically on the order of 1.0 volts and V2 is
about 0.9 volts.) The microcomputer 270 of the CPU section
28 now has a count value that relates to the amount of time
it takes for capacitor 244 to discharge from V1 down to V2.
This process is repeated several times and the result is
averaged. Typically ten iterations may be performed with
the results accumulated and,an average time computed.
As shown particularly with respect to Fig. 5, the CAL
DITHER signal on line 254 is applied to the base terminal
of transistor 212 and is utilized during both the zero
calibration and fixed pulse width calibration times and
incorporates a relatively high value resistor 252. The CAL
DITHER signal allows for the introduction of a deliberately
controlled change in the discharge current in order that
the resultant count will vary slightly such that when the
total counts are averaged together, a finer resolution is
produced than would be the case merely using a fixed
current to get the same count value. An adjustment of one
part in about a thousand is provided during the zero
calibration and fixed pulse width calibration modes because
the finite resolution of the micr=computer 270 timer
otherwise provides discrete timing intervals of 1.5
nanoseconds which would only provide distance measurement
resolution of approximately one yard. In operation, the
zero calibration count in the microcomputer 270 will
typically be about 150 while in the fixed pulse width
calibration mode it will be on the order of 900. The
flight time count during the laser mode of operation can be
CA 02203278 1997-04-21
WO 96/22509 PCT/US95/15438
anything from close to the zero calibration value to about
4500.
For example, during the zero calibration mode, the
count value in the microcomputer 270 might be 150 but there
5 is no way of knowing just how close the count actually is
to 149 to 151. By utilizing the CAL DITHER signal to force
the count over a couple of count boundaries (for example:
150, 150, 150, 151, 151, 152) the resolution of the counter
may be effectively raised by a factor of two without having
10 to utilize additional fine counters. In the embodiment
shown, the resultant resolution is sufficient to maintain
calibration to plus or minus one yard over a range of one
thousand yards or less. Although implementations may vary,
the CAL DITHER signal may be held high for five out of ten
15 pulses and low for the remainder to provide the foregoing
resolution enhancement.
Due to the fact that the actual laser flight time varies
due to noise in the laser pulses and variability in target
aiming, there is generally enough scatter in the measured
20 laser flight time such that it covers more than one clock
boundary and so will automatically average to a higher
resolution through the use of the precision timing section
34 without invoking the CAL/DITHER function in the laser
mode of operation.
With reference additionally now to Figs. 7A, 7B and
7C, the operation of the precision timing section 34 is
shown in the zero calibration, fixed pulse width
calibration and laser measurement function modes of
operation respectively. In its normal state, the voltage
on the top plate of capacitor 244 is clamped at V1, and at
a time T0, the precision timing section 34 will initiate
the TIMER by changing the output state of comparator 236 to
the logic high state. After a very short fixed number of
instructions later shown as T1, the clamp transistor 214
will be turned off and the voltage on capacitor 244 will
begin discharging slowly until that voltage crosses V2 at
time T3 when the output of comparator 236 will change
_ CA 02203278 1997-04-21 -
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state. In essence, during the zero calibration process,
transistor 210 is never turned on thereby determining the
timing conditions of what would effectively be a zero
flight time. Therefore, if there is no charge. current
applied to capacitor 244, T3 - TO zero is the time that
would be in the microcomputer 270 and the timer in whatever
units they operate, which is usually dependent on the CPU
section 28 crystal frequency. In the embodiment shown, the
microcomputer 270 utilizes an 8 MHz crystal and the
internal timer has a 1.5 microsecond resolution resulting
in a count of about 150.
During the fixed pulse width calibration process
(shown particularly in Fig. 7B) at time T4, once again the
microcomputer 270 stops the TIMER and a short time later at
T5 it releases the clamp. At T6, a known pulse width is
applied to the base terminal of transistor 210 which is
precisely derived from the main oscillator 30 as applied to
the CLK input of the shift register 160. The signal
applied to the CLK input of the shift register 160 directly
tracks the main oscillator 30 and the serial data input to
the shift register 160 is a logic line 154 from the CPU
section 28 designated NORM/CAL. When the NORM/CAL signal
is high, the precision timing section 34 is in its-normal
mode of operation and, when it drops to a logic low state,
the fixed pulse width calibration function is initiated.
Thereafter, typically about fifty microseconds later, at
time T6 the NORM/CAL signal on line 154 will be dropped
low. It should be noted that during both the zsro and the
fixed pulse width calibration modes, the logic reset signal
/RESET on line 156 is held low, its active state. In the
logic low state the two flip-flops 158, 162 determine
whether the input signal comes from shift register 160
which generates the fixed pulse width or whether it comes
from the REF and RX(OUT+) signals an relates to an actual
laser flight time. The /RESET signal is generally held low
at all times during the fixed pulse width calibration
process so that any noise on the RX(OUT+) receive line 100
CA 02203278 1997-04-21
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22
will not accidently clock flip-flop 162 and therefore
trigger the precision timing section 34 resulting in an
indeterminate time period measurement invalidating the
calibration. The reset state for the Q outputs of flip-
s flops 158, 162 is low but is high for the Q outputs.
Therefore, the Q outputs can not be directly driven with
the reset circuit and must be driven off the Q outputs in
both cases which introduces a small fixed offset delay
which must be accounted for later. As soon as the NORM/CAL
signal on line 154 is dropped low, which occurs
approximately 50 microseconds after the clamp has been
released, the low signal propagates through the shift
register 160 precisely with the main oscillator 30 clock.
The QO output of the shift register 160 is the first to be
triggered but is not used because it is used to synchronize
with the incoming signal. The Q1 is then the first output
of the shift register 160 to be utilized and on every
positive edge of the clock the zero signal that is applied
into the serial input will propagate one state of the shift
register 160 from Q zero to Q7. Therefore, the Ql output
will go low first, and as soon as that output goes low, the
set line input S forces the Q output of flip-flop 158 to go
high since the Q output of flip-flop 162 is in the low
state. As a result, logic level ones appear at the two
inputs of the transmit gate 204, which turns on the current
switch transistor 210. Exactly six clocks later, the same
thing happens with flip-flop 162 which has its S input
coupled to the Q7 output of the shift register 160. As the
Q output of flip-flop 162 goes high, the output of the
invertor 202 goes low, and the transmit gate 204 will be
turned off. At this point the count pulse will stop
meaning that the fixed width pulse feeding the current
switching circuit at the output of the transmit gate 204 is
precisely six clock cycles. The time difference between
the Q1 and Q7 outputs of the shift register 160 is exactly
750 nanoseconds when utilizing an 8 MHz oscillator 30
applied to its CLK input. The invertor 202 adds an
CA 02203278 1997-04-21
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23
additional delay of about 10 nanoseconds for a total of
delay of about 760 nanoseconds which varies only slightly
with temperature, perhaps one or two nanoseconds, yet still
provides sufficient precision for measurements of less than
one yard resolution.
Transistor 210 is then turned on for a period of time
between T6 and T7 to enable the capacitor 244 to charge
very rapidly and then discharge at the same rate as has
been previously shown with respect to Fig. 7A. As V1
reaches the level of V2 the TIMER signal goes low at Time
T8. The fifty microsecond delay between the unclamping at
T5 and T6 is to allow the clamp transistor 214 to turn off
fully since it is a relatively inexpensive bipolar device.
If a MOSFET were used instead, its turn off would be
virtually instantaneous and the additional delay it
introduced would not be a problem because the microcomputer
270 could not issue the next instruction quickly enough.
Utilizing a bipolar device, approximately 20 microseconds
are required for the discharge to become linear and the
slope of the discharge curve between T7 and T8 is then
___ _ identical to the slope from T1 to T3 in the zero
calibration mode except for the step due to the charging of
capacitor 244. As a consequence, the value of ZEROTIME
equals T3 minus TO and the value of CALTIME value equals
the time due to the CALTIME value not due to the ZEROTIME
value, which is, T8 minus T4 minus the ZEROTIME value or,
T$ minus T3.
In essence then, very small flight times are
effectively disregarded and the value of CALTIME is known.
Therefore, with the zero calibration function and the
addition of a known calibrated pulse width, the time delay
at zero is known together with the time delay for the known
pulse width providing the origin and scale for determining
distance with a constant linear discharge of capacitor 244.
With particular reference additionally to Fig. 7C, the
operation of the precision timing section 34 is shown in
zhe laser measurement mode of operation. The laser
CA 02203278 1997-04-21
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24
measurement operation is essentially the same as the fixed
pulse width calibration mode' except that the NORMAL/CAL
signal on line 154 to the shift register 160 is held high
and the /RESET signal on line 156 is taken high at.time T9
to enable the flip-flops 158, 162 to trigger. At time T10
the timer is started and at T11, (at precisely the same
relationship T11 minus T10 equals T5 minus T4 equals T1
minus TO) the clamp is released. There is normally a fifty
microsecond wait and then the laser pulse is fired when the
microcomputer 270 asserts the /FIRE signal on line 80 to
initiate the firing sequence. Upon firing the laser
emitting diode 20, the laser transmit section sends the REF
signal on line 94 to the CLK input of flip-flop 158 of the
precision timing section 34. This opens the transmit gate
204 which turns on the current source transistor 210,
which, in turn, charges capacitor 244 at a known rate.
When the reflected laser pulse is detected by the
laser receiving diode 24 of the laser receive section 22
(shown in Fig. 3), the RX(OUT+) signal on line 100 is
directed to the CLK input of flip-flop 162. The Q output
- signal of flip-flop 162 is inverted by invertor 202 which
turns off the transmission gate 204 so that the current
source transistor 210 is on for the flight time duration of
the laser pulse to charge capacitor 244 to a level
determined by the timer during that flight time. The
charge applied to the capacitor 244 may be anything from
just a few millivolts (essentially zero distance and,flight
time) to up to two volts (maximum range and flight
distance) depending on the distance to the target. Time T12
represents the firing of the laser as indicated by the REF
signal and T13 represents the receipt of the reflected
laser signal as indicated by the RX(OUT+) signal.
Transistor 210 is turned on at T12 and turned off at T13'
As a consequence, V1 will equal V2 at anytime between T14A
(minimum distance when T12 and T13 are essentially
coincident) and T14B (maximum range of the laser range
finder 10). Times T14A through T14B represent the range of
- CA 02203278 1997-04-21 ,
~'~~9~1.1,5~~~
~J i ~ ~-1 ~.;u I
times (depending on the distance to the target) when the
value of V1 is discharged below the level of V2 and the
comparator 236 output changes state stopping the timer.
The actual laser flight time LASERTIME (or F'LIGHTTIME)
5 then equals T14A (or T14B) minus T10 minus ZEROTIME or, T14
minus T13. The time T8 has to be greater than T3, and T14
is greater than or equal to T3. There is no theoretical
limit on the lower range of the laser range finder 10 and
flight time (and distance) can be measured down to zero due
10 to its linearity. The only factors in the near zero range
are the time it takes transistor 210 to turn on, the
propagation time of the laser beam and the various circuit
gates, but since the time for each of these factors is the
same during calibration as during flight time, they
15 essentially cancel out. The precision timing section 34
can be effectively utilized down to on the order of ten
nanoseconds and still remain perfectly linear. RANGE to a
target is then a constant, "k" times the quantity
FLIGHTTIME - ZEROTIME over CALTIME - ZEROTIME~
20 For each of the values: ZEROTIME, C~'TIME and
"__ _ FLIGHTTIME values are accumulated and are expressed in time
units that derive from the very accurate crystal oscillator
30. Typically, ten pulses may be utilized to establish the
ZEROTIME average, ten pulses to establish the CALTIME
25 average and ten pulses to establish the minimum precision
(or rough) FLIGHTTIME range to the target. Another group
of ten through thirty laser pulse FLIGHTTIMEs may be also
averaged in order to obtain a higher precision distance to
a target as indicated by a "precision flag" which may be
displayed on the LCD display 32 within the laser range
finder 10 eyepiece. Nevertheless, the actual values
derived in these time expansions will, of course, vary with
time, temperature and aging and affects the gain of the
transistors, the leakages, as well as the value of the
resistances and capacitances. Initially the exact values
of these effects are completely unknown but, through the
use of the zero and calibration functions above-described,
CA 02203278 1997-04-21
26
PCT~I~ ~~ 5 / .15 ~+ 3 8
~~f:~i ~ ~; ~ ~ ~ 9'~
the zero problem has been eliminated, and a crystal
reference calibration has been provided for the entire
flight time without having to resort to a complicated
counter circuitry.
Another aspect of the precision timing section 34 is
the automatic set noise control and invertor 168 which
provide, in conjunction with other circuit elements, a
hardware hold off function. Upon firing of the laser and
receipt of the reference signal REF on line 94 at the CLK
input of flip-flop 158, a certain time must elapse, as
determined by the time constant of resistor 164 and
capacitor 166, before the D input goes nigh. Until that
time, all noise pulses and/or early laser pulses on the
clock line are ignored. The purpose for this function is
that, when the laser fires, it generates unintended ground
bounce and noise that may prematurely trigger the receive
flip-flop 162 rather than the real laser return signal
RX(OUT+). For that reason, a hold off period is provided
corresponding to the minimum range of the laser range
finder 10 and, as an example, considering a minimum range
of about twenty yards, the holdoff time is approximately 60
nanoseconds. With a lower sensitivity laser range finder
10 utilized at shorter ranges the function can be
eliminated and it is clearly most useful with a high
sensitivity receiver where the noise from the firing
circuit determines an effective minimum range.
Transistor 174 provides an additional function and
allows the microcomputer 270 to extend the hold off range
by asserting the HOLD OFF signal on line 258. In this
manner, the minimum range of the laser range finder 10 may
be extended out to, for example, sixty or eighty yards,
whatever is the desirable setting. This microcomputer 270
hold off function may be implemented by the mode switch 126
and would allow shooting through branches, twigs,
precipitation or other partial obstructions. By extending
the hold off range out beyond such partial obstructions,
there is insufficient back scatter from the obstructions to
trigger the precision timing section 34 and the measurement
CA 02203278 1997-04-21
".f ° t' t~ (~ ~ / ~ ~ ~ .7 r1 _
~'e z~~ 6 AUG 1~
27
will be made to the desired target instead of the
intervening obstructions. This is accomplished by not
allowing flip-flop 162 to trigger until a set timer -period
has elapsed. Transistor 174 is the switching device
utilized to allow setting of an extension to the hold off
range and gate 180 is used to determine the receive pulse
width in conjunction with the discharge rate of capacitor
194. This allows the microcomputer 270, which has a built
in analog-to-digital ("A/D") convertor, to determine the
residual voltage on capacitor 194 and therefore derive a
measure of the pulse width, (which is a measure of the
return signal power) and thus use an internal lookup table
to correct for that power variation and get a higher range
accuracy. When the logic reset signal /RESET on line 156
is low, transistor 190 clamps capacitor 194 to the +5 volt
rail. During the laser measurement routine, the transistor
190 is turned off. When a pulse subsequently arrives, that
bit turns on transistor 200 and the voltage in capacitor
194 will be discharged via resistor 198 for the duration of
that pulse. The charge on capacitor 194 is then digitized
by the processor to determine the effect of incoming power.
With reference additionally now to Fig. 8, the
automatic noise threshold section 36 of the laser range
finder 10 is shown. The automatic noise threshold section
36 receives the RX(OUT+) signal from the laser receive
section 22 (shown in Fig. 1) on line 100 for input thereto
through resistor 314. Resistor 314 is connected to the
anode of diode 316 which has its cathode connected to the
~'+" input of operational amplifier ("OpAmp") 318 forming a
V3 node 320. V3 node 320 is coupled to circuit ground
through the parallel combination of resistor 322 and
capacitor 324. The output of OpAmp 318 is coupled back to
the "-" input thereof as well as to line 102 through
resistor 326 for supplying the Vthreshold signal to the
laser receive section 22 (shown in Fig. 1) . Line 102 is
connected through resistor 330 to the center tap of
potentiometer 332 which has one terminal thereof connected
to a source of +5 volts through
CA 02203278 1997-04-21 n
~951.15!~~~
~ ~~~,~t~ 1 s au~ 1996
28
resistor 334 and another terminal thereof coupled to
circuit ground through resistor 336.
Lines 308 and 310 from the microcomputer 270 (shown in
Fig. 6) are connected through resistors 338- and 340
respectively to line 102. Additionally, line 312 from
microcomputer 270 is connected to line 102 through resistor
342 as shown.
In operation, the automatic noise threshold section 36
in conjunction with the CPU section 28 (shown in Fig. 6)
provides a simply implemented yet highly effective
threshold adjustment to the laser receive section 22 (shown
in Fig. 3). As shown in Fig. 3, the laser receiving diode
24 utilizes a high-voltage source (of about 50 volts)
supplied via a noise filtering network, comprising low pass
filter network 106, to bias it. The diode 24 responds with
an output current proportional to the incoming laser light
which is generally a short duration laser pulse producing
a short current pulse which is amplified by transistors
118, 120, 122, 124, comprising the active circuit elements
of a transimpedance amplifier 116. The transimpedance
,_ amplifier 116 produces an output voltage pulse proportional
to the incoming laser pulse impinging on the laser
receiving diode 24. The output of the transimpedance
amplifier 116 is capacitively coupled to the "+" input of
comparator 134, which is a high speed comparator. When the
laser pulse input to the "+" input crosses a threshold
determined by the voltage on the "-" threshold pin, a
positive output pulse is produced. =,
To maximize performance, the threshold of the
comparator 134 has to be set for maximum sensitivity in
order detect the weakest possible laser pulse to get the
maximum performance out of the laser range finder 10.
Conventional approaches include using digital controls or
a potentiometer to adjust the threshold. However, these
approaches have the down side that over time and
temperature changes the gain of the receiver will change
CA 02203278 1997-04-21
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29
PCT/US95115438
with the background noise generated by the background light
rendering a fixed threshold as less than an ideal solution.
The automatic noise threshold section 36 of Fig. 8
discloses a circuit that automatically sets a threshold
such that a constant noise pulse firing rate is output from
the detector comprising resistor 314, diode 316, capacitor
324 and resistor 322. In operation, when the threshold pin
of the comparator 134 (Fig. 3) is at a considerably higher
voltage than the input pin, no noise pulses will appear at
the output due to the inherent amplifier and optically
generated noise. As the voltages on the threshold and
input pins are brought closer together, noise pulses will
appear at the output and, when the voltage levels are
nearly coincident, a great deal of noise can be seen. In
essence then, the automatic noise threshold section 36 sets
the noise pulse rate at that point at which, given the
right firmware algorithm, one can still acquire the target
and not be blinded by the noise. The higher the noise that
can be tolerated, and the closer the voltage levels at the
threshold and input pins of the comparator 134, the weaker
the laser pulse that can be detected. The automatic noise
threshold section 36 automatically adjusts that threshold
level to maintain constant noise pulse firing rate.
As shown in Fig. 8, this is accomplished by monitoring
the digital logic receive signal RX(OUT+) on line 100 that
goes to the receive f lip-flop 162 (shown in Fig. 4). The
detector monitors line 100 for the presence of noise pulses
via a detector comprising the aforementioned resistor 314,
diode 316, capacitor 324 and resistor 322. The value of
resistor 322 is typically considerably greater than that of
314, on the order of a 150:1 ratio. The peak amplitude of
the noise pulses is typically at or near the logic
threshold, except for very narrow pulses where the
comparator will not reach full amplitude, however, the
width of these pulses is going to vary randomly because it
depends on the noise signal that is being detected.
Moreover, the spacing of the noise pulses will also vary at
CA 02203278 1997-04-21
p~~lU~ 9 5 /.15 ~ ~ 8
a random rate, but, for any given threshold setting, there
will be a fixed average rate. The average rate is
dependent on the threshold. Therefore, during the time the
pulse is high, capacitor 324 charges via resister 314 and
5 diode 316 at a rate determined by the high on the logic
pulse, resistor 314 and whatever voltage is still existing
on capacitor 324.
Initially, capacitor 324 is charged as follows. Once
the noise pulse terminates, the logic line goes back to
10 zero. There is a residual voltage on capacitor 324, diode
316 will be reverse biased, and the discharge path is now
via resistor 322. (As previously described, the value for
resistor 322 is chosen to provide a relatively longer time
constant, a factor of 150.) When another pulse comes in,
15 capacitor 324 will charge a bit more. What will then
happen is, quite rapidly, (i.e. within a few milliseconds)
the voltage across capacitor 324 stabilizes at a rate that
is proportional to the average firing rate. The reason for
having a large ratio between resistor 314 and resistor 322
20 is because the noise pulses typically may average 50
nanoseconds wide, and the averaged time between them to
maximize the sensitivity of the laser range ffinder 10
should be of the order of two microseconds or so . As an
example, if a 50% voltage were desired, and the high state
25 was occurring for 50 nanoseconds while the low state
average was occurring for one microsecond, a 20:1 ratio
would be produced. Nevertheless, the optimum ratio has
been determined empirically to be about 15:1 as previously
described and is related to average pu~~e widths (typically
30 on the order of 30 nanoseconds in length) and pulse
repetition rates (on the order of 4 microseconds) with a
typical voltage level of 1.5 volts.
Op amp 318 is configured as a unity gain buffer,
although it need not be unity gain, with a voltage V3 at
its "+" input pin on node 320. The input is high impedance
and the output is low impedance in order to drive external
circuitry. The voltage that is derived at the output of
CA 02203278 1997-04-21
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31
the op amp 318 is then fed into a resistor network
comprising resistor 338, resistor 340, resistor 342 and
resistor 330. A summing node of the resistor network on
line 102 goes to the threshold control to provide the
signal Vthreshold to the laser receive section 22 (shown in
Fig 3). Resistor 330 is connected to the center tap of a
potentiometer 332 so that the DC voltage on the other end
of resistor 330 can be controlled.
In combination, the circuit comprises a feedback
network such that, if there are no noise pulses, then V3 is
zero and Vthreshold and drops to a low value. Initially,
Vthreshold will be higher, and the "-" input of comparator
134 (shown in Fig. 3) will be higher than the "+" input,
forcing a logic low on the output as the starting state.
As the level of V3 on node 320 falls, the voltage level on
the "-" pin of comparator 134 starts approaching the level
of the signal from the transimpedance amplifier 116 on the
positive "+". When it approaches the noise zone, noise
pulses start appearing. As soon as noise pulses start
appearing, a charge appears on node 320, so V3 charges.
When V1 and V3 match, the feedback point is reached and the
charging stops. Basically, the voltage on the threshold is
set at such a point that the noise firing rate maintains V3
at that voltage which is necessary to maintain Vthreshold'
Because very small changes in Vthreshold make a very large
change in the noise firing rate, typically, a ten millivolt
change in Vthreshold well change the voltage V3 at node 320
by about a volt. Wha_~:,is produced then, is a fairly high
gain feedback loop, such that Vthreshold will track very
;0 closely the noise firing rate and V3 will stabilize very
accurately and rapidly. This further provides the
capability to adjust the noise firing rate by controlling
the bias and forcing V3 to compensate. The voltage V3 at
node 320 then represents the noise firing rate.
NSET1 line 308 and NSET2 line 310, are two control
lines from the microcomputer 28 such that when held low or
CA 02203278 1997-04-21
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32
high, they adjust the noise rate to obtain the maximum
range to different reflectivity targets. If both lines 308
and 310 are taken high, V3 will drop to compensate to
maintain a constant threshold noise. Similarly,
potentiometer 332 provides an adjustment such that the
threshold point may be set together with the level of V3.
Typically, the V3 point might be set equal to: 0.5, 1.0,
1.5 and 2.0 volts as desirable choices for the average
noise firing rates. As such, since resistor 338 is
approximately twice the value of resistor 340, four voltage
combinations are obtained roughly equally spaced in voltage
by half a volt. Potentiometer 332 is used to set the first
voltage level to .5 or the last one to 2.0 while the
intervals are determined by the logic control lines 308 and
310 set NSET1 and NSET2. Obviously, this approach could be
extended, four combinations provides adequate resolution in
the particular implementation of the laser range finder 10
described and shown. When both lines 308 and 310 are high,
there is a current injected into the node comprising the
Vthreshold line 102, and to compensate for that, V3 must
-- drop, so less current flows through resistor 326 and vice
versa. V3 will follow these values, depending on the
permutations of logic high and low signals on the lines 308
and 310. Resistor 330 is used just to set where this whole
block resides while potentiometer 332 is used to establish
the initial set point. Since the noise characteristics
from unit to unit will vary somewhat, potentiometer 332
eTUables the setting of the initial device characteristics.
Resistor 342 is of a considerably lower value than
resistors 338 and 340 and its value is chosen such that,
when the REFLECTION MODE signal on line 312 is asserted by
being taken high, V3 will drop to zero and will stay there
because it cannot go below zero. At this point, the
feedback loop is saturated and is no longer effective, so
Vthreshold is no longer stabilized. In operation, line 312
will be pulled high by a considerable voltage, on the order
o~ .4 volts, such that it completely desensitizes the
CA 02203278 1997-04-21
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au v ~
33
laser receive section 22 so the laser range finder 10 will
then only respond to a retro'reflector. In this mode of
operation the receiver is detuned and its non-cooperative
range drops from 500 yards down to about 30 or- 40 yards,
such that the laser range finder 10 only latches onto a
retro reflector or survey prism comprising a high grade
reflector that returns the laser energy back to the source.
Possible applications also include determining the distance
to a particular golf hole where a laser reflector is
attached to the pin and the signal might otherwise be
actually returned from trees behind or in front of the
green in a more sensitive mode of operation.
The essence of the automatic noise threshold section
36 is, as previously described, a feedback loop comprising
the detected average noise firing rate forming a feedback
loop that controls the threshold. Use of this circuit has
resulted in an addition of almost 50% to the range of the
laser range finder 10 compared to attempting to manually
set the threshold. By setting the noise firing rate,
noise pulses are being produced deliberately, all the time
_ and the only way to take advantage of that fact is by
implementing a firmware algorithm in the microcomputer 270
discriminates between noise pulses and laser return pulses.
The algorithm operates as follows: During the laser firing
process, on the first pulse that fires, the algorithm gets
a laser pulse, and it places it in a stack of pulses. For
example, the stack may have locations designated 0 through
9, to enable 10 pulses to be maintained in the stack. The
values of the FLIGHTTIME are saved, corrected for power
return, (the microcomputer 270 determines the power level
of the return s igna 1 and corrects the f 1 fight t ime f or power
return) and placed in one of the locations in the stack.
Upon receipt of the next pulse, the microcomputer 270 will
then compare the next pulse with the remaining locations in
the stack. Initially, most of the locations will be empty,
and there will be no match. If no match is found, the
microcomputer 270 puts the pulse in the
CA 02203278 1997-04-21
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34
stack and carries on, merely placing pulses in the stack,
and then when it gets to the top, it goes back and
overwrites the base, so a history of N number of pulses is
developed in the stack. Any time a new pulse comes in, it
compares the entire stack for a match. If N=10, it
searches the preceding ten pulses for a match.
The reason for doing that is, since a high noise
firing rate has been deliberately set to get maximum
sensitivity, many noise pulses are going to have shown up,
but the noise pulses will be of random occurrence and the
chance of a precision match is very low. Because the
tolerance can be set as any other firmware parameter, a
default value will be typically loaded that has been
determined empirically. As an example, a tolerance of~a
few nanoseconds may be set for a match to be assumed to be
a real target and not a ,noise pulse. Utilizing the
algorithm, the process continues, trying to lock on the
target until a match is achieved. The match need only be
two pulses within the preset tolerance (providing very
acceptable results) or, if higher sensitivity were desired,
a match of three through N may be specified, depending on
the reliability needed to guarantee a real target and not
a noise pulse. In an exemplary operation, the first pulse
(pulse 0) could be the real target, followed by eight noise
pulses, and as long as the ninth pulse is again the real
target, the distance to the target can be accurately
determined. The stack can be increased in size up to
whatever memory limit is available in the system, depending
on how far into the noise level the laser range finder 10
must work.
Having found a match, the average of the match values
may then be used to compare all subsequent pulses, rather
than needing to place them in a stack and only pulses that
match up with that initial match average will contribute to
the measurement. If a certain number of pulses elapse
before another matching pulse is received, it may be assumed
that an accidental lock-on to noise has been achieved and
the process restarts. By adjusting the various .
_ CA 02203278 1997-04-21
parameters, a trade off can be made between the time it
takes to get a measurement to how far into the noise the
laser range f finder 10 must work. Because the noise rate
can set to whatever is desired by means of the-automatic
5 noise threshold section 36, it is possible to optimize the
algorithm to provide the optimum acquisition
characteristics against time and against range.
The higher the value of V3, the more noise is coming
out of the receiver, and the more sensitive the laser
10 receive section 22 is running. The probability of a noise
pulse showing up is proportional to the flight time, so
given a very "black" target, the maximum range will be
less, but the maximum flight time is also less, so a higher
noise rate can be tolerated. Therefore, running at a
15 higher gain will provide the best range to a black target.
On the other hand, if the target is very reflective, a high
gain is not required, so the noise rate can be lowered,
which then provides the same probability of a noise pulse
appearing over a longer flight range, and therefore a quick
20 acquisition on a bright white target can be achieved.
Thus, by depressing the mode switch 126, different modes of -
operation of the laser range finder l0 can be selected. As
an example, one mode might be utilized to find the range to
reflective road signs out to a distance of 1000 yards or
25 more. Alternatively, aiming the laser range finder 10 at
something like wet black tree bark, might reduce the
maximum range to only 350-400 yards and so a different
operational mode might be selected which would otherwise
require a relatively long time to hit the road sign, if
30 ever, because there would always be a noise pulse in the
way. The mode switch 126 allows the setting of these
variables to maximize the range of the laser range finder
10, depending on the target quality and a visual indication
of the target quality selected may be provided to the
35 operator on the in-sight, LCD display 32 wherein the first
mode would correspond to the brightest target or most
CA 02203278 1997-04-21
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36
reflective target, and the Nth mode would correspond to the
least reflective target.
