Note: Descriptions are shown in the official language in which they were submitted.
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Distance division multiplexing
BACKGROUND OF THE INVENTION
Technical field
This invention generally pertains to communication of information between a
source and a receiver. More
particularly, it concerns the use of source distance information at the
receiver to ensure maximum bandwidth
of communication and avoid noise and interference from other sources operating
over the same frequencies.
Brief description of the prior art
In his classic paper titled "A mathematical theory of communication" (Bell
System Technical Journal, vol. 27,
pages 379-423,623-656, 1948), Claude E Shannon defined the object of
communication technology as enabling
the transfer of information from a source to a receiver. A perfect receiver
should be accordingly defined as
one that could receive an arbitrary signal f (r, t) from a transmitter at a
relative distance r without noise,
distortion or interference from any other source. Such an ideal receiver is
unachievable by Shannon's theory,
but current technology is especially worse off with respect to the criterion
of interference, because it splits the
available physical bandwidth in some way to keep the signals from multiple
sources separate all the way. The
separation techniques include frequency division multiplexing (FDM) used in
radio and television broadcast,
wavelength division multiplexing (WDM) and mode separation in optical fibres,
spread-spectrum encoding
or code division multiple access (CDMA), or time division multiplexing (TDM)
and its asynchronous variant,
the Ethernet. Recent variations of this theme include blind signal processing
as discussed in the book titled
Adaptive Blind Signal and Image Processing (Wiley, 2002, authors A Cichocki
and S Amari), which uses
statistical analysis to cope with the distortion of the original separation
parameters by the wireless channel,
CA 02575981 2012-11-23
and autocorrelation matching, in which a "prefilter" is applied at the source,
as described by R Liu, H Luo,
L Song, B Hu and X Ling in their paper titled "Autocorrelation - a new
differentiating domain for multiple
access wireless communication", in the Proceedings of ISCAS, 2002. All of
these techniques effectively share
the channel capacity corresponding to the usable physical bandwidth.
Lately, another such idea, using multiple transmitting and receiving antennae
in parallel, is called space
division multiplexing (SDM), as in the articles "Reduced complexity space
division multiplexing receivers"
by C Awater, A van Zelst and R van Nee in the Proceedings of IEEE Vehicular
Technology Conference, May
2000, and "Channel Estimation and Signal Detection for Space Division
Multiplexing in a MIMO-OFDM
System", by Y Ogawa, K Nishio, T Nishimura and T Ohgane in IEICE Transactions
on Communications,
Vol. E88-13, No. 1, January 2005. The usage is debatable, since it concerns
merely using a larger antenna
cross-section to achieve a correspondingly larger channel, i.e. there is no
actual division of space whatsoever,
even though the parallel antennae could be using complementary polarizations,
which, if used to transmit
different channels, would have qualified as a form of space division
multiplexing. However, it would still be
far ambitious than the objects and motivation of the present invention, as
follows.
If only we could string separate cables or fibres between each receiver and
its selected source, the sharing
of the channel capacity would become unnecessary, and the entire capacity of
the cable or fibre link would
become available to each receiver and its respective source. It is desirable
to have a similar capability for
wireless technology, which is being steadily pushed into increasingly higher
frequency bands for bandwidth
as channels compete within the same frequency bands over the same physical
space. The main challenges are
directivity and range selection. The first is partly addressed by using high
operating frequencies so that the
wavelengths are comparable to or less than the receiver dimensions, and partly
by phased array technology,
which enables source direction selection without physically moving an antenna.
There has been no practical
solution for the second, although as range and angle are mutually
complementary as physical dimensions of
space, and on that basis, a similar, receiver-side technology could have
intuitively thought possible.
The present invention is a solution based on a method for enabling a receiver
of electromagnetic or other
waves to determine the distance r to the source of the waves by modifying a
receiver parameter, as described
in copending application titled "Passive distance measurement using spectral
phase gradients",
which issued as U.S. Patent Number 7,180,580.
The method involves varying an
instantaneous frequency selector & at the receiver at a rate a, whereby
frequency shifts aca become induced
in the received waves in proportion to ar, so that r can be computed from a
and bw. This method avoids
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round trip timing (RTT) and coherent phase reference requirements, but it
depends on the phase distribution,
which is already utilized in current technology. For example,
o any kind of modulation as such involves a nonzero bandwidth spread, and
frequency modulation (FM)
especially relates to phase modulation (PM),
o PM itself is also in use, for instance, in data modems as quadrature phase
shift keying (QPSK), and for
the encoding of colour in PAL and SECAM broadcast television formats, and
o in any case, all signal processing, including the autocorrelation
matching and the blind signal processing
methods mentioned above, involve manipulation of the signal phase.
It has riot been obvious, therefore, that this method can be used for source
selection without being impacted
by modulation or signal processing, and without interfering with these
operations. Moreover, in the presence
of a modulating signal, a received carrier is no longer an almost pure
sinusoid, as would appear to be assumed
in the copending application, so that even the inferred distance r(iD) would
vary significantly over the received
modulated carrier bandwidth.
It is in fact generally unobvious how any form of distance determination could
help in signal selection or
source isolation. An independent review of the copending application method
observed, for example, that
timing or coordinate information from the global positioning system (GPS)
could be encoded in transmitted
signals to enable source distance determination without RTT or coherent phase
reference. While the method
could have other applications, it would be specific to signals actually
bearing the encoded coordinate or timing
information and thus less than general. The encoded information would be
generally available only after the
signals are separated, and would be thus useless for the separation itself.
A hitherto unaddressed need exists, therefore, for a method that can separate
signals from multiple sources,
that does not interfere with the signal phase distribution or depend on the
signal form or content, and would
enable the entire physical bandwidth available from a source to be utilized
for communication with only
that source, without interfering with signals from another source. Such a
separation is available for sources
located at different directions from the receiver using phased array antennae
as remarked, but not for sources
along roughly the same direction and differing only in distance. It would be
also desirable to have a method
that can be applied over a large gamut of operating wavelengths, for example,
from radio waves to ultraviolet
frequencies, and even to acoustic waves. It would be additionally desirable
for the method to be also useful
for detecting the presence of multiple sources, i.e. of interference, so as to
enable a receiver to lock on to and
track a selected source.
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SUMMARY OF THE INVENTION
Accordingly, a primary object of the present invention is to provide a very
general mechanism, which would
be largely independent of signal form and content, for separating signals from
multiple sources even when
the sources are located along roughly the same direction from the receiver.
Another object is to enable, at a
receiver, most if not the entire physical bandwidth between a source and the
receiver to be made available for
communication between them, without interfering with communication from other
sources or receivers. A
secondary object is to provide a general means for detecting such interference
and determining the causative
source distance distribution.
Principle of operation
This object, and others which will become apparent, are achieved in the
present invention, within a receiver
receiving an overall waveform comprising a combination of signals Ej F3 (w)
from a multitude of sources sj
at distinct respective distances ri, by applying a succession of
transformations to the received combination
of signals, as will be shortly described. The notations [] and 1[]. will be
used to denote the lower and upper
frequency bounds, respectively, relative to a predefined threshold amplitude
ath, i.e.
IF3(w)12 < lathr if w < L[F3] or co > 1iFI. (1)
Correspondingly, Wi = 711F3] ¨ .C[F31 will denote the respective bandwidths,
so that
(71 ¨ f)[Fj (w)] a 7-1 [Fi(w)] ¨ [Fi (w)]
(2)
A nominal bandwidth W > Wi can be assumed as non overlapping portions of the
spectra would be separable
by filters. The inventive procedure for extracting a specific signal Ft then
comprises the steps of
A. optionally first splitting the combined received signal into n > 1 subbands
of successive widths 131W,
ar,W, i.e. into subbands
n-1
[C L+ =1...n
(3)
0=1 0=1
using a set of subband filters Si.õ so that Ent, Op = 1, and, writing {FA as
short for Ej Fj, we would
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have
n-1
r[SpjFj)] = L[{Fi}] E )304/ and N[Sõ{Fi}1 =- .C[S,{Fi}] +0047;
(4)
A=1.
B. applying to each subband S,{Fi} of the input signal {Fi} a time-varying
sampling or frequency selection
mechanism as described in the copending application, characterized by a
parameter a, independent of
the source, and hence of the subscript j, to cause the subband spectrum to be
linearly shifted to
H (a õ) (w) = F,j(w[1 + i]) whereE.--- SõFj, i.e. by
shifts ii.r.Jj wrj; (5)
C. then applying a selection filter '6,i to the resulting shifted spectral sum
to particularly select
H (a õ)F,i(w) and reject H,Fõi for all j i, i.e.
= 1 (i = j)
Hp ai j E F,õ = H õFõi 7 where 6ii
(6)
0 (i j)
and is approximate because the stop band rejection of real filters cannot be
unity, so that
OH!, HG
(7)
where Gõi Gõ denotes corresponding baseband filters of bandwidths 0,W;
D. applying the reverse mechanism HA-1 H-' (c) = H (- õ) to shift the result
HpFõi back to F,i;
E. and lastly, putting the subbands F,i back together, in reverse to Step A,
to obtain Fi.
Note that Steps D and E can be interchanged, i.e. the subbands can be summed
before applying the reverse
mechanism, if the a, are equal. These steps form successive stages of signal
processing in the receiver. The
essence of the separation, contained in Steps B through D, is summarized by
the following process flow:
H(n,) aõ--
{F} /1õ{F,i} HõGpik{F,j) H,F0 , (8)
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=
and may be alternatively summarized as a product of operations of the form
(9)
corresponding to the orthogonality relation
H1(c) doH(a,) 6ii.
(10)
The utility of the method lies in the fact that the H operations depend only
on a, which are independent of
the signal sources, the latter being distinguished by the indices i or j in
the above equations. The separation
is obtained spectrally via the transformed filters
HõGõi (equation 7), applied in the transformed space
as Hp-low/fp, per equation (9), instead of baseband filters Gõ., which per se
cannot provide the separation.
The design of ow; can be derived from that of G, by frequency scaling it by
a,ri using known principles of
filter design; the estimation of ri for this purpose will be described
shortly.
The utility of subband processing is as follows. Consider that in its absence,
amounting to taking ri = 1
and =
1, only a single parameter a would be applied at the receiver. If ri < ri+i,
the following general
inequality must hold using the short notation Hi for H[FiJ and Gi for G[Fil:
(1+ ari)7li _.=5_ (1+ ar1+)ri+1.
(11)
Solving for a from the first inequality yields
hi - ri+1
ari >
(1 + ordri)Ei+1 -
(12)
which simplifies, for the common case of sources with identical component
bandwidth allocations, to
r _1].
an. > ___________________________________________________________________
(13)
(ori/r = -1
i)r - W ri W
This bound diverges at small r, being positive only if
ri
G> or, equivalently, (Sri > ri¨ (14)
ori
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but no choice of a will suffice if G is less than this value.
For example, at L = 0, the bound will be negative, but the foregoing procedure
cannot possibly separate the
signals because the d.c. (c.v = 0) components will not be shifted at all, as
H(a)Fi (0) = Fj (0[1 arj]) = Fj (0)
by equation (23). The problem is that the lower bound and the spread W 71 - L
are also scaled by H(),
so that the lower part of the signal spectrum tends to remain in the "spectral
shadow" of any nearer sources.
This is not a special limitation of the method - very low frequencies as such
generally pose problems such
as the ratings of reactive elements needed for d.c. isolation, due to which
baseband audio and video systems
are invariably designed with lower frequency bounds well above 0. With
modulated radio frequency signals,
the limiting constraint becomes the physical source separation ori, instead of
L.
For any given .0 and W, there would thus be a sizeable range of ori over which
a complete separation of
signals would be impossible. However, if > 0, partial separation of a
fraction 0 E [0,11 of the bandwidth
becomes possible, with the lower bound on a set by the condition
ctri > 16r, r _11-1
requiring only 5re > Ore¨L.
(15)
I_ re OW
Although 0 signifies a compromise on the full separation of the signal
spectrum, the separation is nevertheless
useful as the separated parts comprise the lower frequency band [L,L+ 0147]
from the nearer source and the
higher frequency band [71 - OW, 71] from the farther source, which will likely
contain much of the information.
In particular, if the signal is preconditioned for separation by
autocorrelation, the separated high frequency
band [71 - 13W, 71] would serve as a strong reference for autocorrelative
separation of the remainder of the
signal bandwidth (G, 71 - OW). With subband processing, however, the
conditions (15) become
rri
apyi >
ri f3,W 1]-1 requiring only 6ri > r=¨
" L
(16)
where Gr, is the lower frequency bound of the /ith subband. These conditions
are weaker than (13) and (14)
by the factor fl,, and assure separability for all subbands p,> 1 even at L =
0, as the subband lower bounds
are raised to L L + 0,W per equation (3).
Thus, with subband processing, near perfect separability can be assured in all
cases, without a fundamental
need for autocorrelation or other techniques that would involve signal
modulation or content. It is generally
desirable, however, to keep the subbands as wide and as few in number as
possible, however, because each
subband entails shifting, filtering and reverse shifting operations, adding
linearly to the complexity of the
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receiver. It would be generally preferable, therefore, to keep n small, and
indeed as close to 1 as possible.
The secondary object of detecting interference and estimating the source
distribution may be achieved by
a simple variation of this procedure, comprising, after Step B, the
alternative steps of
C'. measuring the lower bound G[H(cx,)S,{F.,}1 to compute the distance rmin of
the nearest source as
(p)(G[H(a,)S,{F3}] 1)
(17)
Tniin =
E[SpjFill ,
obtained from the relation
= (1+ aiiri)G[S,JF21],
(18)
identifying the minimum of the -th subband of the shifted combined spectrum
with the shifted lower
frequency bound of that subband ¨ the result will be nonzero if either G> 0 or
p.> 1;
D'. and likewise measuring the upper bound '141/(0,)S,{Fi}1 to compute the
distance r,nax of the farthest
source from the corresponding relation
(p)
= --1 (71[11(2eA)Sp{Fi}l 1) , (19)
r max ap 71[Sõ{F.,}]
so that the spread of sources would be given by the vector of distance
intervals WW1 where
or(P) = 46, ¨ r, > 0. (20)
In the numerators of equations (17) and (19), and 71 are measured from the
spectra, whereas in both the
denominators of these equations and in Steps A-E of the main inventive
procedure, G and 7/ are likely to
be known parameters of design. The bound measurements may be performed in
reverse order. Importantly,
as they concern derivatives of spectral distributions, thee measurements would
be difficult if the spectra are
discontinuous. Smoothening, interpolative and correlative techniques may be
accordingly employed.
The maximum arid minimum spread can be clearly estimated from these subband-
specific values as
5rma. = max{r412x} ¨ min{rnt),,} and &min = min14,1,120 ¨ max{rnt)n},
respectively. (21)
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The likelihood of detecting interference and separating the interfering
sources would clearly improve with
narrower subbands, as the interference might occur within the signal bandwidth
W, and not affect the ends
of the signal spectrum, i.e. 5r(P) > 0 for only some indices p such that 1 <
p. < 72. This would not be an issue
for interfering sources of similar band spreads. However, the technique would
be also useful for eliminating
narrow band noise or other narrow band signals that happens to occur within
the desired signal spectrum,
for this, a large n, or small fl,, would be required.
if the object is to merely detect all such interference, it would be more
productive to use a single, sufficiently
large shift factor a and a single tunable subband filter S(s) of a variable
centre frequency and a narrow
pass-band oW << W, in order to periodically scan the received signal spectrum
for interference. The modified
procedure in this case would be
A*. applying the tunable filter S(W) to the total received signal F(w)
(corresponding to {F3}), for which
G[S (D)F(w)] = ¨ (5W12 arid 7-1[S(3)F(w)1 = D + SW/2, (22)
to obtain the filtered subband S(D)F(w) (S o F)(w) F(0) at cl.) and vanishing
outside of ED oW/2;
B*. applying to the filtered subband (So F)() a time-varying sampling or
frequency selection mechanism as
described in the copending application, characterized by a parameter (1,, to
cause the filtered subband's
spectrum to be linearly shifted to
H(a)(S o F)(w) = E F(cAl + arp E Fpo. + an) i.e. by shifts Ow [Dar
(23)
for each contributing noise or signal source at respective distance r;
C*. and measuring the lower and upper bounds G[H(a)(S o F)(w)1 and N[H (a)(S o
F)(u.))], respectively, of
this shifted subband in order to compute the minimum and maximum contributing
source distances
L[H(a)(S o F)(w)] 1) a_, (LEE, FP[1 ¨ SW/2 arp]
1
TrninP) = a ¨ OW/2 )
w
and P.) =_I (71[H(a)(S o
F)(w)) 1) a_, (HIE, F([1 + ad)] 1 (24)
rma. ) ,
¨ SW/2 ¨ SW/2
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respectively, as a function of the subba.nd centre frequencyrils. We may
likewise compute
ormax (6W) = max rmax() min rmin ("W) and ormin (5W) = min rmax (C.)) ¨ max
rmin () (25)
corresponding to equation (21), but generally representative of the filter
bandwidth 6W chosen, or other
statistics from the r11.11õ(r2i) and rmax (W-) distributions depending on the
application.
Implementation
The above inventive procedures are independent of the physical nature of the
signals and their wavelength
range. An equally general method for inducing the frequency shifts in the
received waveforms, per equation
(23) and orthogonally to their modulated information so as to be suitable for
the objects of this invention, is
provided by the copending application as mentioned. The method concerns the
spectral phase distribution
of a signal, which can be obtained using any appropriate spectrometric means,
such as resonant cavities or
circuits, diffraction gratings for optical signals, and digital signal
processing for electronic media. Specifically,
as stated in the Background, it involves scanning the gradient of this phase
distribution over the signal
spectrum by continuously varying the instantaneous tuning CD of the resonant
cavity or circuit, the intervals
d 7-
/A sin 0 (for any given diffraction angle 0) of the grating, or the sampling
interval T 1/C:i of the digital
processing system, each at the same normalized rate (.7.i'dLii/dt -
j'C1d5k/dt T-1dT I dt equal to ca (or
ca,), where c denotes the wave speed. It is shown in the copending application
that linear frequency shifts
obeying equation (23) result without otherwise affecting amplitude or phase.
The reverse mechanism in Step
D follows from the shift formula w
w(1 ar). The formula also permits negative choices for a, but only
positive values widen the spectrum, which is necessary for the inventive
procedures.
Orthogonality to the signal content comes from the fact that the distance
information in the phase gradient
is an inherently spatial contribution, derived from the spatial contribution k
= r in the expression q5 = k = r ¨ wt
for the total instantaneous phase of a travelling wave, whereas signal content
and modulation ordinarily refer
strictly to the temporal term wt. If r were to vary so as to interfere with
the signal contribution, the variation
would be in the form of a Doppler shift, which would not be new, and can be
corrected for where necessary.
If the r variation were instead random or oscillatory, at frequencies
comparable to that of the signal, it would
as such interfere with the reception, even in absence of other sources. So the
distance information r can be
exploited to separate or extract a set of signals by this method if they can
at all be individually received.
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A modulated signal may be conversely viewed as a jitter in the carrier's
source distance indication, which
makes the shifted modulated carrier frequency uncertain, relative to the
shifted unmodulated carrier, by a
spread of lw,(1 + ari), wc(1 ar2))., where TI and r2 are the minimum and the
maximum values for r,
respectively, revealed by the spectral phase gradient, and w, is the carrier
frequency. Its spectral footprint
is then the interval [G[HF1, H[H Fl] = HIG[F1, 7-1[F]], i.e. H times the
unshifted original footprint of the
modulated carrier, due to the linearity of H. By the principles of Fourier
analysis, these spectral bounds are
simply equivalent to indefinitely stationary sinusoidal components including
the result of modulation, and
involve no extra uncertainty.
Embodiments
In general, a receiver embodying the present invention would thus generally
include
= zero or more optionally tunable input filters {S,L) and at least one
optionally tunable selection filter 6';
= and one or more fixed or variable mechanisms for shifting H() and reverse
shifting H(¨) as explained.
The receiver would additionally include either
= a fixed or variable means for setting either a or 6, or both, in order to
select a desired signal Fi and
reject interfering signal or noise sources, according to Steps A through E; or
= low and high spectral bound detector means for determining and 7-1
particularly of the shifted spectra,
HAF,j} H(a,)Sii{Fi} in Steps C' and D', along with optional means for varying
one or more of
the subband filters S, so as to vary the corresponding subband intervals and
to thus detect interference
within the signal spectrum, according to the alternative Steps C' and D' in
the inventive procedure.
Alternatively, a receiver may use a single tunable subband filter S(iD) and
one set of spectral bound detector
means applied to H (a)(S F) (w) per equation (24) to continually scan the
entire signal bandwidth W using
the modified inventive procedure of Steps A* through C*.
Both of the inventive functions, of separating the signal from a desired
source and of detecting interference
within the signal band, may be implemented within a given receiver, for use
one at a time or in parallel. The
spectral bound detectors may be also applied to the unshifted spectra for
accuracy of measurement, in which
case the same threshold old, (equation 1) must be employed. In principle, the
subband filters should suffice to
ensure that spectral discontinuities within a subband do not matter ¨ it
should be sufficient to scan inward
from the extremities of the spectrum to the first crossing of the threshold
magnitude ath However, as noise
can generate false threshold crossings, one or more of the following schemes
ould be generally necessary: set
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the threshold lath I above a sufficiently high empirically determined value,
compare several successive samples
to skip over narrower noise spikes, or average over several successive frames,
which is common in spectral
measurements. More sophisticated techniques involving smoothening,
interpolation or autocorrelation over
the spectrum may also be used. None of these schemes is usually an option for
Steps C and D for selecting
a desired signal, however.
A basic receiver need not employ subbanding at all, and thus skip Steps A and
E. A more sophisticated
receiver may use subbanding, and would need a multitude of input subband
filters {5,1. With subbanding,
it would be often also useful to use a smaller a for the higher subbands,
while using sufficiently high values
for the low subbands, so as to keep the shifted spectra within the handling
range of the circuits; this would
not be a concern with digital signal processing. The shift parameter a may be
alternatively fixed at a large
enough value for the intended operating distance range. Large values of a can
be achieved using short time
frames in the shifting mechanism, as also described in the copending
application. The desired signal Fi can
then be selected by varying a-, or switching between a set of fixed filters
{ai}. The alternative would be to
use a single fixed selection filter C, and to vary a in order to bring Fi into
the pass band of C. In either case,
the variation may be performed manually through suitably implemented controls
or knobs, or automatically
by scanning the combined shifted spectrum {H(a)F} for a signal matching some
selection criteria, such as a
spread-spectrum code, a specific subcarrier, a signature pattern, etc. that
could be predefined, interactively
set or acquired from a previously selected signal, so as to lock on to that
source.
These selection and detection functions may also be combined in a receiver, by
the use of separate detection
and source separation modules, each containing its own instances of both the
subband filters S and the
frequency spreading shift mechanisms H. In such a receiver, the detection
module may use relatively narrow
subbands to dynamically determine the coarsest subband partitioning of the
signal spectrum to simplify the
operating configuration of source selection module, and to thus ensure better
performance or lower the total
power consumption. Another variation would be to keep the inventive selection
module on standby, so as to
only activate it in the presence of interference. The onset of interference
may be detected automatically using
the inventive Steps C' and D', simplistically without partitioning into
subbands, or more particularly with
subbands, or more accurately using a single scanning subband, as in Steps A*
through C*. An alternative
arrangement could also be employed for the interference detection in order to
activate the inventive source
selection procedure. In the case of audio or video communication, the
inventive signal selection procedure
may even be manually activated or turned off based on perception of
interference.
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If multiple antenna or aperture feeds are available, e.g. as stereoscopic or
array antennae or microphones,
the range separation can be combined with the angular information from the
feeds to determine the source
locations over two or three dimensional space, as opposed to the one
dimensional range distribution of
sources that can be determined using a single feed. In all of these cases, the
determined spatial distribution
of sources can be displayed to the user to allow visual perception of
interference and interactive selection.
More particularly, the inventive interference detection procedure of Steps C'
and D' or Steps A* through
C* is not necessary in these cases. With stereo- or quadraphonic acoustic
feeds, the two or three dimensional
spatial distribution of the sources would be revealed by a diagram of circles,
or spherical surfaces, drawn
with radii corresponding to the peaks in the shifted spectrum and centred on
the geometrical representations
of the feeds or microphones, with intensities proportional to the (analogue)
energy distribution of the shifted
spectrum H(a)SIFil, i.e. IH(a)SIFi 12. Correspondingly, an automatic (non
interactive) source selection
system may use the phase differences between the feeds to discriminate in
direction as well as distance.
Variations
Numerous variations of the inventive procedures are possible and are intended
within the present invention.
For example, prefiltering may be also employed to alter the spectral profile
over the desired band in order
to simplify, or correct for limitations in, the design of the selection filter
Gi. The prefiltering could include
compressing the signal spectrum, using frequency modulation say, and Step D
could likewise be accomplished
by "mixing", i.e. by multiplying with a generated intermediate frequency
signal, or by frequency modulation.
If the same value of a is used for each of the subbands in Step B, Step E
could be performed before Step D,
as mentioned, with the advantage that only one reverse shifting mechanism is
needed, though it must then
handle the combined shifted bandwidth of all of the subbands.
Further, the input signal spectrum may be expanded before Step B using
frequency modulation to limit
noise arising in the subsequent stages. The final stage may likewise comprise
a more complex combination of
mixing and spectral expanding or compacting. Additionally, a receiver needing
to monitor multiple sources
may be designed using a common shift mechanism in Step B and multiple
selectors Gi, each differently
designed and fed the same shift mechanism output in parallel, or using
identical selectors but fed by differently
designed or tuned shift mechanisms, the latter being each fed the same input
combination of signals.
Other objects, features, applications, variations and advantages of the
present invention will be apparent
when the detailed description of the preferred embodiment is considered in
conjunction with the drawings,
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which should be construed in an illustrative and not limiting sense.
Brief Description of Drawings
Fig. 1 illustrates the separability of wave-propagated signals from two or
more sources at different distances
from the receiver, using a distance-dependent frequency shifting mechanism at
a receiver.
Fig. 2 summarizes the simplified inventive procedure for selecting a desired
signal in the scenario of Fig. I.
Fig. 3 demonstrates the problem of spectral shadow that occurs with closely
located sources, or signals
with low frequency content, or with inadequate distance-dependent frequency
shifting.
Fig. 4 illustrates separability of the lower half of the signal bandwidth in
the scenario of Fig. 3.
Fig. 5 illustrates separability of the upper half of the signal bandwidth in
the scenario of Fig. 3.
Fig. 6 summarizes the inventive procedure for selecting a desired signal in
the scenario of Fig. 3.
Fig. 7 is a block diagram for a receiver implementing the inventive procedure
of Fig. 6.
Fig. 8 is a block diagram for a simpler version of the receiver of Fig. 6.
Fig. 9 is a block diagram for a receiver implementing the simplified inventive
procedure of Fig. 2.
Fig. 10 shows the alternative steps in the simplified inventive procedure to
determine the spread of sources.
Fig. 11 shows the modified inventive procedure for measuring the spread of
sources using a scanning filter.
Fig. 12 is a block diagram for a receiver implementing the modified inventive
procedure of Fig. 11.
Fig. 13 illustrates the "scatter plot" approach for displaying the spread of
sources.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT
Fig. 1 illustrates the inventive procedure for separating signals received
from sources at different distances
from the receiver, using a graph of the spectral shift as a function of
distance. Incoming signals of spectra
F(i.,.)) and F'(w), from sources [520] and [530] at distances r and r' = r +
5r, respectively, from the receiver
[600] located at the origin of the graph are assumed to ordinarily occupy the
same frequency band 14/. The
two signals would ordinarily be received together as the combined signal Ei
F3(w) a- {FA [100] and interfere
with each other's reception at the receiver.
By applying Step B of the inventive procedure as given in the Summary, the
receiver causes the spectra
of these component signals to be shifted in proportion to the source distances
using the method described in
the copending application, i.e. by frequency factors (1+ cm) [220] and (1 +
arl) [230]. The component spectra
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then occupy the shifted bands Fi (w1) H (a)Fi(w)-=7 H [320] and F2(w1) H
(a)F2(w) H F2 [330],
respectively. If the shifted component spectra no longer overlap, as shown,
either signal can be separated
by applying a suitable band-pass filter al [420] or o2 [430], according to
Step C, to correspondingly select
either TiFi [320] or HF2 [330], respectively.
This extracted signal, say HF1 C1HE3 Fi, has to be shifted back to its
original band as Fi(w) [120] to
be usable. This shift would be best done using the reverse shift mechanism H-
1(0) H ( -a), according to
Step D. The above steps form the basic inventive procedure, and are summarized
in Fig. 2 as a time sequence
of operations applicable to narrow band sources with sufficient distances
between them. As mentioned in the
Summary, frequency modulation or mixing with intermediate frequency signals
can be additionally applied
in Steps B and D, and the return shift operation H1 of Step D can be replaced
by these methods.
Fig. 3 illustrates the problem of spectral shadow, which arises whenever the
sources are too close (r < ri),
the applied temporal parallax (a) is too small, or the signal contains very
low frequencies (G< W or G 0),
so that equation (13) is not satisfied. The figure shows that under any of
these conditions, the shifted spectra
overlap and cannot be separated using a band-pass filter. if, further, the
sources are of nearly equal strength,
the shifted spectrum of the nearer source, F1(w1) [320], in effect casts a
shadow [322] over the shifted spectrum
F2(w2) [330] of the farther source, i.e. that portions of the latter, F2(w2)
[330], that fall within this shadow
will suffer interference from the nearer source. If the signals are frequency
or spread-spectrum modulated, for
which a receiver typically recovers the carrier coherently using a phase-lock
circuit, the farther or otherwise
weaker source would be likely rejected altogether, regardless of which source
was desired.
Further, Fig. 3 also illustrates the spectral widening property of the H
operators, which exacerbates the
shadow problem. Widening occurs because the lower bound 1321] of the shifted
spectrum would have been
shifted by (1 + ar)L, which is less than the shift (1 + 00?-1 contained in the
upper bound [323] of the shifted
spectrum, so that the shifted bandwidth is itself greater than W, and the
spectral shadow [322] cast by the
source becomes greater than W by the same factor (1 + ar), as shown.
The inventive solution for the spectral shadow problem, as formally treated in
the Summary, is to partition
the incoming combined signal into two or more subbands, to then apply the
procedure of Fig. 2 separately
to each of the subbands, and lastly, recombine the subbands to obtain the
separated signal spectrum. In the
example of Fig. 3, since the shadow [322] covers roughly half of the second
source spectrum [330], separation
can be achieved by partitioning the input signal into two subbands, as
illustrated in Figs. 4 and 5, showing
the results of applying Step B to the lower and the upper subbands,
respectively.
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As shown in Fig. 4, the lower subband So E3 Fi [105] of the combined incoming
signal, obtained from the
lower subband filter So in Step A, separates, under the inventive operation
H(a), into the shifted component
spectra So HFI [325] and S0HF2 [335]. If just separated, the shifted lower
bound [331] of the second signal
will coincide with the shifted upper bound [327] of the first. The lower
subbands become separable because
the lower subband of the first source no longer casts a shadow on the lower
subband of the second though
both are shifted by the same parallax factor a.
Fig. 5 shows the corresponding separation of the upper subband S1E3 F3 [106]
of the combined incoming
signal, obtained from the lower subband filter S1 in Step A, into the shifted
component spectra Si HF1 1326]
and Si HF2 [336]. Again, if just separated, the shifted lower bound [337] of
the second signal will coincide
with the shifted upper bound [323] of the first. The shadow would range from
the shifted lower bound [327]
to the shifted upper bound [323], and fails to cover the shifted subband [336]
of the second source.
Fig. 6 summarizes the complete inventive procedure including the separation
into lower [105] and upper
[106] subbands in Step A, by means of the lower and the upper subband filters
[400] and [402,1 respectively,
to eventually obtain the extracted lower [125] and upper [126] subbands of the
desired signal F1 [120] from
the first source, and recombination of these extracted subbands in Step E. As
mentioned in the Summary,
Step E, recombination, can be performed before applying the reverse shift,
i.e. before Step D, which would
be useful for reducing the number of operations. The figure incidentally also
illustrates that narrower filters
could be used for the source selection, in the place of al [420], and that a
could be made smaller as well.
Fig. 7 is a block diagram of a receiver incorporating the complete inventive
procedure described in Fig. 6.
It shows incoming electromagnetic (or acoustic) waves [610] being collected by
an antenna (or microphone)
[620] to produce the combined input signal {Fi(w)}. This combined signal is
fed to a bank of input subband
filters [630] to produce the combined subband signals SAFJ} as Step A. These
combined subband signals are
then subjected to Step B using a bank of frequency shifting mechanisms [640]
per the copending application,
to get the shifted subband signals H(aõ)S1{F3}, in which the contributions
from the individual sources are
already separated in frequency as shown in the preceding figures. In order to
select the desired source si and
suppress the contributions from the remaining sources, these shifted subband
signals H(ap)S,{Fi} are then
fed to the band-pass selection filter bank [650], as Step C, to obtain the
shifted subbands aiH(a,)4{F3}
H(a,)GiS,{Fj} H(ap)S,F, of the desired signal F, per equations (6) and (7).
These shifted subbands are
then down-shifted by a bank of reverse shifting mechanisms [660] (Step D),
yielding
S,Fi, the subbands of the desired signal, and recombined by a summing device
[670], which can be as simple
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as an operational amplifier (op-amp), to obtain EpSF = Fi, the desired signal.
Fig. 8 is a simpler version of the receiver of Fig. 7, in which the summing
device is applied before down-
shifting, which is only possible when the same value of a is used in each of
the frequency shifting mechanisms
[640]. In this case, the outputs of the band-pass selection filter bank [650]
are immediately recombined by the
summing device [670] to produce the desired signal, except that it is still
expanded and shifted in frequency
as E H(a)S,,F = H(a)F,, arid requires down-shifting by a single reverse
shifting mechanisms [662] to yield
the desired signal as H(¨a) Et, H(a)S,F, = Ft.
Fig. 9 shows an even simpler receiver that treats the entire signal bandwidth
W as one subband, and thus
skips both Steps A and E. Such a receiver would be adequate, as already
explained, when the sources are
well separated from one another and the signal bandwidth W does not include
d.c. It would be generally
sufficient for broadcast radio and also mobile (cellular) telephones, since
the base stations would be typically
spread far apart. The more complex receiver of Fig. 7 would be generally
needed at the cellular base stations,
however, as the mobile (cellular) phones could even be situated side by side.
The related inventive method for detecting interference and estimating the
source distribution, given by
Steps C' and D' in the Summary, is explained in Fig. 10 using the same
combined input signal [100] as in the
preceding figures. After Step B, both the original combined incoming signal
spectrum [100], and its shifted
spectrum, comprising the shifted components [320] and [330] both would be
available to the receiver using
any applicable means of spectral analysis, including digital signal
processing, as typically used for radio or
acoustic signals, and refraction or diffraction, e.g. for optical, microwave
or sonar signals. In the latter case,
it is common practice in related arts like modern astronomy to convert the
resulting spectrum to digital form
for further processing, storage and viewing. It is straightforward, therefore,
to also apply smoothening and
interpolation, to compute autocorrelation of the spectral distributions, and
to average over several successive
frames, as necessary to obtain good estimates of the spectral distributions.
Step C' then consists of seeking, from the low frequency end of the measured
domain, the first crossings
of the obtained distributions above a suitably chosen threshold au, [700], as
indicated by the arrows [710]
and [720], thereby obtaining the values LRFA and C[H(a){Fj}1 as the respective
abscissae. An estimate
of the distance to the nearest source is then computed from the relation
rmin = a_i G[H(a){Fi}]
1) ,
(26)
.C[{F}]
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which is the specialization of equation (17) to a single subband encompassing
the full signal bandwidth W.
Step D' correspondingly consists of seeking, from the high frequency end of
the measured domain, the first
crossings of the obtained distributions above the same threshold ath [700], as
indicated by the arrows [730]
and [740], to obtain the values N[{F3}1 and 71[H (a){Fi}] as the respective
abscissae. The distance rmax to
the farthest source is then estimated using the relation
_1 (NEI (a){F3}]
1),
(27)
rmax
which similarly specializes equation (19) to a single subband encompassing the
full nominal signal bandwidth
W. As remarked in the Summary, these two steps could be performed in the
reverse order, i.e. Step D' before
Step C', since the crossing detections are independent, and for the same
reason, it would be trivial to perform
these steps simultaneously or in random order in a receiver, for example, as
independent threads of execution
in a software implementation.
It would be trivial to extend this procedure for measuring rrhh, and rrhaõ
identically to each of the subbands
Sp,{F(w)}
{Ft,(w)} of the combined received signal to compute the corresponding values
rn and 7-12),
for each subband, and to thereby arrive at the minimum and maximum spread
estimates defined in equation
(21), or other suitable statistics from these measurements.
Scanning with a single, narrow subband filter would be superior for detecting
interfering signal or noise
sources within the signal band W, per the modified inventive procedure, Steps
A* through C* given in the
Summary. This is illustrated in Fig. 11, in which a single subband filter
[450] with a very narrow passband
5W < W is used to scan the received signal spectrum F(a) [100], to obtain the
filtered signal (S o F)(w)
1150] at each instantaneous position of the filter [450] (Step A*). In Step
B*, this filtered signal is subjected
to the frequency shifting mechanism of the copending application to yield the
shifted spectral distribution
1-1(a)(S o F)(w) = Er F (w[1 + ar]) Er F(D[i + ar]) [350]. As Step C*, the
threshold frequency bound
detectors are again applied, as shown by the arrows [710] and [730] to
determine the low and high frequency
bounds of the shifted distribution, respectively, for computing the source
distribution functions 5rmiõ (5W)
and orrhax(5W) per equation (24).
Fig. 12 is a block diagram of a receiver incorporating the scanning procedure
of Fig. 11. In this, the
received signal (or combination of signals) from the antenna [620] is first
subjected to narrow band filtering,
in accordance with Step A*, by a subband filter [450], whose centre frequency
is made to periodically sweep
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over the input band of frequencies by a sweep controller [634]. The resulting
filtered signal is then input
to the frequency shift mechanism [642], per Step B*, and its frequency bounds
are measured, per Step C*,
by the high [732] and the low [712], respectively. The bound values obtained
are used to compute rniir, and
applying equation (24), or other related statistics, by the source
distribution computer [680].
Related to the scanning procedure is the "scatter plot" method mentioned in
the Summary, illustrated in
Fig. 13 for the case of bistatic (stereophonic) antenna (microphone) feeds
[622] and [624] that respectively
provide two input signals FL(w) [102] and FR (w) [104]. These signals are
first scanned simultaneously by the
identical narrow subband filters [452] and [454], per Step A*, and then
shifted by identical frequency shifting
mechanisms to yield the shifted distributions H(a)(SoFd(w) = Er FL (w +ad) Er
FL([1+ arl) [352]
and H(a)(SoFR)(w) = Er FR(w[i+ cxr]) Er FR(iD[i +old) [354], per Step B*.
Next, instead of measuring
the frequency bounds according to Step C*, one draws on a separate graph
circles [552] and [554] representing
the loci of possible source locations, with centres corresponding to the two
feeds and radii proportional to
the shifts. The resulting concentrations of sparse and dense regions resemble
well known two-slit interference
patterns of diffraction theory, since each concentration of (signal or noise)
sources produces multiple dense
regions like [562] and [564]. Fig. 13 also shows that the "scatter plot" is
really a technique for combining
the source distance distribution data from multiple antenna feeds, as the
distribution information from each
individual feed is already revealed by the shifted spectral distributions
[352] and [354].
The difference between the "scatter plot" and a diffractive interference
pattern is that the plot represents
the actual spatial distribution of sources, albeit with multiple aliases,
whereas diffractive interference is only
representative of their spectral distribution. This is because the plot starts
with the spectral distribution,
whereas in diffraction theory, one starts with a spatial distribution of
sources or slits. The method is in this
sense an inverse of diffractive interference.
The invention has been described with reference to the preferred embodiment,
but it will be appreciated
by those of ordinary skill in the arts of general physics, electronics and
communication technologies that
numerous modifications and variations are possible in the light of the above
disclosure. For example, the
invention can be applied to sound and underwater communication, and to
transmission lines or optical fibres.
Indeed, as the filtering, recombining, down-shifting and the key operation of
shifting spectra in proportion to
the contributing source distances can be conceivably applied to signals over
any kind of propagating waves,
provided only that they obey the wave equation, as particularly described in
the copending application for
the shifting operation, the invention could be applied even to matter or
gravitational wave signals.
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As stated in the Summary, Step D could be replaced by a down-converter,
optionally with a modulation-
demodulation stage to scale back the bandwidth by the factor (1 + ar). The
scaling down may be obviated
by moving the modulation-demodulation stage before Steps A or B, so that the
bandwidth is already scaled
down by an estimate of the (1+ar) factor for the desired source. This would
also require narrower subbanding
filters S and source selection filters Gi, which may be useful from the
perspective of ensuring constant or
linear phase over the filter spectra, since phase distortions can affect the
spectral phase gradient and the
linearity of separation assumed in Figs. 1, 3, 4 and 5.
Likewise, the problem of spectral shadow and the inventive use of subbanding
to overcome it have been
illustrated using just two subbands, but it would be clear to those skilled in
the related arts that more than
two subbands may often be necessary and that the lowest subband, especially if
including 0 frequency (d.c.),
may need to be abandoned altogether, as stated in the Summary.
As stated in the Background and in the Summary, the present invention may be
enhanced with direction-
sensitive antenna technology to also provide separation of signals from
sources at almost the same distance
from the receiver, but differing in their directions. The inventive method may
be conversely employed as an
alternative to directional antennae in order to separate sources that are too
close in direction. The present
invention may likewise be combined with content-based separation methods
including, but not limited to,
amplitude, frequency, phase and spread-spectrum modulations, or TDM, and
autocorrelative methods. All
such modifications, generalizations and variations are intended within the
scope and spirit of the invention
as defined in the claims appended hereto.