Note: Descriptions are shown in the official language in which they were submitted.
CA 02670691 2009-06-25
GIGABIT ETHERNET TRANSCEIVER
15 BACKGROUND OF THE INVENTION
The invention relates to systems for, and methods of, reducing the noise
present in the
signals received and processed by devices within a communications system and
to systems for,
and methods of, reducing such noise in communications systems having high
throughputs. The
invention also relates to systems for, and methods of, reducing the power
dissipation in devices
within a communications system and to systems for, and methods of, reducing
such power
dissipation in communications systems having high throughputs. The invention
further relates
to a startup protocol for initiating normal transmission between transceivers
within a high
throughput communications system. A "high throughput" as used within the
context of this
disclosure may include, but is not limited to, one gigabit (GB) per second.
A basic communications system is illustrated in FIG. 1. The system includes a
hub and
a plura.lity of computers serviced by the hub in a local area network (LAN).
Four computers are
shown by way of illustration but a different number of computers may be
contained within the
system. Each of the computers is usually displaced from the hub by a distance
which may be as
great as approximately one hundred meters (100 m.). The computers are also
displaced from
each other. The hub is connected to each of the computers by a communications
line. Each
communication line includes unshielded twisted pairs of wires or cables.
Generally, the wires
or cables are formed from copper. Four unshielded twisted pairs of wires are
provided in each
communication line between each computer and the hub. The system shown in FIG.
1 is
operative with several categories of unshielded twisted pairs of cables
designated as categories
3, 4, 6 and 7 in the telecommunications industry. Category 3 cables are the
poorest quality (and
lowest cost) and category 6 and 7 cables are the best quality (and highest
cost).
Associated with each communications system is a "throughput". The throughput
of a
system is the rate at which the system processes data and is usually expressed
in bits/second.
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Most communications systems have throughputs of 10 megabits (Mb)/second or 100
Mb/second.
A rapidly evolving area of communications system technology enables 1
Gb/second full-duplex
communication over existing category-5 unshielded twisted pair cables. Such a
system is
commonly referred to as "Gigabit Ethernet'
A portion of a typical Gigabit Ethernet is shown in FIG. 2. The Gigabit
Ethernet provides
for transmission of digital signals between one of the computers and the hub
and the reception
of such signals at the other of the computer and the hub. A similar system can
be provided for
each of the computers. The system includes a gigabit medium independent
interface (GMII)
block which receives data in byte-wide format at a specified rate, for example
125 MHz , and
passes the data onto the physical coding sublayer (PCS) which performs
scrambling, coding, and
a vaziety of control functions. The PCS encodes bits from the GMII into 5-
level pulse amplitude
modulation (PAM) signals. The five symbol levels are -2, -1, 0, +1, and +2.
Communication
between the computer and hub is achieved using four unshielded twisted pairs
of wires or cables,
each operating at 250 Mb/second, and eight transceivers, one positioned at
each end of a
unshielded twisted pair. The full-duplex bidirectional operation provides for
the use of hybrid
circuits at the two ends of each unshielded twisted pair. The hybrid controls
access to the
communication line, thereby allowing for full-duplex bidirectional operation
between the
transceivers at each end of the communications line.
A common problem associated with communications systems employing multiple
unshielded twisted pairs and multiple transceivers is the introduction of
crosstalk and echo noise
or impairment signals into the transmission signals. Noise is inherent in all
such
communications systems regardless of the system throughput. However, the
effects of these
impairment signals are magnified in Gigabit Ethernet. Impairment signals
include echo, near-
end crosstalk (NEXT), and far-end crosstalk (FEXT) signals. As a result of
these impairment
signals the performance of the transceivers, particularly the receiver
portion, is degraded.
NEXT is an impairment signal that results from capacitive and inductive
coupling of the
signals from the near-end transmitters to the input of the receivers. The NEXT
impairment
signals encountered by the receiver in transceiver A are shown in FIG. 3. The
crosstalk signals
from transmitters B, C, and D appear as noise to receiver A, which is
attempting to detect the
direct signal from transmitter E. Each of the receivers in the system
encounters the same effect
and accordingly the signals passing through the receivers experience signal
degradation due to
NEXT impairment signals. For clarity of FIG. 3, only the NEXT impairment
experienced by
receiver A is illustrated.
Similarly, because of the bidirectional nature of the communications systems,
an echo
impairment signal is produced by each transmitter on the receiver contained
within the same
transceiver as the transmitter. The echo impairment s'ignal encountered by the
receiver in each
transceiver is shown in FIG. 4. The crosstalk signals from transmitters appear
as noise to the
receivers, which are attempting to detect the signal from the transmitter at
the opposite end of
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the communications line. Each of the receivers in the system encounters the
same effect and
accordingly the signals passing through the receivers experience signal
distortion due to the echo
impairment signal.
Far-end crosstalk (FEXT) is an impairment that results from capacitive
coupling of the
signal from the far-end transmitters to the input of the receivers. The FEXT
impairment signals
encountered by the receiver in transceiver A are shown in FIG. 5. The
crosstalk signals from
transmitters F, G, and H appears as noise to receiver A, which is attempting
to detect the direct
signal from transmitter E. Each of the receivers in the system encounters the
same effect and
accordingly the signals passing through the receivers experience signal
distortion due to the
FEXT impairment signal. For clarity of FIG. 5 only the FEXT impairment
experienced by
receiver A is illustrated.
As a result of these noise impairment signals the performance of the
communication
system is degraded. The signals carried by the system are distorted and the
system experiences
a higher signal error rate. Thus there exists a need in the art to provide a
method of, and an
apparatus for, compensating for the degradation of communication system
performance caused
by noise impairment signals and to provide a method of, and apparatus for,
reducing such noise
in a high throughput system such a Gigabit Ethernet. Aspects of the present
invention fulfil these
needs.
Four transceivers at one end of a communications line are illustrated in
detail in FIG. 6.
The components of the transceivers are shown as overlapping blocks, with each
layer
corresponding to one of the transceivers. The GMII, PCS, and hybrid of FIG. 6
correspond to
the GMII, PCS, and hybrid of FIG. 2 and are considered to be separate from the
transceiver. The
combination of the transceiver and hybrid forms one "channel" of the
communications system.
Accordingly, FIG. 6 illustrates four channels, each of which operates in a
similar manner. The
transmitter portion of each transceiver includes a pulse-shaping filter and a
digital-to-analog
(D/A) converter. The receiver portion of each transceiver includes an analog-
to-digital (A/D)
converter, a first-in first-out (FIFO) buffer, a digital adaptive equalizer
system including a feed-
forward equalizer (FFE) and a detector. The receiver portion also includes a
timing recovery
system and a near-end noise reduction system including a NEXT cancellation
system and an
echo canceller. The NEXT cancellation system and the echo canceller typically
include
numerous adaptive filters.
Characteristics of the communication line, e. g., length, may impact the
ability of the
NEXT cancellation system and echo cancellers to effectively cancel NEXT and
echo noise.
Measurements of typical cable responses, as well as simulation, show that in
order to provide an
adequate level of cancellation of these sources of interference, "long" echo
and NEXT cancellers
are required. The term "long" is used to describe a canceller having a large
number of taps as
necessitated by the characteristics of the cable. For example, FIG. 7 shows
the echo impulse
response for a 100m cable with a characteristic impedance of 85 ohm and 100
ohm terminations.
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Although the nominal characteristic impedance is 100 ohm, manufacturing
standards allow for
a 15% tolerance. The mismatch in impedance may result in a reflection at the
far-end of the
cable, which causes a secondary pulse with a delay of about one microsecond.
Because of the
long delay, cancelling this pulse requires an echo canceller with about 140
taps (125 taps to
cover the one microsecond delay, plus approximately 15 additional taps to
cancel the secondary
pulse).
Often the echo impulse response has additional reflections at intermediate
values of delay.
In addition, structural return loss of the cable may cause continuous
variations of the
characteristic impedance along the cable, which results in a large number of
smaller reflections
at intermediate points. These intermediate reflections mean that the echo
canceller should not
be configured to cancel only the initial impulse and the end reflection but
instead should be
configured to cover the full span of the impulse response. Varying cable
characteristics result
in a wide variability of cable impulse responses. In addition, the response of
a particular cable
may change as a result of its operating environment. For example, a change in
operating
temperature may change the impulse response of the cable. Accordingly, it is
difficult to
precompute the locations at which taps are required and to build these
locations into the design
of the echo and NEXT cancellers. FIG. 8 shows the NEXT impulse response for a
100m cable.
As indicated, the NEXT response can also be long, requiring a large number of
taps in the NEXT
cancellers which make up the NEXT cancellation system. The combination of the
NEXT and
echo cancellers consumes the majority of the DSP operations in a Gigabit
Ethernet.
The large number of taps required to achieve a satisfactory performance in
these systems
results in high power dissipation. This high power dissipation is undesirable
in that it may make
high throughput communication systems, particularly Gigabit Ethernet,
inoperable and
unmarketable. Thus there exists a need in the art to provide a method of, and
an apparatus for,
reducing the power dissipation of communication systems employing a large
number of taps and
to provide a method of, and apparatus for, reducing such power dissipation in
a high throughput
system such a Gigabit Ethernet. Aspects of the present invention fulfil these
needs.
One of the most critical phases of the operation of a Gigabit Ethernet
transceiver is the
startup. During this phase adaptive filters contained within the transceiver
converge, the timing
recovery subsystem acquires frequency and phase synchronization, the
differences in delay
among the four wire paius are compensated, and pair identity and polarity is
acquired. Successful
completion of the startup allows norinal operation of the transceiver to
begin.
In one startup protocol, known as "blind start", the transceivers converge
their adaptive
filters and timing recovery systems simultaneously while also acquiring timing
synchronization.
A disadvantage of such a startup is that'there is a high level of interaction
among the various
adaptation and acquisition algorithms within the transceiver. This high level
of interaction
reduces the reliability of the convergence and synchronization operations
which occur during
startup.
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Thus there exists a need in the art to provide a startup protocol for use in a
high
throughput communications system, such as a Gigabit Ethernet, that uses the
optimal sequence
of operations and minimizes the interaction among the various adaptation and
acquisition
algorithms. Aspects of the present invention fulfil these needs.
SUMMARY OF THE INVENTION
Briefly, and in general terms, the invention relates to systems for, and
methods of,
reducing the noise and the power dissipation within communications systems.
The invention
also relates to a startup protocol for use in communications system.
In a first aspect, the invention relates to a communications system having a
prespecified
threshold error. The communications system includes a communication line
having a plurality
of twisted wire pairs, a plurality of transmitters, one transmitter at each
end of each twisted wire
pair and a plurality of receivers, one receiver at each end of each twisted
wire pair, each receiver
receiving a combination signal including a direct signal from the transmitter
at the opposite end
of the twisted wire pair with which the receiver is associated and a plurality
of noise signals. The
system also includes a plurality of adaptive filters responsive to the
combination signal, each
adaptive filter having a plurality of taps each having a coefficient, each tap
switchable between
an active and an inactive state. The system further includes a control device
for periodically
adjusting the transfer function of at least one of the adaptive filters by
selectively deactivating
the taps while ensuring that the error of the communications system does not
exceed the
threshold error.
By selectively deactivating the taps of at least one of the adaptive filters
the present
invention reduces the power consumption of the filter and thus the overall
power consumption
of the communications system.
In a more detailed aspect, the communications system further includes a
plurality of noise
reduction systems each comprising at least one of the adaptive filters. One
noise reduction
system is associated with each receiver and provides at least one replica
noise impairment sigoal.
The system also includes a plurality of devices, one associated with each
receiver. Each device
is responsive to the combination signal received by such receiver and the
replica noise
impainnent signal provided by the noise reduction system associated with such
receiver for
substantially removing at least one of the noise signals from the combination
signal. In another
facet, the noise signals include a plurality of far-end crosstalk (FEXT)
impairment signats, one
from each of the transmitters at the opposite end of the communications line,
except for the
tcanmnitter at the opposite end of the twisted wire pair with which the
receiver is associated, and
the noise reduction system includes a FEXT cancellation system for providing,
as one of the
replica noise impairment signals, a replica FEXT impairment signal. In yet
another aspect, the
noise signals include a plurality of near-end crosstalk (NEXT) impairment
signals, one from each
of the transmitters at the same end of the communications line, except for the
transmitter at the
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same end of the twisted wire pair with which the receiver is associated, and
the noise reduction
system includes a NEXT cancellation system for providing, as one of the
replica noise
impainnent signals, a replica NEXT impairment signal. In still another aspect,
the noise signals
include an echo impairment signal received from the transmitter at the same
end of the twisted
wire pair with which the receiver is associated, and the noise reduction
system includes an echo
canceller for providing, as one of the replica noise impairment signals, a
replica echo impairment
signal. In another detailed aspect, the control device includes means for
setting the state of each
tap, means for calculating a present error for the system and means for
comparing the present
error to the threshold error. In a further facet, the means for setting the
state of each tap includes
means for specifying a tap threshold for each tap, means for comparing for
each tap the absolute
value of the tap coefficient with the tap threshold, and means for
deactivating those taps having
a coefficient with an absolute value less than the tap threshold.
In a second aspect, the invention is a method of operating a communications
system
having a communication line having a plurality of twisted wire pairs and a
plurality of
transceivers, one at each end of each of the twisted wire pairs. Each
transceiver has a receiver
and a transmitter. Each receiver receives a combination signal that includes a
direct signal from
the transmitter at the opposite end of the twisted wire pair with which the
receiver is associated
and a plurality of noise signals. Each transceiver further includes a
plurality of adaptive filters
responsive to the combination signal. Each adaptive filter has a plurality of
taps each having a
coefficient. Each tap is switchable between an active and an inactive state.
The method includes
the steps of specifying a threshold error for the system and periodically
adjusting the transfer
function of at least one of the adaptive filters by selectively deactivating
the taps while ensuring
that the error of the system does not exceed the threshold error.
In a more detailed facet, the method further includes, for each receiver, the
steps of
generating at least one replica noise impairment signal and combining the at
least one replica
noise impairment signal with the combination signal to produce an output
signal substantially
devoid of at least one of the noise signals. In another facet, the step of
adjusting the transfer
function includes the steps of setting the state of each tap, calculating a
present error for the
system and comparing the present error to the threshold error. In another
facet, one transceiver
of the communications system acts as a master and another transceiver acts as
slave. Each
transceiver has a noise reduction system, a timing recovery system and at
least one equalizer.
The method further includes the steps of executing a first stage during which
the timing recovery
system and the equalizer of the slave are trained and the noise reduction
system of the master is
trained, executing a second stage during which the timing recovery system and
the equalizer of
the master are trained and the noise reduction system of the slave is trained
and executing a third
stage during which the noise reduction system of the master is retrained. In
yet another facet,
one transceiver of the communications system acts as a master and another
transceiver acts as
slave. Each transceiver having a noise reduction system, a timing recovery
system and at least
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one equalizer. The method further includes the steps of executing a first
stage during which the
timing recovery system and the equalizer of the slave are trained and the
noise reduction system
of the master is trained, executing a second stage during which the timing
recovery system of the
master is trained in both frequency and phase, the equalizer of the master is
trained and the noise
reduction system of the slave is trained and executing a third stage during
which the noise
reduction system of the master is retrained, the timing recovery system of the
master is retrained
in phase and the timing recovery system of the slave is retrained in both
frequency and phase.
In a third facet, the invention is a communications system including a
communication line
having a plurality of twisted wire pairs, a plurality of transmitters, one
transmitter at each end
of each twisted wire pair and a plurality of receivers. One receiver is at
each end of each twisted
wire pair. Each receiver receives a combination signal including a direct
signal from the
txansmitter at the opposite end of the twisted wire pair with which the
receiver is associated and
a plurality of far-end crosstalk (FEXT) impairment signals, one from each of
the remaining
transmitters at the opposite end of the communications line. The system also
includes a plurality
of FEXT cancellation systems, one associated with each receiver. Each FEXT
cancellation
system provides a replica FEXT impairment signal. The system further includes
a plurality of
delay devices, one associated with each receiver. Each delay device is
responsive to the
combination signal received by such receiver for delaying the combination
signal. Also includes
are a plurality of first devices, one associated with each receiver. Each
first device is responsive
to the output of the delay device associated with such receiver and the
replica FEXT impairment
signal provided by the FEXT cancellation system associated with such receiver
for substantially
removing the FEXT impairment signals from the combination signal.
By providing a plurality of FEXT cancellation systems which generate replica
FEXT
impairment signals and a plurality of device which combine the replica FEXT
impairment
signals with the combination signals, the invention substantially cancels the
FEXT impairment
signals from the combination signal. Thus signal degradation due to noise in
the
communications system is reduced and the transmitted information may be more
reliably
recovered.
In a more detailed facet, the FEXT cancellation system includes means for
receiving a
signal from each of the receivers at the same end of the communications line
except for the
receiver with which the FEXT canceller is associated. The FEXT cancellation
system also
includes means for generating an individual replica FEXT impairment signal for
each received
signal and means for combining the individual replica FEXT impairment signals
to generate the
replica FEXT impairment signal. In another aspect, when the direct signal
arrives at the receiver
after the FEXT impainnent signals, the delay device delays the combination
signal by an amount
substantially equal to the time delay between the arrival, at the receiver, of
the FEXT impairment
signals and the direct signal. In yet another facet, when the direct signal
arrives at the receiver
after the FEXT impaimient signals, the delay device delays the combination
signal by the greater
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of the following: an amount substantially equal to the time delay between the
arrival, at the
receiver, of the FEXT impairment signals and the direct signal; and
an amount so that such combination signal is in sync with the combination
signals from other
receivers.
In a fourth aspect, the invention is related to a method for reducing noise in
a
communications system. The system includes a communication line having a
plurality of twisted
wire pairs. The system also includes a plurality of transmitters, one
transmitter is at each end of
each of the twisted wire pairs. The system further includes a plurality of
receivers, one receiver
at each end of each of the twisted wire pairs. Each receiver receives a
combination signal
including a direct signal from the transmitter at the opposite end of the
twisted wire pair with
which the receiver is associated and a plurality of far-end crosstalk (FEXT)
impairment signals,
one from each of the remaining transmitters at the opposite end of the
communications line. The
method includes, for each receiver, the steps of generating a replica FEXT
impairment signal and
combining the replica FEXT impairment signal with the combination signal to
produce an output
signal substantially devoid of FEXT impainnent signals.
In a fiffth aspect, the invention involves a method for reducing power
dissipation within
a communications system having a plurality of adaptive filters with a
plurality of taps, each tap
switchable between an active and an inactive state, each having a coefficient.
The method
includes the steps of a) specifying an acceptable error for the system; b) for
each active tap,
setting a tap threshold; c) for each active tap, deactivating those taps
having a coefficient with
an absolute value less than the tap threshold set for the active tap; d)
computing a system error;
e) comparing the computed system error to the acceptable system error; f) if
the computed
system error is less than the acceptable system error, increasing the tap
threshold for each active
tap; and g) repeating steps c) through f) until the computed system error
approaches the
acceptable system error without exceeding the acceptable system error.
In a sixth facet, the invention involves a method for reducing power
dissipation within a
communications system having at least one adaptive filter with a plurality of
taps , each tap
switchable between an active and an inactive state, each tap having a
coefficient. The method
includes the steps of a) computing an initial system error; b) for each active
tap, setting a tap
error threshold; c) for each active tap, deactivating those taps having a
coefficient with an
absolute value less than the tap error threshold set for the active tap; d)
computing a subsequent
system error; e) if the difference between the subsequent system error and the
initial system error
is less than a prespecified value, increasing the tap error threshold for each
active tap; and f)
repeating steps c) through e) until the difference between the subsequent
system error and the
initial system error exceeds the prespecified value.
In a seventh aspect, the invention is a startup protocol for use in a
communications system
having a master transceiver at one end of a twisted wire pair and a slave
transceiver at the
opposite end of the twisted wire pair. Each transceiver has a near-end noise
reduction system,
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far-end noise reduction system, a timing recovery system and at least one
equalizer. The
protocol includes the step of, during a first phase, maintaining the master in
a half-duplex mode
during which it transmits a signal but does not receive any signals,
maintaining the slave in a
half-duplex mode during which it receives the signal from the master but does
not transmit any
signals, converging the master near-end noise reduction system, adjusting the
frequency and
phase of the signal received by the slave such that the frequency and phase
are synchronized with
the frequency and phase of the signal transmitted by the master, and
converging the equalizer
of the slave. Also included is the step of, during a second phase, maintaining
the slave in a half-
duplex mode during which it transmits a signal but does not receive any
signals, maintaining the
master in a half-duplex mode during which it receives the signal from the
slave but does not
transmit any signals, freezing the frequency and phase of the slave,
converging the slave near-
end noise reduction system, adjusting the phase of the signal received by the
master such that the
phase is synchronized with the phase of the signal transmitted by the slave,
and converging the
equalizer of the master. Also included is the step of, during a third phase,
maintaining the slave
in a full-duplex mode such that the slave transmits and receive signals,
maintaining the master
in a full-duplex mode such that the master transmits and receive signals, and
reconverging the
master near-end noise reduction system.
In an eighth facet, the invention is a startup protocol for use in a
communications system
having a master transceiver at one end of a communications line and a slave
transceiver at the
opposite end of the communications line, each transceiver having a near-end
noise reduction
system, a far-end noise reduction system, a timing recovery system and at
least one equalizer.
The protocol includes the step of, during a first phase, maintaining the
master in a half-duplex
mode during which it transmits a signal but does not receive any signals,
maintaining the slave
in a half-duplex mode during which it receives the signal from the master but
does not transmit
any signals, converging the master near-end noise reduction system, adjusting
the frequency and
phase of the signal received by the slave such that the frequency and phase
are synchronized with
the frequency and phase of the signal transmitted by the master, and
converging the equalizer
of the slave. The protocol further includes the step of, during a second
phase, maintaining the
slave in a half-duplex mode during which it transmits a signal using a free-
running clock but
does not receive any signals, maintaining the master in a half-duplex mode
during which it
receives the signal from the slave but does not transmit any signals,
converging the slave near-
end noise reduction system, adjusting the frequency and phase of the signal
received by the
master such that the frequency and phase are synchronized with the frequency
and phase of the
signal transmitted by the slave, and converging the equalizer of the master.
The protocol also
includes the step of, during a third phase, maintaining the slave in a full-
duplex mode such that
the slave transmits and receive signals, maintaining the master in a full-
duplex mode such that
the master transmits and receive signals, reconverging the master near-end
noise reduction
system, adjusting the phase of the signal received by the master such that the
phase is
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synchronized with the phase of the signal transmitted by the slave, adjusting
the frequency and
phase of the signal received by the slave such that the frequency and phase
are synchronized with
the frequency and phase of the signal transmitted by the master.
By partitioning the startup protocol into three stages the convergence of the
equalizer and
the timing recovery system is separate from the convergence of the noise
reduction system.
Accordingly, the interaction among the various adaptation and acquisition
algorithms within the
transceiver is reduced and the reliability of the convergence and
synchronization operations is
improved.
These and other aspects and advantages of the present invention will become
apparent
from the following more detailed description, when taken in conjunction with
the accompanying
drawings which illustrate, by way of example, the preferred embodiments of the
invention.
BRIEF DESCRIPTION OF THE DRAWINGS
FIGURE 1 is a scheniatic block diagram of a communications system providing a
plurality
of computers connected to a hub by communications lines to form a local area
network (LAN);
FIG. 2 is a schematic block diagram of a communications system providing a
gigabit
medium independent interface (GMII), a physical coding sublayer (PCS) and a
plurality of
unshielded twisted pairs of wires, each with a transceiver at each end;
FIG. 3 is a schematic block diagram of a portion of the communications system
of FIG.
2 depicting the NEXT impairment signals received by receiver A from adjacent
transmitters B,
C, and D;
FIG. 4 is a schematic block diagram of a portion of the communications system
of FIG.
2 depicting the echo impairment signal received by receiver A from transmitter
A;
FIG. 5 is a schematic block diagram of a portion of the communications system
of FIG.
2 depicting the FEXT impairment signals received by receiver A from opposite
transmitters F,
G, and H;
FIG. 6 is a schematic block diagram of a communications system including a
plurality of
transceivers, each having a NEXT cancellation system, an echo canceller, a
feed forward
equalizer, digital adaptive filter system including one detector, and a timing
recovery circuit;
FIG. 7 depicts an impulse response for an echo signal passing through a 100m
communications line;
FIG. 8 depicts an impulse response for an NEXT signal passing through a 100m
communications line;
FIG. 9 is a schematic block diagram of a communications system including a
plurality of
transceivers each having a NEXT cancellation system, an echo canceller, and a
FEXT
cancellation system, a digital adaptive filter system including a plurality of
detectors and a skew
adjuster, and a timing recovery circuit;
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FIG. 10 is a schematic block diagram of a symbol-by-symbol detector of FIG. 9,
each
including a plurality of slicers, feedback filters and adders and receiving as
input a soft decision;
FIG. 11 is a schematic block diagram of the NEXT cancellation systems of FIG.
9, each
including a plurality of adaptive transversal filters (ATF) and adders and
receiving as input
transmitted signals from adjacent transmitters;
FIG. 12 is a schematic block diagram of the echo cancellers of FIG. 9, each
including an
ATF and receiving as input transmitted signals from same transmitters;
FIG. 13 is a schematic block diagram of the FEXT cancellation systems of FIG.
9, each
including a plurality of ATFs and an adder and receiving as input transmitted
signals from
opposite transmitters;
FIG. 14 depicts a direct impulse response arriving at the receiver after a
FEXT impulse
response;
FIG. 15 depicts a direct impulse response and FEXT impulse response arriving
at the
receiver at substantially the same time;
FIG. 16 depicts a direct impulse response arriving at the receiver before a
FEXT impulse
response;
FIG. 17 is a schematic block diagram of a communications system in including a
plurality
of transceivers, each having a NEXT cancellation system, an echo canceller,
and a FEXT
cancellation system, digital adaptive filter system including one detector,
and a timing recovery
circuit;
FIG. 18 is a schematic diagram of an ATF, including a cascade of taps, present
in the
NEXT cancellation system of FIG. 11, the echo cancellers of FIG. 12, and the
FEXT cancellation
system of FIG. 13;
FIG. 19 is a flow chart illustrating one embodiment of a power dissipation
reduction
method;
FIG. 20 depicts the mean squared error (MSE) to signal ratio as a function of
time during
the initial convergence of a communications system;
FIG. 21 depicts the taps of an echo canceller which remain active after
convergence of
a communications system having an error threshold of 24 dB;
FIG. 22 depicts the taps of an echo canceller which remain active after
convergence of
a communications system having an error threshold of 26 dB;
FIG. 23 is a flow chart illustrating another embodiment of a power dissipation
reduction
method;
FIG. 24 is a schematic block diagram depicting the master-slave relationship
between the
transceivers of each of the transceiver channels of FIG. 2; and
FIG. 25 is a timing diagram depicting the stages of one embodiment of a
startup protocol;
and
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FIG. 26 is a timing diagram depicting the stages of another embodiment of a
startup
protocol.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIlVIENTS
The discussion in this specification may be considered to relate specifically
to a Gigabit
Ethernet for the purposes of explanation and understanding of the invention.
However, it will
be understood that the concepts of this invention and the scope of the claims
apply to other types
of communications systems than a Gigabit Ethernet.
Communications System Overview:
A communications system incorporating the features of this invention is
generally
indicated at 10 in FIG. 1. The system 10 includes a hub 12 and a plurality of
computers serviced
by the hub in a local area network (LAN). Four computers 14 are shown by way
of illustration
but a different number of computers may be used without departing from the
scope of the
invention. Each of the computers 14 may be displaced from the hub 12 by a
distance as great
as approximately one hundred meters (100 m.). The computers 14 are also
displaced from each
other.
The hub 12 is connected to each of the computers 14 by a communications line
16. The
communication line 16 comprises a plurality of unshielded twisted pairs of
wires or cables.
Generally, the wires or cables are formed from copper. Four unshielded twisted
pairs of wires
are provided in the system 10 between each computer and the hub 12. The system
shown in FIG.
1 is operative with several categories of twisted pairs of cables designated
as categories 3, 4, 5,
6 and' 7 in the telecommunications industry. Category 3 cables are the poorest
quality (and
lowest cost) and category 6 and 7 cables are the best quality (and highest
cost). Gigabit Ethernet
uses category 5 cables.
FIG. 2 illustrates, in detail, a portion of the communications system of FIG.
1 including
one communications line 16 and portions of one of the computers 14 and the hub
12. The
communications line 16 includes four unshielded twisted pairs of wires 18
operating at 250
Mb/second per pair. A transceiver 20, including a transmitter (TX) 22 and
receiver (RX) 24, is
positioned at each unshielded end of each twisted pair 18. Between each
transceiver 20 and its
associated unshielded twisted pair 18 is a hybrid 26. The hybrid 26 is the
interface to the
communication line 16 which allows for fall-duplex bidirectional operation
between the
transceivers 20 at each end of the communications line. The hybrid also
functions to isolate the
transmitter and receiver associated with the transceiver, from each other.
The communications system includes a standard connector designated as a
gigabit media
independent interface (GMII) 28. The GIvIII 28 may be an eight bit wide data
path in both the
transmit and receive directions. Clocked at a suitable frequency, such as 125
MHz, the GNIII
results in a net throughput in both directions of data at a suitable rate such
as 250 Mb/second per
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pair. The GIVIII provides a symmetrical interface in both the transmit and
receive directions. A
physical coding sublayer (PCS) 30 receives and transmits data between the GMII
28 and the
tcansceivers 20. The PCS 30 performs such functions as scrambling and
encoding/decoding data
before forwarding the data to either the transceiver or the GMII. The PCS
encodes bits from the
GM1I into 5-level pulse amplitude modulation (PAM) signals. The five symbol
levels are -2, -1,
0, +1, and +2. The PCS also controls several functions of the transceivers,
such as skew control
as explained below.
Transceiver Circuitry:
Four of the transceivers 20 are illustrated in detail in FIG. 9. The
components of the
transceivers 20 are shown as overlapping blocks, with each layer corresponding
to one of the
transceivers. The GMII 28, PCS 30, and hybrid 26 of FIG. 9 correspond to the
G1VIII, PCS, and
hybrid of FIG. 2 and are considered to be separate from the transceiver. The
combination of the
transceiver 20 and hybrid 26 forms one "channel" of the communications system.
Accordingly,
FIG. 9 illustrates four channels, each of which operate in a similar manner.
The transmitter portion of each transceiver 20 includes a pulse shaping filter
32 and a
digital to analog (D/A) converter 34. In a preferred embodiment of the
invention the D/A
converter 34 operates at 125 MHz. The pulse shaping filter 32 receives one one-
dimensional (1-
D) symbol from the PCS. This symbol is referred to as a TXDatax symbol 36,
where x is 1
through 4 corresponding to each of the four channels. The TXDatax symbo136
represents 2 bits
of data. The PCS generates one 1-D symbol for each of the channels. The symbol
for each
channel goes through a spectram shaping filter of the form 0.75 + 0.25z' at
the pulse shaping
filter 32 to limit emissions within FCC requirements. This simple filter
shapes the spectrum at
the output of the transmitter so that its power spectral density falls under
that of communications
systems operating at 100 Mb/second on two pairs of category -5 twisted pair
wires. The symbol
is then converted into an analog signal by the D/A converter 34 which also
acts as a lowpass
filter. The analog signal gains access to the unshielded twisted pair wire 18
through the hybrid
circuitry 26.
The receiver portion of each transceiver includes a signal detector 41, an A/D
converter
42, a FIFO 44, a digital adaptive equalizer system, a timing recovery circuit
and noise reduction
circuitry. The digital adaptive equalizer system includes a feed-forward
equalizer (FFE) 46, two
devices 50, 56, a skew adjuster 54 and two detectors 58, 60. The functions of
these components,
as related.to the present invention, are explained below. The noise reduction
circuitry includes
a NEXT cancellation system 38, an echo canceller 40, and a FEXT cancellation
system 70.
The A/D converter 42 provides digital conversions of the signals received from
the hybrid
26 at a suitable frequency, such as 125 MHz, which is equal to the baud rate
of the signals. The
A/D converter 42 samples the analog signals in accordance with an analog
satnple clock signal
78 provided by the decision-directed timing recovery circuit 64. The FIFO 44
receives the
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digital conversion signals from the A/D converter 42 and stores them on a
first-in-first-out basis.
The FIFO 44 forwards individual signals to the FFE 46 in accordance with a
digital sample clock
signa180 provided by the timing recovery circuit 64. The FFE 46 receives
digital signals from
the FIFO 44 and filters these signals. The FFE 46 is a least mean squares
(LMS) type adaptive
filter which performs channel equalization and precursor inter symbol
interference (ISI)
cancellation to correct for distortions in the signal.
It is noted that the signal introduced into the A/D converter 42 and
subsequently into the
FIFO 44 and FFE 46 has several components. These components include the direct
signal
received directly from the transmitter 22 at the opposite end of the
unshielded twisted pair wire
18 with which the receiver 24 is associated. Also included are one or more of
the NEXT, echo,
and FEXT impairment signals from other transmitters 22 as previously
described. The signal
including the direct signal and one or more of the impairment signals is
referred to as a
"combination signal."
The FFE 46 forwards the combination signal 48 to a second device 50, typically
a
summing device. At the second device 50 the combination signal 48 is combined
with the
outputs of the NEXT cancellation system 38 and echo canceller 40 to produce a
signal which is
substantially devoid of NEXT and echo impairment signals. This signal is
referred to as a "first
soft decision" 52. The signal detector 41 detects the signals from the second
device 50 and
forwards the signals to the skew adjuster 54. Upon signal detection, the
signal detector 41
initiates various system operations, one of which -- as described below -
includes transitioning
between phases of a startup protocol. The skew adjuster 54 receives the first
soft decision 52
from the second device 50 and outputs a signal referred to as a "second soft
decision" 66. The
skew adjuster 54 performs two functions. First, it compensates for the
difference in length of the
unshielded twisted pairs 18 by delaying the first soft decision 52 so that the
second soft decisions
66 from all of the receivers in the system are in sync. Second, it adjusts the
delay of the first soft
decision 52 so that the second soft decision 66 arrives at the first device 56
at substantially the
same time as the output of the FEXT cancellation system 70. The skew adjuster
54 receives
skew control signals 82 from the PCS 30.
The skew adjuster 54 forwards the second soft decision 66 to a first device
56, typically
a sumrning device. At the first device 56 the second soft decision 66 is
combined with the output
of the FEXT cancellation system 70 to produce a signal which is substantially
devoid of FEXT
impairment signals. This signal is referred to as a"third soft decision" 68.
The first detector
58 receives the third soft decision 68 from the first device 56. The first
detector 58 provides an
output signal, i. e., a "final decision" 72. The first detector 58 may be a
slicer which produces
a final decision 72 corresponding to the analog signal level closest in
magnitude to the level of
the third soft decision 68. The first detector 58 may also be either a symbol-
by-symbol detector
or a sequential detector which operates on sequences of signals across all
four channels
simultaneously, such as a Viterbi decoder.
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In one configuration of the transceiver the first detector 58 is a symbol-by-
symbol
detector. A group of symbol-by-symbol detectors 58, one for each channel, is
shown in FIG. 10.
Each first detector 58 includes a slicer 98, adaptive feedback filter 100 and
an adder 102. The
adder 102 combines the third, soft decision 68 with the output of the adaptive
feedback filter 100
to provide an output which is introduced to the slicer 98. The output of the
slicer 98 in
introduced to the adaptive feedback filter 100. The first detector 58 provides
an output signal
72 which corresponds to the discrete level from the set [-2, -1, 0, 1, 2]
which is closest to the
difference between the third soft decision 68 and the output of the feedback
filter 100. The
adaptive feedback filter 100 corrects for distortion in the third soft
decision 68. This filter 100
uses past slicer 98 decisions to estimate postcursor ISI caused by the
channel. This ISI is
canceled from the third soft decision 68 to form the final decision signal 72.
In another configuration of the transceiver the first detector 58 is a
combination of a
sequential decoder with a decision feedback equalizer (DFE) using the
architecture usually
known as multiple DFE architecture (IVIDFE) sequential detector. The
sequential decoder 58
looks at all signals from all four channels at the same time and at successive
samples from each
channel over several periods of unit time. A sequential decoder receives as
input at least one
signal from each of the first devices 56. The sequential decoder 58, in
general, is responsive to
the sequences of the output signals from the first devices 56 for (1) passing
acceptable sequences
of such signals and (2) discatriing unacceptable sequences of such signals in
accordance with the
constraints established by the code standard associated with the system.
Acceptable sequences
are those which obey the code constraints and unacceptable sequences are those
which violate
the code constraints.
The second detector 60 (FIG. 9) receives the first soft decision 52 from the
second device
50. The second detector 60 is a symbol-by-symbol detector similar to the first
detector 58 (FIG.
10). It provides an output signal 74 which corresponds to the discrete level
from the set [-2, -1,
0,1, 2] which is closest to the difference between the first soft decision 52
and the output of the
feedback filter 100. The second detector 60 produces output signals 74 without
the benefit of
FEXT cancellation, as a result, these .decisions have a higher error rate than
those made by the
first detector 58, which enjoys the benefits of FEXT cancellation. Because of
this fact, these
decisions are called "tentative decisions". It is important to note that the
postcursor ISI present
in the input to the second detector 60 is canceled using the adaptive feedback
filter 100, (FIG.
10) contained within the second detector, whose inputs are the tentative
decisions 74. The
coefficients of this adaptive feedback filter 100 are the same as those of the
adaptive feedback
filter associated with the first detector 58 (FIG. 9).
A third device 62, typically a summing device, receives the first soft
decision signa152
from the second device 50 and the tentative decision signals 74 from the
second detector 60.
At the third device 62 the first soft decision 52 is combined with the
tentative decision signa174
to produce an error signal 76 which is introduced into the timing recovery
circuit 64. The timing
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recovery circuit 64 receives the tentative decision 74 from the second
detector 60 and the error
signals 76 from the third device 62. Using these signals as inputs the timing
recovery circuit 64
outputs an analog clock sync signal 78 which is introduced to the A/D
converter 42 and a digital
clock sync signal 80 which is introduced into the FIFO 44. As previously
mentioned, these
signals control the rate at which the A/D converter 42 samples the analog
input it receives from
the hybrid 26 and the rate at which the FIFO forwards digital signals to the
FFE 46.
As mentioned before, the symbols sent by the transmitters 22 (FIG. 2) in the
communications system cause NEXT, echo and FEXT impairments in the received
signal for
each channel. Since each receiver 24 has access to the data for the other
three channels that
cause this interference, it is possible to nearly cancel each of these
effects. NEXT cancellation
is accomplished using three adaptive NEXT cancelling filters 84 as shown in
the block diagram
of FIG. 11. Each NEXT cancellation system 38 receives three TXDatax symbols 36
from each
of the transmitters at the same end of the communications line 18 as the
receiver with which the
NEXT cancellation system is associated. Each NEXT cancellation system 38
includes three
filters 84, one for each of the TXDatax symbols 36. These filters 84 model the
impulse
responses of the NEXT noise from the transmitters and may be implemented as
adaptive
transversal filters (ATF) employing, for example, the LMS algorithm. The
filters 84 produce a
replica of the NEXT impairment signal for each TXDatax symbol 36. A summing
device 86
combines the three individual replica NEXT impairment signals 92 to produce a
replica of the
NEXT impairment signal contained within the combination signal received by the
receiver with
which the NEXT cancellation system 38 is associated. The replica NEXT
impairment signal 88
is introduced into the second device 50 (FIG. 9) where it is combined with the
combination
signal 48 to produce a first soft decision signal 52 which is substantially
devoid of NEXT
impairment signals.
Echo cancellation is accomplished with an adaptive echo cancelling filter 85
as shown in
the block diagram of FIG. 12. Each echo canceller 40 receives the TXDatax
symbols 36 from
the transmitter at the same end of the twisted wire pair 18 as that of the
receiver with which the
echo canceller is associated. As shown in FIG. 12, each echo canceller 40
includes one filter
85. These filters 85 model the impulse responses of the echo noise from the
transmitter and may
be implemented as ATFs employing, for example, the LMS algorithm. The filter
produces a
replica of the echo impairment signal contained within the combination signal
received by the
receiver with which the echo canceller 40 is associated. The replica echo
impairment signa190
is introduced into the second device 50 (FIG. 9) where it is combined with the
combination
signal 48 to produce a first soft decision signal 52 which is substantially
devoid of echo
impairment signals.
FEXT cancellation is accomplished with three adaptive FEXT cancelling filters
87 as
shown in the block diagram of FIG. 13. Each FEXT cancellation system 70
receives three
tentative decision symbols 74, one from each of the receivers at the same end
of the
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communications line as the receiver with which the FEXT cancellation system is
associated.
Each FEXT cancellation system 70 includes three filters 87, one for each of
the tentative decision
symbols 74. These filters 87 model the impulse responses of the FEXT noise
from transmitters
and may be implemented as ATFs employing, for example, the LMS algorithm. The
filters 87
produce a replica of the FEXT impairment signal 96 for each individual
tentative decision
symbol 74. A summing device 108 combines the three individual replica FEXT
impairment
signals 96 to produce a replica of the FEXT impairment signal contained within
the combination
signal 48 received by the receiver with which the FEXT cancellation system is
associated. The
replica FEXT impairment signa194 is introduced into the first device 56 (FIG.
9) where it is
combined with the second combination signal 66 to produce the third soft
decision signal 68
which is substantially devoid of FEXT impairment signals. It is important to
note that the higher
error rate of the tentative decisions 74 does not degrade the performance of
the FEXT
cancellation system 70, because the decisions used to cancel FEXT are
statistically independent
from the final decisions 72 made by the receiver whose FEXT is being canceled.
The symbols provided by the first detector 58 are decoded and descrambled by
the receive
section of the PCS 30 before being introduced to the GMII. Variations in the
way the wire pairs
are twisted may cause delays through the four channels by up to 50
nanoseconds. As a result,
the symbols across the four channels may be out of sync. As previously
mentioned, in the case
where the fitst detector is a sequential detector, the PCS also determines the
relative skew of the
four streams of I-D symbols and adjusts the symbol delay, through the skew
adjuster 54, prior
to their arrival at the first detector 58 so that sequential decoder can
operate on properly
composed four-dimensional (4-D) symbols. Additionally, since the cabling plant
may introduce
wire swaps within a pair and pair swaps among the four unshielded twisted
pairs, the PCS 30
also determines and corrects for these conditions.
Noise Reduction:
As previously mentioned, FEXT is an impairment that results from capacitive
coupling
of the signal from the far-end transmitters to the input of the receivers, as
shown in FIG 5. The
crosstalk signals from transmitters F, G, and H appear as noise to receiver A,
which is attempting
to detect the signal from transmitter E. A similar situation applies to all
other receivers regarding
the signals from the appropriate transmitters located at the opposite end of
the line.
The FEXT noise experienced by receiver A and originating from transmitter F
can be
modeled as the convolution of the data symbols transmitted by F with a certain
impulse response
that depends on the properties of the cable and models the coupling
characteristics of the
unshielded twisted pairs used by transmitter F and receiver A. A typical
measured FEXT
impulse response 104 is show in FIGS 14-16. A similar description can be given
for all the other
possible receiver-transmitter combinations. Therefore, there are a total of
twelve FEXT impulse
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responses 104 describing the FEXT noise signals from transmitters E, F, G, and
H to receivers
A, B, C, and D. These twelve impulse responses are not identical, although
each has a general
shape similar to that shown in FIGS 14-16.
Although FEXT is an impairment for many communications systems other than
Gigabit
Ethernet, in these systems a given receiver usually does not have access to
the symbols detected
by the other receivers, because these receivers may not be physically located
in the same place,
and/or because they operate at rates that are not synchronized to the data
rate of the receiver
suffering from FEXT. Aspects of the present invention takes advantage of the
fact that in
Gigabit Ethernet transceivers the decisions that correspond to all four
channels are available to
the four receivers and the decisions may be made synchronous.
In operation there may be delays associated with the transmission of signals
across the
communications line. The synchronization of the signals within the system is
crucial to effective
cancellation of the noise. It is important that the replica noise impairment
signals arrive at the
summing devices at substantially the same time as the combination signal
and/or soft decision
signals. With regard to the FEXT impairment signal, because the impairment is
caused by the
transmitters at the opposite end of the receiver there is likely to be a delay
between the time that
the second soft decision signal 66 arrives at the first device 56 and the time
at which the replica
FEXT impairment signal 94 arrives. In some channels, as illustrated in FIG.
14, the group delay
of the FEXT signal 104 could be smaller than the group delay of the desired
signal 106. In this
case the tentative decisions 74 provided by receivers B, C, and D of FIG. 5
arrive at the FEXT
cancellation system 70 of receiver A too late to be effective in canceling the
FEXT impairment.
To compensate for this delay, the invention employs a skew adjuster 54 which,
as
previously stated, delays the first soft decision signal 52 by a time
substantially equal to or
greater than the time delay between the arrival at the receiver of the direct
signal and the FEXT
impainnent signals associated with such receiver. If the output is delayed by
an amount greater
than the time delay, which would result in the situation illustrated in FIG.
16, the adaptive
feedback filter 87 (FIG. 13) within the FEXT cancellation system 70
compensates for the over
delay by delaying the replica FEXT impairment signal 94 so that it arrives at
the first device 56
(FIG. 9) at substantially the same time as the second soft decision signal 66.
The third soft decision 68 resulting from FEXT cancellation allows the first
detector 58
to make more reliable final decisions 72, with a greatly reduced error rate.
Computer simulations
show that a typical improvement achievable with the invention described herein
is approximately
2 to 3dB when the signal to noise ratio at the input of the first detector 58
before FEXT
cancellation is approximately 25dB. This corresponds to a reduction of the
symbol error rate of
a factor 1000 or larger.
If there is no delay associated with the transmission of signals across the
communications
line both the FEXT impairment signa1104 and the direct signal 106 arrive at
the receiver at the
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substantially the same time, as shown in FIG. 15. In this situation, the delay
of the skew adjuster
54 is set to zero. In the alternative, an embodiment of the invention, as
shown in FIG. 17, with
only one detector 110 and one summing device 112 may be used. In this
configuration, the
samming device 112 receives the replica NEXT, echo, and FEXT impairment
signals 88, 90, 94
and the combination signal 48 and produces a first soft decision 52
substantially devoid of
impairment signals. This first soft decision 52 is introduced into the
detector 110 and the third
device 62. The detector 110 may include either a single symbol-by-symbol
detector or both a
symbol-by-symbol detector and a sequential detector. In the case of a symbol-
by-symbol
detector the final decision 72 and second output 114 of the detector 110 are
identical. In the
case of both a symbol-by-symbol detector and a sequential detector, the final
output is provided
by the sequential detector and is introduced to the PCS 30. The second output
114 is provided
by the symbol-by-symbol detector and is introduced to the timing recovery
circuitry 64 and the
third device 62 for use in determining the error signal 76.
Power Dissipation Reduction:
As previously mentioned, the NEXT cancellation system, echo canceller and FEXT
cancellation system use ATFs to effectively cancel the noise from the
combination signal. An
example of an ATF which may be employed is shown in FIG. 18. The ATF 120
includes a
plurality of taps 122 each including a multiplier 124 and an adder 126.
Associated with each tap
122 is a coefficient Cn, where n is 0 though x-1 where x is the number of taps
in the ATF. The
circuitry associated with each tap 122 includes a 1-bit storage (not shown)
that allows for
activation and deactivation of the tap. The values of the coefficients Cn are
adjusted in
accordance with an LMS algorithm as mentioned before. Interposed between the
taps 122 are
registers 128. These registers 128 provide data to the taps 122 at timed
intervals in accordance
with a clock signal.
The impulse responses of an echo and NEXT, as shown in FIGS. 7 and 8,
indicates that
not all taps 122 in the NEXT and echo cancellers 38, 40 are contributing
significantly to the
performance of the communications system. Aspects of the present invention
deterniines what
taps 122 are not contributing significantly to the reduction of the mean
squared error (MSE) of
the system and deactivates these taps, thereby eliminating them from the
filtering computation
and thus reducing considerably the power dissipation of the system.
Furthermore, as shown by
the impulse response of FIGS. 7 and 8, the need to build NEXT and echo
cancellers 38, 40 with
a long span is difficult to avoid. Specific cable responses may differ from
the one depicted in
FIGS. 7 and 8, and accordingly require more or fewer taps 122 then that
required for the cable
of FIGS. 7 and 8. As mentioned before it is difficult to determine a priori
what taps 122 are
needed with a particular cable.
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In accordance with aspects of the present invention, the NEXT , echo, and FEXT
cancellers 3 8, 40, 70 are configured with ATFs 120 which employ a sufficient
number of taps
122 to provide adequate cancellation with the worst-case expected impulse
responses. This may
require 140 taps 122 as in the example of FIG. 7, or even more for longer
cables. To reduce the
power dissipation, the taps 122 are examined after convergence, and those taps
that are found
not to contribute significantly to the performance of the system are
deactivated. When the tap
122 is deactivated, it is removed from the NEXT, echo and FEXT replica
computation and from
the adaptation and its contribution to the overall power dissipation of the
system is substantially
eliminated.
When the system is initially converged, all taps 122 are active, so the NEXT,
echo and
FEXT cancellers 38, 40, 70 are converged along their entire length. After
convergence, the taps
122 are examined to determine which ones can be deactivated using the tap
scanning algorithm
depicted in FIG. 19. At step S 1, an acceptable level of error for the system
Sea is specified. At
step S2, a tap coefficient threshold Tth is set for each active tap. While
each individual tap may
have a unique tap threshold Tth, in a preferred embodiment of the invention
the tap thresholds
for all taps are substantially the same. The initial value of the tap
coefficient threshold Tth is
sufficiently low such that only a few taps 122 are deactivated and the
performance of the system
is not significantly affected. In a preferred embodiment of the invention, the
tap coefficient
threshold Tth is initially set equal to the tap coefficient Cn having the
minimum absolute value.
Alternatively, a reasonable value can be determined by simulation. This
initial value is not
critical, as long as it is sufficiently low to avoid a large degradation of
the performance of the
system the first time the tap scanning procedure is applied.
At step S3, the absolute value of the tap coefficient Cn for each active tap
is compared
to the tap coefficient threshold Tth. If the tap coefficient Cn is less than
the tap coefficient
threshold Tih the tap 122 is deactivated at step S4. This process is repeated
for each tap 122
in the filter 120. Preferably, the determination of whether to deactivate a
tap 122 is done in a
sequential manner starting at the input end of the filter 120. At step S5, the
error for the system
Sec is computed. This error is computed by first computing the MSE for each
active tap 122
by multiplying the absolute value of the tap coefficient Cn by the average
energy signal. The
error of the filter 120 associated with the tap 122 is determined by summing
the individual tap
errors. The error of the system is then determined by summing the individual
filter errors.
In step S6, the computed sysGem error Sec is compared to the specified
acceptable system
error Sew If the computed system error Sec is less than the acceptable system
error Sea, the tap
threshold Tth for each active tap is increased by a small amount at step S7
and steps S3 through
S6 are repeated. As a result, some additional taps 122 are deactivated, but
the number of taps
deactivated is usually not very large because of the small increase in the tap
threshold Tth.
Steps S3 through S6 are repeated until the computed system error Sec
approaches the
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acceptable system error Sea without exceeding the acceptable system error. If
the computed
system error Sec is greater than the acceptable system error Sea, the tap
scanning algorithm
stops.
As an alternative to determining whether to deactivate a tap based on the MSE
of the
communications system, a determination may be made based on the MSE of an
individual
filter. In this embodiment of the invention an acceptable level of error for a
filter Fea is
specified. Individual taps are deactivated, as previously described, and a
computed filter error
FeC is calculated. If the deactivation of a tap does not cause the computed
filter error Fec to
exceed the acceptable filter error Fea the tap remains inactive.
In yet another embodiment of the invention, the contribution of each
deactivated tap to
the MSE of the filter and, in turn, the system is calculated. If the MSE
contribution of the tap
is deternzined to be an acceptable amount the tap remains deactivated. This
method is generally
preferred during the initial startup of the system when the overall MSE of the
system is large
due to the nonconverged state of the filter coefficients within the system.
Once the filter
coefficients of the system have initially converged, the deternunation of
whether to deactivate
a tap is generally made based on the taps contribution to the MSE of the
system as previously
described with reference to FIG. 19.
The final result of the tap scanning algorithm is that, in typical channels of
the
communication system, a large number of taps 122 is deactivated, and power
dissipation is
reduced by a large factor. As an example, computer simulations of the tap
scanning algorithm
when operating on the channel whose echo response is shown in FIG. 7 are
presented in FIG.
20. This figure shows both the master and slave MSE to signal ratios as a
function of time
during the initial convergence of the system. At time t=360,000 bauds, the tap
scanning
algorithm begins, and as a result the MSE to signal ratio increases to the
prespecified target of
24dB. FIG. 21 shows the taps of the echo canceller after convergence with a
threshold of 24
dB and FIG. 22 shows the taps with a threshold of 26 dB. The deactivated taps
are shown as
zeros. In FIG. 21, the total number of active taps for the echo canceller is
twenty-two.
Similarly, the number of active taps 122 for the three NEXT cancellers (not
shown) forming
the NEXT cancellation system is six, two, and zero, respectively. In FIG. 22,
the total number
of active taps for the echo canceller is forty-seven. Similarly, the number of
active taps 122 for
the three NEXT cancellers (not shown) is six, two, and zero, respectively.
For the case of the 24 dB threshold, out of 440 initially active taps, only 30
remain active
after the application of the tap scanning algorithm, while maintaining a 5dB
margin for required
bit error rate. Notice from FIGS. 21 and 22 that those taps 122 which remain
active occur at
sparse locations, and it would have been difficult to statically allocate
these taps during the
design of the NEXT and echo cancellers, because the location of taps is highly
dependent on
the specific cable response.
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CA 02670691 2009-06-25
In another embodiment of the invention, the tap scanning algorithm monitors
the change
in MSE as the tap scanning algorithm progresses. The algorithm is applied
until the change in
MSE -- rather than the MSE itself -- exceeds a prespecified value, for example
1dB. If the
MSE to signal ratio before the scanning is applied is 25dB, the final MSE to
signal ratio is
24dB, and a large number of taps is deactivated. This embodiment of the tap
scanning
algorithm is depicted in FIG. 23. In step S 10, the initial system error is
Se1 computed. In step
S 11, an acceptable system error differential De is specified. In step S 12, a
tap coefficient
threshold Tth is set for each active tap. As with the other tap scanning
algorithm, the initial
value of the tap threshold Tth is sufficiently low such that only a few taps
122 are deactivated
and the performance of the system is not significantly affected.
At step S 13, the absolute value of the tap coefficient Cn , for each active
tap is
compared to the tap coefficient threshold Tth. If the tap coefficient C nis
less than the tap
coefficient threshold Tth the tap 122 is deactivated at step. S 14. This
process is repeated for
each tap 122 in the filter 120. Preferably, the determination of whether to
deactivate a tap 122
is done in a sequential manner starting at the input end of the filter 120. At
step S15, the
subsequent error for the system Ses is computed.
In step S 16, the subsequent error for the system Ses is compared to the to
the initial
system error is Se1. If the difference between the subsequent system error Ses
and the initial
system error Sei is less than a prespecified value, the tap threshold Tth is
increased by a small
amount at step S 17 and steps S 13 through S 16 are repeated. As a result,
some additional taps
122 are deactivated. Steps S13 through S 16 are repeated until the difference
between the
subsequent system error Ses and the initial system error Se1 exceeds the
prespecified value.
In some cases the impulse responses of NEXT, echo and FEXT may change during
normal operation, for example as a result of temperature changes. It is
therefore desirable to
periodically activate previously deactivated taps 122, preferably in a
sequential manner, and
recheck if the absolute value of the tap coefficient Cn is below the tap
threshold Tth. If a tap
coefficient Cn has grown to a value above the tap threshold Tth, the tap 122
remains active,
otherwise it is deactivated. Similarly, those taps 122 that were active may
fall below the tap
threshold Tth, in which case they are deactivated. All this can be
accomplished with a periodic
reapplication of the sequential tap scanning algorithm during normal
operation.
In an alternate embodiment of the invention a select number of taps 122, for
example
ten, positioned at the input end of the filter 120 are ilot subject to
deactivation. Usually there
is a large slew rate in these first few taps 122, which means that their
numerical value could
change significantly if the sampling phase changes. This sampling phase could
change
dynamically as a result of jitter, causing some previously deactivated taps
122 to become
significant. By maintaining a number of taps 122 at the input end of the
filter in an active state,
potential degradations in the presence ofjitter are avoided.
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When the total number of taps is very large, for example 440, the power
dissipation could be large during the initial convergence transient before the
tap
scanning algorithm has had a chance to deactivate the taps. Although the
average
power dissipation of the system is still greatly reduced, the peak power is
not. A
preferred embodiment of the invention compensates for this by converging the
NEXT,
echo and FEXT cancellers in stages. For example, a block of 20 taps is
converged at
a time, and the tap scanning algorithm is then applied to these taps on a per-
block
basis. In cases where the MSE during the initial convergence is large, for
example as
a result of the fact that the initial block of 20 taps may not be large enough
to provide
a lower MSE, it may be better to monitor the sum of the squared values of the
coefficients of deactivated taps 122 as a measure of whether the algorithm can
be
terminated.
Start-UR Protocols:
One of the most critical phases of the operation of the communications system
is the transceiver startup. During this phase the adaptive filters contained
within the
FFE 46 (FIG. 9), echo canceller 40, NEXT cancellation system 38, FEXT
cancellation
system 70, timing recovery system 64 and detector 58 of the receiver portion
of each
transceiver converge. During convergence the actual output of the adaptive
filters are
compared to expected output of the filters to determine the error. The error
is
reduced to substantially zero by adjusting the coefficients of the algorithm
which
defines the transfer function of the filter. Similarly, the timing recovery
system is
converged by adjusting the frequency and phase of the phase lock loop and the
local
oscillator contained within the timing recovery system so that the signal-to-
noise ratio
of the channel is optimized. In addition, the differences in delay among the
four wire
pairs are compensated, and pair identity and polarity, are acquired.
Successful
completion of the startup ensures that the transceiver can begin normal
operation.
In accordance with aspects of the present invention each of the transceiver
channels operate in a loop-timed fashion, as shown in FIG. 24. The
transceivers 20
at the two ends of the each twisted wire pair 18 assume two different roles as
far as
synchronization is concerned. One of the transceivers, called the master 130,
transmits
data using an independent dock GTX CI:K provided through the GMII interface 28
(FIG. 9). This clock signal is fixed in both frequency and phase and is
provided to
the master transceiver 130 of each the four transceiver channels in the
communications
system. In actuality the transmit clock used by the master 130 may be a
filtered
version of GTX_CLK, obtained using a phase locked loop with a very narrow
bandwidth, to reduce jitter. The transceiver 20 at the other end of the
twisted wire
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CA 02670691 2009-06-25
pair 18, called the slave 132, synchronizes both the frequency and phase of
its receive
and transmit clocks to the signal received from the master 130, using the
timing
recovery system 64 (FIG. 9) located in the receiver 24. The slave 132 transmit
clock
maintains a fixed phase relationship with the slave receive clock at all
times. The
receive clock at the master 130 synchronizes, in phase but not in frequency,
with the
signal received from the slave transmitter 22. Thus, after an initial
acquisition period,
the master 130 receive clock follows the master transmit clock with a phase
difference
determined by the round trip delay of the loop. This phase relationship may
vary
dynamically as a result of the need of the master 130 receive clock to track
jitter
present in the signal received from the slave 132.
Slave Transmit Clock Fixed:
The sequence of events during the startup protocol of the present invention is
shown in FIG. 25. The protocol consists of three phases 134, 136, 138 during
which
the receivers are trained, e. g., adaptive filters are converged, timing
synchronization
is acquired, etc., followed by normal operation which begins during phase four
140.
During the first phase 134, the master begins transmitting to the slave using
a transmit
dock signal that is fixed in both frequency and phase. The master trains its
near-end
noise reduction system by converging the adaptive filters contained within its
echo
canceller and NEXT cancellation system (E). At the same time, the slave trains
its
equalizers and far-end noise reduction system by converging the adaptive
filters
contained within its DFE, FFE and FEXT cancellation system (D). While training
its
equalizer and far-end noise reduction system the slave simultaneously acquires
timing
synchronization in both frequency and phase M. It may also at this time
compensate
for the differential delay among the four twisted wire pairs, identify the
four pairs, and
correct the polarity of the pairs.
In one embodiment of the protocol, the transition from the first phase 134 to
the second phase 136, at both master and slave, occurs after a fixed and
prespecified
period of time. In a preferred embodiment, however, the slave transitions from
the
first phase 134 to second phase 136 when it detects that its receiver has
converged the
adaptive filters contained within its DFE, FFE and FEXT cancellation system
(D) and
has acquired timing synchronization ('I'). As previously mentioned, the master
receiver indudes a signal detector 41 (FIG. 9) which detects energy in the
line coming
from the slave. The master transitions from the first phase 134 to the second
phase
136 when it detects this energy from the slave. Therefore, the slave takes the
initiative
in transitioning from the first phase 134 to the second phase 136, and the
master
follows when it detects the signal from the slave.
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CA 02670691 2009-06-25
The convergence of the echo canceller and NEXT cancellation system during
the first phase 134 at the master is done with the objective of allowing the
signal
detector at the master to detect the signal from the slave. Without proper
echo and
NEXT cancellation, the signal detector would be triggered by the echo and NEXT
noise present in the receiver. After the transition has occurred, the master
discards the
echo canceller and NEXT cancellation system coefficients which result from the
converging in the first phase 134. This may be done by resetting the adaptive
filters
in the echo canceller and NEXT cancellation system. It is important to note
that the
correct sampling phases for the four receivers at the master is obtained
during the third
phase 138, therefore the echo canceller and NEXT cancellation system
coefficients
obtained during the first phase 134 may differ from the final values to be
reacquired
in the third phase 138.
During the second phase 136, the slave trains its near-end noise reduction
system
by converging the adaptive filters contained within its echo canceller and
NEXT
cancellation system (E). At the same time, the master trains its equalizers
and far-end
noise reduction system by converging the adaptive filters contained within its
DFE,
FFE and FEXT cancellation system (D). While training its equalizers and far-
end
noise reduction system, the master simultaneously acquires timing
synchronization in
phase only (P). The master may also at this time compensate for the
differential delay
among the four twisted wire pairs, identify the four pairs, and correct the
polarity of
the pairs. During the second phase, the slave saves the timing recovery state
variables
that had been acquired during the first phase 134, and freezes its frequency
and phase.
By doing this, the slave is guaranteed to sample with the correct phase, the
signal
coming to it from the master when the master resumes transmission at the
beginning
of the third phase 138. The slave also freezes the coefficients of the DFE,
FFE and
FEXT cancellation system acquired during the first phase 134.
Similar to the transition from the first phase 134 to the second phase 136,
the
transition from the second phase 136 to the third phase 138 may occur after a
fixed
and prespecified period of time. While the duration of the first, second, and
third
phases 134, 136, 138 is fixed, the duration is not necessarily equal for all
phases. In
a preferred embodiment, however, the master transitions from the second phase
136
to third phase 138 when it detects that its receiver has converged the
adaptive filters
contained within its DFE, FFE and FEXT cancellation system (D) and has
acquired
timing synchronization (P). Like the master, the slave receiver includes a
signal
detector 41 (FIG. 9) which detects energy in the line coming from the master.
The
slave transitions from the second phase 136 to the third phase 138 when it
detects this
energy from the master. Therefore the master takes the initiative in
transitioning from
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CA 02670691 2009-06-25
the second phase 136 to the third phase 138, and the slave follows when it
detects the
signal from the master.
During the third phase 138 the slave freezes the coefficients of the echo
cancellers and NEXT cancellation system and maintains a steady state condition
during
which the operating characteristics of the slave are not adjusted. Similarly,
the master
freezes the coefficients of the DFE, FFE and FEXT cancellation system and the
phase
of its clock signal. The master also retrains its near-end noise reduction
system by
reconverging its echo canceller and NEXT cancellation system (E) during the
third
phase 138. It is important to note that in the third phase 138 the slave
resumes
transmission using the dock recovered from the signal transmitted by the
master, and
therefore the master already knows the correct frequency with which to operate
its
receiver. The "relative sampling phases" of the four receivers, i. e., the
differences in
sampling phases of three of the receivers versus one of them arbitrarily used
as
reference, are also known, because they were acquired during the second phase
136.
However, the "overall sampling phase" of the receivers, i. e., the sampling
phase of the
receiver arbitrarily chosen as reference, is not yet known and has to be
acquired
during the third phase 138. When both master and slave have completed their
training
operations, they exchange messages indicating that they are ready to transmit
valid
data. During phase four 140, all coefficients of the adaptive filters
previously frozen
are unfrozen and the transmission of data is ready to take place.
Slave Transmit Clock Free-Running:
The sequence of events during the startup protocol of the present invention is
shown in FIG. 26. The protocol consists of three phases 144, 146, 148 during
which
the receivers are trained, e. g., adaptive filters are converged, timing
synchronization
is acquired, etc., followed by normal operation which begins during phase four
150.
The startup protocol is initiated by the master when it starts transmitting a
signal to
the slave using a transmit dock signal which is fixed in both frequency and
phase.
During the first phase 144, the master trains its near-end noise reduction
system by
converging the adaptive filters contained within its echo canceller and NEXT
cancellation system (E). As previously mentioned, the slave receiver includes
a signal
detector 41 (FTG. 9) which detects energy in the line coming from the master.
Upon
detection of the signal from the master, the slave trains its equalizers and
far-end noise
reduction system by converging the adaptive filters contained within its DFE,
FFE and
FEXT cancellation system (D). While training its equalizers and far-end noise
reduction system the slave simultaneously acquires timing synchronization in
both
frequency and phase M. It may also at this time compensate for the
differential
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, ~.
CA 02670691 2009-06-25
delay among the four twisted wire pairs, identify the four pairs, and correct
the
polarity of the pairs.
In one embodiment of the protocol, the transition from the first phase 144 to
the second phase 146, at both master and slave, occurs after a fixed and
prespecified
period of time. In a preferred embodiment, however, the slave transitions from
the
first phase 144 to second phase 146 when it detects that its receiver has
converged the
adaptive filters contained within its DFE, FFE and FEXT cancellation system
(D) and
has acquired timing synchronization M. The master receiver also includes a
signal
detector 41 (FIG. 9) which detects energy in the line coming from the slave.
The
master transitions from the first phase 144 to the second phase 146 when it
detects this
energy from the slave. Therefore, the slave takes the initiative in
transitioning from
the first phase 144 to the second phase 146, and the master follows when it
detects the
signal from the slave.
The convergence of the echo canceller and NEXT cancellation system during
the first phase 144 at the master is done with the objective of allowing the
signal
detector at the master to detect the signal from the slave. Without proper
echo and
NEXT cancellation, the signal detector would be triggered by the echo and NEXT
noise present in the receiver. After the transition has occurred, the master
discards the
echo canceller and NEXT cancellation system coefficients which result from the
converging in the first phase 144. This may be done by resetting the adaptive
filters
in the echo canceller and NEXT cancellation system. It is important to note
that the
correct sampling phases for the four receivers at the master is obtained again
in the
third phase 148, therefore the echo canceller and NEXT cancellation system
coefficients obtained during the first phase 144 may differ from the final
values to be
reacquired in the third phase 148.
During the second phase 146, the slave trains its near-end noise reduction
system
by converging the adaptive filters contained within its echo canceller and
NEXT
cancellation system (E). The slave also freezes the coefficients of the DFE,
FFE and
FEXT cancellation system acquired during the first phase 144. At the same
time, the
master trains its equalizers and far-end noise reduction system by converging
the
adaptive filters contained within its DFE, FFE and FEXT cancellation system
(D).
While training its equalizers and far-end noise reduction system, the master
simultaneously acquires timing synchronization in both frequency and phase
(T). The
master may also, at this time, compensate for the differential delay among the
four
twisted wire pairs, identify the four pairs, and correct the polarity of the
pairs. The
slave may run off of any stable dock that has a frequency offset of less than
a
prespecified limit, e. g., for example 200ppm, when compared to the master
transmit
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CA 02670691 2009-06-25
clock. During the second phase 146, the slave transceivers transmit using a
free-
running clock. As a result, the master acquires both frequency and phase
synchronization during the second phase 146. Phase synchronization is a normal
function that the master performs in any form of startup protocol, however,
frequency
synchronization is not a usual master function, and it is only performed
during the
second phase 146 of the startup with the objective of interoperating properly
with
slave transceivers that do not save the timing recovery state variables at the
transition
from the first phase 144 to the second phase 146.
Similar to the transition from the first phase 144 to the second phase 146,
the
transition from the second phase 146 to the third phase 148 may occur after a
fixed
and prespecified period of time. While the duration of the first, second, and
third
phases 144, 146, 148 is fixed, the duration is not necessarily equal for all
phases. In
a preferred embodiment, however, the master transitions from the second phase
146
to third phase 148 when it detects that its receiver has converged the
adaptive filters
contained within its DFE, FFE and FEXT cancellation system (D) and has
acquired
tizning synchronization M. The master begins transmitting a signal to the
slave. The
slave transitions from the second phase 146 to the third phase 148 when it
detects this
signal from the master. Therefore the master takes the initiative in
transitioning from
the second phase 144 to the third phase 148, and the slave follows when it
detects the
signal from the master.
During the third phase 148, the slave freezes the coefficients of the echo
canceller and the NEXT cancellation system and maintains its near-end noise
reduction
system in a steady state condition. Because the timing recovery state
variables had
not been saved at the transition to the second phase 146, the slave also
reacquires
timing synchronization in both frequency and phase (T) during the third phase
148.
The master, during the third phase 148, freezes the coefficients of its DFE
and FEXT
cancellation system and the frequency of its clock signal. The master also
retrains its
near-end noise reduction system by reconverging its echo canceller and NEXT
cancellation system (E). The master also reacquires timing synchronization in
phase
only (P). It is important to note that in the third phase 148 the slave
resumes
transmission using the clock recovered from the signal transmitted by the
master, and
therefore the master already knows the correct frequency with which to operate
its
receiver. The "relative sampling phases" of the four receivers, i. e., the
differences in
sampling phases of three of the receivers versus one of them arbitrarily used
as
reference, are also known, because they were acquired during the second phase
146.
However, the "overall sampling phase" of the receivers, i. e., the sampling
phase of the
receiver arbitrarily chosen as reference, is not yet known and has to be
acquired
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CA 02670691 2009-06-25
during the third phase 148. When both master and slave have completed their
training
operations, they exchange messages indicating that they are ready to transmit
valid
data. During phase four 150, all coefficients of the adaptive filters
previously frozen
are unfrozen and the transmission of data is ready to take place.
Although this invention has been disclosed and illustrated with reference to
particular embodiments, the principles involved are susceptible for use in
numerous
other embodiments which will be apparent to persons of ordinary skill in the
art.
The invention is, therefore, to be limited only as indicated by the scope of
the
appended claims.
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