Note: Descriptions are shown in the official language in which they were submitted.
- P}~N.7730
l~Ts/Fl?rRc)M
1069588 3-6-1975
"Vestigial-sideband transmission system for synchronous
data signal~"
The invention relates to a vestigial-sideband
transmission system for transmission of synchronous data
signals from a transmitter to a receiver via a transmission
channel of restricted bandwidth, which transmitter is
provided with a data signal source, a clock signal source
for synchronizing the data signal source, a carrier source
and a filtering and modulating circuit which is connected
to the data signal source and to the carrier source for
generating a vestigial-sideband amplitude-modulated channel
signal,and which receiver is provided with a selecting
filter for the transmitted channel signal, a circuit for
recovering a reference carrier, a demodulator which is
connected to the reference carrier circuit for coherent
demodulation of the transmitted channel signal, a circuit
for recoverlng a reference clock signal and a regenerator
which is connected to the reference clock signal circuit
for regenerating the synchronous data signals.
Such vestigial-sideband systems use the available
bandwidth of the transmission channel in a particularly
efficient manner and hence are frequently used for the
transmission of synchronous data signals (data signals
whose elements occur according to a clock frequency) via a
telephone channel. ~lring transmission via a telephone
channel or via other channels havlng comparable properties
2~ the channel signal is frequently subjected to disturbing
frequency translations the static and dynamic components
of which are generally referred to as frequency offset and
phase ~lltter, respectively.
The coherent demodulation in the receiver requires
a referonce carrier having a correct phase relation to the
-?- ~
-
PIIN.7730
1069588 3-6-1~75
carrier associated with the transmitted channel signal.
In vestigial-sideband systems, this reference carrier
cannot be recovered by e~tracting the carrier from the
channel signal, because the phase of the extracted carrier
depends on the data signals owing to the presence of a
quadrattlre component at the carrier frequency.
o~ co)ni~g
B A known method of ~4~m-t=~ this difficulty
consists in suppressing the very low data signal frequencies
in the transmitter, so that there is no data signal energy
in a narrow band adjacent to the carrier frequency, and
in transmitting a pilot signal at the carrier frequency.
The disadvantage of this method is that the suppression of
the very low data signal frequencies results in a great deal
of intersymbol interference, so that in the receiver this
su~pression must be cancelled by means of quantized feedback.
Another Icnown method, which avoids the latter dis-
- advantage, consists in transmitting two pilot signals having
suitably chosen frequencies outside the data signal band
and in recovering the reference carrier from the selected
pilot signals. However, this method not only requires
additional bandwidth and power, but also has the disadvan-
tage that the pilot signals lie near the edges of the
available transmission band where the phase distortion due
to the frequency characteristics of the transmission channel
is most serious.
It is an ob~ect of the present invention to provide
a vestigial-sideband transmission system of the type
described in the preamble in which even in the case of
disturbing frequency translations in the transmission
channel the reference carrier and the reference clock
signal can slmply be recovered, both having the correct
frequency and the correct phase, ~rom the transmitted
.
--3~
PI-IN.7730
1~69588 3-6-1975
channel signal itself.
The vestigial-sideband transmission system accor-
ding to the invention is characterized in that the filtering
and modulating circuit in the transmitter is arranged for
generating a vestigial-sideband channel signal which both
at the location of its carrier frequency and at the location
of a frequency which in the complete sideband is spaced
from the carrier frequency by a distance equal to one half
of the frequency of the clock signal is a double-sideband
modulated signal within a frequency band whose width is
smaller by an order of magnitude than the frequency of the
clock signal.
Embodiments of the invention will now be described,
by way of example, with reference to the accompanying dia-
grammatic drawings, in which:
Fig. 1 shows a transmission system according to
the invention having a transmitter for generating a vestigial-
sideband channel signal according to the filter method,
F-ig. 2 shows frequency diagrams illustrating the
operation of the system shown in Fig. 1,
Figs. 3-and 4 show circuits which in the system
shown in Fig. 1 can be used for recovering the reference
carrier and the reference clock signal, respectively,
Fig. 5 shows frequency diagrams illustrating the
generation of a vestigial-sideband channel signal according
to the phase method,
Fig. 6 shows a modified embodiment of the trans-
mitter of Fig. 1 in which the phase method illustrated
with reference to Fig. 5 is used,
~ig. 7 shows a modified embodiment of the trans-
mitters shown in Figs. 1 ancl 6,
Fig. 8 shows frequency diagrams illustrating the
PHN.7730
~69588 3~6-1975
operation of the transmitter shown in Fig. 7, and
Fig. 9 shows a frequency diagram illustrating
another mode of operation of the system shown in ~ig. 1.
Fig. 1 shows a system in which syn~hronolls binary
data signals at a data rate of 2,400 bit/second are trans-
mitted from a data signal source 1 in a transmitter 2 to a
data signal sink 3 in a receiver 4 vla a transmission channel
5 of restricted bandwidth. This transmission channel 5, may
for example, be a telephone channel and may comprise a
plurality of telephone transmission links in tandem, such
as subscribersl lines, systems for carrier commlmication -
via cables or radio, and one or more telephone exchanges
with the associated switching equipment. The telephone
channel 5 has a transmission band from 300 to 3,300 Hz of
which only the central portion from 600 to 2,700 Hz is used
for data transmission. This data transmission is effected
by means of vestigial-sideband amplitude modulation of a
carrier having a frequency of 2,100 Hz, and the transmission
speed is 2,400 Baud.
The transmitter 2 includes a clock signal source 6
.
~or synchronizing the data signal source 1 BO that the ele-
ments of the binary data signal occur in accordance with a
clock frequency of 2,400 Hz. This synchronous binary data
signal is applied to a filtering and modulating circllit 7
to which is also applied a carrier having a frequency of
2,100 Hz and originating from a carrier source 8 for genera-
ting a vestigial-sideband amplitude-modulated channel signal
which is transmitted via the telephone channel 5 to the
receiver 4.
In the receiver 4~ the transmitted channel signal
is s~lpplied via a selecting filter 9 and an equalizer 10
to a demodulator 11. The receiver further includes a circuit
.. . . . ... . . ............ -- .. .. ~ .. , .- . . .. . .
PHN.77~0
~069588 3-~-1975
12 which is collpled to the telepllone chann~l 5 and ser~es
to recover a reference carrier of` Z,100 llz which is applied
to the demodulator 11 for coherent demodul~tion o~ the
transmitted channel signal. To the OUtpllt of the demodlllator
11 is connected a low-pass f`ilter 1~ for separating the ~`
desired demodulated signal from which the original synchro-
nous binary data signal is obtained by means of a regenera-
tor 1L~. For this purpose the receiver 4 comprises a circuit
15 which is coupled to the telephone channel 5 and serves
to recover a reference clock signal of 2,400 Hz which is
applied to the regenerator 14. The regenerated data signal
is transferred to the data signal sink 3 for further pro-
cessing. The circuits 12 and 15 for recovering the reference
carrier and the reference clock signal respectively may be
implemented in various known manners; further particulars
are not given here but can be found, for example, in W.R.
Bennett and J.R.~avey, "~ata Transmission", Néw York,
McGraw-Hill, 1965.
Several methods of generating a vestigial-sideband
ampiitude-modulated signal are known. In the transmitter
of Fig. 1 a highly usual method is employed which consists
in first generating a double-sideband amplitude-modulated
signal with suppressed carrier and then removing the un-
desired sideband in a filter having a suitably chosen
transfer function. For this purpose the filtering and
modulating circuit 7 of Fig. 1 include.s a premodulating
filter 16 which is connected to the data signal source 1
and is in the form of a lowpass filter having a cutoff
frequency equal to about one half` of the clock frequency
(1,200 Hz) a double-balanced amplitude modulator 17 (pro-
duct modulator) which modulates the carrier from the carrier
source 8 with the output signal of the premodulating filter
.
--6--
PJIN.7730
1069588 3_~_1975
16, and a postmodulating filter 18 in the form o~ a lowpass
filter having a cutoff frequency equal to the carrier fre-
quency (2,100 Hz). This postmodulation filter 18 removes
the upper sideband from the double-sideband signal generated
in the amplitude modulator 17 and supplies the desired
vestlgial-sideband channel signal to the telephone channel 5.
The overall transfer characteristic of the trans-
mission system of Fig. 1, inclusive of the filters 16, 18, 9,
13, the equalizer 10 and the telephone channel 5, must comply
with the first Nyquist criterion so that no intersymbol inter-
ference occurs at the nominal regeneration instants. Fre-
quently the filters in the receiver are designed so as to
provide optimum noise suppression and the filters in the
transmitter are designed so as to give, when used with the
0~4~//
B 15 said filters in the receiver, the desired owe~,-t transfer
characteristic. For simplicity, however, it is assumed
for the present that the vestigial-sideband channel signa
at the OUtpIlt of the transmitter 2 already satisfied the
first Nyquist criterion.
According to the invention a vestigia~-sideband
transmission system is realized in which both the reference
carrier and the reference clock signal can simpl~^ be re-
covered, both having the correct frequency and the correct
phase, from the transmitted channel signal itself in that
the filtering and modulating circuit 7 in the transmitter
2 is arranged for generating a vestigial-sideband channel
signal which both at the location of its carrier frequency
and at the location of a frequency which in the complete
sideband is spaced from this carrier frequency by a distance
euqal to one half of the frequency of the clock signal is
a double-sideband modulated signal within a frequency band
whose width is smaller by an order of magnitude than the
--7--
PIfN. 7730
3-G-1975
1~6~588
frequency of the clock signal.
In Fig. 2, frequency diagram a shows by way of
example a spectrum C(f) of the resulting vestigial-sideband
channel signal at the OUtpllt of the transmitter 2 of` Fig. 1.
Both at the carrier frequency of 2,100 Hz and at the fre-
quency of 900 Hz which in the complete lower sideband is
spaced from this carrier frequency by one half of the clock
frequency (1,200 Hz) the said spectrum C(f) has a flat
portion within a frequency band of width, for example,
120 Hz. Furthermore, the spectrum C(f) in the frequency
range from 1,500 to 2,700 Hz has vestigial symmetry with
respect to the value C(2,100) at the carrier frequency of
2,100 Hz and also in the frequency range from 600 to 1,200
Hz it has vestigial symmetry with respect to the value
C(900) at the frequency of 900 Hz. Thus the requirements of
the flrst Nyquist criterion are satisfied, as will also be
seen from the spectrum B(f) of the baseband signal which
is obtained by coherent demodulation of this vestigial-
sideband channel signal by means of a carrier of 2,100 Hz
having the correct phase, which spectrum B(f) is shown in
frequency diagram b of Fig. 2. Owing to the symmetry of
C(f) in the range of 1,500 to 2,700 Hz, in the coherent
demodulation the partial suppression of the lower sideband
in the range of 1,500 to 2,100 Hz is exactly compensated
for by the partial transmission of the corresponding part
of the upper sideband in the range of 2,100 to 2,700 Hz, so
that B(f) in the range of O to 600 Hz is flat (the dotted
lines in frequency diagram b represent the contributions of
the lower and upper sideband), and the symmetry of C(f) in
the range of 600 to 1,200 Hz is found again in the range
of 900 to 1,500 Hz, so that B(f) has a vestigial symmetry
with respect to the value B(1,200) at the frequency of
. PHN.7730
~0~9588 3-6-1975
1,200 Hz, which is exactly equal to one half of the clock
frequency of the synchronous data signals.
To obtain the vestigial-sideband channel signal
having the said spectrum C(f) in the filtering and modulating
circuit 7 of ~ig. 1, the premodulatlng filter 16 is given
a transfer function such that the spectrum of its output
signal corresponds to the spectrum B(f) in frequency
diagram b of Fig. 2. If the elements of the binary signal
consist of rectanglllar pulses having a duration T equal
to the period of the clock frequency of 2,400 Hz, the ampli-
tude characteristic H1(f) of this premodulating filter 16
is made equal to B(f)/S(f), where S(f) = sin( ~ fT)~ ~ fT
is the spectrum of a rectangular pulse of duration T. In
the amplitude modulator 17 a double-sideband signal is
pro.duced having a spectrum M(f) as shown in frequency
diagram c of Fig. 2. From this spectrum M(f) the desired
spectrum C(f) is obtained by giving to the post-modulating
filter 18 an amplitude characteristic H2(f) of the shape
shown in frequency diagram d of Fig. 2, where H2(f) in -.
the range of 1,500 to 2,700 Hz has the same shape as C(f).
In the above discussion of a system.in which the requirements
of the first Nyquist criterion are already satisfied iD
the transmitter 2, it is tacitly assumed that the phase
characteristics of the premodulating filter 16 and the post-
m,odulating filter 18 are linear in the entire frequency
range of interest. This fact must be taken into account
in the practical design of these filters 16 and 18. Possible
deviations from the desired linear ph~se characteristics
of these filters 16 and 18 in thetransmitter 2 can in prac-
tice also be corrected in the receiver 4 by means of the
equalizer 10.
It will now be shown that the resulting vestigial-
_g_
; " ... . ..
,
PHN.7730
l~9S~8 3-6-1975
sideband channel signal really is a double-sideband modula-
ted signal within a freqllency band of width 120 Hz at the
carrier frequency of 2,100 IIz and also at a frequency which
is lower by 1,200 IIz and hence is 900 Hz.
For this purpose, in respect of the band at the
carrier frequency of 2,100 lIz a component of the data ~ignal
having a frequency f of less than 60 Hz will be considered.
In the Olltpllt signal of the amplitude modulator 17 this
component gives two sideband components at the frequencies
(2,100 - f) and (2,100 + f), which sideband components have
equal amplitudes and equal but opposite phase shifts
relative to the carrier of 2,100 Hz. Because the post-
modulating filter 18 has a linear phase characteristic and,
moreover, within a band of wldth 120 Hæ at the frequency
f 2,100 Mz has a flat amplitude characteristic (compare
H2(f) in Fig. 2), the sideband components at the frequencies
(2,100 - f) and (2,100 + f) in the vestigial-sideband channel
~ignzl also have equal amplitudes and equal but opposite
phase shifts relative to the carrier of 2,100 Hz. Conse-
quently within a band of width 120 Hz at the carrier fre-
quency of 2,100 Hz the vestigial-sideband channel signa
really is a double-sideband modulated signal.
In contrast therewith, for the band at the frequency
of 900 Hæ a component of the data signal will be considered
which has a frequency (1200 - f), where f again is. less
than 60 Hz. It is known that in the spectrum of a synchro-
nous data signal having a clock frequency of 2,400 Hz a
component at a frequency f~ will never occur by itself, but
always will be accompanied by components at frequencies
(2~100 - f~)~ (2,~00 + f~), (4,800 - f~), (li,800 + fl), etc.
The amplitudes and phases of these simultaneously occurring
components depend upon the pulse shape which is used for
-10-
. .
pHN.7730
10695~8 3-6-1975
the data signal ele~ents, whilst in general the phases are
either eclual or differ by 1~0, however, in the range of
O to 2,400 Hz they invariably are equal~ In the present
case this means that the component considered which has a
frequency (1,200 - fl) is always accompanied by a component
having a frequency (1,200 + f) and the same phase. Because
the premodulating filter 16 has a linear phase characteris-
tic and moreover within a band of width 120 Hz at the
frequency of 1,200 Hz has an amplitllde characteristic such
that the OUtpllt spectrum at this location is flat (compare
B(f) in Fig. 2), the components which always occur as a pair
at the frequencies (1,200 - f) and (1,200 + f) have equal
amplitudes and equal but opposite phase shifts relative to
an imaginary carrier of 1,200 Hz. Within a band of width
120 Hz at the frequency of 1,200 Hz the baseband signal at
the OUtpllt of the premodulating filter 16 hence really is
a double-sideband modulated signal. The same applies to the
vestigial-sideband channel signal within a band of width
120 Hz at the frequency of 900 Hz, because the amplitude
modulator 17 transposes the band at the frequency of 1,200
Hz in the baseband signal only with frequency-inversion to a
band at the frequency of 900 Hz and furthermore the post-
modulating filter at this location has both a linear phase
characteristic and a flat amplitllde characteristic ~compare
H2~f~ in Fig- 2)-
Owing to the fact that the vestigial-sideband
channel signal at the carrier frequency of 2,100 Hz, and
at the frequency whlch is lower by 1,200 Hz (one half of
the clock frequency) and hence is 900 Hz, locally is a
double-sideband modulated signal, the reference carrler for
coherent demodlllation and the refercnce clock signal for
regeneration can be recovered, both having the correct
PIIN.7730
1069588 3-6-1975
frequency and the correct phase, from the transmitted
channel signal itself` by means oI` the known comparativel~
simple methods of recovering the re~erence carrier from a
double-sideband signal.
Fig. 3 shows by way of example an embodiment of a
very simple circ~lit 12 which may be used in the vestigial-
sideband transmission system of Fig. 1 for recovering the
reference carrier of 2,100 Hz. This circuit 12 includes a
bandpass filter 19 tllned to the carrier frequency of 2,100
- 10 Hz in order to select the do~Ible-sideband portion of the
channel signal at this frequency. BecalIse the dollble side-
band .signal contains no quadratllre component at the carrier
frequency the Olltput signal a(t) of the bandpass filter 1
can be represented by
a(t) = x(t) cos ( U~ ct + e)
where x(t) is representative of the components of the data
signal having frequencies lower than 60 Hz, ~ c = 2 . 2,100
is the carrier radian frequency and e is the carrier phase.
The signal a(t) is sq~lared in a squaring circuit 20 the Ollt-
put signal b(t) of which can be written
b(t) = (1/Z) ~ x2(t) + x2(t) cos (2 ~ ct + 2e)~
The low-frequency part of this signal b(t) is eliminated
by means of a highpass filter 21 and the res~llting signal
is ideally limited in a limiter 22 to obtain an Otltput
signal c(t) of the form
c(t) = A cos (2 W ct + 2e)
where A is a constant. This signal c(t) at twice the carrier
frequency is applied to a freqllency divider 23 to obtain
the reference carrier having the correct frequency of
2,100 Hz and, apart from a phase ambiglity of 180, the
correct phase. The problems to which this ambiguity gives
rise in the coherent demodnlation can be avoided in known
.
-12-
Pll~.7730
~0~9588 3-6-1975
manner, for example by using differential coding in the data
signal source 1 of the transmitter 2.
In the practical embodiment of the circuit 12 of
Fig. 3 the ~quaring circuit 20 generally takes the form of
a full-wave rectiiier to which is connected a narrow band-
pass filter tuned to twice the carrier frequency instead
of the highpass filter 21. In many applications the said
narrow bandpass filter is a phase-locked oscillator which
has an input circult for suppressing amplitude variations;
in this case the frequency divider 23 is connected to the
oscillator without the interposition of the limiter 22. If
the transmitted channel signal suffers from disturbing
frequency offset and phase jitter, the use of a phase-locked
oscillator provides the advantage that in spite of the said
disttlrbing frequency translations the reference carrier
always has the correct frequency and substantially the
correct phase. In order to reduce the inflllence of the dis-
turbing phase ~jitter, which may be considered as an inciden-
tal low-index frequency modulation of all the signal com-
ponents, on the demodulated baseband signal to a small value,
in the implementation of the circuit 12 for recovering the
reference carrier care mllst be taken to ensure that the
linear phase shift which this circuit introduces into the
sideband components of the phase ~jitter is kept as small as
possible.
To recover the reference clock signal a circuit may
be used similar to that shown in Fig. 3 (without the frequen-
cy divider 23) to recover from the double-sideband portion
Or the channel signal at the frequency of 900 Hz a reference
signal at twice this frequency (i.e. at 1,800 Hz), to mix
this reference signal with the signal at twice the carrier
frequency (i.e. at 4,200 ~z) at the Olltput of the limiter 22
-13-
PIIN.7730
1069S88 3-6~1975
in Fig. 3, and finally to select from the mixing products
the component at the difference frequency which corresponds
to the reference clock signal of 2,400 Hz in the correct
phase.
The reference clock signal may also be recovered
from the baseband signal which is obtained by coherent de-
modulation of the transmitted channel signal. In the system
of Fig. 1 this possibility is used, and Fig. 4 shows an
example of a very simple circuit 15 which may be used for
this purpose. In respect of its structllre and operation the
said circllit 15 differs from the circuit 12 of Fig. 3 only
in that the frequency divider 23 is absent in Fig. 4. In
particular, the circuit 15 incllldes a bandpass filter 24
tuned to the frequency of 1,200 Hz for selecting the double-
sideband portion of the demodulated baseband signal at one
half of the clock frequency (compare B(f) in ~ig. 2). This
selected double-sideband portion is squared in a squaring
circuit 25, the low-frequency part of the squared signal is
eliminated by means of a highpass filter 26, whereupon the
reference clock signal having the correct frequency of 2,400
Hz and the correct phase is obtained by means of an ideal
limiter 27. With regard to the practical implementation of
the circuit 15 of Fig. 4 the same considerations apply as to
the circuit 1Z of Fig. 3, however, becallse of the coherent
demodulation substantially no allowance need be made for
the disturbing frequency translations.
Consequently~ for recovering the reference carrier
and the reference clock signal no pilot signals need be used.
However, dllring the data transmission circllmstances may occur
in which temporarily there is little energy in the vestigial-
sldeband channel signal within the frequency bands of width
120 IIz at the frequencies of 900 Hz and 2,100 H~, so that the
-14-
PIIN.7730
~; 1069588 3-G- 1 975
recovered reference carrier and reference clock signal may
temporarily show small flnctlIations- To ensure that under
all circnmstances snfficient energy is available for re-
covery withont appreciable flllctllation, in Fig. 1 two pilot
signals of comparatively low level having Jrequencies of
900 ~Iz and 2,100 Hz respectively are added with correct
phase to the channel signal. The leve~ of the pilot signals
relative to the channel signal is, for example, - 12 dB.
~or this purpose in Fig. 1 a pilot signal of 1,200 Hz is
derived from the clock signal sonrce 6 by means of a frequen-
cy divider 28 and an attemlator 29, which pilot signal is
combined with the baseband signal at the OIItplIt of the pre-
modulating filter 16 in a combining circuit 30. ~Irthermore~
by means of an attemIator 31 a pilot signal of 2,100 Hz is
derived from the carrier source and combined with the channel
signal at the OUtplIt of the f`iltering and modulating circnit
7 in a combining circuit 32. This pilot signal at the frequen-
cy of 2,100 ~z may also be obtained by applying a direct-
voltage signal of sllitable valne to the combining circnit
30. Another known possibility to ensnre that sufficient
energy is always available within the frequency bands from
which the reference carrier and the reference clock signal
are recovered, consists in the nse of a method of data
scram~ling in the data signal sonrce 1 of the transmitter 2,
which method is recommended by the CCITT for some types of
data transmission. In the data signal sink 3 of the receiver
4 the corresponding data descrambling m~Ist then be employed.
Thus by the nse of the steps according to the
invention a vestigial-sideband transmission system is ob-
tained in which in a very simple manner the reference carrier
and the reference clock signal can be recovered at the cor-
rect freqnency and with the correct phase from the vestigial-
-15-
PllN.7730
~06~588 3-6-1975
sideband channel signal itself, whilst the disadvantages
of the known more complicated methods of recovering the
reference carrier and the reference clock signal are
avoided and fllrthermore the margin for impairment of the
transmis~ion channel, stlch as disturbing frequency trans-
lations, is increased.
The premodnlating filter 16 and the postmodulating
filter 18 nsed in the transmitter 2 shown in Fig. 1 may be
realized in analog techniq1les, however, the preference for
a linear phase characteristic in particular greatly compli-
cates their design and implementation. Hence it is far more
attractive to realize the premodulating filter 18 as a binary
transversal filter as described in United States Patent
Specification 3,500,215 and to realize the post-modulating
filter 18 as an analog code filter as descrlbed in United
States Patent Specification 3,521,170, because the desired
amplitude characteristic and the linear phase characteristic
can then be obtained very simple and with a high amount of
m~ltual freedom, whilst fllrthermore these types of filter are
highly suitable for monolithic integration. Fllrther parti-
culars abollt design and implementation of the two types of
filters can be folmd not only in the abovementioned Patent
Specifications but also in P.~euthold "Filternetzwerke mit
digitalen Schieberegistern", Philips Res. Repts., Suppl.
No. 5, 1967, and H.B. Voelcker "Generation of digital sig-
naling waveforms", I.E.E.E. Transactions on Cnmmunication
Technology, vol. COM-16, pages 81-93, February 1968.
In the transmission system of Fig. 1 the vestigial-
sideband channel signal is generated according toa method
f modulation which in single-sideband technology is general-
ly referred to as filter method. However, the modulation
methods Icnown in this technology as phase method and ~eaver
-16-
10~9 588 PIIN 7730
Method can also be used for genel-ating the desired vestigial-
sideband channel signal.
~or the phase method this will be described with
reference to Fig. 5. Frequency diagram a of Fig. 5 again
shows the spectrllm C(f) oi` the vestigial-sideband channel
signal (compare frequency diagram a in Fig. 2). This spectrum
C(f) can be regarded as the sum of a part Ce(f) of even
symmetry with respect to the carrier frequency of 2,100 Hz,
as shown in frequency diagram _, and of a part Co(f) of odd
symmetry with respect to this carrier frequency of 2,100 Hz
as shown in frequency diagram c. These spectra Ce(f) and
Co(f) are obtainable by the phase method. Ce(f) represents
the spectr~lm at the OUtpllt of a product modulator which is
fed with a carrier of 2,100 Hz and a baseband signal having
a spectrum Be(f) = Ce(f + 2,100), whilst Co(f) represents
the spectrum at the OUtpllt of a product modulator which is
fed with a 2,100 Hz carrier shifted 90 in phase and a base-
band signal having a spectrum Bo(f~ = -Co(f + 2,100) like-
wise shifted 90 in phase.
Fig. 6 shows a simple modification of the trans-
mitter 2 shown in Fig. 1 in which the vestigial-sideband
channel signal is produced by the abovedescribed phase
method and in which further the aforementioned binary trans-
versal filters are used as premodulating filters for ob-
taining the desired baseband signals.
In the filtering and modulating circuit 7 of Fig. 6
the synchronous binary data signal from data source 1 is
sllpplied to a shift register 33 the contents of which are
shifted at a shift freqllency equal to an integral multiple
of the clock frequency of 2,400 Hz, which shift freqnency
is obtained by means of a freqllency multiplier 34 connected
to the clock signal source 6, The elements of this shift
-17-
~069 588 3-6-1~75
register 33 are connected to a f`irst summing circuit 36 via
a first set 35 of weighting networl~s and to a second summing
circuit 38 via a second set 37 of weighting networks. In
the manner extensively described in the aforementioned
publications the weighting factors of the weighting net-
works of the first set 35 are dimensioned so that ~t the
OUtpllt of the first summing circuit 36 a baseband signal
having a spectrum B (f) is produced. Similarly the weighting
factors of the weighting networks in the second set 37 are
so dimensioned that at the Otltput of the second summing
circuit 38 a baseband signal having a spectrum Bo(f) shifted
90 in phase is produced. By using the symmetry properties
of Be(f) and Bo(f) with respect to the frdquency f = 0 no
additional steps need be taken to obtain the desired 90
phase shift, To the outputs of the summing circuits 36 and 38
simple first-order RC low-pass filters 39 and 40 respectively
are connected for suppressing the higher-order pass bands
which, as is known, are produced at the shift frequency of
the shift register 33 and at multiples thereof.
In amplitude modulators 41 and 42 which are realized
as product modulators the obtained baseband signals are
used to modlllate carriers of 2,100 Hz relatively shifted
in phase by 90 which are derived, directly and via a 90
phase shifting network 43 respectively, from the carrier
source 8. Thus at the Olltpllt of the modulator 41 a signal
having a spectrum Ce(f) and at the Otltput Or the modulator
~2 a signal having a spectrum C (f) is produced, and by
summing these output signals in a summing circuit 44 the
desired vestigial-sideband channel signal having a spectrum
C(f) is obtained.
In the transmitters shown in Fig. 1 and Fig. 6 the
spectrum of the synchronous data signal is first limited by
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10 69 588 Pl-IN 7730
means of a premodulating ~ ter and then supplied to an
analog product modulator. IIowever, the binary na-tllIe of the
data signal can be used to reverse the order of premodlllation
I`iltering and modulating and the analog modulator can be
replaced by a simple logic circuit to which the binary data
signal and a rectangular carrier to be considered as a
binary signal are applied. Thus in the transmitter a post-
modulating filter only is required. A transmitter of such a
structllre is described in United States Patent Specification
3,611,143 and in P.J. van Gerwen and P. van der Wurf, "~ata
modems with integrated digital filters and modulators",
I.E.E.E. Transactions on Commlmication Technology, Vol.
COM-18, No. 3, pages 214-222, June 1970. However, it is also
shown in these two publications that the aforementioned
reversal is meaningful only if the carrier frequency is an
integral multiple of one half of the cloclc frequency, for
in this case only the distortion due to lower sidebands of
carriers harmonics and to foldover of lower sidebands of
both carrier and carrier harmonics about zero frequency can
be regarded, as linear distortion, which then can be cor-
rected by a linear network. The postmodulating filter can be
designed so that the linear correction also is effected in
it. In this case the postmodulating filter may also take
the form of a simple binary transversal filter so that the
transmitter as a whole is particularly suited for manufac-
ture in monolithic integrated-circuit form.
Althoughin the present case the carrier frequency
of 2,100 Hz is not an integral multiple of one half of the
clock frequency (1,200 Hz), the abovedescribed modulation
technique may still be used to obtain a simple transmitter
constr~lctlon, as will be described more fully with reference
to the modification of the transmitters of Fig. 1 and Fig. 6
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I'I~N.7730
1069588 3-~-1975
which is shown in Fig. 7.
In th~ filtering and modulating circuit 7 of Fig. 7
the synchronous binary ~ata signal from the data source 1
and a rectangular carrier having a frequency of 4,800 Hz
are supplied to a logic circuit 45 in the form of an ex-
clusive-OR gate. This carrier at twice theclock frequency
of 2,400 Hz is derived from the clock signal source 6 by
means of a frequency multiplier 46. The exclusive-OR gate
45 forms the modulo-2 sum of the data signal and the carrier,
which operation is equivalent to amplitude modulation with
carrier suppression. The binary output signal of the OR-
gate 45 is applied to a shift register 47 whose contents
are shifted at a shift frequency which is an integral mul-
tiple of the carrier frequency of 4,800 Hz and which is also
derived from the clock signal source 6 by means of a frequen-
oy multiplier 48. The elements of this shift register 47 are
connected to a summing circuit 50 via a set 49 of weighting
networks.
In the manner extensively described in the last-
mentioned publications the weighting factors of the set 49
of weighting networks are chosen so that the lower sideband
of the 4,800 Hz carrier is largely suppressed and at the
same time the linear modulation distortion is corrected.
In particular, these weighting factors are dimensioned so
that at the OUtpllt of the summing circuit 50 a vestigial-
sideband signal is produced which has a speotrum ~(f) as
shown in frequency diagram a of Fig. 8. This spectrum ~(f)
satisfies the relation ~(f) = C(6,900 - f), where C(f) is
the spectrum of the desired vestigial-sideband channel
signal (compare frequency diagram a of Fig. 2). Here also
a simple first-order RC lowpass filter 51 is connected to
the OUtpllt of the summing circuit 50 to suppress higher-
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PHN.7730
~069588 ~-G- 1975
order pass bands at the shift freqllency and at mtlltiples
thereo r .
In an analog product modulator 52 the vestigial-
sideband signal having a spectrum ~(f) modulates a carrier
oI' 6,900 IIz derived from a carrier source 53. The output
signal from modulator 52 then has a spectrum N(f) as shown
in frequency diagram b of Fig. 8. As will also be seen from
Fig. 8, the lower sideband exactly corresponds to the desired
vestigial-sideband channel signal having a spectrum C(f~
and the upper sideband is spaced from the lower side band by
a distance such that the upper sideband can be eliminated
by means of a lowpass filter 54 having an amplitude charac-
teristic A~f), for example, of the shape shown by a broken
line in frequency diagram b.
Thus, in general, in the case of carrier frequencies
o~ the vestigial-sideband channel signal not equal to an
integral multiple of one half of the clock frequency, for
the first modulation step a carrier frequency is chosen
which is equal to an integral multiple of one half of the
clock frequency, whilst furthermore this multiple and the
carrier frequency for the second modulation step are chosen
so that one sideband of the resul~ing signal exactly corres-
ponds to the vestigial-sideband channel signal having the
desired carrier frequency and the other side-band is spaced
therefrom by a distance large enough to enable it to be
eliminated by means of a simple filter.
The filtering and modulating circuit 7 in the trans-
mitter may also be entirely realized in digital techniques.
For this purpose in the transmitter each element of the
synchronous binary data signal from data source 1 is sampled
once to ascertain whether this element represents a binary
value '~ or a binary value "0". These data signal samples
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PI-IN.7730
3 069588 3-6-1975
constitute the digi-tal inpu-t signal for a digital filtering
and modulating circuit 7 and are processed therein in the
form of code words which represent numbers. The code words
at the output of the digital circuit 7 are converted in a
dlgital-to-analog converter lnto the corresponding amplitude
values of a current or voltage, and the desired vestigial-
sideband channel signal is derived from the resulting quan-
tized signal by a low-pass filter.
In such a digital data transmitter, when carrying
out the operations, the main line of the modulation schedule
used in Fig. 7 can be followed. However, it is not necessary
actually to perform the operation which corresponds to the
first modulation step in order to obtain the code words
which represent the samples of the vestigial-sideband signal
at the first carrier frequency of 4,800 Hz, for, as has
been mentioned hereinbefore, in the spectrum of the data
signal having a clock frequency of 2,400 Hz a component at
a frequency f~ does not occur isolated but is always
accompanied with components having frequencies (2,400 - f ~ 7
(2,400 + f~), (4,800 - f~), (4,800 + f~), etc., the ampli-
tudes and phases of these simultaneously occurring components
being dependent on the pulse shape of the data signal ele-
ment. In the digital data transmitter the pulse shape of
the digital input signal (the data signal s~mples) is that
f a ~irac pulse the spectrum of which, as is known, is
flat throughout the entire frequency range. As a result,
in the spectrum of the digital input signal the simultaneous-
ly occurring components of frequencies f~, (2,400 - f~),
(2,400 + f~), (4,800 - f~), (4,800 + f~), etc. all have the
same amplitude and phase. Hence the vestigial-sideband
signal at the frequency ~f 4,800 Hz can directly be derived
from the digital input signal by means of a bandpass filter.
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,. .... . , . ., ,., .. ,. , , .... . . . -; .
10~9588 PHN 7730
In our co~pending Canadian patent application 234,814, -
which was filed on Septenber 4, 1975, is described how such a
digital data transmitter can be realized with a minim~m of
technical means.
In the explanation given hitherto the data rate
(2,400 bit/sec) is equal to the transmission speed (2,400 Bal~).
me vestigial-sideband transmission system acoording to the ~ ~`
invention obviously is not restricted thereto. Fbr exa~,ple,
data signals at a data rate of 4,800 bit/sec also may be trans-
mitted at a transmission speed of 2,400 Baud. For this purpose,
in the present system the elements of the data signal are divided
in groups each oomprisLng thO elements. In Fig. 1, these groups
can be subjected to a four-level encoding such that the data sig- --
nal elements again occur in acoordance with the 2,400 Hz clock
frequency but have four levels (for example +3, +1, -1, -3)
instead of tWD levels. ~he regenerator in the receiver then is
adapted to this four-level encoding. In the embodirlnts of
Fig. 6 and Fig. 7 the same result is obtainable by converting the --
said groups by means of a series-p æ allel converter into groups
which each ccmprise t~o simLlt~neously occurring elements. Ihus
tw~ parallel data signals at a clock frequency of 2,400 Hz are
obtained which each can be individually processed in the m~nner
shcwn in the said Figures and then can be oo~bined, using dif-
ferent weighting factors (in a ratio of 2 : 1), to form the
ultimate vestigial-sideband channel signal. Finally, in the
digital data transmitter the dIbit encoding which corresponds
to the fcur-level enccding can be applied to these gmups.
FurthermDre it has so fæ been assumed that the
re~uirements of t;he first Nyquist criterion are already
satisfied at the output of the tran~mitter. However, these
requirements~may alternatively be jointly satisfied by the
B
`
l'IIN.7730
1069588 3-G-1975
transmitter al1d the receiver and, may, f`or example, be
evenly divided between the transmitter and the receiver.
For this purpose, for example, the transmitter 2 of Fig. 1
is arranged so that the vestigial-sidebancl channel signal
5 at the OUtpllt now has a spectrum C 9 (f) as shown in the
frequency diagram of Fig. 9. This spectrum satisfies the
relation C9(f) = ~ ), where C(f) is the spectrum shown
in frequency diagram a of Fig. 2. Imparting, for example,
an amplitude characteristic H3(f) = Cl(f~ to the selecting
filter 9 in the receiver 4 of Fig. 1 ensures that the
vestigial-sideband channel signal at the input of the
demodulator 11 again has a spectrum C(f). Although such an
even distribtltion between transmitter and receiver has ad-
vantages in respect of noise suppression, the approach
whlch is descrlbed hereinbefore and in which the receiver
is not involved is to be preferred in practice, because the
design and implementation of a selecting filter having the
required amplitude characteristic H3(f) and at the same
time a linear phase characteristic will be parti~ularly
complex.
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