Note: Descriptions are shown in the official language in which they were submitted.
10~7~15 1
This invention relates to an integrated circuit,
low level , low power, direct coupled class B amplifier. More
particularly, it relates to a class B amplifier of the kind
above indicated, in which the idle current is reduced to a
very low level while at the same time distortion is reduced to
a low level. The amplifier is particularly suitable for use
in hearing aids.
Class B amplifiers such as those used in hearing
aids usually have a three stage direct coupled transistor
preamplifier followed by transistor output stage. Distortion
had long been a problem in these amplifiers. A major reason
has been that the gain of a transistor increases with the
curre~t through it. Therefore, when an appropriate :Lnput
signal is received by -the amplifier, causing one of the ou-tput
transistors to conduct more heavily, the gain of the output
transistor increases. The increase in current and hence in
gain can be by as much as a factor of 500. Unfortunately,
the gain o the preceding transistor ~namely the third stage
preamplifier transistor) does not decrease as much as that of
the output transistor increases (because of differences in
the resistive loads connected to the various transistors in
a direct coupled amplifier), and the net effect is that the
overall gain of the amplifier increases with current. This
results in a distorted wave form at the output. The conventional
solution to this problem has been to bias the output transistors
at a relatively high idle current, so that the changes in
output transistor current andhence in gain with increased
signal are reduced. Typical prior art class B amplifiers, such
as those shown in Canadian Patents Nos. 811,844 and 844,156,
have an idle current of 200 microamperes for each of their two
.~, ~.
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output transistors. When used in hearing aid or similar
applications, in which a single very small battery cell
is used, it would be desirable to have a lesser idle current,
to increase the battery life.
A further difficulty with low level direct
coupled class B hearing aid amplifiers is that their load
is a hearing aid receiver, the impedance of which changes
with frequency. In general, the impedance of the hearing aid
receiver increases with frequency. The gain of the output
transistor in prior art class B amplifiers increases approx- r
imately in direct proportion to load impedance, so the
harmonics of the signal being amplified receive greater ampli-
fication than the fundamental. This produces Eurther dis-tortion.
~ccordingly, it is an object of the present in-
vention to provide an integrated circuit, low level, low
power, direct coupled class B ampli~ier in which the idle or
stand-by current is reduced, and in which distortion is also
reduced. To this end, the invention provides a two channel
amplifier in which the collector of each output transistor is
connected through a substantially purely resistive feedback
loop to the collector oE the corresponding first preampliEier
transistor. This provides ~C nega-tive feedbac]c which drastically
reduces the distortion, enabling operation at an idle current
-typically of 50 microamperés with less distortion than many
prior art class B amplifiers possess at idle currents of 500
microamperes. The resistive part of each AC feedback loop which
is connected to the ou-tput transistor collector is electrically
isolated from the battery voltage, -to permit the output transistor
collectors to swing more than .6 volts above the battery voltage.
In addition, common mode rejection is provided by connecting -the
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emitters of at least one pair (and preferably a~ least two
pairs) of the preamplifier transistors together and through
a resistor to ground.
Further objects and advantages will appear from
the following description, taken together with the accompanying
drawings, in which:
Fig. 1 is a schematic showing a first embodiment
of the invention;
Fig. 2 is a cross-sectional view showing the
electrical isolation of an AC feedback resistor from the
battery voltage;
Fig. 2A is a plan view showing a floating tub
resistor used in the Fig. 1 embodiment;
Fig, 3 is a schematic showing a second ernbodirnent
oE the invention; and
Fig. 4 is a schematic showing a third embodiment
of the invention.
Reference is first made to Fig. 1, which shows an
integrated circuit, low level, low power, direct coupled class
B amplifier ~ having a source 4 of input signal Vin. The
source 4 is typically a hearing aid magnetic microphone, which
is connected in double-ended fashion. Alternatively, the source
4 may be a microphone having a single-ended output, combined with
a phase splitter which transforms the single-ended microphone
signal into a double-ended signal.
The amplifier 2 includes a dual channel preamplifier
6, having identical channels 8, 10. Each channel amplifies a
single-ended half of the signal Vin. The signals from the
preamplifier 6 are further amplified by an output stage 12 and
are then combined in a transformer 14 and are fed to the load 16.
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The preamplifier channel 8 and one-half of the
output stage 12 will now be described in more detail. Since
the second preamplifier channel 10 and the second half of
the output stage 12 are identical to the first halves, the
second halves will be referred to only briefly and are
indicated ~y primed reference numerals.
Preamplifier channel 8 includes three stages,
constituted by transistors Ql' Q2' Q3 respectively, all shown
as NPN transistors. (There will always be an odd number of
lQ preamplifier stages.) One end of the input signal sourcle 4
is connected through capacitor C2 to the base of first stage
transistor Ql' while the emitter of transistor Ql is connected
via resistor R2 to ground. The collector of transistor Ql
i5 connected through resistors R3 and R~ to the positive battery
terminal Vb, and also to the base of second stage transistor
Q2. The collector of transistor Q2 is connected through
capacitor C3 and resistor R12 to the collector of transistor
Ql and is also connected through resistor R7 to the positive
battery Vb and in addition is connected to the base of third
2Q stage transistor Q3. The emitter of transistor Q2 is connected
to ground through a parallel circuit consisting of resistor R5,
and the collector-emitter circuit of a transistor Q5.
The emitter of the third stage transistor Q3 is
connected through resistor R6 to ground. The collector of
transistor Q3 is con~ected through resistors R8 and R10 to
the positive battery Vb, and is also connected to the base of
transistor Q4, which forms one half ofthe output stage. The
emitter of transistor Q4 is connected to ground, while the
collector of transistor Q4 is connected to one end of a
3Q centre tapped primary winding 18 of the transformer 14. (The
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centre tap is connected to the positive battery Vb.) The
secondary winding 20 of transformer 14 is connected to the
load 16.
As is conventional in a class E~ amplifier, when
signal Vin is applied differentially to the preamplifier
inputs (the bases of transistors Ql, Ql'), each channel 8,
10 effectively amplifies a single-ended signal with a 180
degree phase difference between each channel. The amplified
signals are then applied to the inputs of the output stage 12,
i.e. to the bases of transistors Q4, Q4'. As a result, t:he
output transistors Q4, Q4' are turn~d on and off in an alternate
fashion, i.e. one oE the output transistor Q4, Q4' conduc:-ts
~uring one half of the input signal cycle while the other
transistor is turned off, and vice versa during the other half
cycle of the input signal. The half wave signals which appear
at the output transistor Q4, Q4' collectors are then combined
differentially by the transformer 14, which reconstructs
complete sine waves across the load 16. An important
characteristic of the class B output stage is its ability to
deliver high signal power and yet idle at a very low quiescent
current, which is an i~lportant feature for hearing ai.d
applications.
In order to control the idle current of the output
stage transistors Q4, Q4', a DC negative féedback circuit is
conventionally provided to hold the base voltage of the output
transistors Q4, Q4l constant. The DC feedback circuit
consists of a resistive connection from the preamplifier
output back to the base of the first preamplifier transistor.
The resistive connection is from the junction of resistoxs R8,
3Q R10, through resistors R9 and Rl to the base of transistc)r ~1.
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AC signal is prevented from being fed back through this
connection by decoupling with a capacitor Cl, which short
circuits to ground any AC signal which would otherwise be
fed back to the base of transistor Ql through resistor Rl.
The operation of this DC feedback is conventional and is
fully explained in Canadian Patent No. 844,156 to which
reference may be made.
As previously indicated, a major disadvantage of
prior art class B amplifiers, such as those shown in Canadian
Patent 844,156, is the distortion due to large signal conditions
existing at the output. The reason for this is as follows.
When transistor Q4 turns on, the current passing through this
transistor increases from the DC idle current ~which rnay be
as lo~ as 50 microamperes) to a peak current which may be as
much as 25 milliamperes, i.e. a five hundred times increase in
current. Since the gain of transistor Q4 is directly
proportional to the current passing through it, the gain may
therefore also increase by a factor of as much as five hundred.
The load seen by transistor Q3 is the pa~allel
combination of transistor Q4 ancl resistors R8 and R10
(resistor R9 is very large and can to a first approximation
be ignored). The input irnpedance presented by transistor Q4 is
inversely proportional to the current -through transistor Q4 and
decreases by the same order of magnitude as the current
increases, namely by as much as a factor of 500. ~owever,
because of the effect of resistors R8 and R10, the load on
transistor Q3 will not decrease by a factor of 500, but will
instead decrease by a substantially lesser amount. Specifically,
if the impedance presented by transistor Q4 to transistor Q3
is RinQ4, then the load on transis-tor Q3 namely RL3, is:
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R 3 = - ~ 1
R8 + R10 R
inQ4
It will be seen that because of the effect of
resistors R8, R10~ the load on transistor Q3, namely ~ 3, is
not proportional to the input impedance RinQ4 of transistor
Q4. As the input impedance of transistor ~4 decreases (as
the current through transistor Q4 increases), the load ~n
transistor Q3 decreases by a lesser amount. Since the gain
of transistor Q3 is to a first approximation directly proportional
to the load on transistor Q3, thereEore the gain of transistor
Q3 does n~t d~crease as much a9 that of transistor ~ increas~s.
Th:Ls produces distortion, since large siynals are a~lpli~!ie~
more than small signals. The result is a distorted wave form
at load 16.
The distortion problem becomes more severe when
the load which consists of transformer 14 and resistor 16,
is replaced by a typical hearing aid output transducer (a
receiver). Because the gain of the output stage transistors
Q~, Q4' is approximately directly proportional to the load
impedance, and because receivers have an impedance which
increases with frequency, the harmonics which arise from
distortion are amplified more than the fundamental. The
distortion is therfore highlighted.
Typical prior art class B amplifiers have attempted
to solve these problems by employing increased idle currents
in their output stages (to reduce the factor by which the
current changes in output transistors). Idle current6 of
200 microamperes or more per output transistor are co~on. How-
ever,even withthese relatively large idle currents, c]ass B
8 --
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hearing aids ~ith greater than ten percent acoustic distortion
are common.
The invention employs a novel circuit which
permits reduced idle currents in the output transistor stage
(typically as low as 50 microamperes per output transistor) with
less distortion than most prior art class B amplifiers possess
at idle currents of 200 microamperes. In fact, a prototype
hearing aid class B amplifier fabricated according to the invention
possessed distortion of approximately two percent, which was
approximately that possessed by typical prior art hearing aid
class B amplifiers having idle currents of 500 microamperes.
An important feature of the invention is that it
provides a novel AC negative ~eedback circuit, in which the
collector o each output transistor Q~, Q4' is resistivcly coupled,
through resistors Rll and R3, and Rll' and R3', to the collector
circuit of the first stage transistors Ql, Ql' respectively.
It is found that this linearizes the gain of the amplifier,
reducing the distortion due to output dependent non-linearity.
A result of this feedback is that gain becomes more independent
of load impedance, reducing the distortion highlighting occuring
when a Erequency dependent load such as a receiver is usecl.
It is found that when the open loop gain oE the stages Q2,
Q3, Q4, multiplied by the ratio of the values of the resistors
Rll/R4, is much greater than one, then the gain of the combined
circuit (preamplifier and output stage) is much less dependent
on the output stage current. As previously indicated, it is
found that an idle current of about 50 microamperes per output
transistor will result in distortion of about two percent, which
has been found to be acceptable.
The AC feedback loop through resistor Rll is not
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connected to the base of transistor Ql because this would
interfere with the DC feedback loop previously described and
would also lower the input impedance of the transistor Ql stage.
A lowered input impedance for the transistor Q1 stage is
undesirable in hearing aid applications because it reduces gain
and makes tone control difficult. In addition, a connection
through resistor Rll to the first preamplifier stage would have
to be to the base of transistor Ql' (this is known as cross-
coupling)to provide negative feedback. If the transducer fails
to function as a good transformer, the AC feedback loop will be
poorly defined, resulting in unpredictable gain for the circuit
and possibly also in increased distortion.
In order to stabili~e the ampliEier wi-th the
two feedback circwits, compensation in the form of res:is-tor R12
and capacitor C3 is provided. These components introduce a
pole and a zero in locations which greatly improve the stability
of the amplifier.
The amplifier shown is fabricated as an integrated
circuit, typically (for the transistor types shown) on a P-type
substrate. In such circuits, integrated resistors are normally
fabricated as a diffused P-type layer residing in an epitaxial
N-type region or tub. The epitaxial region is normally biased
at the DC supply voltage to prevent the difEused layer to tub
diode from conducting. However, with the circuit of the present
invention, the voltage at the collectors of the output transistors
Q4, Q4' under some operating condictions may swing above and
below the battery voltage, due to the transformer action. When
the diffused layer potential is taken .6 volts above the DC
supply, the diffused layer to tub diode of resistors Rll, Rll'
will conduct and short the output. To prevent this, an
unconventional arrangement may be used in fabricating resistors
Rll and Rll'. The arrangement is shown in Fig. 2, in which the
-- 10 --
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P-type substrate on which the circuit is formed is indicated at
30 and is shown as being grounded (which is conventional), and
resistor Rll is shown as being abricated of P-type material
32 residing in an epitaxial N-type region or tub 34. The
contacts for resistor Rll are indicated at 36, 38. It will
be seen that no contacts are provided for the tub 34; instead
the tub 34 is left to float. In this manner, even when a contact
of resistor Rll is taken above or below the battery voltage,
the diode formed by P-type material 32 and N-type region 34 will
not conduct and short the output. The resistors Rll, Rll' may
be isolated in other ways, such as by use of thick or thin film
deposition techni~ues, or by use o discrete external res:istors.
It will be noted that althou~h resistors Rll, Rll'
clre isolated from the battery voltage, Erom the substrate 30,
and from each other (they are located in separate floating tubs
3~), the remaining resistors of the amplifier 2 may if desired
all be located in a single tub biased at the battery voltage.
This is indicated in Fig. 2, which shows a further tub 50 containing
a resistor 52 having contacts 54. Resistor 52 may be any of the
resistor of Fig. 1 except for resistors Rll, Rll'. All of the
resistors o Fig. 1 may be located in the same tub 50 except for
resistors Rll, Rll'. Tub 50 is biased at the battery voltage Vb
via contact 56 which is connected to the battery.
It is normally desirable to bias at the battery
voltage the tubs in which resistors are located, to supply
the reverse bias leakage current between the tub 50 and substrate
30, so that this current need not be supplied from the resistor
in the tub. If the leakage current were supplied from the tub
resistor, this could result in PNP transistor action; causing
excessive current to flow from the resistor to the substrate.
However, where the resistor voltage can rise more than .6 volts
above the battery voltage (as in the case of resistors Rll, Rll'~,
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this undesired transistor action becomes a certainty. There-
fore the tubs for resistors Rll, Rll' have been isolated
from the battery, and the current gain of any undesired
transistor action has been made as small as possible by
making the tub 34 as small as possible. As shown in Fig.
2A, the boundaries of the tub 34 are located in close prox-
imity to the material 32 which constitutes resistor Rll. In
other words, the area of the junction between tub 34 and
substrate 30 is made as small as possible.
A further feature of the amplifier of the inven-
tion is that it includes common mode rejection. Common mode
rejection is particularly important in a hearing aid, because
typical hearing aid batteries have an output impedance of
between two and fifteen ohms. This may cause a signal to
appear on the battery line which can be of magnitude comp
arable to the signal being ampliEied. The battery line signal
can easily find its way into the conventional signal path
via the preamplifier collector circuits, resulting in instab-
ility or distortion. Since the battery line signal is equal
or common to both channels, whereas the applied signal is 180
degrees out of phase in each channel, it is important that
the amplifier be able to distinguish between common and
difference signals, and to give the common signal less gain
than it would the difference signal. The common mode rejection
or CMR is measurable by the ratio of the difference gain to
the common gain.
Common mode rejection or CMR is provided in the
third preamplifier stage of the Fig. 1 circuit by connecting
the emitters of transistors Q3, Q3' together at node ~0 and
connecting node40 to ground through resistor R6. With this
arrangement, if identical positive-yoing signals appear at
the bases of transistors Q3, Q3', this will tencl to turn the
- 12 -
~(1177S~
transistors on harder, and will also tend to drive their
emitters positive. The net result is that there is little
or no change in the base-emitter voltage of transistors Q3,
Q3' (to a first order approxima~ion~ and therefore the common
mode signals receive little or no amplification. However,
if difference mode signals appear at the bases of transistors
Q3, Q3', for example if a positive-going signal appears at
the base of transistor Q3 and a corresponding negative-going
signal appears at the base of transistor Q3', then the net
change in potential at the node 40 is very small (since the
positive-going change contributed by the emitter of transistor
Q3 is cancelled by the negative-going change contributed by
the emitter of transistor Q3'). ~lence the change in ~,he base-
emi t ter of transistor Q3 is large, resulting :in ~mpli-
fication of the positive going signal at the base of trans-
istor Q3. The result is that common mode signals are amplified
substantially less than difference mode signals, i.e. the
stage possesses common mode rejection.
In contrast, the transistor Ql, Ql' stage does
not possess CMR, since although resistors R2, R2' will reduce
the gain given to common mode signals, they will equa:Lly
reduce the gain given to difference mode signals.
The second preamplifier stage, consisting of
transistors Q2 and Q2' is provided with CMR -through transistor
Q5 and resistor R5. Specifically, the emitters of transistors
Q2, Q2' are connected together to node 42 and are then connected
to ground through resistor R5. The collector-emitter circuit
of transistor Q5 is connected in parallel with resistor R5.
Transistor Q5 is biased to operated as a current source, by
means of diode connected transistor Q6, which is connected
to the base of transistor Q5 as shown. The transistor Q6
collector is connected through resistor R13 to the posit:ive
- 13 -
- ~77S8~
supply Vb, and the current through transistor Q6 sets the
current through transistor Q5.
Ideally, optimum CMR would be provided by
eliminating resistor R5 and simply providing transistor Q5
to connect node 42 to ground (since the current source
constituted by transistor Q5 theoretically has infinite
impedance). However, if the node 42 were connected to ground
only through the current source constituted by transistor Q5,
then the voltage at node 42 would be unconstrained and could
be at any potential, as determined by the current source.
This would destroy the DC gain provided by the second stage
transistors Q2, Q2' and would also destroy -the regulating
act:ion provided by the DC feedback loop through resistors R9
and Rl. IL'hercEore, a finite res:istance conncction Erom nod~
~2 to yround is required to constrain the voltage at the
emitters of transistors Q2, Q2', and for this reason resistor
R5 is provided connected in parallel with transistor Q5.
It should be noted that small resistances may be
inserted for example between the emitters of Q3, Q3' and node
40 and between the emitters of transistors Q2, Q2' and node
~2, without seriously reducing the CMR of those stages.
Reference is next made to Fig. 3, which shows a
circuit closely reseTnbling that of Fig. 1. In Fig. 3,
corresponding references have been used to indicate parts
corresponding to those of Fig. 1.
The Fig. 3 circuit is designed for less gain
than the Fig. 1 circuit, and accordingly, transistors Q2,
Q2' of Fig. 3 have been made PNP transistors. Since the
currents flowing in the emitter-collector circuits of trans-
istors Q2, Q2' are effectively the base currents of trans-
istors Q3, Q3', the power requirements and the gain of the
Fig. 3 circuit are both reduced from those of the Fig. 1
circuit. Common mode rejection is provided in the Fig 3
- 14 -
~L~7758~
circuit by connecting the emitters of transistors Q3, Q3'
together and connecting them to ground through resistor
R14, and by similarly eonnecting the emitters of transistors
Ql, Ql' together and connecting them through resistor R14
to ground.
It is found in the Fig. 3 eireuit that eapaeitors
C3, C3' require substantial eharging current in order to
follow the signal. Under some eonditions the colleetor current
of transistor Q2 is too small to provide adequate charging
current, resulting in distortion. Accordingly, and as
indicated in dotted lines, resistors R20, R20' may be provided
to connect the collectors of transistors Q2, Q2' to ground,
thereby increasing the currents in the Q2, Q2' stage~
R~sistor R21 may be provided to eonneet th~ eolleetors of
-transistors Q2, Q2' tocJether, thereby compensating the gain
increase which results from the increased collector currents
of Q2, Q2' eaused by the addition of resistors R20, R20'
In addition, and if desired, additional resistors
R22, R22' may be provided, as indieated in dotted lines,
between the collectors of transistors Q3, Q3' and the bases
of transistors Q~, Q4' respectively, to further linearize
the operation of transistor Q4 (by reducing the variability
with current of the input impedanee of transistors Q4, Q4').
Apart from these changes, the Fig. 3 eireuit is
essentially identieal with the Fig. 1 eircuit.
Referenee is next made to Fig. 4 which shows a
circuit which is again almost identieal with that of Fig. 1,
and again eorresponding referenee numerals indieate parts
eorresponding to those of Fig. 1. The differenees between
the Fig. 4 eireuit and the Fig. 1 eircuit are that in the
Fig. 4 circuit, -transistor Q5 and resistor R5 have been r
- 15 -
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deleted, removing the CMR for the second stage of the pre-
amplifier. Instead/ CMR has been provided for the first
stage by connecting the emitters of transist:ors Q1, Ql'
together and through resistor R6 to ground. In addition,
compensation resistor R12 has been deleted~
Typical values for the components shown in
Fig. 1 are given in Table I below.
Table I
COMPONENT TYPICAL VALUE
Rl, Rl 27 Kohms
R2, R2 2.3 Kohms
R3~ 3 52 Kohms
R~, R~' 120 ohms
R5 10 Kohms
R6, R6 1.5 Kohms
R7, R7 24 Kohms
R8, R8 400 ohms
Rg Rg' 47 Kohms
Rlo, Rlo 9.1 ~ohms
Rll, Rll 15 Kohms
R12, R12, 8.2 Kohms
R13 36 Kohms
Load Resistor 16 1 Kohms
Cl, Cl' 10~ f
C2 .03~ f
C3, C3' 100 pf
Vb 1.55 volts
Q , Q ,, Q , Q ', Q3 Q3, )npn transistors with an emitter
1 1 2 2 )area mu:Ltiple of 1
Q5' Q~ 3
Q , Q ' )npn transistors with an ~emitter
4 4 )area multiple of 3
Transformer 14 )centre tapped primary l::L turns
)ratio
