Note: Descriptions are shown in the official language in which they were submitted.
This invention relates generally to circuits
for driving inductive loads such as solenoids and, more
particularly, to circuits wherein power consumption is
minimized by operating a power stage in a switching mode.
This application is related to the subject matter
of applicant's United States Patent to Reddy 3,725,678
issued April 3, 1973. This application is also related
to the subject matter of applicant's co-pending application
Serial No. 321,335, filed February 13, ]979.
Description of the Prior Art.
The above-identified Reddy patent discloses
injector drive circuits wherein the current is controlled
to first regulate the voltage across the injector coil
until after the current has built up enough to open or
pull in the injector armature and to thereafter regulate
the current at a holding level in excess of a closing or
drop out current but considerably less than the pull in--
current. These drive circuits render the injector opening
`~ and closing times, and therefore the fuel supplied
therebetween, substantially independent of variations in
the power supply and variations in the unit-to-unit
?~ s mb/~lD
voltage drops across the power stage.
However, in regulating first the voltage and
then the hold current, the power stage experiences a
voltage drop thereacross corresponding to the difference
between the voltage of the power supply and the voltage
- at the injectors. The power stage therefore consumes
and must dissipate power at a level increasing with
the number of injeetors being driven at one time and
the current required to operate each injector. The
heat dissipation and therefore the heat sink and tem-
perature ranges associated with these power stages also
increase accordingly to the point that the heat sink
is often larger than all of the other elements of the
electronic control unit leading up to the injectors.
Moreover, the temperature cycles produced by the dis-
sipation decrease the life while increasing the mounting
costs of the semi-conductor elements comprising the
power stage. It is therefore desirable to reduce the
power dissipated by the power stage.
The United States patent to Paine 3,549,955
discloses a circuit for minimizing power consumption
by operating the power stage fully ON so that there
is a very small voltage drop thereacross until a pull-
in current level is detected and thereafter operating
the power stage in a switching mode where it is either
fully ON or OFF to maintain a hold-in current level
in excess of the drop out level. More specifically,
the current is alternately increased to an upper hold-
in level in excess of the drop out level and is then
allowed to decay slowly through the solenoid coil to
~lower hold-in level still in excess of the drop-out
level. Thereafter, the power stage is switched fully
GN again until the upper hold-in level is again de-
tected.
The United States patent to Ule 3,896,3~
discloses an inductive load driver circuit of the type
.~ s
~isclosed in the Paine patent but wherein power con-
sumption ls further Minimized by returning -the energy
ln the collapsing solenoid field -to energy storage means
in the form of the power supply or a second solenoid.
Citing the Paine patent, the United States
patent to Stewart 4,041,546 discloses a solenoid drive
circuit including capacitor timing means for discon-
necting the driving voltage for fixed intervals between
eaeh application of hold current.
To minimize power consumption, switching-mode
decay rates effected by circuits of the type disclosed
in Paine, Ule, and Stewart patents must be sufficiently
slow to maintain the current above the drop-out level
while the power stage is OFF. Using such circuits to
'control the fuel injector would -ender the turn off time
of the injector, and therefore the fuel delivered there-
by, subject to when in the decay cycle it is desired
to close the injector. For example, if the end of the
injeetor actuation command coincided with the instant
ao that the eurrent exceeded the upper hold-in level, then
the elosing time would be the time rec~uired to deeay
to the lowex hold-in level plus the time required to
deeay, therefrom to the drop-out, level. If the end of
the aetuation eommand eoineidec~ with the instant that
the eurrent fell below the low~3r hold in level/ then
the closing time would be just the time required for
the current to deeay to the drop out level.
Use of circuits of the type disclosed in the
Paine, Ule, and Stewart patents to eontrol a fueI in-
jeetion valve would therefore incur not only the varia-
tions in opening times eliminated by the eircuits of
the above-identified Reddy cases but would also intro-
duee variations in closing times that would offset a substan-
tial portion o~ all the variations eliminated by the
Reddy eircuits.
-- 3 --
. .
ms/~
'
~4 ~ 5
It is therefore desirable to provide an im-
proved circuit wherein the solenoid current is held
above a drop-out level by switching the power stage
between fully ON and OFF conditions to reduce power
consumption and then rendering the closing time inde-
pendent of decay rates associated with such hold level
switching.
OBJECTS
It is an object of the present invention to
provide a new and useful method and dr apparatus for
controlling the current drive to the solenoid coil of
an electromagnetically-actuated device.
It is another object of the present invention
to provide a method and apparatus of the foregoing type
that minimizes the power consumed after the electro-
magnetic device has been actuated from one position
to another and thflt minimizes the time required to re-
turn the electromagnetic device from the second position
to the first.
It is another primary object of the present
invention to provide a method and apparatus for activat-
ing and deactivating a solenoid operated electromagnetic
device wherein current is allowed to flow to the coil
of the solenoid through a slow current decay path when
- 25 it is desired to maintain the devi~e in its activated
position and through a fast decay path when it is de-
sired to allow the device to return to its unactivated
position.
It is another object of the present invention
3~ to provide a method and apparatus of the foregoing type
wherein the slow dec&y path comprises an SCR disabled
by a final momentary turn ON of the power switch nor-
mally building up the load current.
It is another primary object of the present
invention to provide Q method and apparatus for con-
0~5
trolling the current through an inductive load wherein
comparator means respond to sensed levels of load cur-
rent to generate HIG~ and LOW output levels that are
fed back to the reference input of the comparator to
S generate reference voltages representative of first
and second levels of load current and that are also
applied to switch means to connect and disconnect power
to the load so as to maintain the load current between
the first and second current levels.
~It is another object of the present invention
,;~ro v;~e
to ~m~e a method and apparatus of the foregoing type
wherein flip-flop means cause variable current source
means to change the magnitude of current through a
sense resistor coupled to the comparator sense input
lS when the load current first exceeds a peak actuation
level.
SUMMARY OF INVENTION
An inductive load driver circuit comprises
first and second switch means and high and low impedance
load current decay means. For the duration of a load
actuation signal the first switch means are responsive
to the magnitude of the load current to complete and
interrupt a first current path to the load to cycle
the load current between first and second levels. The
second switch means are enabled for the duration of
the load actuation signal to complete a second current
path to the load when the first switch means interrupts
the first current path. When completed the second cur-
~ rent path comprises the low impedance decay means al-
- 3~ lowing the load current to decay slowly from the first
current level to the second level to provide during
;- such slow decay a loa,d current sufficient to maintain
~ e
actuation of the i~rt~to~ load.
The end of the load actuation signal disables
the seooni switch means to interrupt the seconù current
,,
path and thereby require the load current to decay
rapidly through the high impedance decay means. In
one embodiment of the invention, the second switch means
comprise a SCR disabled by a final momentary turn ON
of the first switch means.
Comparator means respond to the current levels
through an inductive load to generate HIGH and LOW level
outputs causing first switch means to connect and dis-
connect power to the load. When the load current ini-
tially exceeds a peak load actuation level, flip-flop
means responsive to the corresponding change in com-
parator output levels change the current through an
auxilliary sense resistor coupled to the sense input
of the comparator means. In one embodiment, ~ one-
shot multivibrator prevents a change in the currentto the auxilliary sense resistor for a fixed period
after the commencement of a load actuation signal.
The subsequent HIGH and LOW comparator output levels
are fed back to the comparator reference input to switch
the reference voltage thereat so as to represent first
and second levels of the current sufficient to maintain
actuation of the load.
These and other objects and features of the
present invention will become more apparent from the
following written description taken in conjunction with
the following figures wherein
FIGURES
FIGURE 1 illustrates in block diagram form
an engine control system in the form of a fuel injection
control system utilizing one or more transduced engine
dependent parameters to control the pulse duration of
fuel injected into an internal combustion engine;
FIGURE 2 illustrates in circuit schematic
form one embodiment of the present invention;
S
FIGURE 3 illustrates in time coordinated form
certain waveforms illustrative of the operation of the
Figure 2 embodiment; and
FIG~RE 4 illustrates in circuit schematic form a
second embodiment of the present invention.
FIGURE l
The circuit of the present application may be
utilized in combination with any inductive device requiring
the precise control of actuation thereof with minimum
expenditure of power. One such application is to control
one or more electromagnetically-actuatable fuel injection
valves to provide single, simultaneous, or grouped fuel
pulses of controllable duration to certain solenoid
controlled fuel inlet passages of an internal combustion
engine.
One such fuel injection system may be of the type
shown in the block diagram of Figure 1. Therein, an
injector drive circuit 10 of the present invention responds
to injector actuation signals provided by a fuel injection
control system 12 to control one or more injectors 14 to
inject a precise quantity of fuel into a suitable fuel
intake passage of internal combustion engine 16. The
injector actuation signals are computed in response to one
or more engine-dependent parameters communicated from
engine 16 to fuel injection control system 12 by one or
more engine trans~ucers, such as RPM transducer 18, an
intake manifold pressure transducer 20, an engine temperature
transducer 22~ or an engine roughness sensor 24.
mb/~o _ 7 _
.
~" ~ ''''' , . ' .
Fuel injection control system 12 may be of the
type disclosed in commonly-assigned U.S. Patents 3,734,068
and RE 29,060 issued to Reddy respectively on May 22, 1973
and December 7, 1976. As disclosed more fully therein,
during one portion o-E an engine cycle, a capacitor is
initiali~ed to a starting value in accordance with one
engine-dependent parameter such as the magnitude of a
speed-dependent trigger signal as developed by RPM
transducer 22. In the next portion of the engine cycle
the capacitor is charged with a ramp voltage, the slope
of which may be modified in accordance with a temperature
signal as developed by temperature transducer 22.
Comparator means then compare the resulting magnitude of
this ramp voltage with a reference signal in the form of a
pressure signal, as developed by a pressure transducer
20 which may be of the type disclosed in applicant1s U.S.
Patent No. 4,131,088, issued December 26, 1978. The
comparator means generates an injector actuation signal Tp
having a duration beginning at the start of the second
portion of the engine cycle and ending when the ramp voltage
exceeds the pressure reference. The injector actuation
~:: signal may be further modified as disclosed in the commonly-
assigned patent to Taplin et al. 3,789,816 using a
roughness signal as developed by a roughness sensor which
may be of the type shown in applicantls U.S~ Patent No.
4,092,955, issued June 6, 1978.
The fuel injection valve 14 may be of the type
disclosed in commonly-assigned U.S. patent 4,030,668 issued
mb~ - 8 -
s
on June 21, 1977 to Kiwior. The fuel injection
valve described comprises electromagnetic coil means
operative when suitably energized to apply a
motion-imparting magneto-motive force to armature
means of a movable actuator means. The actuator
means are moved against the bias of a closing spring
from a closed position whereat valve head means carried
by the actuator means seat on valve seal means to
an open position whereat a radial shoulder element
of the actuator means abuts a radial surface
~b/J~ - 8a -
.
L7'0:~5
fixed with respect -to the val~e body in which the actua-
tor means reciprocate.
FIGURE 2
Figure 2 shows one embodiment of the present
invention, the operation of which is explained in con-
junction with the waveforms shown in Figure 3.
In Figure 3, waveform 3a represents an in-
jector actuation signal Tp, the width of which is gen-
erated to control the opening and closing of one or
more fuel injectors. Waveform 3b illustrates the ap-
plication of either a regulated voltage or an unregulated
voltage s+ to one side of the solenoid coil of an
injector. Waveform 3c illustrates the current through
tne solenoid coil, Io being the current at which the
injector armature begins to move ~rom its closed to
its open position, Ip being a current slightly in excess
of the current at which the in~ector always fully opens,
IHL being a lower level of holding current slightly
in excess of the current level at which the injector
armature begins to drop out, and IHH being a h~igher
level of holding current in excess of the lower holding
level. rl is the efective decay constant associated with
the decay of current from IHH and IHL and l2 is the
effective decay constant associated with the return
of the injector armature from its open to its closed
position.
In the Figure 2 embodiment, inductor Ll repre-
.
sents the solenoid coils of one or more electromagnetic-
cally-actuated fuel injection valves 14. First switch
means in the form of a suitable NPN power transistor
` Ql couples one side of A of coil Ll and a load conductor
30, here connected to the high or B-~ side of a suitable
power supply. Coil sensing means in the form of a small
ohmage current sensing resistor Rl couples the other side
B of coil Ll and a second load conductor 32, here
ms/~
~L~L47~
connected to -the ground side of the power supply,
Connected in series between ground conductor 32 and
coil side A are controlLable coil current decay means'
here in the form of an NPN transistor Q2 and unidirec-
tional current conducting means in the form of diode Dl~
When rendered conductive as will be discussed
shor~-ly, power transistor Ql completes a first current
path to and through coil ~1 and sensing resistor Rl~
such path comprising B+ conductor 30 and the emitter-
to-collector junction of ~1. When rendered non-con-
ductive, power transistor Ql interrupts this first,
current path. Transistor Q2 when rendered conductive
'completes a second'current path to and through coil
Ll and sensing resistor Rl, such second path comprising
the ground conductor 32, the collector-to-emitter j,unc-
tion of transistor Q2, and anode-to-cathode junction
of diode Dl. And when rendered non-conductive~ tran-
sistor Q2 interrupts this second current path.
~ To enable both powe:r transistor Ql and decay
20, transistor Q2 during the presence of an injector actuation
signal TP, at terminal C, the terminal is coupled
respectivel~v to thè base of an NPN input transistor Q3
by an input resistor R2 and to the base of another NPN
,input transistor Q4 by another input resistor R3.' The Q3
coilector is coupled to B+ by series-connected res'istors
: R8 and R9 and, at a node D therebetween, is coupled by
resistor R10 to the non-inverting input of a comparator
- CPl. A Zener diode ~1 having its anode connected to node
-~ D and its cathode coupled to B+ conductor 30 is operative,
for the duration of an injector actuation signal TP at
the ~3 base, to clamp the voltage at node D at a reference
voltage of B+ less the breakdown voltage of the Zener ~1.
' The Q4 emitter is ~rounded at conductor 32,
``~ and tlle Q4 collector is coupled to B+ conductor 30 by
series-connected resistors R4, R5 and at a node E there-
between is coupled to the base of a PNP transistor Q5.
- 10 -
;,~, ms/~
7~ 5
The Q5 emitter is coupled to B-l~ conductor 30~ and the Q5
collector is coupled by resistor R6 to the base of
transistor Q2 and therefrom by resistor R7 to point A.
An injector actuation signal TP saturates transistor Q4
and through transistor Q4 enables transistors Q2 and Q5
to saturate whenever the potential at coil side A is
suf~iciently beLow that o~ coil side B~
Comparator CPl may be a conventional oper~-
tional amplifier such as a 2901 quad comparator of the
type that Produces HIGH and LOW ouput levels, here
of B+ and zero volts, when the voltage at its non-in-
verting input is respectively greater and less thàn
the voltage at its inverting input. In the Figure 2
embodiment, the voltages at the inverting input and
non-inverting input are both referenced to that of
B+ conductor 30 and respectively comprise'a sense voltage
and a reference voltage. As will be explained more fully
shortly,'the voltage at the inverting CPl input drops
below B+ with increasi.ng current`through'coil L1.
' The output of comparator CPl is coupled by
a feedback resistor Rll back to the non-inverting input
and is cou,pled by another resistor R12 to the,base of
a first PNP drive control transistor Q6. The Q6 collector
is connected to ground conductor 32, and the Q6 emitter
is coupled to B+ conductor 30 by series-connected
resistors R13 and R14 to suitably bias, at a node F
therebetween, the base o~ a second PNY drive control
transistor Q7. The Q7 emitter is coupled to B~ conductor
30, and the Q7 collector is connected to the Ql base.
~ Zener diode ~2 having`its anode coupled to the Ql emitter
at coil side A and its cathode coupled to the B-~ conductor
30 through resistor R14 protects both transistors Ql.and
Q7 against the possibly-damaging
, -- 11 --
~ ms ~
~l~7a~s
reverse voltages effected by the inductive kicks produced
when transistor Ql interrupts the first current path to
coil Ll.
Representative of a one circuit location responsive
to the attainment of the peak current the output of
comparator CPl is also connected to the reset input R of
a flip-flop FFl. The set input S of flip-flop FFl is
coupled by a capacitor Cl to the injector actùation terminal
C and in response to a positive-going set input produces
both a HIGH level output at one output terminal Q and a LOW
level output at another output terminal Q*. The LOW level
output Q* is connected to the base of a PNP transistor Q8,
the emitter of which is connected to B-~ conductor 30. The
Q8 collector is coupled to the inverting input of comparator
CPl by series-connected resistorc; R15 and R16, and the node
G therebetween is coupled by a resistor R17 to B+ conductor
30.
The exact values of voltage drop across resistors
R15 and R17 needed to cause comparator CPl to switch from
one of its output levels to the other are determined by
the relative values of resistors R10 and Rll and the voltage
thereacross. These points may be computed assuming a B+
of 14 volts, Rl0 and Rll values of 10K and 100K respectively,
and a 5.6 volt breakdown voltage for Zener ~.1.
With a zero output level from comparator CPl, as
exists to complete the first current path to coil Ll, the
voltage across resistors R10 and Rll is B+ less the breakdown
voltage of Zener Zl. The reference voltage resulting at
the non-inverting input of comparator CPl is therefore
mb/~ 12 -
.
~14~$
equal to [B+ - VZ] -R10~+ Rll' or (14-5.6) 0.9 = 7.6.
With a HIGH output
~i mb/~)b - 12a-
. .
,
s
level ~rom comparator CPl of 1~ volts as exits to
interrupt ~he first current path, the voltage across
resistors R10 and ~11 decreases to 5.6 volts which
when multiplied by the 0.9 ratio of resistance R10/R10 +
Rll results in a second reference voltage at the non-
inverting CPl input terminal of about S.l volts below B+.
In other words, the ma~nitude of feedback re-
sistor Rll cooperates with the magnitude of input re-
sistor R10 and the magnitudes of the two different CPl
output levels ~here Zero and B+~ to vary reference
voltage at the non-inverting CPl input between 6.4 volts
below B+ when the CPl output is ~ero volts and 5~1 volts
below B+ when the CPl output is 14 volts. As will be
discussed shortly~ when resistor R15 is connected in
parallel with resistor R17 to set the current dropping
the voltage from B-~, these reference voltages and the
"hysteresis" between them, determine the peak opening
current level Ip, Figure 3c, and the lower level of
the hold current IHL and the difference between them.
These reference voltages also determined the higher
level of the hold current level IHH and lower hold
current level IHL, and the difference between them,
when resistor R15 is not used to set the current drop-
ping volta~e from B+.
To generate a voltage corresponding to that
produced across sensing resistor Rl, node G is also
coupled to the collector of an NPN transistor Q9, the
emitter of which is coupled by a variable resistor R18
to ground conductor 32. The Q9 base is coupled to B+
conductor 30 by resistor R19 and also to the emitter
of a PNP transistor Q10, the collector of which is
grounded at conductor 32. The Q10 base is coupled to the
current sensing resistor Rl at coil side B and the QlO base-
-~ to-emitter voltage drop is selected to negate the-Q9
base-to-emitter drop.
~J csm/h~
70~l5
As will be discussed shortly, when the
injectors are being opened, ~ransistors ~9 and Q10 co-
operate with resistors Rl5 and Rl7 to develop a very
small current that produces across resistor Rl8 a vol-
tage representing that developed across current sense
resistor Rl at the much larger injector opening drive
current. After the injectors have opened and the cur~
rent thereto is reduced to just hold them open, tran-
sistors Q9 and Ql0 cooperate with resistors R17 and
Rl8 to produce a second very small current that produces
across resistor R18 a voltage representing that produced
across resistor Rl at the much larger injector holding
current. Thus, assuming each injector requires a peak
current of 1.5 amps to open and current of 0.4 amps
to hold it open, a voltage drop of 0.15 volts would
be produced across a sensing resistor Rl of 0.1 ohms
by the 1.5 amp peak opening current lp and 0.04 volts
would be produced by the 0.4 amp holding currentO
Assuming the circuit of Figure 2 drives eight injectors
ak a time, the required total opening current of 12
amps and total holding current of 3.2 amps would produce
respective volt drops across sensing resistor Rl of
about 1.2 and 0.32 volts and corresponding voltage drops
would be produced across resistor R18.
The magnitude o~ the ~18 resistance is ad-
justed so that, with transistor Q8 biased ON, the peak
current through resistance Rl8 produces a voltage
drop across resistance R15 in parallel with a resistance
Rl7 just exceeding the 6.4 volt drop to the non-inverting
input of comparator CPl. Assuming values of resistances
R15 and R17 respectively of 5K ohms and 10k ohms and a 5.6
volt breakdown for Zener ~1~ this 6.4 vo.lt drop-to the non-
inverting CPl input when divided by the 3.3K parallel
resistance of R15 and R17 would produce therethrough
a current of about l.93 milliamps Then to match the
- 14 -
.. csm/,~
, .
~L~4710~
1.2 volt drop produced across resistor Rl by the 12
amp peak opening current with these 1.93 milliamps
across R18, the value of R18 would be set at about 625
ohms.
As will be explained more fully shortly, after
the peak opening current is detected by comparator CPl,
the resulting HIGH output therefrom resets flip-flop
FFl. The HIGH Q* output stops current flow through
resistor R15 and causes just resistor R17 to determine
both the sense voltage at the inverting input of comparator
CPl and the magnitude of the current through resistor R18
associated with the higher and lower levels of the hold
currentO Resistance R17 reduces to 0.51 milliamps the
current required to exceed the now 5.1 volt drop to
- the non-inverting input of comparator CPl and these 0.51
milliamps in turn produce about a 0.32 volt drop across
resistor R18 corresp.)nding to a lower level of hold current
of about 3.2 ampsO
In the operation of the embodiment illustrated
in Figure 2, prior to the beginning of an injector
actuation signal TP, transistors Q3 and Q~, and through
Q4 transistors Q2 and Q5, are biased OFF. Until set
by the beginning of an injector actuation signal TP,
flip-flop FFl produces a HIGH voltage at its Q* output,
thereby biasing Q8 OFF. With B-~ voltage coupled to
- - the non-inverting input of comparator CPl by resistors R9
and R10, that input controls and comparator CPl produces a
HIGH level output, thereby biasing OFF transistors Q6 ana
in turn Q7 and Ql. With no current flowing through
sensing resistor Rl, transistor Ql~ and therefore tran-
sistor Q9 are biased OFF and essentially no current
is drawn aaross resistor Rl70
Concurrent with the beginning of an injec-
tor actuation signal TP, the leading positive-going
edge thereof is differentiated by capacitor Cl to pro-
vide a set pulse to the set input S of flip-flop FFl.
.
- 15 ~
c sm/J~
~'
.
. .
7015
The resulting LOW level produced at the Q* output of
flip-flop FFl biases ON transistor Q8 so that the vol~
tage at node G to inverting input o~ comparator CPl
i9 then that on B+ conductor 30 less that developed
across the parallel combination of resistors R15 and
R17. However, immediately after the beginning of a
TP signal, the voltage at the inverting input is still
close to B-~ while the voltage at the non-inverting CPl
input is dropped to B+ less the Zener breakdown. The
higher voltage at the inverting CPl input then causes
the comparator CPl to produce a LOW output to bias ON
transistor Ql through transistors Q6 and Q7.
As the current to coil Ll builds up toward
the 12 amp peak opening current lp, the current through
~ resistors Rl and RlB also builds. When the R18 current
; exceeds 1.93 milliamps, this current produces a voltage
drop of greater than 6.4 volts across resistors R15
and R17, thereby lowering the voltage at node G to the
inverting input below that at the non-inverting
input. With a greater voltage resulting at its non-
inverting input, comparator CPl produces a HIGH level
output, biasing OFF transistor Q6 and therethrough tran~
sistors Q7 and Ql.
When output of comparator CPl switches to a HIGH
level output, this output resets flip-flop FFl to produce a
HIGH level output at its Q* terminal and in turn to bias
OFF transistor Q8. With transistor Q8 OFF, just re-
sistor R17 develops the volta~e drop from B+ to the
inverting input of comparator CPl and does so with less
current since the resistor R15 no longer is in parallel
with resistor R17-.
- With transistor Q1 biased OFF, the first cu~-
rent path to coil Ll is interrupted to allow the current
therethrough to decay and to thereb~ reverse the voltage
induced thereacross so that side B is positive with
respect to side A. With its collector now at a lower voltage
- 16 -
csm/~ -
than its base~ previously enabled transistor Q5 completes
a biasing circuit to -the 02 base across R7. With its
emitter now at a lower potential than its base, pre-
viously enabled transistor Q2 now completes the second
current path to provide replenishment current from
ground conductor 32 through diode Dl, coil Ll, and
sensing resistor Rl.
The ef~ective impedance to the current de-
caying through this second path comprises approximately
0.3 ohms representing the parallel equivalent of an
internal resistance of 2.3 ohms for each of eight
parallelly-connected injectors plus .016 ohms produced
by 12 amps across the 0.2 volt drop of 0.1 volt each
across transistor Q2 and diode Dl. Dividing this ef-
fective 0.316 ohm impedance into the 1.67 millihenry
equivalent of an inductance of eight parallelly-con-
nected injectors of 13.25 millihenries each fixes the
L/R decay time constant T 1 of this circuit at about
5.3 milliseconds and thereby fixes the perio~d rec~uired
to decay from one known level to another.
Thus, the coil current decays at essentially
this rate from 12 amps to the lower holding level IHL
of about 3.2 amps where the volt:age across resistor
R17 at the inverting input o~ the comparator CPl is less
than the now 5.1 volt drop to the non-inverting CPl input.
To develop S.l volts across the lOK resistor R17 requires
about 0.51 milliamps which produces a 0.32 volt drop
across R18.
~hen the injector current decays just below
- 30 3.2 amps, the voltage drop across R17 to the inverting
CPl input decreases below the 5.1 volts dropped to the
noninverting CPl input so that the voltage at the in-
verting input of the comparator CPl is greater than at the
non-inverting input. With the inverting input then
controlling, comparator CPl again produces a ~ero level
output biasing ON transistor Ql through transistors 06 and
Q7 and increasing the reference voltage at the CPl non-
inverting input to a 6.4 volt drop from B+.
.
~s csm/~ - 17 -
.
The injector current then increases agaln
until 6.4 volts is again effected to the inverting
input across now just resistor R17. This drop is
effected by 0.~4 milliamps across resistor R17 which
produces a drop of about 0.~ volts across resistor R18.
A drop of 0.4 volts across resistor R18 corresponds
to a similar drop across resistor Rl and therefore to
an injector current of 4 amps thereacross.
The injector current thereafter is cycled
between the lower and the higher levels of holding
current of 3.2 and 4 amps respectively until the end
of the injector actuation signal TPo At that instant
the injector current would be at some unknown value
between 3.2 and 4 amps. If the second current path
were still enabled, the coil current would require some
unknown time up to one millisecond to decay below the
low holding level of 3.2 amps. Since an uncertainty
of up to a millisecond in the closing time of the in-
jector would detract from the precision required for
fuel injection, the second current path is thus imme-
.diately disabled with the end of the injector actuation
pulse to require the.coil replenishment current to be
supplied through a much faster decay circuit comprising
Zener ~2. Assuming a breakdown voltage o 33 volts
for Zener ~2 r an effective resistance of about 8 ohms
is presented by this Zener to a 4 amp holding level
` ~ current. Com~ining these effective resistances with`
the 0.3 ohm the equivalent internal resistance of the
: eight 2.3 ohm injector coils results in a total efféc~
,` 30 tive resistance of 8.3 ohmsO Dividing the 1~67 milli~
henry effective inductance of eight 13.25 millihenry
injector coils results in a L/R decay time constant
2 of about 0.2 milliseconds. This 0.2 millisecond de-
cay constant ~2 is. more than 20 times faster than the
decay constant Tl effected ~y the second current path
and effectively eliminates variations in closing times
as a factor degrading precision of fuel injection.
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FIGURE 4
In the alte~native embodiment of the invention
illustrated in Figure 4, the current sense resistor
Rl is located on the B-~ side of each injector so that
the voltage across the sense resistor is measured with
respect to s+ rather than with respect to ground. Also,
-to reduce the power loss and added heating associated
with driving the decay transistor Q2, this device is
replaced by a silicon controlled rectifier SCR 1, which
does not require a continuous drive but does require
`additional control circuitry to effect turn on and turn
off. Also, to assure that the injectors pull-in in the
presence of increases in the supply voltages to levels
where the .rise time of current might be faster than
the mechanical response time of the injectors, the Figure
4 embodiment also comprises circuitry for holding the
peak opening current Ip for at least a minimum fixed
period.
Again, inductor Coil Ll represents the sole-
noid coils of one or more electromagnetically-actuated
fuel injector valves 14 to be d~.iven singly, simultane-
ously, or in groups with pbwer supplied between a B+
conductor 30 and a ground conductor 32. First switch
means in the form o Darlington connected NPN transis-
tors Qll, such as a RCA expitaxial TA 8997, couples.
-:: one side A of coil Ll and ground conductor 32, and a
small ohmage-current sensing resistor Rl couples the
other side B of coil Ll to the B+ conductor 30. Coil side
A is also coupled directly to the anode of a siIicon
controlled rectifier SCR l, the cathode of which is
coupled to the B+ conductor ~0. Coil side A is also
. coupled to the gate o~ SCR l by a pair of series con-
nected capacitors C2 and C3 and to ground 32 by a re-
sistor 21.
Whenever transistor Qll is biased OFF, during
a TP signal the resulting back-emf-induced voltage rise
at coil side A is communicated by capacitors C2 and
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C3 -to the gate o~ SCR 1 to turn ON SCR 1 un~ll reverse
biased OFF again by a subsequent turn ON of Qll. To
insure that SCR 1 does not come ON when an injector
actuation signal TP is not present, the node H between
capacitors C2 and C3 is coupled to the collector of
an NPN transistor Q12 and is grounded therethrough
during a TP* signal, the complement of the TP signal~
To be able to alternately-complete and in-
terrupt a first current path from coil side A through
transistor Qll to ground 32, the Qll base is coupled
to ground 32 by a resistor 22 and also to B+ 30 by a
resistor 23 connected in series to the Qll base by the
emitter-to-col].ector junction of a PNP transistor Q13.
The Q13 base is connected to the emitter of a PNP tran-
sistor Ql~ and to B+ 30 by a resistor R24. The Q14
collector is grounded at 32, and the Q14 base is coupled
by a resistor R25 to B+ conductor 30 and by a resistor R26
to the output of a comparator CP2.
Comparator CP2 has an inverting input and
a non-inverting input, here comprising the reference
and sense inputs, respectively. The CP2 output is
coupled by a resistor R27 to the CP2 non-inverting
input, which is coupled to grour~d 32 by series connected
resistors R28 and R29 having a rlode J therebet~een.
Coupled to the node J are the collectors of a pair of
PNP current source transistors Q15 and Q16 each of which
comprise an emitter follower circuit with an NPN tran-
sistorO ~he Q15 emittex is coupled to B+ R30 by a fixed
` resistor R30 and a variable resistor R31, and similarly
the Q16 emitter is coupled to B~ 30 by a fixed resistor
R32 and a variable resistor R33. The Q15 and Q16, bases
are both coupled by a resistor R34 to ground 32 and
by the emitter-to-base junction of transistor Q17 to
coil side B. The Q17 collector is coupled to B+ conductor
30. With the base-to-emitter drop of transistor Q17
`~ matching the base-to-emitter rises of transistor Q15 and
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Q16, the Q15 and Q16 emi-tters therefore follow the
voltage at poin-t B to provide currents through resistors
R30-R31 and R32-R33, each varying directly wi-th the current
through sensing resistor Rl. Summed at node J, these
Q15 and Q16 currents develop a voltage across resistor
R29 varying with the injector currentO To selectively
bias OFF transistor Q15 and stop current therethrough
to resistor R29 after the pull-in level Ip of injector
current is attained, the Q15 emitter is coupled to
ground conductor 32 by the series connection of a resistor
R35 and the emitter-to-collector junction of an NPN tran-
sistor Q18.
The output of comparator CP2 is also coupled
by a capacitor C4 and a diode D2 to the base of an NPN
transistor Ql9 of a flip-flop FF2 also comprising a
second NPN transistor Q20. (Another circuit location
responsive to the attainment of the peak current to
which flip-flop FF2 might be coupled by capacitor C4
and Diode D2 is the emitter of Darlington transistor
Q13. This circuit location provides more current`and
power than the uncommitted emitter in the 2901 quad
comparator CP2.) The Ql9 collector is coupled by a
resistor R36 to the Q18 base, and the Ql9 and Q20 col-
lectors are coupled by respecti~ve resistors R37 and
R38 to B+ conductor 30 and by resistors R39 and R43 to
the Q20 and Ql9 bases respectively. The Ql9 and Q20
emitters are grounded at 320
Injector actuation terminal C is coupled by
a capacitor C5 to the node between diodes D4 and
D5 connected ~orwardly in series from ground 32 to the
Q20 base. Terminal C is also coupled by resistors R40
and R41 to the bases of NPN transistors Q21 and Q22
respectivelyO The Q21 collector is series-connected
to ground 32 by a capacitor C6 and a resistor R42 having
a node K therebetween coupled by a resistor R43 to the
inverting CP2 input. The Q22 collector is coupled to
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s+ conductor 30 by a resistor R4~ and by a resistor R~5 to
the Q12 base. An injector actuation signal TP biases ON
transistor Q22 and biases OFF transistor Q12 through
the Q22 collector-to-emitter junction. When an injector
actuation signal Tp is not present, the resulting HIGH
Q22 collector voltage produces the TP* signal biasing
ON transistor Q12.
Also coupled to the Q22 collector is the
base of an NPN transistor Q23, the collector of which
is coupled by a resistor R47 to B~ conductor 30 and
the emitter of which is coupled to capacitor C6 and the
collector of a transistor Q21. ,When the TP signal is
present, resistor R46 and the Q22 collector-to-emitter
junction are coupled within the base of a PNP transistor
, Q24 to ground 32, thereby biasing ON transistor ~24 and
coupling the breakdown voltage of a Zener diode ~4 to the
inverting input of comparator CP2.
Even though the SCR 1 gate signal is effectively
grounded by transistor Q12 by the fall of the TP signal,
~ 20 the SCR 1 is not cut off until reverse biasedO To pro-
``' vide a short (approximately 50 microsecond) reverse bias toSCR 1 upon the fall of the TP signal, the Q21'collector
voltage rising with the fall of the TP signal is coupled
by capacit~r C6 and resistor R~3 to the inverting CP2 input
causing a positive spike thereat producing a momentary
- LOW level output turning ON transistor Qllo The momen-
tary turn ON of transistor Qll quickly diverts the
current through SCR 1 to quickly turn it OFF.
An additional feature afforded by the Figure
, 30 ~ embodiment are means to assure that the coil current
is not dropped below the peak pull-in current before
a minimum time has elapsed from the beginning of an
injector actuator signal. ~uch protection is desirable
where the combination of a higher-than-normal B+ supply
and/or a slower responding injector would otherwise
cause the coil current to exceed the peak pull-in-current
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and then drop back to the holding level before the
injectors have in fact opened. To avoid this pos-
sibility, the comparator CP2 is caused to maintain a
LOW level output for at least a minimum period such
as 1.5 milliseconds after the beginning of an injector
actuation command. This minimum period is generated
by timing means in the form of a one-shot monostable
comprising a resistor R50 coupling B~ conductor 30 to
a node N between a capacitor C7 and a diode D6 connected
in series between the collector of flip-flop transistor
Q20 and the base of an NPN transistor Q25~ The Q25
collector is coupled by a resistor R51 to B+ conductor
30 and by a resistor R52 to the base of an NPN transistor
Q26, the em tter of which.is grounded at 32. The Q26 col-
lector is cou~led to the output of comparator CP2 by
capacitor C4 and to the node between diodes D2 and D3
connected in series between ground 32 and the base
of flip-flop transistor Ql9.
.As will be discused more fully shortly, the
beginning of an injector actuat:ion signal TP causes
flip-flop transistor Q20 to ground one side of capacitor
C7. This momentarily biases OFF one-shot transistor
Q25 and biases ON transistor Q26 to ground the CP2 out-
put. If, auring the short l.S millisecond period that
one-shot capacitor C7 charges through resistor R50,
the sense voltage at the non-inverting CP2 input exceeds
the reference voltage at the inverting CP2 input, the
resulting HIGH level output o comparator CP2 would be
diverted to ground through transistor Q26 and would not
be communicated to the Ql9 base until after the short
one-shot period~ Thus inhibited for the one-shot period,
transistor Ql9 would therefore not bias OFF transistor Q18
to downshift the reference voltage to the non inverting
comparator CP2 until both the one-shot period had elapsed
and the coil current exceeded the peak opening current
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level.
In operation o~ the Figure 4 embodiment, prior
to the receipt of terminal C of an injector actuation
signal TP, all transistors are biased OFF with the
exception of flip-flop transistor Ql9, the one-shot
transistor Q25 and transistor Q12 which is biased ON
by the Q22 collector voltage TP* thereby grounding the
SCR 1 gate. With the Q24 base coupled to B+ conductor 30
by resistors R44 and R46, transistor Q24 is biased OFF
so that the inverting input of comparator CP2 is clamped
to ground 32 by resistors R42 and R43. The non-inverting
CP2 input there~ore controls to produce a HIGH CP2 output
biasing OFF power transistor Qll through control
transistors Q13 and Q14~ No current therefore flows
through Coil Ll.
- With the presence of an injector actuation
: pulse TP, the rise thereof is communicated to the Q21
and Q22 bases by resistors R40 and R41, biasing ON
transistors Q21 and Q22. The Q12 base is grounded
through transistors Q22 and resistor R45 to bias OFF
transistor Q12, thereby enabling SCR 1 by removing the
ground to the SCR 1 gate. The rise of the TP actuation
signal is also communicated by capacitor C5 and diode
D4 to the base of flip-flop transistor Q20, biasing
ON transistor Q20 and therethrough grounding the bias
to flip-flop transistor Ql9. The B+ voltage at the Ql9
:~ collector biases ON shunt transistor Q18 to reverse bias
: . current source transistor Q15 and through transistor
Q18 and resistor R35 the current that would otherwise
flow through current souxce transistor Q15~
With its base now grounded through resistor
R46 and transistor Q22, transistor Q24 is biased ON
- to communicate the breakdown voltage of Zener ~4 to
the inverting input of comparator CP2. With no voltage
: drop.develop:ed. across resistor R29 until current begins
to flow through coil Ll~ the Zener breakdown voltage at
the inverting CP2 input produces a LOW level output of
comparator CP2 biasing ON control transistor Q13 and Q14
to develop a voltage across resistor R22 biasing ON power
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transistor Qll.
As the coil current begins to build up, the
voltage across sensing resistor Rl increases lowering
the potential at coil side B. The Q16-Q17 current
source-emitter follower circuit causes the voltage drop
across resistors R32-R33 to correspond closely to that
across sensing resistor Rl. Thus, increasing with the
current through sensing resistor Rl, coil Ll, and tran-
sistor Qll, the Q16 current at node J develops a voltage
across resistor R29 increasing with that across sensing
resistor Rl.
Variable resistor R33 was previously set so
that the QlG current caused the voltage across resistor
R29 to just exceed the breakdown voltage of Zener ~4
at a value of coil current slightly in excess of the
solenoid pull-in current Ip under worst-case conditions.
Representative switching points may be computing as-
suming representative circuit values shown in the Table
- o~ Component Values below. That is, a minimum pull-in cur~ !
rent Ip of 12 amps, a low level of hold current IHL
o~ 3.2 amps, and HIGH and LOW level CP2 outputs
corresponding respectively to 14 volts and zero volts.
Then, when the CP2 output is LOW, as is ini-
tiall~ the case until the 12 amp pull-in current is
exceeded, the sense voltage at the non-inverting CP2
- input is that developed across resistor R29 multiplied
by the 0.9 ratio of R27/R27 + R28 (i.e., 0.9 = 100/
(100`+ 10). Thus, to exceed the 4.3 volts re~erence
at the inverting CP2 input, about 4.8 volts must be
developed across the 3.6K resistance of R29, requiring
`~ about 1.33 milliamps thereacross. To provide this 1.33
milliamps current when the minimum pull-in current o~
12 amps is just exceeded, resistors R32-R33 must be
set to develop the same-1.2 volts developed across sense
resistor Rl. To develop 1.2 volts with 1.33 milliamps,
` the resistance of resistors R32 and R33 must be set at
` 900 ohms.
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If more than the 1.5 millisecond delay of
the one-shot has elapsed when the 4.3 volt reference
at the inverting input of comparator CP2 is exceeded
for the first time, the resulting 14 volt HIGH output
is communicated by capacitor C4 and diode D2 to the
base of flip-flop transistor Ql9, thereby biasing ON
. transistor Ql9 and therethrough biasing OFF shunt
. transistor Q18. The voltage at the non-inverting sense
; input of comparator CP2 is now the voltage developed
across resistor R29 the 14 volt CP2 output less the voltage
developed across resistor R29 multiplied by 0.9. Thus,
to fall below the 4.3 volt reference at the inverting
CP2 reference input when the CP2 output is 14 volts~
. about 3.3 volts must be de~eloped across R29, requiring
about 0.9 milliamps therethrough. These 0.9 milliamps
are provided through current sources transistors Q15 and
Q16 in developing a 0.32 volt drop across resistors R30-
R31 and R32-R33, corresponding to the lower hold level
current of about 3.2 amps. This requires that the
resistors R30-31-32-33 provide an èquivalent resistance
of about 350 ohms, meaning that R30 and R31 be about
570 ohms since resistors R32 and R33 were previously
~ set at: 900 ohms O
: TABLE OF REPRESENTATIVE VALUES
The following is a table of representative
values and designations of components that may be used
to embody the cireu1ts 111ustrated ln Figures 2 and 4
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TABLE OF COMPON~NTS
RESISTORS (Ohms) Zener (Breakdown Voltage)
Rl 0 1 R21 4.7K R41 10K ~1 S.6 IN4734A
5 R2 10K R22 100 R42 10K ~2 27 IN4750A
R3 10K R23 58 R43 10K ~3 43 IN4755A
R4 560 R24 2.2K R44 6.8K ~4 4.3 IN4731A
R5 10K R25 4.7K R45 10K
R6 33 R26 l.lK R46 22K Capacitors (Microfarads)
R7 50K R27 100K R47 2.2
R8 2.2K R28 lOK R48 22K Cl 0.001
R9 6.8K R29 3.6K R49 C2 0.0068
10 R10 10K R30 330 R50 50K C3 0.0068
C4 0.05
Rll 100K R31 0-lK R51 10K C5 0.05
R12 6.8K R32 330 R52 10K C6 0.05
R13 560 R33 0-lK C7 0.05
R14 lK R34 3.3K
R15 5K R35 2.2K Transistors and Diodes
15 R16 10K R36 22K All NPNs - MPSA05-Motorola
R17 10K R37 4.7K All PNPs - MPSA55-Motorola
R18 0-lK R38 4.7K Qll 2N6387-RCA
Rl9 33K R39 22K SCR 1 - C122D - G.E.
R20 R40 10K Diode Dl - MR754 - Motorola
2C Other Diodes IN4004
CONCLUSION
Having described several embodiments of the
invention, it is understood that the specific terms
and examples are employed herein in a descriptive sense
only and not for the purpose of limitation. Other em-
bodiments of the invention, modification thereof, and
alternatives thereof will be obvious to those skilled
in the art and may be made without departing from our
invention. We therefore aim in the appended claims to
cover the modifications and changes as we would in the
true scope and spirit of our invention.
WHAT WE CLAIM IS:
