Note: Descriptions are shown in the official language in which they were submitted.
t~ RD-11045
POWER SUPPL~ FOR MAGNETRON AND
THE LIKE LOADS
Background of the Invention
~ he present invention concerns power supplies
and, more particularly, a self-resonant power supply
of the flyback type which does not require a
high-voltage rectifier for supplying operating
energy to a microwave oven magnetron and the like
loads.
Magnetron microwave generators are becoming
more widely used in food preparation appliances,
such as microwave ovens and the like. The power
supply utilized in many presently available microwave
ovens typically utilizes a high-reactance voltage
step-up transformer and a voltage doubler. Typically,
a capacitance is in series between the transformer
secondary winding and the load, and a voltage-
doubling diode is across the anode-cathode circuit
of the magnetron to provide a voltage-doubled,
half-wave current supply for the ma~netron. A
rectified sinewave portion of operating current is
applied to the magnetron at a repetition rate
equal to the line frequency, e.g. 60 Hertz (Hz.~.
These relatively-low-frequency power supplies are of
relatively great weight and require addi-tional
structural strength in the microwave appliance to
; protect against physical damage during shipment and
use. Additionally, the typical magnetron power
supply is o relatively great manufacturing cost.
; It is desirable to not only reduce the cost and weight
of the magnetron power supply, but also to more
easily control the amount of energy being supplied
to the miGrowave-power-generating magnetron to pxovide
:
RD-11015
l 1~2~3
great~r control of the food preparation sequences
achievable therewith.
Brief Summary of the I_vention
In accordance with the invention, a power
supply for energizing a magnetron and the like
self-recti~ying load, through which a current flows
only when a predetermined minimum voltage of a
single polarity is exceeded thereacross, comprises
~ a transformer having a primary winding in series
; 10 between a controlled-current-path through a switching
device, and a source of operating voltage, as may be
provided by rectifying the power line voltage and
the like. A secondary winding of the transformer
connects across the load. An electrical capacitance
across the inductance of the primary winding is of
a magnitude sufficient to resonate the transformer
winding at a frequency greater than the frequency of
a driving signal applied to a controlling element
of the switching device. A device having
unidirectional curLent-flow characteristics is
connected in parallel with the controlled-current-
flow circuit of the switching device to prevent
application of voltages of improper polarity across
the switching device during half-cycles of oscillatory
voltage, present at the primary winding of the
transformer due to the resonance effect. The
frequency and/or duty cycle of the controlling signal,
to the controlled switching device, is varied to
vary the amount of current drawn by the load device
such as a magnetron and -the like, and thus control
the amount of power consumed (and the microwave
power generated -thereby).
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~ 1~29~3
In one presently preferred embodiment, the
resonating eapacitanee is provided by filament bypass
capaeitanee eoupled from the magnetron filament~
itself connected to an end of the high voltage
secondary of the transformer, to eleetrical ground
potential. In other presently preferred
embodiments, a resonating capacitance is connected in
parallel with the controlled-current-cirueuit af the
switehing device and is of magnitude selected to
to resonate with the induetance appearing at the
primary winding of the high voltage transformer;
the total eapaeitanee across the winding induetanee
may be the sum of the resonating eapacitance
aeross the primary winding and the load eapaeitance
reflected from the secondary winding baek to the
primary winding.
Accordingly, it is an object of the present
invention to provide a resonant power supply for
energizing a load eonsumer power only if a minimum
voltage thereacross is exeeeded.
This and other objeets of the present invention
will beeome apparent upon consideration of the following
detailed deseription, when taken in eonjunetion with
the drawings.
Brief Deseription of the Drawin~
Figure 1 is a sehematie diagram of a
mierowave oven magnetron and of a power supply,
therefor, in aeeordanee with the prineiples of the
present invention.
Figure 2 is a sehematie diagram of the equivalent
eireuit of a portion of the power supply of Figure 1
-- 3 --
~ 1~2,9~3 RD-11045
and useful in understanding the operation thereof~
Figure 3 is a set of interrelated current and
voltage waveforms Erom the simplified circuit of Figure 2,
and useful in understanding operational principles of the
present invention;
Figure 4 is a schematic diagram of a base drive
circuit suitable for use in the power supply of Figure l; and
Figure 5 (located on the first sheet of drawings)
` is a second presently preferred embodiment of a power supply
for supplying operational power to a microwave oven magnetron
and similar loads.
Detailed Description of the Invention
Referring initially to Figure l, a power supply
lO for energizing a load, such as microwave oven
magnetron'll and the like, provides a voltage
VM across the load magnetron,, with positive polarity
at a magnetron anode lla coupled to electrical ground
potential, and with negative polarity at one of a
pair of leads llb of a magnetron filament llc
serving to heat a magnetron cathode lld for emission
of electrons therefrom. The magnetron filament leads
llb are connected to the ends of a secondary winding
12a of a filament transformer 12 having its primary
winding 12b connected to the power line voltage,
typically on the order of 115 volts A.C. at 60 Hz.
As is well known, upon application of an electrical
potential between magnetron anode lla and magnetron
cathode lld, of a magnitude greater than some minimum
potential, t~pically on the order of three to four
kilovolts (kV.), the magnetron draws anode current
IM and produces microwave power which is output
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~ ~6~3
from the generator 11 to, inter alia, cook food and
the like in a microwave oven and the like. Presently,
a typical power supply for supplying operating current
to magnetron 11 would include a 60 Hz. high-voltage
step-up transformer having a high-voltage capacitor
in serie~ between one end of the high-voltage secondary
winding and the magnetron, and a high-voltage doubler
diode in parallel with the magnetron. This form of
power supply (not shown) typically operates at the
60 Hz. line frequency and is characterized by a
relatively heavy and expensive transformer, as well
as relatively expensive high-voltage capacitor and
diode components.
Power supply 10 operates at a frequency
typically two to three orders of magnitude greater
;
than the line requency, e.g. between about 20 kHz.
and about 100 kHz., whereby the weight of a transformer 14,
utilized for voltage step-up purposes, is reduced.
Power supply 10 does not require either a high-voltage,
voltage-doubler capacitor or a high-voltage diode.
` ~ A primary winding 14a of high-voltage transformer 14
is connected between a source of voltage od magnitude
-ttV and the controlled-current-flow circuit of a
switching device 15. The operating voltage of
magnitude tty may be obtained by rectification of the
115 volts 60 Hz. line voltage and may thus be of
magnitude on the order of 170 volts D.C. peak. In
my preferred embodiment, control]ed switching device
15 is a transistor having: a collector electrode 15a
coupled to the remaining end of transformer primary
winding 14a; an emitter electrode 15b coupled to
electrical ground potential; and a base electrode 15c,
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"
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~ ~29~3
into which a flow of current controls the current
flowing through the collector-emitter circuit of
transistor 15, and hence through primary winding l~a,
during at least a portion of a power supply cycle.
A base drive circuit 17 receives one, or more,
operating potentials (+V) of relatively low magnitude,
on the order of 5-15 volts D.C., -to provide an output
17a coupled between the base and emitter electrodes
of switching device 15 for providing the current
controlling signal thereto. The input 17b of the base
drive circuit receives a relatively low-power signal
serving to establish the timing characteristics of
the base electrode drive to the switching device and
therefor the magnitude of microwave produced by
magnetron generator 11.
A clamping diode 19 has its cathode electrode
connected to the junction between primary winding 14a
and switching device collector electrode 15a, and has
its anode electrode connected to switching device .
emitter electrode 15b and electrical ground 20. A
snubbing network 21, comprised of a resistance 22 in
series of an electrical capacitance 23, is connected
in electrical parallel across diode 19 and the
controlled-current-flow circuit (from collector 15a
to emitter 15b of device 15).
In the embodiment of Figure 1, magnetron
filament leads llb are coupled through electromagnetic-
interference-suppressing bypass capacitors Cfl and
Cf2, each having a capacitance chosen to provide a
total capacitance C across secondary winding 14b to
resonate the secondary winding 14b at a resonant
frequency greater than the operating frequency, which
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I :~62g~3
is between about 20 kHz. and about 100 kHz. Transformer
14 is a voltage step-up transformer having Nl primary
winding turns and N2 secondary winding turns,
where N2 is greater than Nl.
Referring now to Figures 1, 2 and 3,
operation of the high voltage portion of power supply
10 may be better understood by considering the
equivalent circuit (Figure 2) of the load magnetron,
as reflected to the primary winding side of transformer 14.
~ 10 As previously mentioned hereinabove, current flows
: through the magnetron only if the magnetron anode is
positive with respect to the magnetron cathode and only
if the voltage from anode to cathode of the magnetron
exceeds some minimum voltage. Thus, the magnetron
~; appears to be a series circuit including a diode lle
having its anode connected to the magnetron anode and
having its cathode connected to the anode of a high-
voltage zener diode llf, of zener voltage equal to
~: the minimum magnetron voltage VMagl, and having its
cathode connected to the cathode of the magnetron.
The magnetron circuit capacitance appears from anode
to cathode of the magnetron and in parallel with the
series diode-zenar diode circuit. When reflected
~rom the secondary winding 14b to primary winding 14a,
the magnetron.equivalent circuit appears as an
equivalent capacitance C' in parallel with the magnetron
~ ~ diode circuit, including series diode lle and series
.~ zener diode 11f, all in electrical parallel connection
:~ with the mutual inductance LM f the transformer. The
; 30 magnitude C' of the reflected resonating capacitor is
equal to the resonating capacitance C times the
square of ~he ratio of turns of the secondary winding
: - 7 -
;
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~ ~2~3 2
to the turns of the primary winding, i.e. C' = C(Nl/N2) .
The reflected minimum magnetron zener voltage VMag'
is equal to the minimum magnetron voltage VMag
times the step-down ratio of the turns of thé primary
winding to the turns of the secondary winding
Mag VMag (Nl/N2). ~ primary winding self-
inductance Lp (of magnitude much smaller than the
mutual inductance LM) is in series between the
mutual inductance of the primary winding and the
operating voltage potential ++V at terminal 27. The
opposite end of mutual inductance LM is connected
to the collector of transistor 15, at which point
are connected the cathode of diode 19 and one end
of snubber network 21. The remaining end of
network 21, diode 10 and the emitter electrode 15b
of switching device 15 are connected to the common
(or ground potential) texminal 20.
Operation of the resonant fly-back power
supply is as follows: ~ssume initially that the
minimum magnetron voltage VMag is equal to
~: Vin(N2/Nl); therefor VMag = Vin- Prior to some time
to, capacitance C' has been charged to the input
voltage Vin. The voltage Vc' is equal to Vin
(Figure 3, waveform e)~ There is no current I2
flowing through the mutual inductance LM (Figure 3,
wave~orm ~). When base drive circuit 17 suppl]es a
current signal of sufficient magnitude into base
electrode 15c, transistor 15 is driven into the
heavily-conducting condition, whereby a substantially
short circuit appears between collector electrode
15a and emitter electrode 15b. In this condition,
substantially the entire operating v`oltage Vin
- 8 -
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2~3
appears across the transformer primary winding 14a.
Input current lin flows from opexating potential
terminal 27 sequentially through the primary winding
inductance Lp and the mutual inductance LM. The
voltage across the mutual inductance is such that the
magnetron is reverse-biased and therefore diode lle
is also reverse-biased, whereby the magnetron current
IM' (being the magnetron current IM reflected through
the transformer to primary winding 14a) is substantially
zero, as the magnetron does not conduct. The current
Il (Figure 3, waveform a) increases linearly, between
time to (when switching device 15 is initially placed
in the on condition), and time tl (when device 15 is
turned off, as by removing drive to base electrode
1 c). During the same time interval, the current
I2 through magnetizing inductance LM is also linearly
increasing and is of magnitude substantially iden-tical
to the current Il flowing into switching device
collector electrode 15a, as the current Ic' (Figure 3,
waveform c) flowing through the reflected resonating
capacitance C' is substantially zero and the reverse-
biased magnetron current IM' (Figure 3, waveform d)
is also substantially zero. The shape of the I2
waveform obtains from the condition that (dI2/dt = Vin/LM).
During the time interval to-tl, the voltage Vc',
across reflected resonating capacitor C', remains
substantially equal to the operating potential Vin
(waveform e of Figure 3), while, as previously
mentioned hereinabove, the collector voltage V~
(waveform f of Figure 3) across switching device 15
is substantially equal to zero volts.
g _
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~ ~62~3
At time tl, switching device 15 turns off
and the energy stored in mutual inductance LM is
transferred to the secondary winding. Current Il falls
to zero, as device 15 is now in the open, or non
conducting, condition. Current ~2~ through mutual
inductance LM, cannot abruptly change~ As the
magnetron load is not conducting, the mutual
inductance current I2 must flow as capacitance
current Ic, into the equivalent reflected capacitance
C'. Thus, at time tlj current Ic' suddenly jumps from
an essentially zero current flow to a peak current flow
proportional to the value of the current I2 flowing
in the mutual inductance immediately prior to the
turning-off of transistor 15. The current Ic' flows
into capacitance Cl and sinusoidally charges the
capacitance toward a peak voltage Vc' (Peak) =
LMI2 (peak)/C' (Figure 3, waveform e). The
capacitor voltage Vc' continues to charge in the
negative direction until this voltage, which is also
the voltage across reflected magnetron cathode-anode
circuit, reaches the equivalent minimum voltage
(~VMag') at the magnetron cathode. Magnetron diode lle
now conducts and a flow of magnetron current IM'
commences. At this time t2' essentially all of the
mutual inductance current I2 is drawn by the magnetron
load, and the capacitance charging current Ic' (Figure 3,
waveform d) falls to zero. The decreasing magnitude
of mutual inductance current ~2 (Figure 3, waveform b)
is the decreasing magnitude of the magnetron load
current IM and, during the time interval from time t2
(when the magnetron begins to conduct) until a time
t3 (when the magnetron current reaches zero) is a
- 10 -
RD-11045
substantially linearly decreasing current given by
the condition (dIM'/dt = VMay'/LM). During the time
interval t2-t3, the voltage Vc' across the equivalent
capacitance is held essentially at the magnetron
equivalent zener voltage, which, as previously stated
hereinabove, is equal in magnitude to the magnitude
of the input voltage Vin. The voltage VA across the
open collector-emitter circuit of transistor 15 is
equal to the sum of the input voltage Vin plus the
equivalent capacitance voltage Vc'. Vc' is now equal
in magnitude to the input voltage Vin, by the initial
assumption for the turns ratio of the trans~ormer.
Thus, the maximum collector-emitter voltage which
device 15 must sustain is equal to twice the supply
potential (Vin), as seen in Figure 3, waveform f.
At time t3, the energy stored in mutual
inductance LM falls to zero, whereby the mutual
inductance current I2 is equal to zero. The magnetron
current IM' also is essentially of zero magnitude and
the magnetron ceases conduction. However, the
voltage Vc' across equivalent capacitance C' is still
equal to the equivalent magnetron load voltage ~VMag',
whereby the equivalent capacitance now pumps stored
charge back into the mutual inductance, with the
mutual inductance current I2 having a negative
polarity (current flow in the direction opposite to
the direction of arrow I2). The equivalent capacitance
voltage Vc' rises toward zero volts, and, as the
current I2 now flowing through the mutual inductance
must remain continuous, the effective capacitance and
mutual inductance "ring". The effective capacitance C'
begins to sinusoidally charge toward a voltage Vc'
;
-- 11 --
9 ~ 3 RD-11045
equal to the input voltage Vi . The device collector-
emitter voltage VA (which is equal to the input
voltage Vin minus the capacitance voltage Vc') falls
until, at time t4, voltage VA is equal to zero volts
(when capacitance voltage Vc' is equal to +Vin volts).
At time t4, all of the currents Il,I2,IM' and Ic' are
essentially equal to zero, while the equivalent
capacitance voltage Vc' is equal to the input voltage
Vi and the device collector-emitter voltage VA is
essentially zero volts. Switching device 15 is
again turned on at time t4 (see Il and I2 curves
(broken lines 30-31)) to restart the cycle, at which
time the capacitance voltage Vc' is indeed equal to
the input voltage Vinl, as was the initial assumption.
The cycles are repeated for as long a time interval
as magnetron ouput is desired.
In the event that magnetron 11 does not draw
current after time tl, as might be caused by lack of
power to filament llc, the current stored in the
magnetizing inductance is pumped into the equivalent
` capacitance C' and a sinusoidal oscillation starts.
However, as soon as the voltage Vc' across the
equivalent capacitance reaches a value~equal to -Vin~,
. j
i catching diode 19 is forward biased and draws current
from the mutual inductance-effective capacitance
circuit to ground potential, at terminal 20, until
the effective capacitance voltage returns to a value
of +Vin and another cycle of power supply may commence.
If the load magnetron conducts during a subsequent
power supply cycle, operation is as described hereinabove,
while if the magnetron still does not conduct, the
beginning of an oscillatory current condition again
12 -
RD-110~5
2 ~ ~ 3
occurs and catching diode 19 again conducts until
the capacitance voltage is equal to the input voltage.
It will be seen that: transformer 1~ should
have a relatively low primary leakage inductance Lp
to avoid potentially destructive high-voltage spikes
from appearing across the power transistor at
turn-off; catching diode 19 functions to protect
the collector-emitter circuit of device 15 from
application of negative polarity voltages VA thereacross;
and snubber circuit 21 protects against high positive
voltage spikes at the transistor collector when the
transistor is turned off. Similarly, the required
peak voltage and current ratings of switching
device 15 are determined from the peak magnetron
current, operating potential and transistor turn-on
and turn-off characteristics. In the event of failure
of the magnetron to conduct, switching device 15
is protected from negative collector voltages, with
respect to the emitter electrode 15b thereof, by
means of diode 19, which would then be forward
biased and would be rated to conduct a peak
faultcurrent equal to the maximum current Il conducted
by switching device 15 during the energy-storage portion
of the cycle, e.g., in the time interval to-tl. It
will also be seen that the magnetron output power
can be varied by varying the time interval to ~ tl,
to control the magnitude of I2 (peak) in mutual
inductance LM, and therefore the peak magnitude of IM.
A larger or shorter time interval to-tl will increase
or decrease, respectively, the amplitude of I2
(peak) and the peak value of IM.
- 13 -
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~ ~2~3
In the foregoing, illustrated as MODE 1
of power supply operation, the reflected magnetron
voltage VMag' (being the magnetron voltage VMag
reflected back through the transformer to the primary
winding and therefore equal to the magnetron voltage
: divided by the turns ratio of the transformer) is
: assumed essentially equal to the magnitude Vin of
the operating voltage. However, all magnetrons will
not have essentially identical operating voltages VMag,
; 10 at which voltage current conduction therethrough
occurs. It is desirable to have a single power
supply configuration which can be utiliæed without
adjustment in conjunction with loads having somewhat
greater, or somewhat lesser, voltages at which
current conduction occurs. In MODE 2, and MODE 3,
: respectlvely, the minimum load conduction voltages are
respectively greater thanl and less than, the
nominal design value, i.e., the MODE 1 value, and
accordingly the value of equivalent magnetron
.. : 20 zener voltage VMag', as reflected through transformer 14
to the primary winding thereof, is respectively
greater than, and less than, the value of operating
voltage Vin.
In MODE 2, the same initial conditions
(capacitance C' charged to -Vin volts and mutual
inductance current I2 equal zero) are assumed as
in MODE 1. The portion of each cycle from the initial
; transistor 15 turn-on time t5 to cessation-of-magnetron- :
current-flow time t8 occurs in substantially the same
manner as the operation during the time interval
to-t3 of MODE 1. The only difference is that, due
to the greater negative voltage (-Vhi) needed to begin
: '
~ ::
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i ~2~3
magnetron current conduction, the MODE 2 time
interval between t6, when transistor 15 turns off,
and time t7, when the magnetron voltage has reached
the minimum conduction voltage and current flow can
begin, is somewhat longer than the associated tl-t2
time interval in the MODE 1 case. In MODE 2, the
charge stored .in equivalent capacitance C' causes the
voltage Vc' thereacross to ring from the negative
voltage -Vhi to the input voltage Vin, starting at
time t8 after the magnetron turns off. At time tg/
the voltage Vc' across the capacitor has become equal
to the supply voltage Vin and forward biases catching
diode 19. The catching diode 19 must conduct over
the time interval from time tg to time tlo to allow
the remaining current stored in mutual inductance I~
to flow through the now forward-biased catching diode 19
as current Il of negative magnitude (area 33 of
waveform a of Figure 3). Thus, during the time
interval from time tg to tlo, the current I2 in mutual
; 20 inductance LM will decrease linearly with time, with
a slope given by the condition (dI2/dt = Vin/LM).
There will again be no current flow in the mutual
: inductance at time tlo. The effective capacitance C'
will again be charged to the input voltage Vin, and
the transistor 15 can again be turned on, at time tlo,
to initiate another cycle of the flyback power supply.
In both MODE 1 and MODE 2 the initial conditions are
identical and switching device 15 is very lightly
~` stressed as the device turns on with zero circuit
current flow therethrough and turns off with essential
zero voltage thereacross; only the energy in the
transformer leakage inductance must be dissipated by
- 15 -
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~ ~ 6~
switching device 15, catching diode 19 and snubber
circuit 21.
In MODE 3, the switching device 15 is turned
on at the beginning of a cycle at time tlo. A zero
magnitude current flow exists at tlo Eor currents
Il, I2, IM' and Ic'; the ef~ective capacitance
voltage Vc' is equal to the input voltage Vin, and,
therefore, the voltage VA across the switching
device has an essentially zero magnitude. The
mutual inductance current I2 increases linearly until
de~ice 15 is turned off at time tll, when a pulse
of capacitance current Ic' occurs, as the voltage Vc'
across the capacitance falls. At time tl2 the voltage
across the magnetron reaches the magnetron conduction
voltage, which for the low-voltage case is the negative
voltage -V1O. It should be understood that the time
interval between tll and time tl2 is somewhat less
than the time interval between time tl and time t2 in
the MODE 1 case, as the voltage must reach a smaller
negati~e magnitude. In the time interval from time
tl2 to time tl3, magnetron current IM' flows and, as
in the MODE 1 and MODE 2 cases, is a linearly
decreasing current ramp, reaching essentially zero
magnitude at time tl3. At time tl3, the charge stored
in -the equivalent capacitance ini~iates the resonance
oscillation, and the voltage acrosss the capacitance
; starts to rise. However, since the negative magnitude
of capacitance voltage Vc' was less negative than in
the MODE 1 and MODE 2 cases, there is not sufficient
energy stored in the equivalent capacitance C'
to ring the capacitance voltage Vc' back to the
,
~ - 16
~ RD-11045
~ ~2~3
positive input voltage +Vin to assume priox initial
conditions by a time tl4 when all the stored charge
has flowed from equivalent capacitance C', and at
which time a turn-on signal drives switching device 15
in-to saturation. Thus, at time tl4, the voltage VA
between the collector and emitter electrodes of the
device is a non-zero voltage VA' and the switching
device must conduct an initial spike 34 of current I
of magnitude sufficient to recharge the effective
capacitance C' to the initial condition wherein
the voltage Vc' thereacross is raised, as at 34a in
waveform e of Figure 3, to the equal magnitude to
the input voltage Vin to begin the next flyback
power supply cycle. This large switching-device
current pulse increases the switching losses and
indicates that the minimum load magnetron voltage V1O,
of MODE 3, should be the equivalent magnetron voltage
utilized in designing the turns ratio of transformer 14,
whereby all magnetrons in a production run would have
at least that minimum voltage and the flyback power
supply will always operate in one of MODEs 1 and 2,
and not in MODE 3.
Referring now to Figure 4, a presently
preferred embodiment of base drive circuit 17 is
illustrated. First and second sources of potential
~` supply a positive potential rail 40 with a voltage
of positive polarity and magnitude +V, while a
negative potential rail 41 is supplied with an
operating potential of negative polarity and another
operating potential -V', which may be of the same or
different magnitude as the magnitude V of the
voltage on positive supply rail 40. One terminal 17a'
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~ ~2~3
of base drive circuit output 17a is connected to
ground potential connection 20 at the emitter electrode
15b of the control transistor and to a ground po-tential
bus 42, wlthin base drive circuit 17. The base drive
circuit 'output terminal 17a'' connected to switching
device base electrode 15c is connected through a
resistance Rl to negative supply bus 41. A first
diode string Dl, which in the illustrated embodiment
consists o~ six series-connected diodes, is connected
between the two terminals of base drive circuit output 17a.
A capacitor C2 is connected between positive potential
bus 40 and ground bus 42, while another capacitor C3
is connected between the ground bus and the negative
potential bus 41, with each of the capacitors C2 and C3
providing a low-impedance, energy-storage filter for
the positive and negative supplies, respectively.
A driver transister 45 has its collector
electrode 45a connected to positive supply bus 40 and
has its emitter electrode 45b coupled to the controllable
switching device base electrode 15c, via the parallel-
connec-ted combination of a resistance R2 and a speed-up
capacitor C4. The base electrode 45c of driver
transistor 45 is connected via a second string of
diodes D2, comprised of five diodes in the presently
illustrated embodiment, to ground bus 42. Base
electrode 45c is also connected to the collector
electrode 47a of a first current-source transistor 47,
having its emitter electrode 47b connected to positive
supply bus 40 via the parallel combination o an
emitter resistor R3 and an emitter capacitor C5. The
base electrode 47c of current source transister 47 is
connected to positive supply bus 40 via a third diode
- 18 -
~ 9~3 RD-11045
string D3 having a number of diodes chosen to determine
the output current of the first current source, in
conjunction with the emitter resistance R3. Base
electrode 47c is connected, via a paralleled resistance
R4 and a speed-up capacitor C6, to the collector
electrode 50a of a transistor 50. The emitter
electrode 50b is connected to negative supply bus 41
and a base electrode 50c is connected through a
resistance R6 to the collector electrode 52a of a
phototransistor 52 which forms part of an optoelectronic
coupler 54. The phototransistor collector electrode 52a
is also connected, via a collector resistor R7, to
positive supply bus 40. The emitter electrode 52b
of the phototransistor is connected to negative supply
bus 41. The optoelectronic coupler 54 also includes
a light-emitting device (LED) 56 connected acros~
base drive circuit input terminals 17b. A one-shot
m~nostable multivibrator 58 with a current limited
output is formed by transistors 60, 61, 62, 63 and 64.
The emitter electrodes 60a and 64a of NPN transistors
60 and 64, respectively, are connected to negative
supply bus 41, while the emitter electrode 61a of NPN
transistor 61 is connected to ground bus 42. The
emitter electrode 62a of PNP transistor 62 is connected
directly to positive supply bus 40, while the emitter
electrode 63a of PNP transistor 63 is connected to
positive supply bus 40 through a parallel combination
of an emitter resistance R8 and an emitter capacitance C7.
The base electrode 60b of first multivibrator
transistor 60 is coupled by a resistance Rg to the
phototransistor collector electrode 52a. The
collector electrode 60c of transistor 60 is coupled to
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a first terminal of a timin~ capacitance C8; the
remaining terminal of C8 is connected to the base
electrode 64b of transistor 64, and to a parallel
combination of a pull-down resistor Rlo and a reverse-
voltage-protection diode D4, between the base electrode
64b and the negative supply bus 41. Transistor
collector electrode 60c is also connected, via a
collector resistance Rll to positive supply bus 40,
and to the anode of a diode D5, having its cathode
connected to the base electrode 61b of transistor 61.
Base electrode 61b is connected via a pull-down
resistance R12 to ~round bus 42. The collector
electrode 61c of transistor 61 is connected via a
series pair of resistances R13 and R14 -to positive
potential bus 40. The base electrode 62b of
transistor 62 is connected to the junction between
resistances R13 and R14. The collector electrode 62c
is connected via a resistance R15 to the collector
electrode 64c of transistor 64. The junction between
transistor collector electrode 62c and resistance R15
is connected both to the base electrode 63h of a
second current source transistor 63, and to the cathode
end of another diode stack D6. The current source
- transistor collector electrode 63c is connected to a
turn-off node A which is connected to negative
potential bus 41 via a paralleled pair of series
resistance dividers, respectively comprised of resistances
; R16 and R17, and resistances R18 and Rlg. The junctions
` between resistances R16 and R17, and between resistances
R18 and Rlg, are respectively connected to the respective
base electrodes 66a and 68a of transistors 66 and 68.
The emitter electrodes 66b and 68b of respective
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RD-11045
32~3
transistor 66 and 68 are both connected to negative
supply bus 41. The collector electrode 68c of transistor
68 is connected to the junction of transistor collector
electrode 47a, transistor base electrode 45c and the
anode of diode stack D2. The collector 66c of transistor
66 is connected to the junction of resistances Rl and
R2, capacitance C4, the anode electrode of diode
stack Dl and base-drive circuit output terminal 17a''.
In operation, it is initially assumed that
there is no flow of current into LED 56 of opto-
isolator 54, whereby phototransistor 52 is in the
cut-off condition. The magnitude of resistances
R6, R7 and Rg are chosen such that, with phototransistor
52 off transistors 50 and 60 are saturated. The
saturation of transistor 50 pulls the collector
electrode 50a thereof to the negative supply bus; the
:~;
negative jump in voltage is coupled to the first
current source base electrode 47c, by the paralleled
resistance R4 and speed-up capacitor C6. Diode
stack D3 is ~orward-biased and the voltage drop
therethrough holds base electrode 47c to a voltage
~ below the positive potential on bus 40. Transistor 47
;~ conducts and the voltage across its emitter resistance
is equal to the nu~er of diodes drops in diode
stack D3 less the base-emitter voltage of transistor
47. In the illustrated embodiment, the voltage
across emitter resistance R3 is approximately two
diode voltage drops. The number of diode voltage
drops and the magnitude o~ emitter resistance R3
determine the current into transistor emitter electrode
47b. Paralleled capacitance C5 is a pulse-shaping
- 21 -
RD-11045
~ ~2~3
capacitor allowing current source transistor 47 to
not only turn on qulckly but to also have a higher
current at the current-source ou-tput (collector
electrode 47a) at the beginning of the conduction
period of current source transistor 47. The current
from source transistor 47 turns on driver transistor
45 and current at the emitter electrode 45b thereof is
supplied to the base electrode 15c of the controlled
switching device 15, via base-current-determining
resistor R2 and its paralleled speed-up capacitance C4.
The magnitude of current flowing into base electrode 15c
~ is sufficient to place device 15 in the highly-
; conducting condition, as at time to of Figure 3.
As previously mentioned hereinabove,
- transistor 60 was also placed in the saturated condition
by the removal of current flow through LED 56, whereby
the voltage at collector electrode 60c is pulled down
to the negative voltage on negative supply bus 41.
;~ Transistors 61, 62, 63 and 64 are turned off, whereby
current does not flow into node A and transistors
66 and 68 remain in the off condition.
At time tl, when switching device 15 is to
be turned off and the resonant fly-back action of
the power supply begun (to cause a pulse of current
to flow to the load magnetron) a current pulse is
introduced at base drive input terminals 17b and through
LED 56. The resulting pulse of light is coupled to
phototransistor 52, which saturates. The collector
electrode 52a is pulled down to the negative voltage
(-V') at negative supply rail 41. Transistor 50 is
turned off, turning off current source transistor 47,
which in turn turns off transistor 45. Resistor Rl,
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RD-11045
~ 1~29~
connected between base electrode 15c of the
switching transistor and the negative supply bus,
draws some of the switching transistor 15 stored
charge therefrom to commence turning-off of the
switching transistor.
It is desirable to rapidly turn switching
transistor 15 off, to prevent excessive energy
dissipation therein.
Therefore, the appearance of negative
voltage at phototransistor collector 52a, at time tl,
turns off multivibrator input transistor 60. The
voltage across timing capacitor C8 was previously
about zero volts and cannot change instantaneously.
Accordingly, the voltage at input transistor collector
electrode 60c and the base electrode 64b of transistor
64, both ]ump to a positive ~oltage. Simultaneously,
collector electrode 64c is switched to the negative
supply bus operating potential oE -V' ~volts, current
flows through resistance R15 to forward-bias diode
stack D~, and turns on current source transistor 63.
The magnitude o~ current flowing into node A, from
current-source transistor collector electrode 63c,
is determined by the number of forward-biased diocles
in diode stack D, and the magnitude of emitter
resistance R8. Pulse-shaping capacitor C7 acts to
provide rapid turn-on of the current source, as well
as to provide a higher-current at the beginning of
the conduction period of transistor 63. The flow of
current into node A turns on both of transistors 66
; 30 and 68. Transistors 68 and 66 respectively pull the
base electrode 45c of driver transistor 45, and the
base electrode 15c of the switching transistor 15,
- 23 -
RD-11045
9 ~ 3
both to the negatlve supply voltage on rail 41,
whereby driver transistor 45 ceases pumping charge into
the switching transistor and the charge stored in the
base circult of the switching transistor is rapidly
removed therefrom by transistor 66. Th.us, essentially,
as soon as a pulse of current is received by LED 56,
transistor 15 i5 turned off in as rapid a manner as
possible, initiating the resonant fly-back action of
the power supply.
As previously mentioned, upon receipt of a
current pulse by LED 56, transistor 60 is cut off and
the collector electrode voltage thereof jumps to a
voltage more positive than the negative voltage on
negative supply bus 41. The voltage across timing
~`~ capacitor C8 was initially zero volts, but gradually
increases toward +V volts. When series diode D5 is
. forward-biased transistor 61 is turned on, turning off
transistor 63. Thereafter, transistor 64 is eventually
brought out of conduction at some time after the turn-off
- 20 time tl at which the turn-off pul.se appeared at the
photocoupler. When transistor 64 reaches cut-off,
current flow through resistance Rl5 ceases and
current-source transistor 63 is cut-off, rèmoving
drive at node A and returning translstors 66 a d 68
to the cut-off condition. The time interval, established
by the magnitude of timing capacitor C8, during which
current source transistor 63 operates, is selected
to achieve rapid turn-off of switching device 15. At
the cessation of operation of current-source transistor 63
the reverse current flow through the base-emitter
junction of switching device 15, caused by the connection
of resistance Rl between the negative potential bus ~1
- 24 -
RD-11045
and switching device base electrode 15c, is sufficient
to keep switching transistor 15 in the cut off
condition, until the current flowing through LED is
again removed to supply a turn-on current to the
switching transistor base electrode, at the start of
the next cycle o~ the fly-back power supply
(e.g., at time t5 oE Figure 3).
Referring now to Figure 5, wherein like
elements have like reference designations, in the event
that the capacitance C' across transf~rmer secondary
winding 14b is insufficient to reflect back as an
equivalent capacitance resonating with the mutual
inductance across primary winding 14a of the transformer,
an additional resonating capacitance 80 is utilized
across the controlled-curren~ collector-emitter
electrode circuit of switching device 15. Thus, the
resonating capacitance can be: the equivalent
capacitance C' reflected from the primary winding from
the secondary winding (including the load); capacitance
80 effectively across the primary winding; or a
combination o~ the two. Switching device 15' may be
; the transistor of the Figure 1 embodiment, having a
controlled-current path controlled by the current flow
at the output 17a' of drive circuit 17', or may be
a device, such as a gate-turn-off switch, power FET
and the like, which has current ~low in the path in
series with the transformer primary winding controlled
by a voltage applied to a device input by the output 17a'
of drive circuit 17'.
In this embodiment, the voltage across
capacitance 80, in the time interval between time to
and time tl, while switching transistor 15 is saturated,
- 25 -
RD-11045
t 162983
is essentially zero volts. During this time interval
the current in the mutual inductanee of transformer 14
linearly increases and, at time tl, switching device 15
is turned off. The current flowing through primary
winding must be continuous and therefore flows into
the resonating capaeitance e.g., eapaeitor 80,
eapacitor C' or the parallel combination of capacitors
C' and 80, eausing ringing to eommenee. A high voltage
pulse is generated across secondary winding 14b to
drive the magnetron load into conduction, clamping
the primary winding voltage. As the energy stored in
transformer 14 is depleted by conduetion of the load,
the transistor colleetor voltage attempts to swing
negative and is clamped substantially to zero volts
by eatehing diode 19, at whieh tlme another eyele of
the power supply is commeneed by application of a
next subsequent turn-on eurrent, and/or voltage,
drive pulse to the switching dev:ice from base drive
- eircuit 17'.
The present invention has been described
with reference to several presently preferred
embodiments thereof. Many variations and modifications
will now become apparent to those skilled in the art.
It is my intent, -therefore, to be limited only by
seope of the impending elaims and not by the speeifie
detailed embodiments deseribed herein.
'
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