Note: Descriptions are shown in the official language in which they were submitted.
~2~75~
The present invention relates to radio signaling equipment and in
particular to apparatus for enhancing the probability of detection of a
signal at a receiver.
In known radar apparatus, it is common to use a buffs modulated
carrier wave as a transmitted signal and to provide an identically coded
sequence of signals which can be delayed and correlated with signals no-
Elected from a target and received at a receiver to give the required
target range information. The modulation of the carrier wave is achieved
either by phase shift keying techniques in which the keying operations
are controlled by the output ox a digital code generator, or injection
phase locking the carrier wave oscillator to a stored replica of the
previous transmitted pulse in an interrupted carrier wave coherent radar
system.
In known telecommunications apparatus, it is common to use a binary
signal representation of data for transmission and to code each l-signal
and each 0-signal of the representation with a separate code sequence.
In this way each l-signal and each 0-signal of a given duration is sub-
divided in a predetermined way into a number of much shorter duration
l-signals and 0-signals. The effect of this is to widen or spread -the
frequency spectrum of the transmitted signals so that reception and de-
coding by an unwanted recipient is difficult. In the telecommunications
apparatus it is also common to use a buffs modulated carrier wave as
the transmitted signal where the buffs modulations are controlled by
the coded data signal.
It is an object of this invention to provide receiver apparatus for
a radar or telecommunications system in which the buffs coded signal
is received and detected in-a manner which is relatively insensitive to
the frequency of the return signal (to Doppler frequency shifts have no-
natively little effect on the detection probability) and in which the
signal-to-noise ratio of the received signal is enhanced significantly
above that achieved in alternative Doppler independent receiver circuits.
75~
According to the present invention, a radio signaling equipment
includes receiver apparatus, for receiving signals phase-modulated by a
predetermined code sequence of code signals having a predetermined
regular digit-signal rate, the said apparatus including:-
a decoding means for applying a sequence of phase-shifts, complementary
to those given by -the code sequence, to the received signals;
a first signal path and a second signal path having a common input
connected to an output of the decoding means;
a multi-tap delay means connected in at east one of the signal paths
for progressively integrating signals in one of these signal paths, the
time delay between adjacent taps on the multi-tap delay means being an
integral multiple of the said bit duration;
` and signal mixer means connected to receive at one input the integrated
signals from the multi-tap delay means and at a second input signals
from the second signal path, for deriving an auto-correlation signal the
magnitude of which is dependent on the relative phases of the signals so
received.
me multi-tap delay means may be an acoustic delay line having a
plurality of transducers equispaced along its length, the distance between
any two adjacent transducers being an integral multiple of -the said bit
duration at the appropriate acoustic wavelength.
The transmitter apparatus associated with the receiver apparatus
described above may include a pulse generator circuit for generating pulses
at the desired pulse repetition frequency (PRY) of the radio signaling
equipment, a surface wave acoustic delay line, two input transducers, one
at each end of the surface wave acoustic delay line an integral number of
acoustic wavelengths at an intermediate frequency tiff apart and each
connected to an output of the PRY generator, and an output transducer
- positioned on the delay line between the two input transducers and so-
pirated from each by an integral number of acoustic wavelengths at the
said IF.
~2~55
The PRY generator may be an IF surface wave acoustic oscillator,
such as is described in Ultrasonics - May 1974 at page 115 in the paper
by MY Lewis "Surface Acoustic Wave Devices and Applications", driving
a step recovery diode impulse generator. The output of the delay line
may be connected to a random or pseudo-random buffs modulating circuit
synchronizing clock pulses for which may be derived from the output of
the step recovery diode impulse generator.
An embodiment of the invention as applied to an interrupted carrier
wave phase coherent proximity fuzz radar will now be described by way of
example only and with reference to the accompanying drawing which shows
the transmitter and receiver circuits of a proximity fuzz radar in scheme r
circuit diagram form.
In the drawing the IF stages of a proximity fuzz radar carried
for example in the warhead of a Surface-to-Air missile (SAY), comprise
a surface acoustic wave oscillator 1 providing signals at 50 MHz which
are frequency divided by a frequency divide by 10 circuit 2 to give a
5 MHz reference signal. A step recovery diode impulse generator circuit
3 is connected to the output of the frequency divide by 10 circuit 2 so
as to produce sharp rising pulses corresponding to each positive peak
of the 5 MHz signal hence determining the pulse repetition frequency (PRY)
of the radar system. The output of the step recovery diode impulse generator
circuit 3 is connected to two input transducers 4 and 5 at the extremities
of a lithium niobate surface acoustic wave delay line 6 and also to the
clock input of a pseudo-random code generator circuit 7. The input trays-
dupers 4 and 5 on the lithium niobate surface acoustic wave delay line 6
are separated by an integral number of acoustic wavelengths at a frequency
of 100 MHz. Each pulse from the step recovery diode impulse generator
circuit 3 causes the transducer 4 and the transducer 5 to resonate at 100
MHz for 60 no, to 30% of the interval between successive pulses from the
step recovery diode impulse generator circuit 3. Between the input trays-
dupers 4 and 5, also an integral number of acoustic wavelengths from each,
is an output transducer 8. The output of the transducer 8 is connected
to one input of a buffs modulator circuit 9 another input of which
is connected to the output of the pseudo-random code venerator circuit 7.
Ire pseudo-random code generator circuit 7 produces a 1023 bit pseudo-random
sequence of binary signals each bit of which is produced a-t its output at
a time corresponding to the receipt of a clock pulse from the step recovery
diode impulse generator circuit 3. The phase of each 60 no 100 MHz signal
at the output of the transducer 8 is determined in the buffs modulator
circuit by the polarity of the binary signal received coincidentally from
the pseudo-random code generator circuit 7. For example, the buffs
modulator circuit 9 may give a 0 phase shift to one of the 60 no 100 Miss
signals when a 0-signal is received from the pseudo-random code generator
circuit 7, and a phase shift to another 60 no 100 MHz signal on receipt
of a l-signal.
The output of the buffs modulator circuit 9 is connected to one
input of an up-mixer circuit 10 another input of which is driven from a
local oscillator circuit 11 which may be a surface acoustic wave device.
The up-mixer circuit lo maintains the phase of the IF buffs modulated
pulses but increases their frequency to a suitable microwave frequency
for transmission. The output of the up-mixer circuit 10 is amplified in
a radio frequency power amplifier 12 and fed via a transmit-receive (TRY)
switch 13 to an aerial 14 for transmission. The transmitted signal thus
consists of a coded sequence of buffs modulated microwave pulses which
will be reflected by a target 15 and received by the aerial 14 at a time
dependent on the Range of that target 15. The TRY switch 13 is operated
by pulses derived from the step recovery diode impulse generator circuit
3 such that for the first 60 no following a pulse from the generator circuit
3 the transmitter is connected to the aerial 14 and for the remaining 140
no between successive pulses from the step recovery diode impulse generator
3 the aerial is connected to the receiver. This is described as 30% duty
cycle operation.
~2~75S
The receiver circuit is shown in the drawing. In the receiver a
preamplifier circuit 16 has an input connected via the TRY switch 13
to the aerial 14 and an output connected to a down-mixer circuit 17
which has another input connected to the output of the local oscillator 11.
The down-mixer circuit 17 converts the received microwave frequency signals
to the 100 M~lz IF maintaining the phase relationship between successive
pulses. The IF signal is amplified in conventional IF amplifier stages
18 and fed to one input of a demodulator circuit 19. Another input of
the demodulator circuit 19 is connected via a range delay circuit 20 to
the output of -the pseudo-random code generator circuit 7. The demodulator
circuit 19 operates in a similar manner to the buffs modulator circuit
9 to the phase of each 60 no 100 MHz signal received from -the IF stages
18 is determined in the demodulator circuit 19 by the polarity of the
delayed binary signal received coincidentally from the pseudo-random code
generator circuit 7. Louvre, the demodulator circuit 19 applies phase
shifts to the receive signals which are complementary -to those applied
by the buffs modulator circuit 9 in the transmitter. Thus, if the
example given above of the operation of the buffs modulator circuit 9
was the case, the demodulator circuit 9 would give a or phase shift of
one received signal when a 0-signal is received from the pseudo random
code generator circuit 7 and a 0-phase shift to another receive signal on
receipt of a l-signal. When the target is at a range corresponding to
the time delay given to the binary code sequence by the delay circuit 20
all the received signal applied to the demodulator 19 will be converted
in the demodulator circuit 19 to a common phase sign].
The output of the demodulator circuit 19 is split into two channels.
One channel is fed directly to one input of a phase sensitive detector 21
and the other channel is fed to an input transducer on a 80 tap quart
delay line 22. Each of the 80 taps on the quartz delay line 22 are communed
together to form an output of the delay line 22 which is connected to a
further input of the phase sensitive detector 21. The output of the phase
sensitive detector 21 is connected via a low-pass filter 23, an integrator
so
circuit 24 and a threshold circuit 25 to an output of the system which
in this case is connected to a detonation circuit (not shown).
The 80 tap delay line 22 and the phase sensitive detector I form
an auto-correlator which together with the circuits 23 to 25 form an
arrangement in which each received pulse is first auto-correlated with
a one-bit delayed version of the preceding pulse in order to overcome
carrier frequency variations due, for example, to the Doppler effect.
In this case however the one-bit delay is replaced in the auto-correlator
by the 80 tap delay line 22 which in total provides a 80 bit delay so that
the phase sensitive detector 21 compares the phase of the sty pulse with
the integrated phase of the previous 80 pulses. This is a pre-detection
integration technique which provides significant signal-to-noise enhance-
mint in the receiver.
It will be appreciated that in interrupted carrier wave systems
the problem was to provide a transmitter which could be completely switched
off during receive periods, so that -the signal-to-noise ratio of the received
signal was not impaired, and yet when switched on for the next transmit signal
to have its phase coherently preserved in relation to the previous transmission.
In the apparatus described above however a carrier-wave oscillator is not
used and the requisite quiet receive periods and phase coherence is achieved
by the impulse generator circuit 3 and the delay line 6 to simulate a near
perfect interrupted carrier-wave coherent system.
The three transducer configuration of the lithium niobate surface
acoustic wave delay line 6 is a commonly used technique for reducing the
insertion 1QSS which is associated with two transducer type delay lines.
Typically the insertion loss associated with conventional delay lines is
reduced by 3 dub by employing this 3 transducer technique. Phase coherence
between successive pulses of the transmitted signal is ensured because of
the exact reproducibility of the response of the transducers 4 and 5 to
the sharp pulses produced by the step recovery diode impulse generator 3.
The transmitter therefore produces the 30~ duty cycle microwave pulse train
coded with buffs modulation according to the 1023 bit pseudo-random
I
sequence generated by the pseudo~randomcode generator circuit 7. Between
each transmitting burst a period 140 no is allowed in which pulses may be
received by the receiver after reflection from the target 15. When the
target is at the specified range the demodulator circuit 19 reduces all
the phase coded signals received to common phase and after the auto-
correlation process described for enhancing the slgnal-to-noise ratio of
the signal prior to detection by a predetection integration in the multi-
tap delay line 22 the output of the low-pass filter 23 and the integrator
24 will gradually build Up to a level which will eventually exceed the
reference level applied to the threshold circuit 25. The output of the
threshold circuit 25 is then used to detonate the SAM warhead.
It will now be appreciated that the arrangement described if the
example embodiment provides not only the post-detection integration afforded,
but also gives predetec-tion integration by the use of the rnulti-tap delay
line 22~ The combination of post detection and predetection integration
gives more processing gain than a system employing only post detection
integration and is less sensitive to Doppler frequency shifts than a
system employing only predetection integration. In the embodiment a 80
tap delay line was used but the number of taps used in any particular
application must be carefully chosen because in general an N-bit IF pulse
integrator which in effect the tap delay line is) is quite frequency son-
sitive and the range of frequencies that it will successfully operate upon
- YET wide to the 3 dub point, where N is the number of taps or bits to
be integrated and T is the bit period. Thus in applications where Doppler
frequency shifts are expected the number of taps in the predetection integrator
whilst being chosen to be as high as possible to increase the predetection
integration must not be so high as to render the system sensitive to Doppler
frequency shifts.
It will be noted that the system described in the example makes ox-
tensile use of surface acoustic wave devices. The oscillator 1, the local
oscillator If and the delay lines 6 and 22 aureole surface acoustic wave
devices. Because of this extensive use of surface wave acoustic devices
L7S~
the size and hence weight of the radar fuzz system can be considerably
less than would be possible with conventional devices due to the 10
velocity reduction of acoustic surface waves as compared with electron
magnetic waves of the same frequency. Because the surface acoustic wave
oscillator 1 which defines the pulse reputation frequency is based on
the same substrate as the 80 tap delay line the surface acoustic wave
functions are essentially insensitive to temperature changes. This is
a very significant consideration and means that the substrate can be,
for example, lithium nioba-te with its high coupling constant, to low
coupling losses but its high temperature coefficient which normally
makes its choice undesirable. Aluminum nitride on sapphire is another
possibility enabling radio frequency surface acoustic wave components
for the system to be fabricated without the use of an electron beam.
These devices are planar structures making them very easy to integrate
into miniaturized electronic packages such as -the system described here.
Known interrupted carrier wave coherent radar systems have sense-
tivities limited by inadequate suppression of the carrier during the
receiver interval but the system described hereinabove has a -transmitter
which is completely quiescent at the receive times and a carrier suppression
of greater than 120 dub is possible.
The method of demodulation of the IF pulses in the receiver gives
scope for optimum predetection integration using the surface acoustic
wave tap delay line 22 as described above. In general the greater the
percentage of time spent in predetection integration compared with post-
detection integration the more sensitive is the receiver. The predetection
integration time to the number of taps on the delay line is limited by
Doppler shift in theory as described above, although in practice it is
the maximum size of the surface wave substrate which can be obtained which
provides the limitation.
Although a specific example has been described with respect -to a
radar fuzz the system of combined predetection and post detection integration
described above has application to communications. Slow data for trays-
mission may be spread over a wide bandwidth which is a well-known anti-jam
.. .. .. . . ..
technique in communications. The improved signal-to-noise ratio
provided by the receiver in the above system is of course extremely
desirable in communications systems where unknown Doppler shifts are a
problem.
Many variations and modifications of -the above example will now
suggest themselves to one slcilled in the art. The fixed delay line 20
which determines the activation range can be replaced with a selector
for selecting different outputs of the psuedo-random code generator
circuit 7 and applying them to the demodulator 19. For example if this
pseudo-random code generator circuit consists of a shift register with
appropriate feedback to produce the psuedo-random sequence then the
selector might be arranged to select different stages of the shift
register according to the desired range making use of a digital selector
to address the appropriate shift register stage. This ability to derive
the range function digitally is an important aid in for example altimetry
where range tracking is desired. Another use of the digitally selectable
range is to combat multi path problems in a communications system where
a programmable correlator is needed in the receiver. These features can
be incorporated in the apparatus according to the invention without the
severe penalty which occurs in conventional receivers, or the loss in
processing gain which occurs in the Doppler insensitive post-detection
processors.
Although surface acoustic wave delay lines have been used here
there is no reason in principle why they may not be replaced by electron
magnetic delay lines, for example the 80 tapped delay line 22 could in
principle be a tapped delay on a surface electromagnetic wave device.
Again operation can be at IF and up converted to radio frequency
as described above but with improvements in surface acoustic wave technology
and integrated circuits it will almost certainly be possible to eliminate
the need for IF if the required radio frequency is below 2 GHz.
