Note: Descriptions are shown in the official language in which they were submitted.
~ ;l
~ ~ ~ 44 ~ 9
-1-
A LOW-PROFILE BROADBAND MICROS~IP ANTENNA
,~
This invention was made with partial Government support
under a contact from the U.S. Air Force. The U.S.
Government has certain rights in the invention.
TR~'TTNTc~l!T~ FIRT.n
The present invention relates generally to antennas,
and more particularly relates to microstrip antennas.
BACRGROUND OF T~R I N V ~ ON
In many antenna applications, for example such as for
use with aircraft and vehicles, an antenna with a broad
bandwidth is required. For such applications, the so-called
"frequency-independent antenna" ("FI antenna") commonly
has been employed. See for example, V.H. Rumsey, Fre~l]ency
Independent ~nte~nas, Academic Press, New York, NY, 1966.
Such frequency-independent antennas typically have a
radiating or driven element with spiral, or log-periodic
structure that enables the frequency-independent antenna to
transmit and receive signals over a wide band of
frequencies, typically on the order of a 9:1 ratio or more
(a bandwidth of 900~). For example, European Patent
Application No. 86301175.5 of R.H. D~ mel entitled "Dual
Polarized Sinuous Antennas", published October 22, 1986,
publication No. 0198578 (See
, ,~, .,
WO93/l1582 ~ f~ PCT/US92/09439
_ 2-
aiso U.S. Patent No. 4,658,262 dated April 14, 1987),
discloses frequency-independent antennaswith a
log-periodic structure called "sinuous."
In a conventional frequency-independent antenna,
5 a lossy cylindrical cavity is positioned to one side of
the antenna element so that when transmitting, energy
effectively is radiated outwardly from the antenna only
from one side of the antenna element (the energy radiating
from the other side of the antenna element being
10 dissipated in the cavity). However, high-performance
aircraft, and other applications as well, require that the
antenna be mounted substantially flush with its exterior
surface, in this case the skin of the aircraft. This
undesirably requires that the cavity portion of the
15 frequency-independent antenna be mounted within the
structure of the aircraft, necessitating that a
substantial hole be formed therein to accommodate the
cylindrical cavity, which typically is at least two inches
deep and several inches in diameter. Also, the use of a
20 lossy cavity to dissipate radiation causes about half of
the radiated power to be lost, requiring a greater power
input to effect a given level of power radiated outwardly
from the frequency-ind~pendent antenna.
In recent years the so-called "microstrip patch
an~enna" has been developed. See for example, U.S. Patent
Reissue No. 29,911 of Munson (a reissue of U.S. Patent No.
3,921,177) and U.S. Patent Reissue No. 29,296 of
Krutsinqer, et al. (a reissue of U.S. Patent No.
3,810,183). In a typical microstrip patch antenna, a thin
30 metal patch, usually of circular or rectangular shape, is
placed adjacent to a ground plane and is spaced a small
distance therefrom by a dielectric spacer. Microstrip
patch antennas have generally suffered from having a
narrow useful bandwidth, typically less than 10%.
_3 ~ ~ ~4~ 5~ ~
Our U.S. Patent No. 5,313,216 issued May 17, 1994
recites a multi-octave spiral-mode microstrip antenna which
overcomes many of the prior art limitations. This spiral-
S mode antenna approaches the bandwidth of frequency-
independent antennas and is nearly flushly mounted above a
ground plane. However, multi-mode operation of a spiral-
mode microstrip antenna requires the spiral to be of
circumference at least m~, where m is the highest desired
mode and ~ is the wavelength. Thus, the spiral diameter can
become undesirably large, especially at lower frequencies.
Microstrip patch array antennas have also been known in
the art. See, for example, Munson, R.E., Conformal
Microstrip Antennas and Microstrip Phased Arrays, I~E
lS Transactions on Antennas and Propagation, p. 74. (Jan.
1974). The Munson article discusses an array of rectangular
elements. However, known microstrip arrays, including the
Munson design, generally are electrically large (i.e. the
antenna is relatively large in comparison with the
wavelength of the operating frequency), having individual
elements of approximately one-half wavelength in diameter
and spaced from one another a distance slightly grater than
their diameters.
U.S. Patent No. 4,766,444 of Conroy et al. relates to a
conformal "cavity-less" antenna having an array of single-
arm spiral elements driven in unison and which are aligned
linearly along an outwardly-curved surface. A lossy hex-
cell structure spaces the spiral elements away from the
ground plane and takes the place of the typical cavity. The
resulting antenna is disclosed as being suited for use as an
interferometer and tends to suffer from having an narrow
useful bandwidth. Again this is an electrically large
array.
Accordingly, it can be seen that a need yet
',, '' ' 1'!3 msc
CA 021244~9 1998-0~-0~
remains for an antenna which has a low profile, has a broad
bandwidth relative to prior antennas, and is small in physical
size. It is the provision of such an antenna that the present
invention is primarily directed.
SUMMARY OF THE lNv~NllON
Briefly described, the present invention comprises a compact
broadband microstrip antenna. In a first embodiment, the
invention comprises a compact microstrip antenna comprising: an
array of antenna elements arranged in a closed loop pattern and
being non-resonant over an operating band, said array having a
diameter smaller than 0.7mAc /~, wherein m is a number indicating
a spiral mode produced by said antenna and wherein Ac is the
wavelength at the center frequency of the operating band, said
elements adapted to be driven out of phase from one another for
generating at least one spiral mode; a ground surface; and a
substrate positioned to one side of said antenna elements for
spacing said antenna elements a selected distance from said
ground surface, said selected distance being greater than or
equal to 1/60 and less than or equal to 1/2 of said wavelength
at said center frequency, and said substrate having a relative
dielectric constant which is greater than or equal to 1.0 and
less than or equal to 2Ø
Preferably, the closed array comprises a circular
arrangement of four or more elements, each element being made
from a thin metal foil. Preferably, the substrate has a relative
dielectric constant which is greater than or equal to 1.0 and
less than or equal to 2Ø Also, the thickness of the substrate
CA 021244~9 1998-0~-0~
is carefully selected to get near maximum gain at a particular
wavelength, with the substrate having a thickness typically in
the range of 0.1 to 0.30 inches for microwave frequencies of 2
to 18 GHz. The substrate thickness for other frequencies is
determined by the frequency scaling method. Further, preferably
the array has a diameter smaller than mAc /~, wherein m is a
number indicating a spiral mode produced by the antenna and
wherein m is a number indicating a spiral mode produced by the
antenna and wherein Ac corresponds to the wavelength at the
center frequency of the operating band. Also, a loading material
can be positioned adjacent the antenna elements.
With this construction, an antenna is provided which can be
mounted externally to a structure and which can be conformed to
the surface thereof. Also, the antenna exhibits a fairly broad
bandwidth, typically on the order of 300~. This design is based
on the discovery by the applicants that the ground plane of a
microstrip antenna is compatible with the spiral modes of the
antenna. In this regard, the individual elements of the closed
array are electrically driven out of phase with one another in
a manner to cause the aggregate antenna to generate a beam
pattern according to a desired spiral mode or modes, for example,
modes m=l and m=2.
In a second embodiment, the invention comprises a microstrip
antenna for mounting to one side of a ground plane or other
surface, the antenna comprising one or more antenna elements
positioned to one side of a magnetic substrate for spacing the
antenna elements a selected distance from the ground plane. The
- - CA 021244~9 1998-0~-0~
magnetic substrate is chosen to have a relative permittivity
which is roughly equal to its relative permeability. This allows
the antenna to generate multiple spiral modes effectively,
without the ill-effects of having a substrate with a high
dielectric constant.
In a third embodiment, the present invention comprises a
compact spiral-mode microstrip antenna for mounting to one side
of a ground plane comprising: one or more antenna elements; and
a substrate positioned to one side of said antenna elements for
spacing said antenna elements a selected first distance above the
ground plane in a first radiation zone and for positioning said
antenna elements a second selected distance above the ground
plane in a second radiation zone, said first selected distance
and said second selected distance being different from one
another, said first and second selected distances being chosen
for suppressing radiation in one of said first and second
radiation zones and for fostering radiation in the other of said
first and second radiation zones.
The antenna of the third embodiment is adapted for operating
in a particular mode, for example mode m=2. To this end,
radiation in the radiation zone for the m=1 mode is suppressed
with a relatively close spacing of the antenna element relative
to the ground plane. The mode m=2 is fostered by having a
sufficiently large spacing between the antenna element and the
ground plane in the m=2 radiation zone. This takes advantage of
the fact that an antenna radiates in radiation zones roughly
corresponding to circles having circumferences equal to m~, where
~ is the wavelength and m is the radiation mode or spiral mode.
~ CA 021244~9 1998-0~-0
-6a-
Thus, an antenna tends to radiate in the first radiation zone for
mode m=1 and radiates at a second, outer radiation zone for mode
m=2. By selectively varying the spacing between the ground plane
and the antenna element in these various radiation zones, the
radiation in the m=1 mode can be suppressed, while fostering
radiation in the mode m=2. Of course, it is possible to reverse
this so that the spacing suppresses radiation in the mode m=2 and
fosters radiation in the mode=1 region, although in many
instances there is no need to do this because it is possible to
eliminate the mode m=2 radiation by truncating the antenna
element so that there is no radiation zone which is large enough
to support mode m=2 radiation.
In yet another embodiment, the present invention comprises
a multioctave microstrip antenna system comprising: a spiral-
mode antenna element including at least two metal foil arms
formed in a geometric pattern and adapted to generate at least
one spiral mode when excited; a conducting ground surface
positioned to one side of said antenna element; a substrate
positioned to one side of said antenna element for spacing said
antenna element a selected distance from said ground surface,
said selected distance being greater than or equal to 0.02AC and
less than or equal to .2AC, where Ac is the wavelength at the
geometric mean frequency between the minimum and maximum
operating frequencies, said substrate having a relative
dielectric constant which is greater than or equal to 1.0 and
less than or equal to 2.0; and a feed network with a near-perfect
impedance match with said spiral mode antenna element for
exciting at least one spiral mode.
~ CA 021244~9 1998-0~-0
-6b-
These arrangements are quite compact and efficient. Also,
the capability of selectively operating in one mode, or in
several modes, allows the antenna to be useful in beam steering
and null steering.
Accordingly, it is a primary object of the present invention
to provide a compact antenna which has a fairly broad bandwidth
performance, while having a low profile.
It is another object of the present _ =
WO93/11582 ~ PCT/US92/09439
_ -7-
provide a microstrip antenna which has an i~proved
bandwidth.
It is another object of the present invention to
provide an antenna having a small aperture.
5It is another object of the present invention to
provide an antenna capable of beam and null steering.
Other objects, features, and advantages of the
present invention will become apparent upon reading the
following specification in conjunction with the
l0 accompanying drawing figures.
BRIEF DESCRIPTION OF TH~ DRAWING PIGURES
Fig. l is a plan view of a microstrip antenna in
15 a preferred form of the invention.
Fig. 2A is a schematic, partially sectional side
view of the antenna of Fig. l.
Fig. 2B is a schematic, partially sectional side
view of a portion of the antenna of Fig. 2A.
20Fig. 3 is a schematic view of a feed for driving
the antenna of Fig l.
Figs. 4A and 4B are plan views o: modifie~. forms
of the antenna of Fig. l, depicting sinuous antenna
elements.
25Figs. 5A and 5~ are plan views of modified forms
of the antenna of Fig. l, depicting log-periodic tooth
antenna elements.
Fig. 6 is a plan view of a modified form ~f the
antenna of Fig. l, depicting a rectangular spiral antenna
30 element.
Figs. 7 and 8 are plan views of modified forms of
the antenna on Fig. l, depicting Archimedean and
equiangular spiral antenna elements, respectively.
Figs. 9A and 9B and l0A and l0B are schematic
35 illustrations of mathematical models used to analyze the
the~retical basis of the antenna of Fig. l.
Fi~s. ]lA an~ l]~ are graphs of experimental
WO93/11582 ~<~ 7 ~ PCT/US92/09439
laboratory results of the disruptive effect of the
dielectric substrate (when the dielectric constant is
great) on the radiation pattern of an antenna as shown in
Fig. 1.
Fig. 12 is a graph of laboratory results
comparing antennas according to the present invention with
a prior cavity-loaded spiral antenna.
Fig. 13 is a graph of laboratory results for the
antenna of Fig. 1 showing the effect of positioning the
10 antenna element on antenna gain at various spacings from
the ground plane for three different operating frequencies.
Fig. 14 is a graph of antenna radiation patterns,
specifically, spiral mode patterns (for n~l, n=2, etc.).
Fig. 15 is a schematic plan view of an antenna
15 according to another preferred form and having closed
array elements.
Fig. 16 is a side sectional view of the antenna
of Fig. 15.
Fig. 17A and 17B are graphs of radiation patterns
20 for modes m.l and m-2, respectively.-
Fig. 18 is a schematic plan view of analternative embodiment in which concentric circular arrays
of elements are arranged.
Fig. 19 is a schematic illustration of a tunable
25 multiple-resonance-frequency microstrip antenna switched
by PIN diodes.
Fig. 20 is a schematic illustration showing that
a substrate material used in a spiral-mode microstrip
antenna with equal relative permittivity and permeability.
Figs. 21A and 21B show mode-2 antennas with a
non-constant spacing above the ground plane.
~ ~4~ ~
DT'T~TT.T'n DT'.C:~T~TpTTON
Sections 1-3 below illustrate some of the principles of
the present invention, particularly including the principle
of how the antenna and its elements are mounted and spaced
above a ground plane. The sections that follow numbered
sections 1-3 provide the remainder of the disclosure of the
present invention, including how the antenna is comprised of
phased array elements, or uses a magnetic substrate
material, Ol~ has a non-constant spacing between the antenna
element(s) and the substrate as a function of radius.
1. The Physical Structure of the Mountin~ of the Antenna
Referring now in detail to the drawing figures, wherein
like reference characters represent like parts throughout
the several views, Figs. 1, 2A and 2B show a multi-octave
microstrip antenna 20, according to a preferred form of the
invention and shown mounted to one side of a group plane GP.
The antenna 20 includes an antenna element 21 comprising a
very thin metal foil 21a, preferably copper foil, and a thin
dielectric backing 21b. The antenna element foil 21a shown
in Figs. 1, 2A and 2B has a spiral shape or pattern
including first and second spiral arms 22 and 23. Spiral
arms 22 and 23 originate at terminals 26 and 27 roughly at
the center of antenna element 21. The spiral arms 22 and 23
spiral outwardly from the terminals 26 and 27 about each
other and terminate at tapered ends 28 and 29, thereby
roughly defining a circle having diameter D and a
corresponding
msc
W093/ll~82 PCT/US92/09439
circumference of ~D . The antenna element foil 21a is
formed from a thin metal foil or sheet of copper by any of
well known means, such as by machining, stamping, chemical
etching, etc. Antenna element foil 21a has a thickness t
5 of less than 10 mils or so, although other thicknesses
obviously can be employed as long as it is thin in terms
of the wavelength, say for example, 0.01 wavelength or
less. While the invention is disclosed herein in
connection with a separate ground plane GP, it will be
10 obvious to those skilled in the art that the antenna can
be constructed to include its own ground plane, making the
antenna suitable for mounting on non-conductin~ surfaces,
e.g., on engineering plastics and composites.
The thin antenna element 21 is flexible enough to
15 be mounted to generally nonplanar, contoured shapes of the
ground plane, although in Figs. 2A and 2B the ground plane
is represented as being truly planar. The antenna element
foil 21a is uniformly spaced a selected distance d (the
standoff distance) from the ground plane GP by a
20 dielectric spacer 32 positioned between the antenna
element 21 and the ground plane GP. The dielectric spacer
32 preferably has a low dielectric constant, in the range
of 1 to 4.5, as will be discussed . more detail below.
The ~i~]ectric spacer 32 is generally in the form of a
25 disk and is sized to be slightly smaller in diameter than
the antenna element 21. The thickness d of the dielectric
spacer 32 typically is much greater than the thickness of
the dielectric backing 21b of the antenna element 21. The
thickness d of spacer 32 typically is in the neighborhood
30 of O.Z5" for microwave frequencies. However, the specific
thickness chosen to provide a maximum gain for a given
frequency should be no greater than one-half of the
wavelength of the frequency in the medium of the
dielectric spacer.
WO93/1l582 ~ PCT/US92/09439
A loading 33 comprising a microwave absorbing
material, such as carbon-impregnated foam, in the shape of
a ring is positioned concentrically about dielectric
spacer 32 and extends partially beneath antenna element
5 21. Alternatively, a paint laden with carbon can be
applied to the outer edge of the antenna element. Also,
the antenna element can be provided with a peripheral
shorting ring positioned adjacent and just outside the
spiral arms 22 and 23 and the peripheral shorting ring
10 (unshown) can be painted with the carbon-laden paint.
First and second coaxial cables 36 and 37 e~tend
through an opening 38 in the ground plane GP for
electrically coupling the antenna element 21 with a feed
source, driver or detector. The coax cables 36 and 37
15 include central shielded electric cables 42 and 43 which
are respectively connected with the terminals 26 and 27.
The outer shieldings of the coaxial cables 36 and 37 are
electrically coupled to each other in the vicinity of the
,
antenna element, as shown in Fig. 2B. As shown
_
20 schematically in Fig. 3, this electrical coupling of the
shielding of the coasial cables can be accomplished by
soldering a short electric cable 44 at its ends to each of
the coa~ial cables 36 and 37.
Preferably, as shown in Fig. 3, the coaxial
25 cables 36 and 37 are connected to a conventional RF hybrid
unit 46 which is in turn connected with a single coax
cable input 47. The function of the RF hybrid unit 46 is
to take a signal carried on the input coax cable 47 and
split it into two signals, with one of the signals being
30 phase-shifted 180~ relative to the other signal. The
phase-shifted signals are then sent out through the
coa~ial cables 36 and 37 to the antenna element 21. By
providing two signals, phase-shifted 180~ relative to each
- other, to the two antenna element arms, a voltage
35 potential is developed across the terminals 26 and 27
WO93/ll~82 PCT/US92/09439
~ ~ 4 4 ~ g
-l2-
corresponding to the waveform carried along the coaxial
cables 36, 37 and 47, causing the antenna to radiate
primarily in a n.l mode (although some components of
higher-order modes can be present). As an alternative, a
5 balun may be used to split the input signal into first and
second signals, with one of the signals being delayed
relative to the other. A balun can be used to feed the
antenna for operating in the n~l mode (single beam
pattern). The RF hybrid circuit can be used for
10 generating higher-order modes, e.g., n~2. For generating
these higher-order modes, 4, 6, or 8 antenna element arms
are used in conjunction with a corresponding number of
feed terminals.
The dissipative loading 33 can be done away with
15 by using a substrate with a very low relative dielectric
constant, preferably close to unity (1.0), and by using a
feed network with a near-perfect impedance match with the
arms of the spiral to excite the desired spiral modes
(with the effect of the ground plane incorporated in the
20 impedance matching).
Fig. 4A shows an alternative embodiment of the
antenna of Fig. 1, with the spiral arms 22 and 23 of
Fig. 1 being replaced with sinuous arms 52 and 53. While
a two-arm sinuous antenna element is shown in Fig. 4A, a
25 four-arm sinuous antenna element can be provided if
higher-order modes are desired, as shown in Fig. 4B.
Fig. 5A shows a modified form of the antenna
element 21 in which the spiral arms 2~ and 23 are replaced
with log-periodic toothed arms 56 and 57. The toothed
30 antenna element illustratively shown in Fig. 5A includes
toothed arms which have linear segments which are
perpendicular to each other, i.e., the ~teeth" of each arm
are generally rectangular. Alternatively, the teeth can
be smoothly contoured to eliminate the sharp corners at
35 each tooth. Also, the teeth can be curved as shown in
Fig. 5B.
WO93/11582 ~ 2 ~ ~ t~ ~ PCT/US92/09439
~l3-
Fig. 6 shows another modified form of the antenna
element of Fig. 1 in which the spiral arms 22 and 23 are
replaced with rectangular spiral arms 58 and 59. Each of
the spiral arms is in the form of a spiraling square, as
5 compared with the rounded spiral of the antenna element of
Fig. 1. Figs. 7 and 8 show that the spiral pattern of
Fig. 1 can be provided as an "Archimedean spiral~ as shown
in F~g. 7 or as an "equiangular spiral" as shown in Fig. 8.
10 2. Theoretical BA-cis of ~h~ Mol~nti~g ArrAnqem~nt
The following discussion represents the results
of a theoretical study by applicants establishing the
viability of the invention. Esperimental verification of
15 the theoretical basis will be provided in the section
immediately following this one.
The basic planar spiral antenna, which consists
of a planar sheet of an infinit~ly large spiral structure,
~ diates on both sides of the spiral in a symmetric
~nner. When radiating in n.l mode, most of the radiation
occurs on a circular ring around the center of the spiral
whose circumference is approsimately one wavelength. As a
result, -one can truncate the spiral outside this active
region without too much disruption to its pattern, or
25 dissipative loss to its radiated power.
WO93/11582 ~ 14 - PCT/US92/09439
Figs. 9A and 9B depict an infinite, planar spiral
backed by a ground plane. The spiral mode fields in
Region ~ can be decomposed into T~ and TM fields in terms
of vector potentials ~Q and Ae as follows:
F~ = z'F~ TE Solution (1)
A~ = z A~ ylt TM Solution
In Region 1 where modes propagate in the +z direction, we have
q~ = ei"~ ¦ g~(k ) Jn (kpp)e 2~ k dk (3)
k21 = [k~--k2]lQ
kl ~)(to~o) (4)
2 ~ and the explicit expressions for the fields in region 1, where e = 1, are given by:
A ~F~
E
= ci"~¦;g (kp)t 2~2 [_ kp Jn ~ (kpp)k2~ rA(kpp)l kpdkp (5)
SUBSTITUTE ~3~FET
WO 93/11~82 ~ ~ 2 ~ PCI/US92/09439
- 15-
A d2 ~r d~
e ;~J,eO p dq5 dz e dp
jn~ ,~e (k ) i ze ¦ e ze J (k p) + F'ek Jn (kpP)¦ kp d kp (6
E, e =--( + k~ ) ~e
= ~ J g (k ) e Z~ ¦--k2 + k2 ¦ J (k p)k dk (7)
Ae d~e F d2yl
pe p d~ dpdz
= e i n~ ¦ g (k ) e jkZ eZ ¦ eJn J (k Fe~zelcp ¦ (8)
~,~ F~ a ry~
~e ~ dp j~:~p d~ dz
io P I ~ p n p Jc~ p ~ p ¦ p p
F~ d2
H e = j--( 2 + ke) ~J~
WO 93/l 1582 ~ ~ 3 ~ I f 5 1 6 PCr/US92/09439
ej n ~ l g (k ) e 2 ~ ¦ - k2 + k2 ¦ J (k p) k d k (10)
O p ~e ~ ~ p p p
5 ~n Region 2, modes propagating in both +z and -z directions exist and therefore
the vector potentials are
Fi = zF2 ~i TE solution (11)
A2i = zA2 ~p2i TM solution (12)
where
~i =ejn~l g(k ) J (k p)c s /cpdkp (13)
15 The explicit expressions for the fields in Region 2 are as follows.
Ep2=e ~ J g (kp)[--A2+ e s +A~ ei~s2] J ' (k )dk
+e ~ ~ g(kp)[--F2+ e s ~F~eiiJSl J P J (k )dk (14)
E~,2=e ~ J g (kp)[A2 c ~ --A- ei~JS] ~ P J k
+ej ~ ~ g(kp)[F2 e s + F2 e s ] k2 J ~ (k p)d k (1~)
Es2=-- ~ g(kp)Jn(k p)[A2+c s +A2-c s ]kp d kp ~ (16)
Hp2 = Cj ~ ~ g(kp)[A2+ e s +A2-c s ] P J ~k p) d k
+cj~~~ g(ko)[-F2 ~ ~ +P2 ~I P J ~(k p)dl~ (17)
H~2 = ei ~ ~ g(k )l--A2+ e ~ --A~ejissl k2 J ~(k ) d k
~U~S ~ Te SH~ET
WO93/11582 212 4 ~ 5 ~ PCT/US92/09439
+ eJn~ ¦ g (k )I+F+ e Z _F2 e Z I P J (k p) dk (18)
o P J ~)~Op n ~ p
H = I g(k )J (k p)[F+e J 2 +F- ik2zJk3 dk (19)
~ j~O O p n p 2 2 p p
By matching the boundary conditions at z.0 (where
tangential E and H are continuous in the aperture region)
and z--d (where tangential E vanishes) and by requiring
the fields satisfy the impedance conditions
Fl = jrl Hl, E2 = jrl H2 . E2 = ~ H2 (20)
we obtain the necessary and sufficient conditions for the
spiral modes as follows:
A,=A2 -A2
Fl=F2 +F2
--A+ e Z + A2 e 2 =o
F2+ e Z + F2 C ~ =~
F = --jr~AI
F2+ = -i~A2
_ (21)
F2 = J 11 A2
There are si~ unknowns in the above seven
30 equations. However, the seven equations are not totally
independent, and can be reduced to the following five
independent equations.
Fl = F2 + F2~
F2 e 2 + F- e i 2d =
WO93/l1582 ~ 18- PCT/US92/094~9
Fl = --Jr~A
F+-- -- A
2 Jrl 2
F2 = j rl A 2
s
Equations (22) have si~ parameters in the five equations.
Let, say Al, be given, then we can solve for all the other
five parameters. Thus the spiral radiation modes can be
10 supported by the structure of an infinite planar spiral
backed by a ground plane as shown in Figure 1. This
finding is the design basis of the multi-octave
spiral-mode microstrip antennas disclosed herein.
In practice, the spiral is truncated. The
15 residual current on the spiral beyond the mode-l active
region, therefore, faces a discontinuity where the energy
is diffracted and reflected. The diffracted and reflected
power due to the truncation of the spiral, as well as
possible mode impurity at the feed point, is believed to
20 degrade the radiation pattern. Indeed, this is consistent
with what we have observed.
To esamine the effect of a dielectric substrate
on the spiral microstrip antenna, we study the simpler
problem of an infinite spiral between two media, as shown
25 in Figs. lOA and lOB.
Region 1 is usually free space (~1 . ~o) where
radiation is desired. Region 2 is an infinite dielectric
medium with ~1 and ~O. Following the method of Section I,
we express the fields in both Region 1 and 2 in terms of
30 electric and magnetic vector potentials Fe and Ae.
The esplicit espressions for fields in Region
1 or 2) are
A~ F~ ~y~
E = _ _
P~ dp~z p d~
=ej n~J g (kp)e ~ kpJn'(kpp)~zz~ ~ --Jn(k p)] kpdkp
~n
wo 93/11582 ~ . . 9 PCT/US92/09439
E =-- --+ F
~ e ~ p d$~ dZ t dp
= e J n~ J g (k ) e Z~ ¦ . Jn (kpp) + F~k Jn ' (kpp)l k d kp (24)
E~ t = ( 2 + k~
=--~ g (kp ) c ¦--ke2 + ke ~ Jn(k p) k d k (25)
H =-- _ +_ d y~t
p~ p d ~ tp dp d z
= ~ g (kp ) e ¦--JA (kp p) -- P J ~(k p) ¦k dk (26)
H =--A _ + P, d v~
~ ~ ~p j~p d~ dz
n~ ) e-j~'ztz ¦--A k J ' (IC P)-- t Z~ J (k P)l k dk (27) p p n p J ~u p n p p p
3 0 z~ J Go/~ ~ d z
=--ejn9bl g(kp)e Z~ ¦--k2e+k2e ¦Jn(k p)kpdkp (28)
3 5 Continuity of the tangential E field at z = 0 in the aperture region requires
WO 93/11582 ,. ~. PCr/US92/09439
2 0
-A~ A2 z2 P
~Jt~P Z 1 ~2
Fljn F2 jn
P P
Al n A~ --A2 n k 2
i~)Clp i~2P
F,k =F2-~ (29)
Eq. (29) can be reduced to
--Alk~ 1 A2A~2
~1 C2
Fl F2 (30
- The impedance condition
2o El=jrllHl (3
requires
Al jrll F,
-
--Fl = jrllAl (32
which can be reduced to
Fl= jrllAl (33)
3 0 Similarly,
E2= jll2H2 (34
requires
F2=irl2A2 (3
3 5 Eqs. (30), (34)and (3~) are constraints on Al, Fl, F2, A2, which we sl-mmarize
as follows:
SUB~ TE SHEET
WO93/11582 2 ~ PCT/US92/09439
- 21 -
-A~ A2~2
= _
. ~2
F,=F2
S F = - j~ A (36)
F = fr~
The four equations in (36) can not be satisfied simultaneously unless
~ 2r2 (37)
_ = _
k,l ~.2
or
~1 k2 (38)
_ = _
k~l k~2
15 or
We see that Eq. (39) can be satisfied only if
kl=k2 or tl=~2
This means that the m=l spiral mode cannot be
supported by the dielectric-backed spiral shown in Figure
25 2 without significant components of higher-order modes.
This finding e~plains why earlier efforts to design a
broadband spiral microstrip antenna failed.
3. Esperimental Res~lts Verifying ~he Theoretical Basis of
3 0 th~ Mountin~ ArrA~yement
The effect of the presence of
high-dielectric-constant material on the performance of
the antenna was studied in two ways: with and without a
35 ground plane. To investigate the case of no ground plane,
SUE~5TITUT~ EET
WO93/11582 PCT/US92/09439
~ 2~-
both calculations and measurements were used. The basic
conclusion was that patterns degrade in the presence of a
dielectric substrate; the higher the dielectric constant,
and the thicker the substrate, the more seriously the
5 patterns degrade. Even though dielectric substrates cause
pattern degradation, it is possible to design spiral
microstrip antennas with acceptable performance over a
narrower frequency band.
The case of dielectric substrates between the
10 spiral and the ground plane was studied for materials of
relatively small dielectric constant, the greatest being
4.37, and little degregation was found at these
frequencies. The studies were conducted using the
configuration of Figure 1 with a substrate of 0.063 inches
15 of fiberglass, and for a substrate of 0.145 inches of
air. In both of these configurations, the electrical
spacing is the same (within 10%).
On the other hand, Figs. llA and llB show some
disruptive effect on the mode-l radiation patterns at 9
20 and 12 GHz for an antenna with ~,4.37 (fiberglass) and a
substrate thickness of d-l/16 inch. When the substrate
thickness d is reduced to 1/32 inch, the effect of the
dielectric becomes larger, especially at lower
frequencies. However, VSWR (voltage standing-wave ratio)
25 remains virtually unaffected by the presence of the
dielectric. We have thus demonstrated, both theoretically
and e~perimentally, the disruptive effect of dielectric
substrates on antenna patterns.
In many practical applications, the spiral
30 microstrip antenna is to be mounted on a curved surface.
To examine the effect of conformal mounting of the spiral
microstrip antenna on a curved surface, we placed a 3-inch
diameter spiral microstrip antenna on a half-cylinder
shell with a radius of 6 inches and a length of 14
35 inches. The truncated spiral was placed 0.3-inch above
WO93/l1582 2 1 2~ 3 PCT/US92/0g439
and conformal to the surface of the cylinder with a
styrofoam spacer. A 0.5 inch-wide ring of microwave
absorbing material was placed at the end of the truncated
spiral, with half of the absorbing material lying inside
5 the spiral region and half outside it. The ring of
absorbing material was 0.3-inch thick, thus filling the
gap between the spiral antenna element and the cylinder
surface.
The VSWR measurement of the spiral microstrip
10 antenna conformally mounted on the half-cylinder shell was
below 1.5 between 3.6 GHz and 12.0 GHz, and was below 2.0
between 2.8 GHz and 16.5 GHz. Thus, a 330% bandwidth was
achieved for VSWR of 1.5 or lower, and a 590% bandwidth
for VSWR of 2.0 or lower was reached.
The measured radiation patterns over ~ on the
y-z principal plane with ~.90~ yielded good
rotating-linear patterns obtained over a wide frequency
bandwidth of 2-10 GHz. Measured radiation patterns on the
x-z principal plane (~.0~) over e are of the same
20 quality. Thus, the spiral-mode microstrip antenna can be
conformally mounted on a curved surface with little
degradation in performance for the range of radius of
curvature studied here.
Recently, a researcher has reported a theoretical
25 analysis which indicated that poor radiation patterns are
due to the residual power after the electric current on
spiral wires (not "complementary") has passed through the
first-mode radiation zone which is on a centered ring
about one wavelength in circumference. (H. Nakano et al.,
30 ~A Spiral Antenna Backed by a Conducting Plane Reflector",
IEEE Trans. Ant. Prop., Vol. AP-34, pp. 791-796 ~1986)).
Thus, if one can remove the residual power from radiation,
it should be possible to obtain excellent radiation
patterns over a very wide bandwidth.
One technique for removing the residual power is
to place a ring of absorbing material at the truncated
WO93/11~82 PCT/US92/09439
2 ~ 24-
edge of the spiral outside the radiation zone. This
scheme allows the absorption of the residual power which
would radiate in "negative" modes, which cause
deterioration of the radiation patterns, especially their
5 axial ratio. This scheme is shown in Figs. l and 2A by
the provision of the loading ring 33.
Performance tests were conducted for a
configuration similar to that shown in Figure l, except
that the spiral was Archimedean as shown in Fig. 7, with 2
l0 separation between the arms of about l.9 lines per inch.
The experimental results demonstrate that for a spacing d
(standoff distance) of 0.145 inch, the impedance band is
very broad -- more than 20:l for a VSWR below 2:l. The
band ends depend on the inner and outer terminating radii
15 of the spiral. The feed was a broadband balun made from a
0.141 inch semi-rigid coaxial cable, which made a feed
radius of 0.042 inch. It was necessary to create a narrow
aperture in the ground plane in order to clear the balun.
The cavity's radius was 0.20 inch, and its depth 2
20 inches. This aperture also affects the high frequency
performance.
Other tests were performed using a log-spiral
(equiangular spiral) 0.3 inch above a similar ground plane
and balun. Both spirals, incidentally, were
25 "complementary geometries~.
The diameter of each spiral (the Archimedean and
the equiangular) was 3.0 inches, with foam absorbing
material (loading) e~tending from 1.25 to 1.75 inches from
center. If this terminating absorber is effective enough,
30 the antenna match can be extended far below the
frequencies at which the spiral radiates significantly.
More importantly, at the operating frequencies, the
termination eliminates currents that would be reflected
from the outer edge of the spiral and disrupt the desired
35 pattern and polarization. These reflected waves are
WO93/11582 ~/1$~ g PCT/US92/09439
~ ~ .
sometimes called "negative modes" because they are mainly
polarized in the opposite sense to the desired mode.
Thus, their primary effect is to increase the axial ratio
of the patterns.
For an engineering model, the Archimedean and
equiangular antennas operate well from 2 to 14 GHz, a 7:1
band. It is expected that the detailed engineering
required to produce a commercial antenna would yield
excellent performance over most of this range. The gain
10 is higher than that of a 2.5" commercial lossy-cavity
spiral antenna up through 12 GHz, as shown in Figure 12.
(We believe that the dip at 4 GHz is an anomaly.) The
increased gain of antennas of the present invention over a
lossy-cavity spiral antenna is in part attributable to the
15 relative lack of loss of radiated power from the underside
of the spiral mode antenna elements. The spiral mode
antenna element radiates to both sides, with radiation
from the underside passing through the dielectric backing
and the dielectric substrate relatively undiminished.
20 This radiation is reflected by the ground plane (sometimes
more than once) and augments the radiation emanating from
the upper side.
Fig. 12 also shows gain curves for a ground
plane spacing of 0.3 inch. The Archimedean version of
25 this design demonstrates a gain improvement over the
nominal loaded-cavity level of 4.5 dBi (with matched
polarization) over a 5:1 band. The gain of the 0.145 inch
spaced antenna is lower because the substrate was a
somewhat lossy cardboard material rather than a light foam
30 used for the 0.3 inch e~ample.
We have found that a decrease in thickness causes
the band of high gain to move upwardly in frequency,
subject to the limitation imposed by the inner truncation
radius. Figure 13 shows gain plotted at several
35 frequencies as a function of spacing for a "substrate" of
WO93/11582 4~ PCT/US92/09439
air. At low frequencies, the spiral arms act more like
transmission lines than radiators as they are moved closer
to the ground plane. They carry much of their energy into
the absorber ring, and the gain decreases.
For these types of antennas, we have found that
efficient radiation generally can take place even when the
spacing is far below the quarter wave "optimum". We have
observed a gain enhancement over that of a loaded cavity
for frequencies that produce a spacing of less than l/20
lO wavelength. If one is willing to tolerate gain
degradation down to 0 dBi at the low frequencies, as found
in most commercial spirals, the spacing can be as small as
l/60th wavelength.
We investigated several configurations of edge
15 loading, most notably foam absorbing material and magnetic
RAM (radar absorbing materials) materials. For the foam
case, we compared log-spirals terminated with a simple
circular truncation (open circuit) and terminated with a
thin circular shorting ring. There was no discernable
20 difference in performance. The magnetic RAM absorber was
tried on open-circuit Archimedean and log-spirals with
spacings of 0.09 and 0.3 inches. The results show that
the magnetic RAM is not nearly so well-behaved as the
foam. In addition to the gain loss caused by the VSWR
25 spikes, the patterns showed a generally poor axial ratio,
indicating that the magnetic RAM did not absorb as well as
the foam. In our measurements, the loading materials were
always shaped into a one-half-inch wide annulus, half
within and half outside the spiral edge. The thickness
30 was trimmed to fit between the spiral and the ground
plane, and in the very close configurations it was mounted
on top of the spiral.
This disclosure presents an analysis, supported
by experiments, of a multi-octave, frequency-independent
35 or spiral-mode microstrip antenna according to the present
W O 93/11582 2 ~ PC~r/US92/09439
-~1
invention. It shows that the spiral-mode structure is
compatible with a ground plane backing, and thus expla;ns
why and how the spiral-mode microstrip antenna works.
It is shown herein, both theoretically and
5 experimentally, that a high dielectric substrate has a
disruptive effect on the radiation pattern, and therefore
that a low-dielectric constant substrate is preferred in
wideband microstrip antennas. This finding may explain
why earlier attempts to develop a spiral microstrip
10 antenna have generally failed. It is also shown herein
experimentally that a conformally mounted spiral
micro rip antenna can achieve a frequency bandwidth of
6:1 or so.
"Spiral modes", as that term is used herein,
15 refers to eigenmodes of radiation patterns for structures
such as spiral and sinuous antennas. Indeed, each of the
spiral, sinuous, log-periodic tooth, and rectangular
spiral antenna elements disclosed he-ein as examples of
the present invention exhibit spiral modes. A
20 "spiral-mode antenna element" is an antenna element that
exhibits radiation modes similar to those of spiral
antenna elements. A mode can be thought of as a
characteristic manner of radiation. For example, Fig. 14
sh~ws .s~me typica] spiral modes for a prior spiral
an~nn~, and particularly shows modes n=l, n=2, n=3, and
n=5. Here, the axis perpendicular to the plane of the
antenna points to zero degrees in the figure. The "spiral
mode" antenna elements disclosed herein as part of a
microstrip antenna radiate in patterns roughly similar to,
30 though not necessarily identical with, the patterns of
Fig. 14. As shown in Fig. 14, the spiral mode radiation
pattern for n=l is apple-shaped and is preferred for many
communication applications. In such applications, the
donut-shaped higher order modes should be avoided to the
35 extent possible (as by using only two spiral ar~ or
su~ressed in some manner.
W093/ll582 PCT/US92/09439
2~ zg- -~
"Multioctave", as that term is used herein,
refers to a bandwidth of greater than 100%.
"Frequency-independent~, as that term is used herein in
connection with antenna elements and geometry patterns
5 formed therein, refers to a geometry characterized by
angles or a combination of angles and a logarithmically
periodic dimension (excepting truncated portions), as
described in R.H. Rumsey in Frequency Independen~
Antennas, supra.
To obtain near maximum gain at a given frequency,
the stand-off distance d should be between 0.015 and 0.30
~f a wavelength of the waveform in the substrate (the
~ielectric spacer). With regard to the relative
dielectric constant of the substrate, applicants have
15 found that materials with ~ of between 1 and 4.37 work
well, and that a range of 1.1 to 2.5 appears practical. A
higher dielectric constant (5 to 20) leads to gradual
narrowing of bandwidth and deterioration of performance
which nevertheless may still be acceptable in many
20 applications. This and other design configurations, which
operate satisfactorily for a specific frequency range, can
be changed so that the antenna will work satisfactorily in
another frequency range of operation. In such cases the
dimensions and dielectric constant of the design are
25 changed hy the well known "frequency scaling" technique in
antenna theory.
4. The Spiral-Mode Circular Array
Referring now to Figs. 15 and 16, the closed
array of the present invention is considered. As shown in
these figures, an antenna 60 is mounted above a ground
plane GP and includes a somewhat stiff, comformable
backing 61. The backing 61 is a unitary structure,
35 preferably made of printed circuit board material. The
WO93/l1582 ~r 1 ~ f ~ 9 PCT/US92/09439
q_
backing 61 is spaced above the ground plane GP by a
dielectric spacer 62 in accordance with the principles set
forth in the above numbered sections 1-3. A closed array
or series of patch elements 63, 64, 65, 66, 67, 68, 69,
5 and 70, is formed atop the upper surface of the backing 61
by conventional techniques, such as by photoetching.
Preferably, the array is circular, although what is
essential is that the array be "closed", i.e., is
generally of the form of a loop. While eight elements are
10 depicted in Fig. 15, a greater or lesser number of
elements can be used. In Fig. 16, the vertical dimensions
of the patch elements and of the backing are exaggerated
somewhat to make these elements more visually discernible
in the figure. The patch elements 63-70 are connected to
15 unshown electrical means for driving the individual
elem~rlts, the driving means being adapted to drive the
individual patch elements in a phased manner. The
electrical circuitry used to phase signals delivered to
the individual patch elements is well-known. In general,
20 the signal is split up into several signals and delayed or
phase-shifted an appropriate amount, by a network of
"hybrids" sometimes called a "processor", before being
delivered to the patch elements. Of course, the
inAividual patch elements 63-70 are electrically coupled
25 with the driving means in a manner similar to that shown
in Fig. 2B, i.e., through the use of cabling or in another
suitable manner.
The structure just described is extremely compact
and is well-suited for being used on the surface of an
30 object, for esample, on the surface of an airplane. The
antenna 60 with the array of individual antenna elements
63-70 has a small overall dimension for a bandwidth of 30
to 300%, depending on the diameter of the array. The
applicants have found that this arrangement allows the
35 antenna to be made substantially smaller than prior
antennas at a sacrifice of some bandwidth and some gain,
W093/ll582 PCT/US92/09439
2 ~ J --30
and that the smaller the diameter of the circular array,
the smaller the bandwidth. As compared with the spiral
arm antennas disclosed in the above-referenced co-pending
U.S. patent application, the present invention allows the
5 diameter of the antenna to be reduced by up to 2/3 or so.
When compared with other prior antennas, such as the
antenna arrays disclosed in the Munson IEEE paper, the
reduction in physical size is even more dramatic. This
reduction in size is achieved at a sacrifice of bandwidth
10 and perhaps even gain. However, for many applications, 30
to 50% bandwidth is sufficient; yet such a bandwidth
cannot be obtained by conventional microstrip patch
antennas. Thus, the spiral-mode circular array fills the
need for a conformable, low-profile, antenna with a
15 moderately wide bandwidth in the 30% to 300% range while
the array diameter can be only 1/2 to 1/3 the spiral
diameter.
The basic concept of a spiral-mode circular
phased array is shown in Figure 15. The circular array is
20 on a x-y plane which is treated as a horizontal plane
parallel to the earth. The array elements are on a circle
of radius a, and can be represented as either magnetic or
electric current elements, denoted by Jmn for the nth
element of mode m.
The current Jmn must have a polarization,
amplitude, and phase as follows:
mn
Jm=pJm c N n=1,2,3...N; formodem (41)
where p = cos ~ ~ + sin ~ y, p being a u~it radial vector in t~e cylindrical
coor~in~?s The patter~ of t}lis a2~ay r~m~inS t~e same if t~e polar~ sns of t~e
cum~t sources are c~anged to ~ at is, if
~5
mn
Jm=~J ~ j N ~ n=1,2,3...N; formodem (42)
S~ TITVT~. ~;nc-~
WO93/11~82 ~ L7~ ~ PCT/US92/09439
-3i-
When m~l, the radiatior -attern of this circular
array is apple-shaped as shown Fig. 17A. When m.2 or
higher, the radiation pattern is that of the doughnut
shape shown in Fig. 17B. Thus, this circular array can
5 provide the spatial coverage shown in Figures 17A and
17B. Now if two or more of these modes are combined, the
resultant pattern has a narrower steerable beam, as well
as one or more steerable nulls for noise or interference
reduction.
This multi-mode circular array alternatively can
be realized, as in the co-pending patent application, by a
multimode planar spiral, for which the radiation current
band theory is well known. However, the pl- ~r spiral
requires a much larger aperture, because its radiation
15 occurs on a circle whose circumference is m~ in length.
For esample the m-l mode of a planar spiral radiates on a
circumference of one wavelength (1~), and the m.2 mode
radiates on a 2~ circumference. Thus, for higher mode
numbers, the planar spiral can be unattractively large.
In the multi-mode circular array disclosed
herein, radiation occurs on the circle of radius a, where
the array elements are located. Theoretically, the array
radius a can be arbitrarily small. In reality, the
tolerance of the array becomes increasingly stringent as
25 the array diameter is reduced to below about 0.3~ for mode
1 and 0.6A for mode 2. By a simple array factor analysis,
one can show that the asial ratio deteriorates at angles
away from the antenna asis (z asis) and that the asial
ratio increases as the array size (in wavelength)
30 decreases.
As has been pointed out, a major advantage of
this spiral-mode circular array is its ability to radiate,
especially for higher-order modes (m > 2), on a smaller
aperture. For e~ample, to radiate an m~3 mode, a planar
35 sporal needs to have a circumference of more than 3~ (a
di~leter of 0.955~). For the mode-3 circular array, a A
WO93/l1582 PCT/US92/09439
~ J' '~ ~ _ 3~_
circumference (0.318A in diameter) is acceptable.
However, it has been observed that the tolerance
requirements on the feed networ~ becomes more and more
stringent for smaller apertures.
5. Bandwidth Covera~e of the Array Arrangement
Two techniques can be employed to expand the
bandwidth of the array to 10:1 or more:
(a) Concentric circular arrays,
as shown in Fig. 18, wherein four concentric
circular arrays are shown, only two of which are needed
for the breadboard model; and
(b) Element broadbanding.
The individual microstrip patch antenna is known
for its narrow bandwidth, typically 10% and often 3 to
6%. By increasing its effective cavity, the bandwidth of
a microstrip antenna can be increased. For example, with
a substrate of 0.318 cm, and a related permittivity of
20 2.32, the bandwidth at 10 GHZ is about 20%. In addition,
by having the patch elements closely spaced with each
other, the impedance bandwidth of the array can be made
much larger than that of the individual array elements.
By employing a dissipative loading similar to that of the
25 planar spiral or the circular array of loaded loops, a
bandwidth of 3:1 can be reached with a loss no more than
that of the cavity-loaded spiral antenna.
Although dissipative loss, perhaps on the order
of 2 dB, is an undesirable feature it is more than
30 compensated for by a higher gain from the antenna patterns
and the anti-jamming capability against noise. As a
result, the signal-to-noise ratio of the antenna disclosed
herein should be equivalent to the single-element low-gain
antennas with broad apple or doughnut beams.
CA 021244~9 1998-0~-0~
To broaden the tunable frequency bandwidth, one
can switch the effective length of a mic;ostrip antenna
with PIN diodes as shown in Fig. 19. This technique of
switching the effective length of a microstrip structure
5 has been esperimentally investigated and analyzed in some
instances. The high temperature limits for this
diode-switching device are yet to be determined.
6 . U~; no ~agnetic ~l~h~tr~te To R~l-Ce ~nt~nn~ r~; ~e
In a manner similar to that in the above-noted
co-pending patent application, we have determined that if
the substrate between the antenna element and the ground
plane is a magnetic material with egual relative
15 permittivity and permeability, the spiral modes can
radiate effectively. As has been shown in the co-pending
patent application, with a substrate having hiqh relative
permittivity (say, greater than 5) the antenna pattern
begins to deteriorate. However, when its relative
20 permittivity and permeability are equal, the substrate is
compatible with the spiral modes and therefore good
radiation patterns for each mode can be generated without
other unwanted modes that can disrupt the pattern. This
is depicted in Fig. 20 wherein antenna element(s) 72 is
25 positioned atop a magnetic substrate 73 having
substantially equal relative permittivity and
permeability. A loading material 74 is placed about the
periphery.
Now, if the relative permittivity and
30 permeability of the magnetic substrate are chosen to be a
higher number, say, 10, then the wavelength in the
substrate will be only 1/10 (10%) of that in free space.
This allows the antenna size to be reduced to 1/10
(one-tenth) of its size when using a honey-comb substrate
35 (relative permittivity and permeability being close to
unity).
W O 93/11582 A' ~ r ~ PC~r/US92/09439
1 2
-~4 -
Fig. 20 shows that a magnetic material is used as
the substrate 73 for the spiral-mode microstrip antenna.
By carrying out an analysis similar to that in Section 2,
we have demonstrated that if the relative permittivity ~r
5 equals the relative permeability ~r~ the structure shown
in Fig. 20 is compatible with the spiral modes. In other
words, when ~r_~r, the substrate is not expected to
disrupt the spiral modes as the ordinary dielectric
substrates do. (For an ordinary dielectric material,
10 ~r~l, while Er is a number larger than l; thus ~r'~r.)
Now if we use as substrate a material with ~r_~r
(~r-~r being highly unlikely), we can reduce the physical
size of the antenna by the factor
~ ~r ~r~ or approximately ~r (since ~r_~r). For example,
15 if we use a material with ~r~~r_10, we can reduce the
size of the antenna (both the thickness of the substrate
and the diameter of the frequency-independent element) by
a factor of 10. That is, we can reduce its size to 1/10
of its size when using a substrate with its permittivity
20 near that of free space (~r~ 1).
At present, no ready-made material with equal
relative permittivity and permeability appears to be
commercially available. However, custom materials can be
constructed by mi~ing grains of two materials to achieve
25 equal, or nearly equal, relative permittivity and
permeability. The size of the grains must be small in
comparison with wavelength (in the material), and must be
uniformly distributed to achieve homogeneity on a
macroscopic scale. For example, two different types of
30 cubes, one more dielectric and the other more magnetic,
and with their linear dimensions being identically equal
to 0.1 wavelength (in the material), can be alternately
spaced to appro~imate a homogeneous material of equal
relative permittivity and permeability.
W093/11582 PCT/US92/09439
_3~
Another method of making custom magnetic material
for substrate of equal ~r and ~r is to place electrically
thin dielectric and magnetic sheets parallel to the ground
plane alternately in a stack. (Sheets placed
5 perpendicular to the ground place should have similar
effects.) The stack then appears macroscopically to be
homogeneous with equal ~r and ~r . For example, sheets
with Er-3-jO.l and ~r~l can be alternately stacked with
sheets with ~r~ 1 and ~re3-jO.l to achieve this effect
10 (the imaginary part jO.l is related to the dissipation of
the material and is chosen to be small, jO.l is a
practical choice; other small numbers are acceptable.)
7. Varyin~ ~ffective Sllh~trate Thi~n~s Tn a mode-2
~nt~nn~
The physical size of a mode-2 an;enna, which
generally has a larger and more comple~ feed network, can
be reduced by varying the effective thickness of the
20 substrate. A simple coax feed at the center excites a
transmission-line wave propagating away from the center
along the spiral structure, thereby forming spiral modes.
In the region covered by a circle with a circumference
sliqhtly over one wavelength, the substrate is
25 sufficiently thin so that m-l radiation is minimal.
Outside this region the effective thickness of the
substrate is increased so that radiation of mode-2 is
effective.
The advantage of this mode-2 antenna is not only
30 a reduction in physical size, including that of its feed,
but also a reduction in cost, improvement in reliability
and greater structural simplicity.
As shown in Fig. 12, the gain of the spiral-mode
microstrip antenna drops sharply when the spacing between
35 the antenna element and the ground plane is decreased to
WO93/11582 PCT/US92/09439
~ 3~
below, say, 0.02 wavelength. This phenomenon is taken
advantage of in the following mode-2 antenna.
Figs. 21A and 21B show two versions of a simple
illustrative design in which the center conductor of a
5 coaxial line 76 is fed through a ground plane GP to the
center of a spiral structure 77. The two spiral arms
within the mode-l radiation region (where the
circumference is less than 1.1 wavelength) join at the
center with the center conductor of the coaxial line.
10 Also, the fine Archimedean spiral arms as shown in the
mode-2 region (outside the circumference of 1.1
wavelength) are broadened in the mode-l region. The
specific pattern of the broadening of the arms is not
critical as long as it transforms the impedance (usually
15 50 ohms of the coa~ cable at the center into the impedance
of the spiral microstrip structure.
Radiation in the mode-l region is minimized by
choosing dl, the spacing between the spiral element 77 and
the ground plane, to be electrically small (less than say,
20 0.02 wavelen~th).
However, as the wave moves outwardly from the
center of the spiral structure and enters the mode-2
region (where the circumference is greater than about 1.1
wavelength), effective radiation takes place because the
2S spacing d2 between the spiral element 77 and the ground
plane GP is now greater than about 0.05 wavelength. The
fact that the radiation occurs in the mode-2 region means
that the radiation pattern should be that of mode-2.
In Fig. 21A, the spacing between the spiral
30 element 77 and the ground plane abruptly changes from dl
in the mode-l region to d2 in the mode-2 region. In this
version, radiation in mode-2 is effective. However, the
abrupt increase in spacing for substrate thickness from d
to d2 causes undesired reflections.
~ CA 02124459 1998-05-05
,3?-
As shown in Fig 21~, the reflection betweenmode-l and mode-2 reqions is r~duc-d by employing a
tapered section to effect a gradual increa~e in substrate
thickness from dl to d2 However, th- mod--2 radiation is
5 not as effective at frequencie at which mode-2 regions
begins in the tapered transition region, since the smaller
substr~te thickness in the transition region suppresses
radiation
The taper between dl and d2 shown in Fig 21B can
10 be linear or of some other smooth curve, the selection of
which is a tradeoff among several considerations,
including technical performance as well as production cost
and ruggedness
It is well known that the eff-ct of the ground
15 plane on the mode-2 radiation is g-nerally negati~e
Therefore it i~ d~irable, whene~er po~sibl-, to reduce
the size of the ground plane and/or to make it convesly
cur~ed so that, for ~sampl-, tb- ground plane i~ a large
conducting -~phere and th- spiral is positioned outside it
The patch elements can compris- lossy components
for impedance matching.
While the invention has been disclosed in
preferred forms by way of esamples, it will be obvious to
one skilled in the art that many modifications, additions,
25 and deletions may be made therein without departing from
the spirit and scope of the invention as set forth in the
following claims
