Note: Descriptions are shown in the official language in which they were submitted.
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A POWER-CONSERVING LINEAR BROADBAND RF AMPLIFIER
Field of the Invention
This invention relates to radio frequency (RF) amplifiers and, more
particularly, to RF
amplifiers for use with limited battery supply voltage, such as employed in
hand-held
radiotelephone transceivers.
Background of the Invention
Hand-held radiotelephone transceivers are currently available for operation in
either
the 900 MHz or 1.8 gHz bands. Since the size and weight of a hand-held unit
must be kept to
a minimum, it is desirable to employ as small and low voltage a battery as
possible, typically
2.7 volts, and to employ circuits which reduce the battery current drain in
order to maximize
useful operating time. Such transceivers, however, also use a form of vector
or phase
modulation which requires that the RF amplifier stage preceding the final
power amplifier be
operated in class A linear mode to prevent phase shift error. Such a stage
typically employs
an emitter-follower at its output which must supply 1 milliwatt drive power
into the low
impedance (usually 50 ohms) presented by the single-ended filter leading to
the final power
amplifier. To maintain the RF amplifier stage in its linear, class A region,
where the peak-to-
peak excursion of the driving voltage cannot be permitted to clip the signal,
it has heretofore
been necessary to make sure that the transistors of the emitter-follower
output neither saturate
nor cut off. Accomplishing these goals has heretofore required that these
transistors draw a
large do standby current equal, theoretically, to at least half the peak-to-
peak output current.
In practical applications the value of standby current is much larger.
Also, because of the use of a low voltage battery, the RF amplifier stage is
limited in
the amount of voltage swing that can be delivered to the base of the emitter-
follower output
transistor, typically not more than about 300 millivolts peak-to-peak. Since
the load to be
driven is single-ended, a coupling arrangement such as an air core RF
transformer or a phase
shifting circuit is required to convert the differential output to a single
ended output.
Unfortunately, both types of coupling arrangements tend to operate best in
only one
frequency band. It is thus very difficult to have the same RF amplifier
circuit work at both the
900 MHz and 1.8 or 1.9 gHz bands.
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It would be extremely advantageous to provide an RF amplifier capable of
delivering
the required output power to a single ended load without requiring the use of
transformers,
without requiring the use of a large standby current to maintain amplifier
linearity and which
operates efficiently at more than one RF band so that a hand-held
radiotelephone transceiver
could in fact be operated at more than one, such as both the 900 MHz and 1.8
or 1.9 gHz
bands.
Summary of the Invention
The foregoing and other features of the present invention are achieved in the
illustrative embodiment of an RF amplifier suitable for use in a hand-held
radiotelephone
transceiver powered by a low voltage battery and in which the collector swing
of its input
signal from the previous mixer or multiplier stage is limited to a small
fraction of the battery
voltage. The limited swing, differential signal from the preceding stage is
applied to the
emitters of a pair of input transistors whose bases are coupled together. The
first of the input
transistors is diode-connected in series with a first constant current source
which provides the
bias to both input transistors. The second input transistor has its collector
coupled to the base
of an emitter-follower output transistor whose emitter is connected to a
second constant
current source. The differential voltages applied to the emitters of the two
input transistors
produce a large difference voltage at the base of the emitter-follower output
transistor. The
emitter of the emitter-follower output transistor provides a single-ended
output capable of
delivering the required power to a load without the need for a coupling
transformer and
having sufficient power to drive the load in Class A without requiring
excessive standby
current.
In accordance with one aspect of the present invention there is provided an RF
amplifier for driving a single-ended load impedance from a differential ac
input signal whose
peak-to-peak excursion does not exceed a small fraction of the battery voltage
without the use
of a coupling transformer, comprising: a pair of input transistors having
their bases
interconnected; means for coupling said differential input signal to the
emitters of said input
transistors; a first current source; means connecting the first of said input
transistors in circuit
with said first current source so that said differential input signal
effecting a change in base
voltage alters conduction in the second of said input transistors; a second
current source;
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an emitter-follower output transistor having its emitter coupled to said load;
means
connecting the collector of the second of said input transistors to the base
of said emitter-
follower output transistor; and means connecting the emitter of said emitter-
follower output
transistor to said second current source.
In accordance with another aspect of the present invention there is provided a
method
of driving a single-ended load from a Gilbert multiplier supplied by a low-
voltage battery,
said multiplier providing a differential RF voltage whose peak-to-peak value
is a small
fraction of said voltage, comprising: applying said differential RF voltage to
the emitters of a
pair of common-base configured transistors adapted to be powered by a battery
to effect
corresponding changes in their base voltage, said pair of transistors being
configured as a do
current mirror wherein a first of said transistors establishes the do current
through the second
of said transistors; and coupling the collector voltage of said second of said
transistors to the
base of an emitter-follower transistor adapted to be powered by said low-
voltage battery to
drive said load.
Description of the Drawings
The foregoing and other features of the present invention may become more
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apparent when the ensuing description is read together with the drawing in
which:
Fig. 1 is a simplified schematic diagram of a prior art mixer or multiplier
stage
followed by an RF amplifier stage having a pair of emitter-followers for
driving a low
impedance load through an RF transformer; and
Fig. 2 is a simplified schematic diagram of a portion of a transceiver showing
an
RF amplifier stage in accordance with an illustrative embodiment of the
invention which
employs only a single emitter-follower output and requires no RF transformer.
General Description
Fig. 1 shows a prior art circuit of the type employed in a hand-held
radiotelephone transceiver for operation at either 900 MHZ or 1.8 gHz
employing a
low voltage battery supply Vcc, typically 2.7 volts. At the left the rectangle
100
encloses an RF mixer or analog multiplier stage and the rectangle 101 at the
right
encloses a pair of emitter-followers that provides a differential output to a
low
impedance load L through a coupling transformer T 1. The circuit may be used,
for
example, at 900 MHZ with transformer T1 properly tuned for operation at that
frequency or, with a different transformer T1, it may be used at the higher
1.9 gHz
band since the same transformer cannot be used in both bands without serious
loss of
efficiency.
When configured for use as a mixer, a modulated intermediate frequency signal
is presented to terminals c, d and a local oscillator signal is applied _at
terminals a, bto
provide an RF output at terminals e, f at the right. When configured for use
as a
multiplier, an RF signal is applied at terminals c, d and a control voltage is
applied at
terminals a, b to provide an amplitude modulated RF signal at output terminals
e, f.
The circuit comprising transistors B22, B23, B24, B25 and including
transistors B 16,
B 17, B20 and B21 and resistors R19, R20, R30 and R31 have become known as a
"Gilbert" mixer or multiplier which is described in B. Gilbert, "A Precise
Four-
Quadrant Multiplier with Subnanosecond Response", IEEE Journal of Solid-State
Circuits, Vol. SC-3, pp. 365-373, December, 1968.
The circuit at the left-hand side of Fig. 1 is substantially a Gilbert mixer
circuit.
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Briefly, transistors B 16 and B 17 are current sources having their bases
connected to
the bias rail that provides a voltage reference establishing the emitter
currents for
transistors B20 and B21, respectively. Transistors B20 and B21 are in series
with or
"stacked" atop transistors B 16, B l7are an emitter-coupled pair. Resistors
R19 and
R20 connect the emitters of transistors B20 and B21 to provide linearizing
degeneration to overcome the variation in small signal emitter resistance with
emitter
current (illustratively, 26 mv/Ie) of transistors B20, B21. The values of
resistors R19,
R20 are chosen to be approximately ten times the value of the small signal
emitter
resistance.
Because the emitters of transistors B20, B21 are connected to constant current
sources, the ac collector current of either transistor B20 or B21 is equal to
the
differential ac input voltage applied at terminals c, d to their bases,
divided by the total
series resistance of resistors R19 and R20. The signal applied at terminals c,
d
modulates the amplitude of the current available to flow through transistors
B22
through B25 which are "stacked" above transistors B20, B21. When, for example,
the
ac signal applied at terminal d is positive-going with respect to terminal c,
transistor
B20 tends to carry more current and transistor B21 less current. Since the
current
through transistor B 16 cannot increase, a path is provided for the extra
current from
transistor B20 by interconnected resistors R19 and R20 so the current through
transistor B 17 may remain constant. The center point acg between resistors R
19 and
R20 is effectively at ac ground potential. . - - -
The modulated current from transistor B20 is applied to the junction of the
emitters of transistors B22 and B23 while the modulated current from
transistor B21 is
applied to the junction of the emitters of transistors B24 and B25. Signals
applied at
terminals a, b steer the modulated current from transistor B20 either to the
"outboard"
pair of transistors B22 and B25 or to the inboard pair of transistors B23 and
B24 to
produce differential voltage drops across resistors R30 and R31. When the
input signal
applied at terminal b is relatively positive with respect to terminal a, the
collector
current of transistor B20 tends to be steered through transistor B22 and
collector
resistor R30 while the collector current of transistor B21 tends to be steered
through
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transistor B25 and collector resistor R31. Likewise, with opposite polarity
signals at
terminals a, b , the collector current of transistor B20 tends to be steered
through
transistor B23 and collector resistor R31 while the collector current of
transistor B21
tends to be steered through transistor B24 and collector resistor R30.
Accordingly the
voltage drops across resistors R30 and R31 produce signals at terminals e, f
that are
four quadrant differential "products" or "mixes" of the signals at terminals
a,b and c,d.
These differential (ac) signals drive the emitter-follower output transistors
B26 and
B27 which apply their differential (ac) output currents to the left-hand
winding of
transformer T1, the center point of which, acg, is at an effective ac ground.
The circuit of Fig. 1 illustrates the "headroom" problem encountered with low
voltage battery sources arising from the finite collector-emitter drops of
series-
connected transistors. With a series stack of three transistors, e.g., B 16,
B20 and B22,
each of which has a base to emitter voltage drop, V,~ , of 0.9 volts, there is
very little
room for signal swing at the collector of transistor B22 and in practice the
peak-to-
peak value of this voltage swing is limited to about 300 to 325 millivolts.
That is the
maximum output voltage drive that the mixer 100 can make available at
terminals e, f
to the bases of emitter-follower transistors B26 and B27.
As mentioned above, it is a usual requirement that the RF amplifier be able to
deliver 1 milliwatt of ac output power to a single-ended 50 ohm load at
terminal x or y,
which equates to a 600 millivolt peak-to-peak swing. However with a voltage
swing of
only 325 millivolts peak-to-peak at terminal a or f, there is not enough
voltage swing to
produce 1 milliwatt in a 50 ohm load at terminal x or y to ground:
z
Pi E
R
(.7071 x .162 )z ~ 0.0002 watts.
To provide 1 milliwatt ac output power from either terminal x or y, the prior
art circuit
of Fig. 1 required that the current be doubled to 12 milliamperes (ma) by
reducing the
25 load resistance to 25 ohms or less. Accordingly, the 325 millivolt signal
at terminals e,
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f is applied to a pair of emitter-follower transistors B26 and B27.
Transistors B 18,
B 19 and resistors R 17 and R 18 provide a constant current source to the
emitters of
transistors B26, B27. To maintain transistors B26 and B27 in class A mode each
must
carry a quiescent, do standby current that is at least 12 ma in order not to
be cut off on
the negative half cycle of the ac drive signal. Actually, best linearity is
obtained when
the standby current is of the order of 15 ma do in each of transistor B26 and
B27 or a
total of 30 ma do even when no signal is present. This represents a large
current drain
on the battery supply. Moreover, since the load L is single ended, the
differential
signals at the emitters of transistors B26, B27 must be applied to a
transformer, such as
transformer T1, whose input winding receives the differential signals from
transistors
B26, B27 and whose single output winding drives the load. As mentioned above,
RF
transformers tend to be efficient couplers at only a narrow band of
frequencies and it
would be impractical to use the circuit of Fig. 1 for a multiband transceiver
such as one
that could operate at either 900 MHZ, 1.8 gHz or 1.9 gHz.
The problem is solved by circuitry 200 of Fig. 2 whose terminals e, f are
intended to be coupled to the correspondingly designated terminals e, f of the
Gilbert
mixer of Fig. 1, replacing the prior art circuit 101. It should be understood
that the
circuits of both Fig. 1 and 2 may be fabricated using any known VLSI
technology. The
limited swing, differential-input signal at terminals e, f from the preceding
stage is
applied via capacitors C1 and C2 to the emitters of a pair of input
transistors B5, B6
whose bases are coupled together in a common-base configuration: Input
tralTsistor B6
has its base and collector connected together to operate as a diode in circuit
with a first
constant current source Il. The do current through diode-connected transistor
B6
establishes the do current through transistor BS in a current-mirror
configuration, e.g.,
where resistors R6 and RS and transistors B6 and BS are identical, transistors
B6 and
BS have identical collector currents since their base-emitter voltages are the
same.
Input transistor BS has its collector coupled to resistor R4 and to the base
of emitter-
follower output transistor B3 whose emitter is coupled to a second constant
current
source I2. The differential voltages applied to the emitters of input
transistors B5, B6
produce a large difference voltage at the base of output transistor B3. The
emitter of
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output transistor B3 provides a single-ended output at terminal Z capable of
delivering
the required power to load L without the need for a coupling transformer and
which
operates in Class A without requiring excessive standby current.
The Thevenin equivalent circuit looking into terminals e, f of Fig. 1 is an ac
generator having an open circuit peak- to-peak voltage of 650 millivolts and
an
internal resistance approximately equal to the sum of the values of resistors
R30 and R
31, it is capable of delivering approximately 1 ma output at terminals e, f .
This current
is injected into the emitters of transistors B5, B6 of Fig. 2. When the ac
signal applied
to terminal f goes positive with respect to ground, it produces a
corresponding increase
in the base voltage of transistors B5 and B6. This occurs because diode-
connected
transistor B6 is maintained in conducting state by constant current source I1.
The rise
in base voltage of transistor B5 increases the current through its collector,
increasing
the voltage drop across collector resistor R4 at the base of emitter-follower
output
transistor B3, reducing the voltage drive to emitter-follower B3. At the same
time that
the ac signal at terminal f was assumed to be positive-going, the ac signal at
terminal a
is negative-going since the Gilbert mixer provides a differential signal. The
negative-
going ac signal at terminal a is applied to the emitter of transistor B5
increasing current
through the collector emitter path of B5 and thereby further raising the
voltage drop
across resistor R4 and further decreasing the drive voltage to the base of
emitter-
follower output transistor B3. Since the input signals at terminals e, f are
differential
signals, whenever one of which is positive going the other of which is
negative going,
the process just described converts the small voltage swing differential input
signal
applied at terminals e, f to a large voltage swing across resistor R4 and, in
the process,
converts the differential input to a single-ended output, providing a large
voltage drive
to the single-ended output transistor B3. The ac voltage swing across resistor
R4 is
substantially equal to the ac input current applied at terminals e, f ,
multiplied by the
current gain of B5, multiplied by the value of resistor R4.
In the illustrative embodiment the components had the following properties:
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Item Value
Vcc 2.7 volts
C1, C2 3 pF
R4 SOOS2
R5, R6 20052
R15, R16 15052
R19, R20 225 52
R30, R31 32552
L 125 SZ
What has been described is deemed to be illustrative of the principles of the
invention. It should be apparent that whilst, in the illustrative and
preferred embodiment,
constant current sources have been shown at I1 and I2, either or both thereof
may be
replaced by resistances of suitable size and, further, that the signal on
terminal f is
shown as being applied via capacitor C1 to the emitter of diode-connected
transistor B6,
it is also possible with somewhat different results, to connect the signal
through
capacitor C1 to the collector of transistor B6. Also, unipolar transistors may
be
substituted for bipolar transistors in which case the term emitter will be
replaced by the
term source, the term base will be replaced by the term gate, and the term
collector
replaced by the term drain in the foregoing description. Numerous other
modifications
may be made by those skilled in the art without, however, departing from the
spirit and
scope of the invention.
