Note: Descriptions are shown in the official language in which they were submitted.
CA 0223~101 1998-06-12
METHOD AN~ APPARATUS FOR SCRAMBLING AN~ DESCRAMBLING
OF VIDE~ SIGNALS WITH EDGE FILL
This is a divisional of Canadian Patent
Application Serial No. 2,082,675 filed November 12,
1992.
BACKGROUND OF THE INVENTION
Field of the Invention
This invention relates to signal processing of time
10 domain electronic signals, such as video information
signals. In particular, the invention relates to
improvements in scrambling and descrambling such signals
to prevent unauthorized use thereof, including several
improvements in security and concealment.
15 Description of Related Art
Commonly assigned U.S. Patent No. 5,058,157 issued
October 15, 1991 to John O. Ryan -
- - discloses a method and apparatus for encrypting
(scrambling) and decrypting (descrambling) information
20 signals normally arranged as a succession of lines of
active information, with each line having a line timing
reference, such as color video (television) signals. The
active video portion of each line is time shifted with
respect to the horizontal sync portion of that line using
25 a predetermined slowly varying time-shifting function.
The time-shifting information is conveyed to the
decryption site by encoding the instantaneous value of the
time-shifting waveform for the beginning of each field in
the vertical blanking portion of that field. To provide a
30 reasonable maximum time-shifting range, portions of the
trailing edge of the active video in the preceding line
and portions of the leading edge of the active video in
the current line are discarded. During decryption, the
original line timing and colorburst signals are discarded
35 and new signals are generated which are time displaced
CA 0223~101 1998-06-12
from the active video portion by the original amount
before encryption. This provides a secure video type
information encryption and decryption technique compatible
with all video tape formats and transmission systems, and
5 which is free of picture impairments caused by the
interaction of the scrambling algorithm and the
chrominance consecutive line averaging systems used in
color-heterodyne recording.
The type of time shifting performed may comprise any
10 one of a number of slowly varying functions, such as a
sinusoidal waveform or a linearly changing ramp signal.
The rate of change in the signal, i.e. the "wobble", is
relatively slow when compared to the line rate of the
input signals to be processed. For video type signals, a
15 sinusoidal waveform having a frequency of no more than
about 20 Hz is used. The absolute amount of time shifting
performed is preferably limited to a maximum value which,
in the case of NTSC video signals, does not exceed a total
of 4 microseconds (plus or minus 2 microseconds in each
20 direction).
The instantaneous value of the time shifting waveform
function at the beginning of each field is conveyed along
with the field information, typically during the vertical
blanking interval. For example, with respect to a sinu-
25 soidal time shifting function, the starting amplitude ofthe waveform during a given :f ield is transmitted during
the vertical blanking interval as a single byte of
information which, when combined with a separately
provided authorization key, enables a descrambling circuit
30 to synthesize the scrambling waveform function.
Decryption is performed by restoring the original timing
relationship between the horizontal sync (and colorburst)
and the active video portion of the corresponding line.
This is done by generating new line timing reference
35 signals (horizontal sync and colorburst) which bear the
same timing relationship to the active video portion as do
the original line timing reference signals before
CA 0223~101 1998-06-12
encryption. The resulting descrambled signals still
contain time base errors, but these errors are within the
capture or correction range of the follow-on television
monitor/receiver.
Figures lA and lB herein correspond to Figures 3A and
3B of the above cited U.S. Patent No. 5,058,157, and
illustrate the manner in which the scrambled signals are
descrambled at the reception site, i.e. the descrambler.
With reference to Figure lA, three successive lines of
10 NTSC video are shown which have been time shifted
successively by increasing amounts. (The active video
portions of each of the lines in Figures lA and lB are
only fractionally illustrated.) The topmost line
represents a line N having no time shifting between the
15 active video portion and the end of horizontal blanking,
and the time between the beginning of a horizontal sync
portion and the active portion is designated as tl. The
next line N+1 has undergone time shifting in the delay
direction so that the time between the beginning of the
20 horizontal sync portion and the beginning of active video
portion is t2, greater than tl. Line N+2 has undergone
even more time shifting in the delay direction by an
amount labeled t3 which is greater than t2. These three
successive lines represent llnes from the upper portion of
25 a raster image. The line timing reference part of each of
lines N, N+1 and N+2 are all temporally aligned in Figure
lA; the leading edge of the horizontal sync portion of
each line is exactly aligned with the lea~ing edge of the
horizontal sync portion of the other lines. The same is
30 true of the location of the colorburst portions (hatched
areas). The active video portions, however, are
deliberately misaligned in lines N+1 and N+2 with respect
to line N.
Figure lB illustrates the signals for the same three
35 lines after descrambling, i.e. decryption. As can be seen
in this figure, the leading edges of the horizontal sync
portions of the three lines are no longer precisely
CA 0223~101 1998-06-12
aligned, but are rather staggered; however, the distance
between the leading edge of the horizontal sync portion
and the beginning of active video is the same for all
three lines, i.e. the value t1. Similarly, the colorburst
5 portions (hatched areas) of the three lines are no longer
temporally aligned, but are rather staggered in the same
fashion as the horizontal sync portions. Relative
positioning of the active video portion of the three lines
remains the same.
Although the descrambled signals are still relatively
misaligned, the precise timing relationship tl between the
leading edge of horizontal sync and the beginning of
active video ensures that each line of information, as
processed by the follow-on television receiver or monitor,
15 can be properly displayed, provided that the timing error
in a given line does not exceed the capture range of the
television receiver or monitor synchronization circuitry.
The time shifting applied to the original signals during
encryption is relatively slowly varying (20 Hz for NTSC
20 TV) compared to the video line rate.
Figures 2A and 2B herein are the same as Figures 4A
and 4B of U.S. Patent No. 5,058,157. These figures show
in block diagram form a scrambler system capable of
providing the above-described scrambling. As seen in
25 Figures 2A and 2B, input video to be scrambled is coupled
to an input terminal 11 of a video input processor unit
12. Processor 12 functions to normalize the incoming
video signal relative to gain, DC offset and bandwidth and
provides a stable low impedance buffer unit for the video
30 appearing on output terminal 13. In addition, the
incoming vertical and horizontal sync portions are
separated from the input video by processor unit 12 and
supplied as input to a sync/timing generator and phase
locked loop 15.
The signals from processor unit 12 appearing on
output terminal 13 are coupled to a conventional NTSC
decoder and anti-alias filter 16 in which the luminance
CA 0223~101 1998-06-12
component Y and chrominance quadrature components I,Q are
separated for three channel parallel processing in the
digital domain. The Y output of unit 16 is coupled to an
analog-to-digital converter 18 in which the luminance is
5 converted from analog to digital form at a preselected
clock rate by means of an input sample clock signal
supplied on clock output line 19. The input of converter
18 is coupled to an input portion of a dual-ported
luminance memory unit 20. This is then the Y channel
10 memory connected to the Y channel D/A converter 22.
Memory unit 20 is configured as a memory in which a word
is written from A/D converter 18 into every memory cycle
and a word is read from memory unit 20 to a digital-to-
analog converter unit 22 every-memory cycle.
Read/write control signals and multi-bit address
signals are supplied to the luminance memory unit 20 from
a memory controller unit 24. The output of luminance
channel memory unit 20 is coupled to the input of a
digital-to-analog converter 22, in which the multi-bit
20 digital words output from memory 20 are converted into
analog samples at the clock rate by clock signals supplied
from unit 15 on clock input line 23. The output of
converter unit 22 is coupled to the input of an NTSC
encoder and low pass filter unit 25 in ~hich the luminance
25 signal is combined with the I and Q chrominance components
and renormalized with respect to band-width and DC offset.
The I, Q chrominance quadrature components are processed
in an essentially identical manner to that described above
for the luminance component Y in respectively units 18',
30 20' and 22' and 18", 20", and 22", which function in the
same manner respectively as do units 18, 20 and 22.
Sync timing unit 15 generates the input clock signals
used to provide the sample cloclc for A/D converter unit
18, the read and write clock signals from memory unit 20,
35 and the cloc}c signals for D/A converter unit 22.
Preferably, unit 15 is comprised of a discrete phase
detector, a number of sampling gates, and error amplifier
CA 0223~101 1998-06-12
and a crystal clock oscillator.
The above described units are coupled to a user
interface device 32, such as a keyboard terminal, via
controller unit 34 and a plurality of control registers
5 36.
The above described device and the associated
scrambling method have several shortcomings.
First, the device is relatively expensive and
complicated in that there are three sets of A/D converters
10 and associated memories, one for each of the Y, I, and Q
components. Thus, there are three independent channels
for digital processing, each channel requiring relatively
expensive components, thus increasing the cost and
complexity of the scrambling device.
Secondly, the method of scrambling as depicted in
Figures lA and lB while reasonably secure has the
potential defect that in the process of moving the active
portion of the video to the right as shown in the
drawings, the leading and trailing edges of the horizontal
20 sync signal have both been moved to the right also. This
displacement of the normally well known position of
horizontal sync within the horizontal blanking interval
could be detected by a clever pirate, i.e. unauthorized
user, to determine the amount of wobble (time
25 displacement) in each line. The pirate would be able at
least in theory to descramble the signal to determine what
the amount of wobble and reverse the process, thus
obtaining a descrambled and viewable signal. Thus, the
method as depicted in Figures lA and lB is lacking in the
30 very high degree of security desirable for a commercial
scrambling system.
Another shortcoming of the above-described scrambling
system is that while providing security, i.e. generally
preventing unauthorized use, the scrambled signal when
35 viewed on a normal television set is not completely
concealed. That is, a determined viewer who is willing to
watch a television picture which is in effect horizontally
CA 0223~101 1998-06-12
jumping back and forth can still watch the program and
understand at least partly what is going on. This is
undesirable for transmission for instance of adult type
material where it is desired to prevent children from
5 watching even the scrambled picture. This is especially
problematic because it has been determined by
experimentation that such adult type material, i.e.
depictions of sexual activity, is particularly easy to
follow on the picture by a viewer even though the picture
10 is scrambled. This is another way of saying that the
scrambling while relatively secure does not provide an
adequate level of concealment for all program material.
Another problem associated with the above described
device is one common to comb-type NTSC decoders in which
15 the composite video is subjected to a one-line delay.
Simple addition of the delayed video to the same video
before the delay causes the chrominance portion of the two
signals to cancel, leaving only luminance. Similarly and
simultaneously, subtraction of the delayed signal from the
20 undelayed signal (or vice versa) causes the luminance
portion to cancel, leaving only chrominance. This problem
is not specific to a scrambling device but is typically
encountered in video processors which perform NTSC
decoding and is the reduction of vertical detail,
25 resulting in smeared vertical edges in the picture. This
is due to the two-line summation process of the Y, I and Q
components in which fine grain picture detail tends to be
lost when the composite video is converted to digital and
then in the digital domain a luminance/chrominance
30 separation is performed. It is known that this problem
can be overcome by complicated and expensive circuitry
which takes the incoming composite video signal in the
analog domain, using a band pass or high pass filter to
isolate the chrominance component before separation. The
35 band pass filtered signal is then delayed and subjected to
the subtraction process. The band pass filtering removes
the vertical luminance edges because they are low
CA 0223~101 1998-06-12
frequency in nature. Thus the chrominance separation is
performed only on the high frequencies and having done
this, the resultant separated chrominance has no luminance
component. Finally the luminance signal is isolated by
5 subtracting the finished, high frequency chrominance
signal from the incoming composite video so there is no
loss of vertical detail. This process is effective but
when done digitally requires two A/D conversions: one for
the band passed (or high passed) chrominance and one for
10 the broad band composite video. It would be desirable to
eliminate or simplify this process in order to reduce the
number of components needed and reduce the amount of
processing on the signal.
Thus, the method and apparatus disclosed in the above
15 cited patent while adequate is still subject to
significant improvement in both security, concealment, and
complexity.
It is to be understood that the above cited patent is
commonly assigned with the present invention and that the
20 above description is not an admission that the subject
matter disclosed and claimed in the'above cited patent is
necessarily prior art with respect to the subject matter
of the present disclosure and claims.
SUMMARY OF THE INVENTION
A scrambler and descrambler are provided in
accordance with the invention to overcome the above
described shortcomings of the method and apparatus
disclosed in the above cited patent and also to provide
improved security and concealment and greater flexibility
30 i.e. additional applications~
In accordance with the invention, first the problem
of the easily detected edge of horizontal blanking in the
scrambled signal is overcome by filling in the gap between
the position where active video would nominally start or
35 end and where it actually starts and ends due to the
scrambling process of Figure lA. This gap is filled with
CA 0223~101 1998-06-12
signals which are undetectable compared to the eXpected
active vldeo, preventing the pirate from ~uilding a device
which would be able to track electronically the transition
at the gap. This "edge fill" process recognizes that for
5 four adjacent (successive) pixels, the luminance will
generally be relatively static and the chrominance will
describe one complete cycle in these four pixels. Thus ~f
at the edge of the active portion of the video, i.e. the
edge of the picture, four adjacent pixels are repeated in
lO sequence, this provides a continuous unbroken chrominance
sine wave with exactly the phase and amplitude of those
four sampled pixels and a luminance signal which mimics
that of the four pixels. This in effect generates a
continuous signal which matches the single set of four
15 pixels for as long as is desired, i.e. the sequence of
four pixels may be repeated as long as desired to fill the
gap. A number of pixels other than four may also be used.
In accordance with the invention this edge fill is
provided in the digital domain by stopping the normal
20 incrementation of the address counters for the eight most
significant of the ten bits which define the location of
each pixel, while the two least significant bits continue
to run, i.e. to increment normally. Thus the eight
"frozen" most significant bits define a set of four
25 pixels, and the two running least significant bits cycle
through those four. Thus at the right edge of the picture
where the active video has been moved to the left due to
the scrambling process (see Fig. lA) opening a qap at the
right edge of the picture, if the address is subject to its
30 normal incrementation until the edge of the picture
is reached, at that point the eight most signlrlcant bits
in the pixel value are stopped and the two least
significant bits are allowed to run. This generates a
repetitive set of four pixels defined by the changing of
35 the two least significant bits. This "pseudo-active"
video is stretched as long as desired, defining the
desired pseudo-active video fill at the edge of the
CA 0223~101 1998-06-12
-- 10 --
picture, i.e. at the end of the active video portion of
one line. Similarly, at the left edge of the picture, the
eight MSB's are set to the address of the first four
pixels of the active video and the two LSB's allowed to
5 run. This "anticipates" the video content of the left
edge of the picture in a manner identical to that
described above; when the left-edge gap has been filled,
the eight MSB's are released or "unfrozen" and the address
increments normally.
In a refinement of this edge fill process, to prevent
a clever pirate from detecting that at the edge fill each
pixel would be identical, a low level and low frequency
noise signal is added, i.e. a random noise overlay, so as
to slightly disturb the steady state which is the repeated
15 pixel data. This insertion of what amounts to edges, i.e.
variation, in that portion of the active video would
prevent the pirate from distinguishing the edge fill from
the active picture by observing where there is no change.
This random noise is provided on luminance and also both
20 portions of the chrominance (for Y, I and Q). Otherwise,
an extremely clever pirate could decode all three signals
and look for a steady state on any one of them. This
random noise is provided in either the digital or analog
domain in various embodiments of the invention.
Also in accordance with the invention, the problem of
the relative complexity of the three channel NTSC digital
decoder as described above and shown in Figures 2A and 2B
is overcome by a simplification which only separates
luminance from chrominance, thus using only two channels
30 instead of three; thus Y, I and Q instead become only
luminance which is Y and chrominance designated C.
Luminance is then caused to "wobble" (time shift) in terms
of the location of the beginning of active video in each
line directly, with the separated chrominance portion
35 identically wobbling and then processed by a heterodyne
circuit which frequency stabilizes the chrominance. Thus
the processing is done in only two channels - luminance
CA 0223~101 1998-06-12
and chrominance, which saves substantially on the amount
of expensive circuit elements provided and improves the
tracking of chrominance with respect to colorburst.
Additionally, advantageously the reduction of the amount
5 of encoding and decoding for NTSC signals reduces the
generation of undesirable artifacts in the picture.
In accordance with another aspect of the invention, a
sync "wiggle" is provided as a concealment overlay to the
main timeshifting scrambling system. This means that
10 there is a further modification of the video signal which
obscures the picture when viewed prior to descrambling,
thus preventing an unauthorized person from seeing
anything recognizable on the picture if he attempts to
view the picture without having descrambled it. This
15 concealment overlay involves in one embodiment time
shifting in a pseudo random fashion the location of the
horizontal sync signal in each line. In another
embodiment, the sync signal location is wiggled in a more
complex fashion by using two non-fixed, i.e. genuinely
20 random patterns, hence moving the location of the sync
signal in a more complex fashion.
In another version, the sync wiggle also includes a
vertical sync wiggle, i.e. the locations of the vertical
sync signals in the vertical blanking interval are wiggled
25 in pseudo-random or other fashion in addition to the
horizontal wiggle, thus adding a vertical rolling effect
to the appearance of the scrambled picture when displayed
on a television screen. The resulting pa'terns tend to
roll through the picture vertically, obscuring the picture
30 in two dimensions. In one embodiment the vertical wiggle
is performed in a four field sequence in which in the
first field the vertical sync is reinserted so as to be
advanced, i.e. move forward in time. In the second field
the vertical sync is removed. In the third field the
35 vertical sync is reinserted so as to be moved far back,
i.e. to the right in time as far as possible. In the
fourth field in the sequence again the vertical sync
CA 0223~101 1998-06-12
signal is removed. This process confuses both newer type
television sets which have line counters which attempt to
track the moving vertical sync and hence will become
confused developing a jump, and also, will confuse the
5 older type television sets in which the absence of the
sync pulse will cause uncontrollable rolling again
completely obscuring the picture.
In accordance with another aspect of the invention,
the above-described problem of loss of vertical detail in
10 NTSC decoders is overcome by recovering the missing
vertical detail from the pre-existing chrominance channel.
This is performed by providing as described above the two
separated signals chrominance ("chroma") and luminance
("luma"). The chroma signal also contains the "missing"
15 luma vertical detail. The chroma channel data in the
digital domain will in any case be converted to analog
form. Then in all portions of the video picture outside
the vertical blanking interval the chroma information is
low pass filtered to remove the chrominance itself,
20 leaving the missing vertical detail information, which is
simply added back into the luma analog signal. This
restores the missing vertical detail without any extra,
expensive digital processing or extraneous A/D conversion
steps.
In accordance with another aspect of the invention,
the predetermined slowly varying time shifting function
(the wobble) is provided by digitally generating a low
frequency, randomly frequency modulated sine wave. This
is done by operating a digital counter from a randomly-
30 varying clock source, and applying the output of the
counter as an address to a programmable read-only memory
(PROM) which holds the sine wave function. Thus at each
step the PROM outputs a digital word representing one
point on the sine curve. This data is then applied to a
35 digital analog converter providing an analog output
signal. In accordance with the invention the above
described sine wave would in most cases provide excessive
CA 0223~l0l l998-06-l2
- 13 -
amounts of information which it would not be possible to
process. Thus in accordance with the invention instead
the sine wave is sampled at the field rate, i.e. 60 Hz,
and only the samples are transmitted. Then the decoder
5 may reconstruct the same sine wave from the sampled data.
The predetermined slowly varying time shifting
function, as described above~ varies at about 20 Hz or
less which is less than half the 60 Hz sample rate,
thereby allowing perfect reconstruction of the original
10 sine wave in accordance with well-known sampling theory
requirements. Thus in accordance with the invention the
digitally generated sine wave from the PROM is latched
once per video field, i.e. at 60 Hz, into a latch which
holds the digital word to suitable precision over the
15 timed duration of the entire video field. Then once in
each video field the data is read out from the latch and
applied to a D/A converter, thereby generating an analog-
domain version of the sampled sine wave. This wave form
is smoothed by a conventiona] filter and applied to the
20 circuitry which controls the position of the picture in
the output video signal.
Simultaneously, the latched digital data is
transmitted to the decoder, which then performs in the
same order the similar function in extracting the digital
25 word, latching this word and holding it over one video
field and applying the data to a digital to analog
converter, thus providing the output function which allows
descrambling of the signal. The step approximation then
is smoothed through an identical RC filter to that in the
30 encoder, thereby restoring an analog type sine wave
matching that in the encoder. The decoder has then merely
to regenerate horizontal sync, horizontal blanking, and
color~urst signals in accordance with the
recovered/reconstructed sine wave and insert them into the
35 received scrambled video wave form to complete the
descrambling process.
In accordance with another aspect of the invention,
CA 0223~l0l l998-06-l2
- 14 -
for use with a dedicated video system (non network
compatible) the time-shift wobbling of the active video
portion of the line is allowed to result in a narrowed
horizontal blanking interval. This allows maintenance of
5 the full width of the active video and hence an improved
picture at the picture edges. Although such a system
would not be compatible with for instance broadcast or
cable television, it would be suitable for use with
dedicated system such as a video cinema application where
10 transmission is not required, but where it is desired to
retain the full width of the picture. Thus it is possible
to reduce the duration of the horizontal blanking interval
to provide space for the active portion of the video i.e.
the picture to "wobble into" without having to lose any
15 portion of the signal thereby. If the width of a
horizontal blanking interval is thus reduced, there is
still the necessity to transmit an adequate length
colorburst portion of the signal. This is performed in
accordance with the invention by locating a portion of the
20 colorburst on the front porch of the horizontal blanking
interval and the remainder of the colorburst
conventionally on the rear porch of the horizontal
blanking interval. In a variation, the colorburst can be
started on the front porch and allowed to continue all the
25 way through the sync pulse.
Also in accordance with the invention a video
inversion picture concealment method is provided wherein
on a line-by-line basis particular lines of the picture
are inverted in random fashion, i.e. the light portions
30 are dark and the dark portions are light. The pattern of
which lines are inverted or not inverted changes in random
fashion at a predetermined rate so as to offer the desired
degree of concealment. Such video inversion is also
relatively secure in the sense that it would be difficult
35 for a pirate to determine in real time whether any
particular line is inverted or not and hence it would be
difficult to defeat this concealment technique.
CA 0223~101 1998-06-12
Also provided in accordance with the invention is a
vertical wobble in the active video in the sense of time
displacement of the location of the vertical blanking
interval in a slowly varying fashion in succession video
5 fields, analogously to the previously described horizontal
wobble.
BRIEF DESCRIPTION OF THE FIGURES
Figures lA and lB show scrambled and descrambled
signals as disclosed in U.S. Patent No. 5,058,157.
Figures 2A and 2B show a block diagram of a scrambler
unit as disclosed in U.S. Patent No. 5,058,157.
Figure 3 shows a flowchart of the scrambling process
in accordance with the present invention.
~ igure 4 shows a block diagram of a scrambling device
15 in accordance with the present invention.
Figure 5 shows a block diagram of random noise
overlay circuitry in accordance with the present
invention.
Figure 6 shows a block diagram of the input board
20 portion of the circuitry of Figure 4.
Figure 7A shows a block diagram of the control board
portion of the circuitry of Figure 4.
Figure 7B shows the wide band oscillator portion of
the circuitry of Figure 7A.
Figure 8 shows a block diagram of the RAM board
portion of the circuitry of Figure 4.
Figure 9 shows a block diagram of the output board
portion of the circuitry of Figure 4.
Figures lOA, 10B, 10C, 10D, and lOE show spectra
30 depicting a heterodyne function performed by the circuitry
of Figure 9.
Figure ll shows in block diagram form another version
of the heterodyne circuitry of Figure 9.
Figure 12 shows a flowchart of the descrambling
35 process in accordance with the present invention.
Figure 13 shows a block diagram of a descrambler in
CA 0223~101 1998-06-12
- 16 -
accordance with the present invention.
Figures 14A, 14B, 14C depict waveforms relating to
the descrambler of Figure 13.
Figures 15A, 15B, 15C show the use of pre-burst in
5 the scrambler of Figure 4.
Figures 16A, 16B show block diagrams of descramblers
using digital HBI synthesis and interpolation in
accordance with the invention.
Figure 17 shows a vertical wobble, i.e. scrambling
10 function, in accordance with the invention.
DETAILED DESCRIPTION OF THE INVENTION
SCRAMBLER OPERATION
Throughout the description following, specified
parameters ("eight-bit", "ten-bit", "0-909 counters" etc.)
15 refer to the preferred embodiment of the invention in the
particular case of a 4-times subcarrier frequency sampled
NTSC-standard unit. The principles described herein are
generally applicable to other standards (such as PAL), and
other sampling rates by applying detail modifications in
20 accordance with principles well known to those versed in
the art.
Figure 3 is a flowchart showing scrambling in
accordance with the invention as performed in the
scrambling device. Beginning at step 42, the incoming
25 analog video signal is digitized and written into a
conventional random access memory. In parallel, the
conventional random-number generator gene-ates a randomly
varying number in step 44. Then the randomly generated
number is converted to an analog waveform and the waveform
30 is used to make the wobbling time base for purposes of
video encryption in step 46.
Step 46 generates an analog waveform in conjunction
with the randomly frequency modulated sine wave which is
used to generate a wobbling time base. In an alternative
35 embodiment, the wobbling time base can be generated
digitally, identically to the process described for the
CA 0223~101 1998-06-12
digital descrambler (Fig. 16A). The 1:1880 counter 588
referenced there provides precisely the desired address
bus. In step 48, for the digitized video previously
written into memory, the luminance Y is separated from the
5 chrominance C by an addition and subtraction process and
both of the signals Y, C are read out of memory with the
same randomly wobbling timing. This provides both
luminance and chrominance which are wobbling. Since there
is intrinsically a one line delay in this separation
10 process of step 48, the vertical blanking interval video,
which is not scrambled but needs to match time-wise with
luminance and chrominance, is delayed by one line in step
50 so as to maintain its time alignment with the luminance
and chrominance portion of the signals.
This provides three signals: luminance, chrominance
and vertical interval. The luminance signal is wobbling,
the chrominance signal is wobbling, and the vertical
interval signal is stable in terms of time. There is need
then to restore the phase stability of the chroma signal,
20 as is done in the heterodyne process of step 52. Then the
composite video signal is reconstructed in step 60 by
putting the chroma back on the luma, reblanking, then
generating sync. Thus the wobbling luma and the phase
stable, heterodyned wobbling chroma are combined and in
25 the vertical interval the output signal is switched over
to the stable vertical interval output signal from step
54. In the horizontal interval there is a need to
synthesize a sync pulse which wiggles for purposes of
concealment. This is done in step 58 where there is
30 synthesis of a position modulated horizontal sync for
concealment overlay. This synthesized sync is then added
into the composite video signal in step 60.
There is also a need to transmit the information
needed for descrambling to the decoder device (not shown).
35 Thus the randomly varying number from step 44 is latched
into one line of the vertical blanking interval in step
56. This data is formally encrypted by conventional means
CA 0223~101 1998-06-12
- 18 -
in order to prevent a pirate (unauthorized person) from
extracting the randomly varying number.
Figure 4 shows a block diagram of a scrambler for the
process of Figure 3. Beginning in the upper left hand
5 portion of Figure 4 the conventional input video signal is
input to frame input buffer 66- The conventional clamping
and AGC (automatic gain control) processes are performed
in block 72. Conventional genlock is performed on the
input video signal in block 68 by operating a crystal
10 oscillator at four times the subcarrier frequency which is
phase-locked to the incoming colorburst signal. The
output of the genlock circuitry 68 is then the write clock
signal. The incoming video signal is applied to a video
analog-to-digital converter 74, outputting from A/D
15 converter 74 a digitized video signal.
This digitized video signal is then applied to a
digital system which includes a one horizontal line buffer
76 providing a one video line delay. Both the input and
output of the buffer 76 are provided to adder block 78 and
20 summed therein digitally. The output of adder block 78 is
the Y (luminance) signal. Similarly the input and output
of buffer 76 are subtracted in subtractor block 80, the
output of which is the C (chrominance) signal. Thus the
input video signal is separated into a two channel signal
25 i.e., chrominance and luminance. The outputs of adder 78
and subtractor 80 are still stable in terms of time. The
output of adder 78 is provided to buffer 82 which is also
one video line in length. The output of ~ubtractor 80
(which is the chrominance signal) is provided to buffer 84
30 which is also one video line long.
Each of buffers 76, 82, and 84 are so-called "ping-
pong RAM's" i.e., dual banked random access memories
providing double buffering. Thus each of these buffers
includes two random access memory banks, one of which is
35 written into and on the next video line the second one is
written into while the first one is being read out from.
In alternate embodiments, any suitable "real-time" FIFO-
CA 0223~101 1998-06-12
~ -- 19 --
type memory or shift register may be used.
Write address block 70 receives the write clock
signal from genlock circuitry 68- Thus the write address
black is locked in time to the input write clock signal,
5 and thereby to the input video signal- That write clock
signal is four times the subcarrier frequency and is
stable so that the luminance and chrominance are written
in a stable synchronous fashion to the buffers 82 and 84
respectively.
In order to write to the buffers 82 and 84 three
steps are taken. First, it is necessary to apply the
signal data which it is desired to write. Secondly, one
must supply addresses of the location in the buffers to
which the data is to be written to. Third, it is
15 necessary to tell the buffers when the writing is to take
place. Thus the write address block 70 also provides the
write timing. There is a 10 bit wide address bus from
write address bloc~ 70 and also a clock line connecting
block 70 to each buffer 82 and 84. The write address
20 system i.e., both the address bus and the clock, are
stable with respect to the input video signal. The read
address (which is provided from read address circuitry 94)
and its corresponding clock signal are wobbling in time as
described below.
Thus when the contents of the luminance buffer 82 and
the chrominance buffer 84 are output with the wobbling
address signal from read address block 94, the resultant
is a wobbling video signal. At this point luminance and
chrominance from respectively buffers 82 and 84 are
30 wobbling time-wise in the digital domain. These two
signals are fed respectively to luma digital to analog
(DJA~ converter 104 and chroma digital to analog (D/A)
converter 98 along with the matching cloc~ signals and the
address bus data. Thus the output of chroma D/A converter
35 98 and luma D/A converter 104 are analog signals.
The chroma signal output by chroma D/A converter 98
is heterodyned to achieve phase stability; this function
CA 0223~101 1998-06-12
- 20 -
is performed in the heterodyne block 100 as explained in
detail below.
Now referring to the center left hand portion of
Figure 4, a randomly varying number is generated by random
5 number generator 88 which outputs a frequency modulated
sine wave in the digital domain. This is latched once per
video field and provided to data digital to analog (D/A)
converter 90 thus providing a stepped approximation of the
sine wave. This stepped approximation of the sine wave is
10 smoothed and drives a phase lock loop (PLL) 92 SO that the
frequency provided out of phase lock loop 92 is tracking
the sine wave, i.e., tracking the phase of the phase lock
loop, thereby generating a read clock signal which
includes the wobble in terms of time. This read clock
15 signal is then applied to a counter in read address block
94. This read address block 94 outputs a running address
bus which is applied to buffers 82 and 84 as described
above. Read address block 94 is essentially a counter.
Thus the read address signal from block 94 is wobbling in
20 time, unlike the write address signal from block 70 which
is stable in time. The output of both write address block
70 and read address block 94 are on 10 bit wide buses and
the output signals of these two counters 70 and 94 are
ramping i.e., counting up.
Thus data is written simultaneously to both buffer 82
and buffer 8 4 and both buffer 82 and buffer 8 4 are read
simultaneously. The write address bus signals provided
from write address block 70 are on a 10 bit wide bus and
the data i.e., the addresses, are counting up from O to
30 909 which is the digital length conventionally assigned to
one video line, in an NTSC system with a 4FSC sampling
rate. Similarly, the read address from read address
counter 94 is counting from O to 90g, but the timing
thereof varies with respect to the write address by the
35 amount of the wobble which typically varies from +2 to -2
microseconds. Thus another way of describing the wobble
is that if one looks at the instant in which the write
CA 0223~101 1998-06-12
~ddresc has a value OFh, the read address at that same
time would have a different value and might not reach OFh
until 2 microseconds later or perhaps 2 microseconds
~earlier.
Thus the chroma analog signal from block 98 and the
luma analog signal from block 104 are wobbling in time
when read respectively in digital form from buffers 84 and
82. As described above, the chroma signal from block 98
must be heterodyned to maintain its relative phase. That
10 is, it is desired to maintain stability of the relative
phase with respect to colorburst of the chroma signal, but
still to allow the amplitude and phase modulation
envelopes to wobble. This is done as described in further
detail below by using the read clock signal from phase
15 :lock loop 92 which also includes a wobble which is
identical to that in the chroma signal, and applying the
read clock signal to a double balanced modulator circuit
in heterodyne circuit 100. Then if the difference is
t:aken between the two signals, the wobble on the read
20 clock signal is subtracted from the wobble on the chroma
signal, resulting in a phase-stable chroma signal with its
envelope wobble unaffected.
Also, the digitized video output from buffer 76 is
also delayed one line as applied to vertical blanking
25 interval D/A converter 106. The vertical blanking
interval data is not wobbling but is stable. Thus the
vertical and horizontal blanking interval signals are
regenerated in vertical blanking interval and horizontal
blanking interval regenerator 108.
Then all three signals from blocks 100, 104, and 108
are combined together in video adder 102 to reform
composite video with the vertical interval data inserted
during the proper time. Also inserted is the encryption
data from encrypt data block 96, which is typically
35 inserted in the region of line 20 of the vertical blanking
interval. Then the output of the video adder 102 is
provided to output driver 110 (which is a conventional
- CA 0223~101 1998-06-12
- 22 -
amplifier) providing output analog video signal as shown.
The various blocks shown in Figure 4 in the preferred
embodiment of the invention in the scrambling device are
embodied in circuitry conventionally located on various
5 printed circuit boards which include integrated circuits
and discrete components. In the preferred embodiment the
scrambling device includes four such printed circuit
boards, the first of which is the input board which
includes input buffer 66, genlock circuitry 68, clamping
10 AGC circuitry 72, and video A/D converter 74. The second
board is the RAM (random access memory) board which
includes buffer 76, adder 78, subtractor 80, buffer 82,
and buffer 84. The third board is the control board which
includes write address circuitry 70, randomly varying
15 number generator 88, data D/A 90, phase lock loop 92, read
address circuitry 94, and encrypt circuitry 96. The
fourth board is the output board which includes chroma D/A
g8, heterodyne circuitry 100, video adder 102, luma D/A
104, VBI D/A 106, VBI/HBI regeneration 108, and output
20 driver 110. Each of these boards is discussed below in
further detail.
Figure 5 shows the random noise overlay generation
circuitry as discussed above which provides the added
concealment to the edge fill. There are four different
25 parameters in the edge-fill region that must be randomized
for full security: (1) the luminance ("Y"), (2) the in-
phase chrominance component ("I"), (3) the quadrature
chrominance component ("Q"), and (4) the envelope or
timing of the inserted noise ensemble. It should be
30 pointed out that any or all of these may be omitted for a
simpler but less secure implementation. Alternatively,
the entire system may be implemented as shown but using
fewer than four independent noise generators (that is,
sharing noise sources), again with reduced effectiveness.
As shown in Figure 5, random noise generator #1 122,
and the 2MHz LPF 124 generate random luminance. Random
noise generator #2 138 and its associated balanced
CA 0223~101 1998-06-12
- 23 -
modulator 140 generate a random "I" chroma componentS;
:random noise generator #3 128, the 90-degree phase shifter
L36 and the related balanced modulator 130 generate a
:random "Q" chroma component; the two are combined in
5 the first summing stage 132 and band-passed 134 to form a
totally random chroma signal- The random luma and random
chroma are combined in the second summing stage 126 and
qated 118 on and off so as to generally fill the "edge-
fill" region of the frame. The transition from edge-fill
10 noise to standard video and back again, at the left and
right sides of the frame, respectively, must be
sufficiently random in time and gentle in amplitude so as
not to permit detection; to this end, random noise
generator #4 112 generates a random timing function using
15 generator 114 which when filtered 116 is applied to the
noise gate 118. The edges of the gating waveform are
softened by the 200-nanosecond shaping filter 116 to avoid
detectability of the gating function itself, and the
resultant gated composite noise waveform is simply added
20 l:inearly 120 at a suitable low level to the wobbled video
s:ignal. The output of the circuit of Figure 5 is provided
in the video output board (see below, Figure 9) at some
suitable point, for instance into the luma blanking switch
st:age 414.
Note that the filter characteristics described above
are indicative only; other cutoff frequencies, bandwidths
and risetimes could be used as the application warrants.
SCRPM~3LER INPUT BOARD
Figure 6 shows in detail the circuitry of the input
30 board as discussed above which includes (with reference to
Figure 4), input buffer 66, genlock 68, clamping AGC
circuitry 72, and video A/D circuitry 74. Each of the
blocks shown on input board circuitry in Figure 6 is
conventional and well known in the video area. The
35 clamping and AGC circuitry 72 of Figure 4 is shown on
Fi~ure 6 as including conventional AGC circuitry 140,
CA 0223~101 1998-06-12
amplifier 144, back por~h clamp 146, amplifier 144, second
back porch clamp 142 for the AGC circuitry 140, sync tip
AGC circuitry 148, white peak AGC circuitry 150 and 5 MXz
LeF 154 The processed video is provlded to the four
S times subcarrier frequency A/D converter 74 which
outputs digital video to the RAM board discussed below
The genlock circuitry 68 includes the
voltage control crystal oscillator 158 which is at four
times the subcarrier frequency. This four times
10 subcarrier frequency is then divided by four in divider
160 and provided to a subcarrier phase detector 152 which
compares the output frequency from divider 160 to the
colorburst of the incoming signal from amplifier 144.
Thus this ensures that the voltage control oscillator 158
15 does operate in exact synchronicity with the incoming
colorburst.
Thus the output of voltage control oscillator 158 is
the reference frequency which is four times the subcarrier
frequency. Also the output of divider 160 is the
20 reference frequency subcarrier signal. Also provided as
part of the input board are conventional horizontal timing
one shots 164 which are for various internal timing
purposes.
The lower portion of Figure 6 shows the digital
25 circuitry for generating the timing pulses for the
horizontal reset signal which is provided to the RAM board
as described below, as well as vertical timing for various
internal purposes ("housekeeping").
SCRAMBLER CONTROL BOARD
Figure 7A shows the control board which includes
(referring to Figure 4) the write address block 70,
randomly varying number generator 88, data D/A 90, phase
lock loop 92, read address block 94, and encryption
circuitry 96. Beginning at the upper left hand portion of
35 Figure 7A, the reference 4 times subcarrier frequency and
horizontal reset signals are received from the input board
CA 0223~101 1998-06-12
- 25 -
,-ircuitry of Figure 6. These input signals are then
provided to the write counter 200 which generates the
,table (non-wobbled) write address on a ten bit bus as
shown. The output of the write counter is also provided
5 l_o write EPROM 202 which at the proper point in the video
line in response to the count from counter 200 outputs
respectively signals for generating the sync gate, burst
gate, and the line 20 yate signals, and for resetting the
counter itself.
An output of the write EPROM 2 02 is also provided to
read counter 2 04 for initial synchronization purposes.
Read counter 204 outputs a count to the read EPROM 206
which in response then generates the phase lock loop
gating pulse. Blocks 208 through 230 and including also
15 258 generate the wobbled read address (RADR) which is used
by the RAM board as described below for purposes of
scrambling.
The lower line of circuit blocks of Figure 7A provide
the sync overlay concealment function i.e, the wiggled
20 sync signal as a fixed pseudo-random pattern. The output
CS designates composite sync. Beginning at the left hand
portion of the bottom portion of Figure 7A, the input
signal is the vertical blanking interval pulse. This
signal is used to reset a line counter 262 which is
25 incremented by the write beginning of line (WBO~) command
which comes from WBOL comparator 210.
The line counter 262 is used because in this
particular embodiment the sync wiggle is provided in a
fixed pseudo-random pattern on various lines of each video
30 field. Thus one runs a line rate address count which is
provided to a sync pattern EPROM 264 and line-by-line the
sync pattern EPROM describes the off-set i.e., amount of
wiggle that is desired on the horizontal sync pulse in
each line. That offset value from EPROM 264 is applied to
35 sync/burst logic and one shots 266 to generate the actual
sync pulses. Circuit 266 is also controlled by two
switches one of which is the sync overlay switch which
CA 0223~101 1998-06-12
- 26 -
determines if the sync overlay is functioning or not, and
also a sync width switch which determines whether the
width of the inserted sync signal is less than nominal.
The output of the one shots in block 266 is the CS
5 (composite sync) signal which is provided to the output
board as described, and the burst gate signal which is
provided to the blanking portion of the blkg/invert logic
in order to "unblank" the colorburst. The sync width
switch connected to circuitry 266 allows the reduction of
10 width in the horizontal sync signal in order to
accommodate the wiggle of sync. This is therefore non-
network compatible video, i.e., non-NTSC standard video
which in fact can be used successfully by most video
equipment including standard television sets, but may
15 upset or be affected by various types of transmission
equipment.
The blanking and invert logic 270 receives the output
signal from the sync/burst logic one shots 266 for
purposes of performing inversion of video for further
20 concealment under control of the attached invert switch.
Thus the "invert" line which is one output of block 270
indicates that a given video line is inverted or not. CB
refers to composite blanking which is provided as a
cGntrol line to the output board for determining when to
25 blank and when not to blank. The logic for video
inversion as shown in block 270 is that in order to avoid
clues to a pirate as to the presence of video inversion,
when inverting it is desired that the co]orburst remain
noninverted. Thus the inverted line must be in its non-
30 inverted position during the horizontal blanXing interval.Thus block 270 is gated by both vertical blanking interval
and also by the horizontal blanking pulse.
With reference to the concealment by use of overlays,
the sync/burst logic and one shots 266 are controlled by a
35 sync overlay switch as shown. It is possible to drive
this switch by a random number generator (as described
ab~ve with reference to Figure 5) thus providing a random
CA 0223~101 1998-06-12
form of sync concealment. This "sync wiggle" has been
found to operate well at 330 Hz. This of course does
not provide any pro~lem in removal of same by the
descrambler which invariably regenerates new sync for each
5 b:Lanking interval in any case. Also in accordance with
the invention it is a further modification to "double-
w:Lggle" the horizontal sync using two dissimilar
frequencies to provide further concealment as in Figure 5.
Furthermore it is also possible at the same time to vary
10 the location of the vertical sync signals i.e a vertical
sync wiggle which would add further concealment.
The second to last row of circuit blocks in Figure 7A
begins with the random clock generator 240 which provides
(i.n response to write address signals) random numbers to
15 frequency modulated counter 242 which then causes the sine
EE'ROM 244 to output an 8 bit data word to vertical latch
246. The sine EPROM 244 is controlled by a switch PK
which enables or disables the EPROM and thereby turns the
basic wobble "on" or "off." The 8 bit data word from the
20 sine EPROM 244 is also encrypted in encryptor 271 and
provided to a parallel to serial converter 272. Thus the
8 bit words output by the sine EPROM 244 are put into
serial form and inserted as data on line 20 or
thereabouts as the application suggests of the vertical
25 blanking interval of each video field for transmission to
the descrambler for descrambling purposes. Then the
descrambler (as described below) removes the 8 bit data
words decrypts them and applies them to an identical set
of' circuitry for purposes of descrambling.
Frequency modulated counter 242 applied to sine
EE~ROM 244 generates a randomly FM'd sine wave output
typically in the region of 3-15Hz. The output of vertical
latch 246 when enabled at line 19 of the VBI is then
provided to digital to analog converter 248 to output an
3~ analog signal which is then smoothed by a conventional RC
low pass filter 250 with a time constant of the order of
10 milliseconds and provided to comparator 252 the
CA 0223~l0l l998-06-l2
- 28 -
second input of which is connected to a ramp generator 258
which provides in response to write address line 7 a ramp
or "sawtooth" wave form at a rate of four times horizontal
frequency.
Thus comparator 252 generates a set of moving edges
which are moving in time exactly as it is desired for the
picture to wobble- The 4H phase detector 254 compares
those moving edges to a phase lock loop (PLL) gate signal
from EPROM 206, thereby locking the read clock and the
10 read address to those edges by means of error amplifier
256. The output of the error amplifier 256 is the
amplified output of phase detector 254 which is provided
to voltage controlled crystal oscillator 258 which is
designated as the "read" oscillator and is operating at
15 four times the subcarrier frequency (fsc). The output of
voLtage oscillator 258 is the read clock (RDCK). This
ret~d clock signal is thus wobbled i.e., operating at
ex~ctly four times subcarrier frequency but displaced from
it; nominal location by up to + 2 microseconds as
20 controlled ultimately by the action of random number
generator 240.
Thus read oscillator 258 Frovides a wobbled read
clock signal RDCK, which is in contrast to the analogous
oscillator 158 which provides a stable reference signal of
25 4 t:imes subcarrier frequency reference signal. Thus these
two oscillators 158 and 2 58 one of which ( 158) is stable
and one of which (258) is wobbling provide timing signals
applied to respectively the write counter 200 and the read
counter 204 of Figure 7a. Both counters 200, 204 are
30 divide by 910 counters (since there are 910 cycles of four
times subcarrier per NTSC video line); thus counters 200
and 2 04 are both running at the video line rate. This is
conventional except that the read counter 204 is wobbling.
The outputs of counters 200, 204 are provided respectively
35 to the write address bus WADR which is stable and the read
address bus RADR which is wobbled. Each of these buses
are 10 bit width buses as shown.
CA 0223~101 1998-06-12
- 29 -
With regard to the remainder of the circuitry shown
in Figure 7A, as discussed above if the active portion of
the video on each line is moved to the right a gap is
developed on the left edge which must ~e filled in; thus
5 there are two points of interest in time at the left edge
of each video line. The first point of interest is when
it is needed to have the video (at the beginning of the
gap) and the second time is when the video will be
available (the end of the gap.) In between those two
10 defined times it is necessary to provide an "edge fill"
signal resembling the active video. It will be seen that
these two times correspond respectively to the beginning
of the active line for the write cycle and the beginning
of the active line for the read cycle. It will be seen
15 that an identical situation occurs on the right side of
the picture when the picture is moved to the left. In
this case the two times of interest are (a) when the read
video has been exhausted and (b) when the video is no
longer required. These two times correspond respectively
20 to the end of the active line for the read cycle and the
end of the active line for the write cycle.
The problem is that the read system and the write
system are asynchronous, meaning that they cannot remain
in the proper time relationship. Thus an interface is
25 provided in which the address or addresses that are the
desired beginning and end of the lines i.e., "BOL" and
"EOL" for the write system and for the read system are
defined. Then the write end of line (WREOL) comparator
208, the read end of line (RDEOL) comparator 209 and the
30 write beginning of line ~WRBOL) comparator 210 and the
read beginning of line (RDBOL) comparator 212 compare the
actual addresses coming out of the read counter 204 and
the write counter 200 to the preset values corresponding
to the desired read & write BOL & EOL. When these
35 addresses match the preset values, that says that each
counter has reached the point where it is desired to start
filling in the active video, or has reached the point
CA 0223~101 1998-06-12
- 30 -
where it is no longer necessary to continue to fill in the
v:ideo because now the actual active video is being
provided (on the left edge) or the beginning of horizontal
b:Lanking interval has been reached (on the right edge).
Block 218 is the "fill end-of-line" one-shot
circuitry, and below that is the "fill beginning-of-line'~
one-shot circuitry 222. For the left edge, the write
beginning of line comparator 210 defines the left edge of
the active video where it is desired to start the filling
10 process. The read beginning of line comparator 212
determines when it is possible to stop the filling i.e.,
edge fill process. Thus both the outputs of comparators
210 and 212 are provided to the fill beginning of line one
shots 222 and the output of one shots 222 is a pulse which
15 is high only when is necessary to fill the left edge of
the line. Thus the end of that pulse is moving with the
wobble. On average, half the time that pulse is not
provided because it is not necessary to fill in the left
edge of the line because the picture has been moved to the
20 left instead of to the right.
For the right edge of the picture, the fill end of
line one shots 218 similarly are controlled by the write
end of line comparator 208 and the read end of line
comparator 209, and provide an analogous output signal
25 looking for the end of each video or an active portion.
Thus the "fill end-of-line" one-shots 218 generate a
single pulse that is high when it is desired to fill the
end of the line. The output of one shot circuits 218 and
222 are displaced from one another by the width of
30 horizontal blanking; the inner edges correspond to the
edge of formal blanking and the outer edges correspond to
the edge of the moving active portion of video.
The portion in between the two pulses is the region
in which a stable colorburst must be generated which
35 matches in amplitude and phase the moving colorburst which
intrinsically results from the wobbling read cycle. The
"fill burst" flip flop is set by the trailing edge of the
CA 0223~l0l l998-06-l2
- 31 -
EOL pulse and reset by the leading edge of the BOL pulse.
The "end-of-line" tri-state 228 looks at the address
defined as the read end-of-the-line (that is, the address
which is to be used to fill the right-side gap period,)
5 and similarly the "beginning-of-line" tri-state 224
provides an analogous signal for the left-side gap. For
the "fill burst" region the address used is that of the
clenter of colorburst. Thus under control of the command
signals from the one shots 218 and 222, an address is
10 either provided at the end of line or the address of the
middle of the colorburst or the address of the beginning
O e the line as provided on the bus which is connected to
the eight most significant bit read address select
circuitry 2 30.
Note that all of the busses in the central portion of
Figure 7A are only eight bits wide, because it is the two
least significant bits of the ten bit address system which
are allowed to run (as described above). Thus the select
circuitry 230 selects between three fixed addresses
20 corresponding to the end of line, the colorburst, and the
beginning of line. The burst (colorburst) tri-state 226
as shown is controlled by the output of the fill burst
f]ip-flop and also by the burst address. The effect of
setting the eight MSB's of the address buss to the address
25 corresponding to the center of colorburst is to fill the
entire read-cycle HBI with a continuous sine wave exactly
matching the input colorburst, regardless of the wobble
st:ate. The desired output colorburst can then simply be
gated out. Thus the select address provided on read
30 address bus RADR switches between the actual running
address count from the read counter 204 and the static
st:ates which are the output of the select circuit 230,
which runs normally during active video but is frozen at
the end of line, or beginning of line, and in the middle
35 of the burst. Thus the ten bit read address is wobbled
and at the desired intervals stops to perform the fill
process.
CA 0223~101 1998-06-12
- 32 -
The four times subcarrier frequency voltage
controlled crystal oscillator 258 of Figure 7A also
described with other embodiments in copending and commonly
owned U.S. patent application serial no. 07/860,643
5 entitled "Wide Frequency Deviation Voltage Controlled
Crystal oscillator", inventor Ronald Quan, attorney docket
no. M-1854, incorporated herein by reference. Figure 7B
o~ the present disclosure is one embodiment of the wide
frequency deviation voltage controlled crystal oscillator.
10 In Figure 7B the output signal ("OUT'~) corresponds to the
wobbled read clock (RDCK) of Figure 7A and the voltage
control input (VCONT~OL) corresponds to the output of error
amp 256 of Figure 7A.
With reference to Figure 7B, first crystal 313 is
15 connected in series with resistor 312. The series
combination of resistor 312 and crystal 313 is driven by a
first driving transistor 325. Current supplies 327 and
328 connect the emitters of transistors 325 and 326 to a
negative supply voltage VEE and the collector of transistor
20 325 to a positive supply voltage Vcc. The phase control
circuit includes a varactor (voltage controlled variable
capacitor), diode 320 along with capacitors 321, 322 and
323 and an inductor 324. The phase imposed by the phase
control circuit is varied by adjusting the VCONTROL which
25 changes the capacitance of the varactor diode 320. Diodes
32~3, 329' limit the amplitude of the oscillations in the
ci:rcuit .
A second transistor 325', a second crystal 313' and a
resistor 312' are coupled to common base amplifier
30 trcmsistor 326. First transistor 325, first crystal 313 and
first resistor 312 are also coupled to transistor 326 emitter.
The emitter of transistor 325'is
connected to a current source 327' to negative supply
voltage VEE~ and a collector of transistor 325' is
35 connected to positive supply voltage Vcc. Crystals 313 and
313' are driven in-phase with each other. The varactor
diode 320 has a relatively low ratio (i.e. 2:1) of maximum
to minimum capacitance.
CA 0223~101 1998-06-12
- 33 -
The resonant fre~uencies of crystals 313 and 313',
respectively, are selected such that they are spaced at a
predetermined interval (e.g. 3 KXz). The value of
resistors 312 and 312~ is typically about 15~ to 300 ohms.
5 Unity-gain buffer 330 providès the output signal.
SCRAMBLER RAM BOARD
Figure 8 depicts in detail the RA~ board which
includes certain blocks of Figure 4 including buffer 76,
adder 78, subtractor 80, Y buffer 8 2, and C buffer 84. As
10 shown in Figure 8, there is an input latch 3 40 (not shown
in Figure 4) receiving the video from the input board and
supplying same to ping-pong RAM buffer 342, 344. Adder 78
and subtractor 80 as shown in Figure 8 are the same as in
Figure 4. The buffer 82 for the Y luminance channel of
15 Figure 4 in Figure 8 is shown as a ping-pong RAM including
dual RAM banks 350 and 352 each of one horizontal line
(lH) length. Similarly, the buffer 84 for the chrominance
channel of Figure 4 is shown in Figure 8 as being ping-
pong RAM 360, 362.
Both the luma and chroma channel which are desired to
be wobbled must switch from the write address system to
the read address system because the write address and the
write clock are stable as coming in, i.e. WADR and WRCK,
whereas the read address RADR and read clock RDCX are both
25 unstable. The selector circuits 354 and 364 steer the
clocks and address busses respectively for the luma
channel and the chroma channel so that at any given video
line one of the buffers in each pair of buffers 3S0, 352,
and 360, 362 is writing and the other is reading. The
30 horizontal blanXing interval reset signal is provided to
the lH flip-flop 356 for control of the select circuitry
354 so that the buffer pairs alternate reading and writing
appropriately.
SCRAMBLER OUTPUT BOARD
The output board portion of the block diagram of
CA 0223~101 1998-06-12
~- 3~ -
Figure 4 includes the chroma D/A converter ~8, the luma
D/A converter 104, the vertical blanking interval D/A
converter 106, the heterodyne circuit 100, the video adder
102, the output driver 110, the VBI/HBI regeneration 108.
As can be seen in Figure 9 showing the output board,
the luma DAC (digital to analog converter) 104, VBI DAC
106, chroma DAC 98, and output driver 110 are the same
blocks as in Figure 4. Additionally, Figure 9 shows the
line 20 data at the upper left hand portion of the figure
10 which is provided as discussed above from the control
board of Figure 7A going to a line 20 inject circuitry 400
which is then provided to luma blanking switch 414.
Additionally, the video invert signal also provided as
shown in the lower right hand portion of Figure 7A is
15 provided in the upper left hand portion of Figure 9 to the
luma DAC 104 which also receives the luma digital signal
from the RAM board output bus as shown. Additionally, the
digitized vertical blanking interval signal from the RAM
board (which is time stable) is provided to the VBI DAC
20 106 and the chroma digitized signal from the RAM (which is
wobbled) is provided from the RAM board also to chroma DAC
98; the chroma DAC is controlled by the source video
invert signal as is the luma DAC.
The VBI switch 406, controlled by the VBI signal,
25 switches in the vertical blanking interval as desired in
the appropriate portion of the signal. The output of VBI
switch 406 is then filtered by a conventional inverse sine
X/X type filter to compensate for sampling-induced high
frequency roll off. The output of the filter 410 is then
30 provided to summing amplifier 412. The output of summing
amplifier 412 is provided to the luma blanking switch 414.
The "fill burst" action replaces the entire HBI signal
with continuous burst at the output of the RAM board. The
action of the luma blanking switch is to re-insert H
35 blanking and H sync, and to get the continuous burst to
form the expected colorburst, thereby regenerating the
desired HBI format. The output of luma blanking switch
CA 0223~101 1998-06-12
414 is then filtered by low pass filter 416 for removing
the extraneous sampling sidebands frequencies above about
5 MHz. The output of low pass filter 416 then is provided
to the output driver amplifier 110.
Similarly, the output of the chroma DAC 98 is
connected to vertical blanking interval switch 420 for
switching out the chroma signal during vertical blanking.
The output of the vertical blanking switch 420 is then
subject to low pass filter 422 to remove the chroma
10 frequencies above about 2 MHz and then also subject to the
black clipper 412 and hence follows the same path as
described above for the luma DAC.
With regard to restoration of the vertical detail
from the chroma channel output, this is performed by the
15 output board circuitry of Figure 9. As shown, this takes
place except during vertical blanking under control of
vertical blanking switch 420 which switches out chroma
during vertical blanking. This chroma signal during the
active video portion of the line is filtered by low-pass
20 filter 422 and black clipped at summing amplifier 412 and
recombined with luma, thus restoring the missing vertical
detail. This is because the missing vertical detail
appears in the chroma channel, so low pass filtering
removes the chroma content, leaving only the vertical
25 detail, and adding it back to the luma channel restores
the missing vertical detail. Sync shaper 426 and blanking
shaper 428 convert the logic-level CS (composite sync) and
CB (composite blanking) signals to analog signals of the
required levels and having the standard rise and fall
30 times, i.e., 140 microseconds for NTSC.
The lower part of Figure 9 is the heterodyne circuit
100 of Figure 4. As shown, analog chroma data from chroma
DAC 98 is provided to an inverse sine X/X filter 424 to
restore losses in high frequencies due to sample and
35 holding in A/D. This filtered chroma signal (which is
wobbling in time) is not however at the nominal 3.58 MHz
subcarrier frequency. Thus this signal is provided to
CA 0223~101 1998-06-12
- 36 -
balance modulator 438 for further processing.
Subcarrier frequency voltage controlled crystal
oscillator 450 is part of a loop including subcarrier
frequency phase detector 446 which drives error amplifier
5 448 which in turn drives subcarrier frequency voltage
control oscillator 450. The output of frequency control
oscillator 450 is divided by 2 by divider 452 to provide a
frequency one-half of the subcarrier frequency. The
output of divider 452 is provided to two band pass filters
10 436, 454; the first filter 436 passes only the fifth
harmonic of half of the subcarrier frequency i.e., 5/2
Fsc. The second band pass filter 454 is passing only the
third harmonic, i.e. 3/2 Fsc. Band-pass filter 454 then
outputs the stable carrier 3/2 FSC signal which is applied
15 to balance modulator 456 which mixes this with the
divided-by-4 Read clock signal (RDCK) which is wobbled and
is equal to (four times the subcarrier frequency/4).
This RDC~ signal is divided by four at divider 458
thus outputting the wobbled subcarrier frequency, which at
20 balance modulator 456 is modulated with the 3/2 of the
subcarrier frequency. The output of balance modulator 456
is then filtered at band-pass filter 460 to select the 5/2
of subcarrier frequency (upper sideband) which contains
the wobbled subcarrier frequency. In the figure, this is
25 labeled "wobbled carrier" and is then applied to balance
modulator 442.
The upper arm of the heterodyne circuit as shown
accepts the fifth harmonic of the stable subcarrier
frequency divided by 2 from band-pass filter 436 and
30 modulates that at balance modulator 438 with the wobbled
chroma from filter 424. The output of balance modulator
438 is then filtered by band-pass filter 440 (having a
pass band about 3 MHz wide) to select the 7/2 of the
subcarrier frequency. The output of band-pass filer 440
35 is then 7/2 of subcarrier frequency (upper sideband) which
contains wobbled chroma, which when mixed in balance
modulator 442 with the wobbled carrier provides a stable
CA 0223~101 1998-06-12
- 37 -
chroma signal at 3.58 MHz (to the chroma blanking switch
430) via lower sideband output of balance modulator 442.
The object of this heterodyne circuit is that the
amount of wobbling (jittering) of the video line is well
5 known via the master clock, via the read clock (RDCK)
timing signals. That is, this master clock's signal is
actually tied to the changes in frequency in proportion to
the changes in the scrambled chroma frequency, i.e., the
wobbling. Thus this read clock signal can be used as a
10 form of cancellation to remove the wobbling from the
chroma signal in terms of the frequency. As shown, the
burst signal which helps control subcarrier frequency
phase detector 446 is the colorburst signal from the
output video. The video output colorburst thus matches
15 the input video source colorburst.
The operation of this heterodyne circuit is shown
with further reference to the frequency spectra of Figures
lOA through lOE. Starting with Figure lOA, the input
program chroma signal (prior to scrambling) is shown
20 distributed over a spectrum centered at 3.58 MHz i.e., the
subcarrier frequency. Upon scrambling in Figure lOB, the
wobbling chroma which is provided from the chroma DAC 98,
is shown "jittering" (wobbled) by ~F and having a center
frequency of 3.58 MHz + ~F. The master clock at the same
25 time is "jittering" (wobbling~ by exactly the same amount
at four times the subcarrier frequency, i.e., centered at
li.32 MHz with a jitter of 4 times ~F, as shown in Figure
lOC. This is because the wobble in the chroma is exactly
one-quarter of the master clock.
As shown in Figure lOD, by band-pass filtering the
heterodyne circuit selects 7/2 of subcarrier frequency
which confirms wobbling chroma and the 5/2 of subcarrier
frequency, both of which include the same amount of
wobble, i.e., + ~F.
Thus by subtracti~g out (modulating and selecting the
lower sideband) the 5/2 of subcarrier frequency chroma
from 7/2 subcarrier frequency (both of which include the
CA 0223~101 1998-06-12
- 38 -
~F wobbling), one arrives at the output of the low-pass
filter which is a stable 3.58 MHz and which is the desired
stable chroma signal.
Figure 11 shows a different version of the heterodyne
5 circuit in accordance with the invention as applied to the
above-described scrambling system. As shown, the master
clock varies in frequency by + 4 times ~F due to the
scrambling process, in order to cause the desired wobbling
in the video thereby scrambling the video. The "program
10 in'l video is digitized by the A/D converter in block 470
and separated into the Y (luminance) and C (chrominance)
channels, each of which is processed to be wobbled by the
master clock by +4 ~F. After the master clock varies both
Y and C components i.e., wobbles them in a time-varying
15 way, the C chrominance component has color frequencies
that are undesirably not stable (~Ijittered~). Thus the
object of the heterodyne circuit is to stabilize the
scrambled chrominance component frequency so that the
television receiver can view the color with use of a
20 simple low-cost descrambling device.
It is known that the master clock is at 4 times
subcarrier frequency +4 ~F. As shown, after the digital
processing, both the Y and t~e C signals are converted
back to analog by D/A converters in block 470, thus
25 outputting the so-calleduY'~ jittered (wobbled) signal and
the ~c~,,jittered (wobbled) signal which is the undesirably
unstable carrier frequency. The heterodyne circuit at the
lower left-hand portion of the figure applies the master
clock signal (which is also designated the RDCK signal)
30 which is divided by 4 at divider 474 and which is then
multiplied by a stable 3/2 times the subcarrier frequency
by balance modulator 476. As noted, the subcarrier
frequency is 3.58 MHz. The upper sideband of the output
of balance modulator 476 is selected by band-pass filter
35 478 to obtain 3/2 of the frequency subcarrier plus the
subcarrier frequency + ~F. At the same time the C'
(wobbled chroma) component which is subcarrier frequency +
CA 0223~101 1998-06-12
- 39 -
~F is first filtered at band pass filter 482 by the
inverse sine X/X filter 424 of Figure 9. The output of
band-pasS filter 482 is then multiplied by a stable S/2
subcarrier frequency signal at modulator 484 and the
5 output of modulator 484 is filtered at band pass filter
486 to pass the upper side-band to provide 7/2 times the
subcarrier frequency + ~F.
As noted above (see Figure 9), the stable 3/2
subcarrier frequency and 5/2 subcarrier frequency are
10 provided from a phase-locked voltage controlled oscillator
that is locked to incoming stable video color frequency,
i.e., the reference subcarrier frequency. The outputs of
band-pass filters 478 and 486 are multiplied by balance
modulator 480 and then filtered at band-pass filter 488 so
15 that the lower sideband output is a chroma signal at
frequency subcarrier which is free of the +~F wobble. As
shown in Figure 9 (but not in Figure 11), the burst from
the output low pass filter 432 is sent back to the phase
detector 446 to phase lock the subcarrier frequency
20 voltage control oscillator 450 to incoming video color
frequency. As shown in Figure 11, the output chroma
signal from band-pass filter 488 is then added by a video
adder to the wobbled luminance signal Y', the output of
which is low-pass filtered at 490 to provide the output
25 video which includes the wobbled luminance signal and a
wobbled chroma signal ~ith stable subcarrier frequency.
In conventional heterodyne color stabilizers the
master clock/4 is really the colorburst from the input
video. This could have been done here as well in a
30 similar way by taking the wobbled colorburst from inverse
Sin X/X filter 424; but the color chroma stabilization
would not have been as effective and thus chroma would be
more unstable. For the best chroma stability RDCK (unique
to this system) is used as described above.
With regard to the above-described heterodyne
circuit, its applicability is in addition to use in
scrambling. For instance, it is suitable for use with any
CA 0223~101 1998-06-12
- 40 -
sort of video processing which involves time-base errors.
DESCRAMBLER
The wobbled video output signal from the encoder or
~scrambler circuitry of Figure 4 is transmitted
5 conventionally by coaxial cable, satellite, broadcast
television, cable television or otherwise to a descrambler
(decoder), which typically is located in a home and the
descrambled output of which is connected to a conventional
home television set or monitor. One of the objects of the
10 present invention is to provide a system which is highly
secure, offers adequate concealment, and yet compatible
with a low cost and reliable decoder. This is because
there are thousands or tens of thousands of decoders made
and used and hence, is essential that they be relatively
15 low cost and require little servicing since they are
located at the home. Note that this is not the case with
the scrambler which is typically located at a head-end and
of which there are relatively few (one per TV channel) in
any one television system.
Figure 12 is a flowchart of the descrambler signal
processing. In step 500 the scrambled video is received
and the encrypted random number (which is the decoding
seed) extracted. From this extracted number, in step 502
the random number is decrypted and converted to an analog
25 wave form. Then in step 504, this analog wave form
generates a wobbling time base which tracks the input
video, i.e., includes the information necessary to
indicate exactly how the video is wobbling. From that
data it is possible to synthesize in step 506 the required
30 wobbling horizontal sync, blanking, and colorburst. In
step 508 the input signal's complete horizontal blanking
interval is replaced with a complete synthesized wobbling
horizontal blanking interval made up from the synthesized
sync, blanking, and colorburst which tracks the video,
35 thus allowing the signal to be viewed on a conventional
television receiver.
CA 0223~l0l l998-06-l2
- 41 ~-
Figure 13 shows in block diagram form one embodiment
of the decoder for performing the processing of Figure 12.
In the upper left hand portion the scrambled video input
signal is provided to an input buffer 520. In the
5 descrambling data path, the data is extracted by data
extractor 522 and then conventionally decoded by decryptor
524. The data has been converted from digital to analog
form in block 526, smoothed by low pass filter 528 and
provided to comparator array 530.
Meanwhile phase lock loop 534 is locked to the
horizontal line rate of incoming horizontal sync, to drive
an analog ramp generator 536. The comparator array 530
then compares the horizontal ramp with the varying DC
(direct current) signal coming out of low pass filter 528
15 to provide a moving edge at the point at which they cross,
i.e. where the comparison is made, from which edge it is
possible to time scale all elements of the horizontal
blanking interval.
This comparator data is then used to generate burst,
20 horizontal sync, and blanking pulses using colorburst
regenerator 542 and horizontal blanking interval
regenerator 544 which are applied to video switch 548.
Video switch 548 switches between the active video which
(with one exception) is not to be processed by the
25 descrambler and the horizontal blanking interval which is
processed by the lower portion of the circuitry of Figure
13. The video switch 548 is driven by the regenerated
horizontal blanking.
The input video from buffer 520 is processed by the
30 descrambler only to the extent that the video is re-
inverted wherever it has been previously inverted by the
scrambler in order to restore the original video. This
now fully non-inverted video is provided to video switch
548, the output of which then is provided to output driver
35 550 for the video output to the TV receiver or monitor.
The analog ramp generator 536 generates a series of
waveform ramps as shown in Figure 14A which are clocked
CA 0223~101 1998-06-12
with horizontal sync from PLL 534- As shown in Figure 14A
each ramp has the duration adequate to cover the entire
regenerated HBI including the wobble -- that is, around 20
microseconds. Thus the comparator compares the ramps with
5 a reference voltage which is shown as a horizontal line in
Figure 14A. The comparator provides as output the square
pulses shown in Figure 14B each of which is wobbling in
time synchronously with the wobble present in the input
video, as shown by the horizontal arrows at the leading
10 edge of each horizontal pulse in Figure 14B.
As shown in Figure 14B, there is one such square
pulse output of the comparator for each video line. Thus
this edge moves in time synchronously with the wobble.
Then using the single wobbling edge for each line as shown
lS in Figure 14B, it is possible fully to reconstruct the
horizontal blanking interval, as shown in Figure 14C by
the vertical arrows which indicate six edges which are:
(1) the leading edge of horizontal blan~ing; (2) the
leading edge of the horizontal sync pulse; (3) the
20 trailing edge of the horizontal sync pulse; (4) the
leading edge of colorburst; (5) the trailing edge of
colorburstiand(6)the end of the horizontal blanklng interval.
In accordance with one embodiment of the invention, this
is done by providing an array of six different comparators
25 each with an offset to the preceding one. Alternatively a
single comparator would generate the first edge and then a
sequence of timed one-shots would provide the other five
edges of the horizontal blanking interval.
The chief task of the descrambler, in addition to
30 removing the inversion of the active portion of the video,
is to generate a horizontal synchronization pulse which
moves in exact synchronism with the time induced wobble
and a colorburst which moves in exact synchronism with the
time-induced wobble. The restoration of the sync pulse is
35 relatively straightforward, but the restoration of the
colorburst is more difficult as shown with reference to
Figure lB. In line N the colorburst occurs in time
CA 0223~101 1998-06-12
- 43 -
advanced as regards to location of the colorburst in line
N+l and the colorburst in line N+2 is retarded relative to
the location of the colorburst in line N+l. Thus the
descrambler must provide a sine wave which matches in
5 amplitude and phase a colorburst which has not yet
occurred for certain particular lines.
Thus the circuitry must anticipate the location of
colorburst for particular lines. This is performed in the
descrambler in one embodiment by using the colorburst to
10 ring a crystal filter in the burst regenerator 542 of
Figure 13, so that the filter rings at the same amplitude
and phase for an entire line thus generating a continuous
wave having the same amplitude and phase as the
colorburst. This typically requires two cascaded
15 oscillating crystals to provide a filter that rings well
enough (that is having a high enough Q) so that the output
has not dropped away to zero prior to the end of the video
line.
One improvement in accordance with the invention over
20 use of these two cascaded crystals is to provide a non-
standard form of the colorburst, by dividing the
colorburst into two portions (or more) in the scrambler.
Figure 15A shows a conventional RS-170A horizontal
blanking interval with colorburst ("burst") on the back
25 porch. Figure 15B shows in accordance with the invention
that instead a first portion of the burst is a pre-burst
portion provided on the front porch of the horizontal
blanking interval of each video line, with the remaining
of the colorburst located conventionally on the back porch
30 of HBI. Thus there would be no need to make the crystal
filter ring for more than about 5 microseconds i.e., not
even the full duration of horizontal blanking. This has
the advantage of allowing use of a simpler and less
expensive descrambler, but is a non-standard format due to
35 the need for the provision of the pre-burst. This,
therefore, is a non-network transparent, i.e., non-NTSC
compatible system suitable for use with for instance a
CA 0223~101 1998-06-12
- 44 -
video cinema application.
An alternative approach (shown in Figure 15C) is to
superimpose a continuous "colorburst" upon the entire HBI
-- that is, start "burst" at the beginning of blanking and
5 let it run all the way through, adding the sync pulse
linearly.
DESCRAMBLER WITH DIGITAL SYNTHESIS OF HBI
The above described descrambling process involves
synthesizing (regenerating) horizontal sync, blanking, and
10 colorburst that track the video "wobble", and replacing
the incoming standard sync, blanking and burst with them
to form a video signal having a unified time-base
variation (the "wobble"), which the TV receiver can track
in order to present a stable, "descrambled" picture.
In another embodiment those sync, blanking, and burst
signals are generated digitally in the descrambler. The
subsequent insertion into the analog signal, and all of
the video signal processing (clamping, AGC, inversion,
etc.) remain in the analog domain as with the above
20 described "analog" embodiment of Figure 13.
The following circuitry is for NTSC; PAL works
similarly but with different numerical values, as will be
apparent to one skilled in the art. The horizontal offset
required for each line's horizontal blanking interval
25 (HBI) is mathematically calculated based upon some
suitable interpolation algorithm from the field-rate data
byte sent in the vertical interval. That offset is
applied as a preset or preload to a "divide-by-1820"
counter running at 8 times subcarrier frequency, nominally
30 organized so that the counter counts out the entire line -
- that is, the count of 1820 takes 63.555 microseconds.
It will be seen that, if the counter is preset with a
value of, say "10", the counter will finish with its count
to 1820 in a time 349 nanoseconds sooner than if it had
35 not been so preset. If the counter is in fact designed to
count to 1880, and provision made for preloads ranging
CA 0223~101 1998-06-12
- 45 -
from 0 to 120, the net effect is that the line-time as set
by the counter can be varied by +2 microseconds, in
increments of 35 nanoseconds.
In practice, with the present scrambling process, the
5 line-to-line variation of line length is no more than 10
nanoseconds; thus the counter need only achieve a count of
1820 +/- 1, or 1821 with a preset ranging from 0 to 2.
(It will be seen that the time offset accumulation over
240 lines in a single frame, at 10 nanoseconds/line, is
10 2.4 microseconds).
Then referring to the block diagram of Figure 16A,
the top row of blocks is the analog video processing
corresponding to the similarly numbered elements of Figure
13. In the second row, an oscillator 578 operating at
15 8*Fsc is phase-locked to incoming colorburst by the
subcarrier PLL 576. Its output is divided by eight at
divider 580 to produce a 3.58 MHz signal which is gated by
burst gating 582 to form the new colorburst, as well as
providing a clock for the 1:1880 counter 588.
In the third row, the data byte in the vertical
interval is separated from the incoming video and
decrypted in block 584, and supplied to the line-offset
calculator 586 (a microprocessor). The calculator 586
calculates in real time the line-by-line offset required
25 to fit the vertical-rate data byte, and supplies that
number (still in real time) to the divide-by-1880 counter
588. The calculator 586 can be simple since at most it
only has to calculate one number to eight-bit precision
every 63.555 microseconds; moreover, it will generally
30 have at least four lines (or 245 microseconds) in which to
work. In an alternative embodiment of Figure 16B
(otherwise similar to that of Figure 16A), in order to
reduce the required clock speed without degrading the
fineness of time resolution, the system runs at 4* Fsc
35 instead of 8* Fsc, and the counter is preloaded with only
the 7 most significant bits of the offset word. This
limits the shift to a minimum increment of 70 nanoseconds;
CA 0223~101 1998-06-12
- 46 -
the last bit (LS8) which defines the 35 nanosecond shi~t
is used to invert the cloc~ in an XOR gate 587. The
inversion causes the "trailing edge't to ~e the active edge
instead of the "leading edge", into latch 589, and thereby
5 shifts the latch output by the desired 35 nanoseconds.
Referring to both Figures 16A and 16B, the ll-bit
output of the counter 588 is conventionally decoded at
edge decoder 590 to provide six timing edges corresponding
to leading and trailing edges of the desired sync,
10 blanking, and burst qate pulses; it will be seen that
these edges are moving, as an ensemble, with the declared
"wobble" due to the line-rate varying preset to the
counter. The timing edges are conventionally applied to
three R-S flipflops 592 to generate the actual pulses. In
15 practice, additional "house-keeping" pulses can be
similarly decoded and formed as required.
In the fourth row, the sync signal is separated at
sync stripper 594 from the video and separated further
into horizontal and vertical sync pulses at separation
20 block 596. The horizontal sync is used to reset the 1880
counter 588; the vertical is used to reset a divide-by-
525 counter 5~8, which is clocked by horizontal sync and
used to count lines in the frame in line nll~her decoding
block 600 for various house-keeping purposes -- in
25 particular, to inhibit the HBI regeneration process in
block 544 during the 22 lines of the vertical interval.
VERTICAL SYNC TIMESHIFT SCRAMBLING
The object of this embodiment is to cause the picture
wobble in the vertical as well as horizontal. The
30 implementation requires only that the present one-line
memory (RAM) which drives the adder 78 and subtractor 80
on the RAM board be extended to something like twenty-one
lines, with provision to select the output of any of the
twenty-one lines randomly. In practice, then, compared
35 with the video out of the eleventh of the twenty-one
memory stages, video from the first is advanced by ten
CA 0223~101 1998-06-12
- 47 -
lines and video from the last is delayed by ten lines;
twenty lines pea~c-to-peak out of 240 active lines per
field compares directly to 4 microseconds horizontal
motion out of 52 microseconds active picture width. A
5 second randomly frequency modulated digital sine-like
signal (analogous to the one which varies the read address
for horizontal wobble) is used to select the output of
different pairs of lH buffer delays to be applied to the
adder and subtractor for Y/C separation.
The number of lH memory buffers can be varied for
different applications and any suitable rate of variation
can be used; in particular the rate of variation can be
~ randomly controlled as with the horizontal scrambling of
the Figure 4 system, in which case a second byte of data
15 would be added to the vertical interval to describe the
vertical variation, analogous to the first byte used to
describe the horizontal variation. The second byte would
of course be encrypted like the first byte.
Figure 17, shows such a system with a 5-line
20 variation and replaces ln toto the block 76 labelled "lH
BU~FER" of Figure 4. Figure 17 shows portions
corresponding to the circuitry of Figure 4 including video
A/D 74, adder 78 and subtractor 80. Five lH buffers 602,
604, 606, 608, 610 replace the single lH buffer 76 of
25 Figure 4. The buffers 602, ..., 610 are selected
according to a randomly varying number generated by random
number generator 612, which as described above generates a
frequency modulated digital sine wave-like signal to
select one of buffers 602, ..., 610 for each video field
30 thus randomly varying the amount of vertical timeshifting.
In any case, for the selected buffer, the adder 78 and
subtractor 80 are driven with the selected buffer's input
and output, analogously to the circuit of Figure 4. Thus
at all times the adder 78 and subtractor 80 are presented
35 with two video signals differing by exactly one line (lH)
and thus the Y-C separation proceeds uninhibited.
Descrambling requires simply that the vertical sync
CA 0223~101 1998-06-12
- 48 -
signal be wobbled to match the picture, analogously to the
above-described horizontal wobble; TV sets, monitors or
projectors which use line-counting vertical deflection
systems would be modified to accept a time-varying line-
5 count, while the older multivibrator-based TV units would
require no modification.
Edge-fill provisions similar to those described above
a:re used to fill the top of the frame when the picture is
shifted downward and the bottom of the frame when the
10 picture is shifted upward; minor variations of the above
described edge fill circuitry accomplish this.
The above description of the invention is
illustrative and not limiting; further modifications will
be apparent to one of ordinary skill in the art in the
15 light of this disclosure and the appended claims.
