Note: Descriptions are shown in the official language in which they were submitted.
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DIGITAL TELEVISION RECEIVER WITH ADAPTIVE FILTER CIRCUITRY
FOR SUPPRESSING NTSC CO-CHANNEL INTERFERENCE
The present invention relates to digital television
systems, such as the digital high-definition television
(HDTV) system used for terrestrial broadcasting in the
United States of America in accordance with the Advanced
Television Sub-Committee (ATSC) standard, and more
particularly, to digital television receivers with
adaptive filter circuitry for suppressing co-channel
interference from analog television signals, such as those
conforming to the National Television Systems Committee
(NTSC) standard.
BACKGROUND OF THE INVENTION
A Digital Television Standard published 16
September 1995 by the Advanced Television Subcommittee
(ATSC) specifies vestigial sideband (VSB) signals for
transmitting digital television (DTV) signals in
6-MHz-bandwidth television channels such as those
currently used in over-the-air broadcasting of National
Television Subcommittee (NTSC) analog television signals
within the United States. The VSB DTV signal is designed
so its spectrum is likely to interleave with the spectrum
of a co-channel interfering NTSC analog TV signal. This
is done by positioning the pilot carrier and the principal
amplitude-modulation sideband frequencies of the DTV
signal at odd multiples of one-quarter the horizontal scan
line rate of the NTSC analog TV signal that fall between
the even multiples of one-quarter the horizontal scan line
rate of the NTSC analog TV signal, at which even multiples
most of the energy of the luminance and chrominance
components of a co-channel interfering NTSC analog TV
signal will fall. The video carrier of an NTSC analog TV
signal is offset 1.25 MHz from the lower limit frequency
of the television channel. The carrier of the DTV signal
is offset from such video carrier by 59.75 times the
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horizontal scan line rate of the NTSC analog TV signal, to
place the carrier of the DTV signal about 309,877.6 kHz
from the lower limit frequency of the television channel.
Accordingly, the carrier of the DTV signal is about
2,690122.4 Hz from the middle frequency of the television
channel.
The exact symbol rate in the Digital Television
Standard is (684/286) times the 4.5 MHz sound carrier
offset from video carrier in an NTSC analog TV signal.
The number of symbols per horizontal scan line in an NTSC
analog TV signal is 684, and 286 is the factor by which
horizontal scan line rate in an NTSC analog TV signal is
multiplied to obtain the 4.5 MHz sound carrier offset from
video carrier in an NTSC analog TV signal. The symbol
rate is 10.762238 megasymbols per second, which can be
contained in a VSB signal extending 5.381119 MHz from DTV
signal carrier. That is, the VSB signal can be limited to
a band extending 5.690997 MHz from the lower limit
frequency of the television channel.
The ATSC standard for digital HDTV signal
terrestrial broadcasting in the United States of America
is capable of transmitting either of two high-definition
television (HDTV) formats with 16:9 aspect ratio. One
HDTV display format uses 1920 samples per scan line and
1080 active horizontal scan lines per 30 Hz frame with 2:1
field interlace. The other HDTV display format uses 1280
luminance samples per scan line and 720 progressively
scanned scan lines of television image per 60 Hz frame.
The ATSC standard also accommodates the transmission of
DTV display formats other than HDTV display formats, such
as the parallel transmission of four television signals
having normal definition in comparison to an NTSC analog
television signal.
DTV transmitted by vestigial-sideband (VSB)
amplitude modulation {AM) during terrestrial broadcasting
in the United States of America comprises a succession of
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consecutive-in-time data fields each containing 313
consecutive-in-time data segments. The data fields may be
considered to be consecutively numbered modulo-2, with
each odd-numbered data field and the succeeding
even-numbered data field forming a data frame. The frame
rate is 20.66 frames per second. Each data segment is of
77.3 microseconds duration. So, with the symbol rate
being 10.76 MHz there are 832 symbols per data segment.
Each segment of data begins with a line synchronization
code group of four symbols having successive values of +S,
-S, -S and +S. The value +S is one level below the
maximum positive data excursion, and the value -S is one
level above the maximum negative data excursion. The
initial line of each data field includes a field
synchronization code group that codes a training signal
for channel-equalization and multipath suppression
procedures. The training signal is a 511-sample
pseudo-noise sequence (or "PN-sequence") followed by three
63-sample PN sequences. This training signal is
transmitted in accordance with a first logic convention in
the first line of each odd-numbered data field and in
accordance with a second logic convention in the first
line of each even-numbered data field, the first and
second logic conventions being one's complementary
respective to each other.
The data within data lines are trellis coded using
twelve interleaved trellis codes, each a 2/3 rate trellis
code with one uncoded bit. The interleaved trellis codes
are subjected to Reed-Solomon forward error-correction
coding, which provides for correction of burst errors
arising from noise sources such as a nearby unshielded
automobile ignition system. The Reed-Solomon coding
results are transmitted as 8-level (3 bits/symbol)
one-dimensional-constellation symbol coding for
over-the-air transmission, which transmissions are made
without symbol precoding separate from the trellis coding
procedure. The Reed-Solomon coding results are
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transmitted as 16-level (4 bits/symbol)
one-dimensional-constellation symbol coding for cablecast,
which transmissions are made without preceding. The VSB
signals have their natural carrier wave, which would vary
in amplitude depending on the percentage of modulation,
suppressed.
The natural carrier wave is replaced by a pilot
carrier wave of fixed amplitude, which amplitude
corresponds to a prescribed percentage of modulation.
This pilot carrier wave of fixed amplitude is generated by
introducing a direct component shift into the modulating
voltage applied to the balanced modulator generating the
amplitude-modulation sidebands that are supplied to the
filter supplying the VSB signal as its response. If the
eight levels of 4-bit symbol coding have normalized values
of -7, -5, -3, -1, +1, +3, +5 and +7 in the carrier
modulating signal, the pilot carrier has a normalized
value of 1.25. The normalized value of +S is +5, and the
normalized value of -S is -5.
In the earlier development of the DTV art it was
contemplated that the DTV broadcaster might be called upon
to decide whether or not to use a symbol precoder at the
transmitter, which symbol precoder would follow the symbol
generation circuitry and provide for preceded filtering of
symbols. This decision by the broadcaster would have
depended upon whether interference from a co-channel NTSC
broadcasting station were expected or not. The symbol
precoder would complement the symbol postcoding
incidentally introduced in each DTV receiver by a comb
filter used before the data-slicer in the symbol decoder
circuitry to reject artifacts of NTSC co-channel
interfering signal. Symbol preceding would not have been
used for data line synchronization code groups or during
data lines in which data field synchronization data were
transmitted.
Co-channel interference is reduced at greater
distances from the NTSC broadcasting stations) and is
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more likely to occur when certain ionospheric conditions
obtain, the summertime months during years of high solar
activity being notorious for likelihood of co-channel
interference. Such interference will not occur if there
are no co-channel NTSC broadcasting stations, of course.
If there were likelihood of NTSC interference within his
area of broadcast coverage, it was presumed that the HDTV
broadcaster would use the symbol precoder to facilitate
the HDTV signal being more easily separated from NTSC
interference; and, accordingly, a comb filter would be
employed as symbol postcoder in the DTV receiver to
complete matched filtering. If there were no possibility
of NTSC interference or there were insubstantial
likelihood thereof, in order that flat spectrum noise
would be less likely to cause erroneous decisions as to
symbol values in the trellis decoder, it was presumed that
the DTV broadcaster would discontinue using the symbol
precoder; and, accordingly, the symbol postcoder would
then be disabled in each DTV receiver. Without the
broadcaster being aware of the condition, actual
co-channel NTSC interference can be substantial for
portions of the reception area for a broadcast, owing to
freakish skip conditions, owing to cablecast leakage,
owing to inadequate intermediate-frequency image
suppression in NTSC receivers, owing to magnetic tape used
for digital television recording having remnant previous
analog television recording, or owing to some other
unusual condition.
The current ATSC DTV standard does not authorize
the transmitter to use symbol precoding. The suppression
of co-channel interfering analog TV signal is presumed to
be carried out in the trellis decoding process, after the
data-slicing procedures associated with symbol decoding.
This procedure avoids the problem of determining whether
or not precoding is done at the transmitter. However,
co-channel interfering analog TV signal undesirably
introduces errors into the data-slicing processes, which
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places more burden on the error-correction decoding
procedures, trellis decoding and Reed-Solomon decoding.
These errors will reduce the broadcast coverage area,
which may lose revenue for the commercial DTV
broadcaster. So, providing for the suppression of
co-channel interfering analog TV signal before
data-slicing is still desirable, despite symbol precoding
at the DTV transmitter not being authorized by the current
ATSC DTV standard.
The term "linear combination" thereto refers
generically to addition and to subtraction, whether
performed in accordance with a conventional arithmetic or
a modular arithmetic. The term "modular combination"
refers to linear combination carried performed in
accordance with a modular arithmetic. That type of coding
that re-codes a digital symbol stream through differential
delay and linear combination of the differentially delayed
terms, exemplified by the symbol postcoding used in
prior-art HDTV receivers, is defined as "symbol re-coding
of first type" in this specification. That type of coding
that re-codes a digital symbol stream through its modular
combination with delayed result of the modular
combination, exemplified by the symbol precoding used in
prior-art HDTV transmitters, is defined as "symbol
re-coding of second type" in this specification.
The problem of co-channel interference from analog
television signals can be viewed from the standpoint of
being a sometime j amming problem at the receiver, to be
solved by adaptive filter circuitry in the receiver. So
long as the dynamic range of the system channel is not
exceeded, so that the co-channel interference can capture
the system channel by destroying signal transmission
capability for DTV modulation, the performance of the
system can be viewed as a superposition of signals
problem. The filter circuitry in the receiver is adapted
for selecting the digital signal from the co-channel
interference caused by the analog television signals,
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relying on the pronounced correlation and anti-correlation
properties of the analog television signals to reduce
their energy sufficiently as to capture the system channel
from them.
Insofar as the co-channel interference from analog
television signals is concerned, it enters the system
channel after the DTV transmitter and before the DTV
receiver. The use or non-use of symbol precoding at the
DTV transmitter has no effect on the co-channel
interference from analog television signals. At the DTV
receiver, so long as the co-channel interference is not so
large as to overload the receiver front-end and capture
the system channel, it is advantageous to precede the
data-slicing circuitry with a comb filter for reducing the
energy of higher-energy spectral components of the
co-channel interference, thus to reduce the errors
occurring during data-slicing. The DTV broadcaster should
adjust his carrier frequency, which is nominally 310KHz
above the lower limit frequency of the television channel
assignment, so that his carrier frequency is optimally
offset in frequency from the video carrier of a co-channel
NTSC analog TV signal that is likely to interfere. This
optimal offset in carrier frequency is exactly 59.75 times
the horizontal scan line frequency fH of the NTSC analog
TV signal. The artifacts of the co-channel interference
in the demodulated DTV signal will then include beats at
59.75 times the horizontal scan line frequency fH of the
NTSC analog TV signal, generated by heterodyne between the
digital HDTV carrier and the video carrier of the
co-channel interfering analog TV signal, and beats at
287.25 times fH, generated by heterodyne between the
digital HDTV carrier and the chrominance subcarrier of the
co-channel interfering analog TV signal, which beats are
quite close in frequency to the fifth harmonic of the
beats at 59.75 times fH. The artifacts will further
include beats at approximately 345.75 times fH, generated
by heterodyne between the digital HDTV carrier and the
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audio carrier of the co-channel interfering analog TV
signal, which beats are quite close in frequency to the
sixth harmonic of the beats at 59.75 times fH. The nearly
harmonic relationship of these beats allows them all to be
suppressed by a single properly designed comb filter
incorporating only a few symbol epochs of differential
delay. The use of an NTSC-rejection comb filter before
data-slicing in the DTV receiver incidentally performs
symbol re-coding of first type, to modify the symbols
obtained by data-slicing.
The data-slicing operation that follows this symbol
re-coding of first type in the DTV receiver is a
quantizing process that is not destructive of the symbols
resulting from the symbol re-coding of first type, insofar
as the transmission of data is concerned, since the data
quantization levels are designed to match the symbol
levels. The quantization discriminates against the
co-channel interfering analog TV signal remnants that
remain after the filtering associated with symbol
re-coding of first type and that are appreciably smaller
than steps between symbol code levels, however. This is a
species of the capture phenomenon in which phenomenon a
stronger signal gains at the expense of a weaker one in a
quantizing process.
Insofar as the transmission of data is concerned,
the digital data symbol stream flows through the full
length of the system channel. When symbol re-coding of
second type is done as symbol precoding at the DTV
transmitter, the additive combination of the
differentially delayed data symbol streams is done on a
modular basis that does not boost transmitter power or
increase average intersymbol distance to help further in
overcoming jamming analog TV signal. Instead, the
principal mechanism for overcoming jamming analog TV
signal is its attenuation vis-a-vis DTV signal, as
provided by the comb filtering at the DTV receiver,
causing the remnant analog TV signal in the comb filter
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response to be suppressed by the quantizing effects in the
data-slicer that immediately follows the comb filter.
The order of performing symbol re-coding procedures
of first and second types has no appreciable effect on
signal transmission through the system channel under such
circumstances, since neither coding scheme destroys signal
transmission capability for the symbol stream. The order
of performing symbol re-coding procedures of first and
second types has no appreciable effect on the capability
of the digital receiver to suppress co-channel interfering
analog TV signal, as long as symbol re-coding of the
second type is not interposed between symbol re-coding of
the first type and the subsequent data-slicing. These
insights provide the general foundation on which the
invention is based.
Co-channel interference accompanying multiple-level
symbols in a digital receiver, such as a digital
television receiver, is suppressed by using a first comb
filter to reduce the energy of the co-channel interference
before data-slicing. The first comb filter is supplied a
stream of 2N-level symbols each having a symbol epoch of a
specified length of time, which stream of 2N-level symbols
is susceptible to being accompanied by artifacts of
co-channel interfering analog television signal, and
supplies a response in which those artifacts of co-channel
interfering analog television signal are suppressed should
they obtain.
The first comb filter incidentally carries out a
symbol re-coding procedure of first type that introduces
error into the symbol decoding results generated by the
data-slicing. Suppose the first comb filter delays the
stream of 2N-level symbols by a prescribed number of
symbol epochs to generate a delayed stream of 2N-level
symbols, then linearly combines the stream of 2N-level
symbols and the delayed stream of 2N-level symbols to
generate first linear combining results as the response of
the first comb filter. This response, which has
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(4N-1)-level symbols, is supplied to a first data-sliver.
In the context of the invention, this symbol
re-coding procedure of first type carried out before
data-slicing by the first data-sliver is viewed as a
precoding procedure. A second comb filter carries out a
symbol re-coding procedure of second type after the
data-slicing, implementing a postcoding procedure to
compensate for the symbol re-coding procedure of first
type and generate corrected symbol decoding results. The
symbol re-coding procedure of first type re-codes an input
symbol stream through differential delay and first linear
combination of the differentially delayed terms. The
symbol re-coding procedure of second type re-codes
partially-filtered symbol decoding results recovered by
the first data-sliver. This symbol re-coding procedure of
second type utilizes second linear combination of the
partially-filtered symbol decoding results with delayed
result of the second linear combination and is performed
in accordance with a modular arithmetic. One of the first
and second linear combinations is subtractive, and the
other is additive. The results of the second linear
combination are postcoded symbol decoding results.
The postcoding done subsequent to comb filtering
and data-slicing has a basic problem that must be solved
in order for the postcoding to operate properly. One
aspect of this problem is that once error occurs in the
partially-filtered symbol decoding results, the error is
fed back with delay, tending to propagate the error during
the generation of postcoded symbol decoding results.
Other aspects of this problem concern how to initialize
the conditions in the delayed feedback circuitry and how
to re-initialize the conditions in the delayed feedback
circuitry once error propagation occurs.
This problem arises when re-coding of the second
type is used for postcoding because the feedback used in
such re-coding is accumulative and provides a sort of
integration over time. When re-coding of the second type
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is done during precoding and re-coding of the first type
is done during postcoding, the re-coding of the first type
provides a sort of differentiation over time that quickly
suppresses response to the initial conditions of the
re-coding of the second type. One does not have to
concern oneself with the initial conditions of
accumulation or integration. When re-coding of the first
type is done during precoding and re-coding of the second
type is done during postcoding, error caused by incorrect
initial conditions of accumulation or integration in the
re-coding of second type propagate themselves during
postcoding. The resulting running error in the final
decoding results is a systematic error, rather than a
random error, so generally speaking the running error will
not be able to self-correct itself by chance.
SUMMARY OF THE INVENTION
An aspect of the invention is a method of symbol
decoding a stream of 2N-level symbols each having a symbol
epoch of a specified length in time, which stream of
2N-level symbols is susceptible to being accompanied by
artifacts of co-channel interfering analog television
signal, N being a positive integer. The method generates
selected symbol decoding results by steps including a step
of comb filtering the stream of 2N-level symbols to
generate a comb filter response with (4N-1)-level precoded
symbols from which the artifacts of co-channel interfering
analog television signal, if any, are suppressed. This
step of comb filtering includes substeps of delaying the
stream of 2N-level symbols by a prescribed number of
symbol epochs to generate a delayed stream of 2N-level
symbols and linearly combining the stream of 2N-level
symbols and the delayed stream of 2N-level symbols, in
accordance with one of additive and subtractive
procedures, to generate first linear combining results as
the comb filter response with (4N-1)-level precoded
symbols. There is a step of data-slicing the comb filter
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response with (4N-1)-level precoded symbols to generate
precoded symbol decoding results. There are steps of
delaying the selected symbol decoding results by the
prescribed number of symbol epochs to generate delayed
selected symbol decoding results, and linearly combining
the precoded symbol decoding results with the delayed
selected symbol decoding results to generate second linear
combining results. The linear combining done to generate
second linear combining results is done in accordance with
an opposite of the additive and subtractive procedures
from the one used in the substep of linear combining done
to generate first linear combining results, and is
performed according to a modular arithmetic. There are
steps of determining when symbol coding descriptive of
synchronization data occurs in the stream of 2N-level
symbols, of regenerating the synchronization data without
error when the symbol coding descriptive of said
synchronization data occurs in the stream of 2N-level
symbols, and of generating the selected symbol decoding
results so as to correspond to the synchronization data
without error when the symbol coding descriptive of
synchronization data occurs in the stream of 2N-level
symbols and so as to correspond to the second linear
combining results at least during selected times when
symbol coding that is not descriptive of the
synchronization data occurs in the stream of 2N-level
symbols.
An aspect of the invention is a combination of
circuitry, as described in this paragraph, which circuitry
is included within a digital television receiver. The
combination includes digital television signal detection
apparatus for supplying a stream of 2N-level symbols each
having a symbol epoch of a specified length in time, which
stream is susceptible to being accompanied by artifacts of
co-channel interfering analog television signal. The
combination includes first and second delay devices, each
exhibiting a delay of a prescribed first number of said
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symbol epochs. The combination includes first and second
linear combiners, one of which is an adder and the other
of which is a subtractor, the second linear combiner
operating in a modulo-2N arithmetic. The first delay
device is connected to respond to the stream of 2N-level
symbols with a first delayed stream of 2N-level symbols,
thereby to generate a first pair of differentially delayed
streams of said 2N-level symbols. The first linear
combiner is connected for linearly combining the first
pair of differentially delayed streams of the 2N-level
symbols, which are received as first and second respective
input signals of the first linear combiner. Responsive to
these input signals the first linear combiner generates a
first stream of (4N-1)-level symbols as its output
signal. A first data-slicer is included in the
combination for generating first precoded symbol decoding
results by decoding the first stream of (4N-1)-level
symbols supplied as respective output signal from the
first linear combiner. The second linear combiner, for
linearly combining respective first and second input
signals received thereby for supplying a respective output
signal therefrom, is connected to receive the first
precoded symbol decoding results as a respective first
input signal thereof. The second delay device is
connected for delaying a respective input signal thereof
to generate the second input signal of the second linear
combiner. The combination further includes data
synchronization circuitry for determining when symbols
used for data synchronization appear in the stream of
2N-level symbols and circuitry for generating ideal symbol
decoding results when symbols used for data
synchronization are determined to appear in the stream of
2N-level symbols. The combination further includes a
plural-input first multiplexer connected for supplying a
respective output signal therefrom to the second delay
device as the respective input signal thereof, for
receiving the ideal symbol decoding results as a first of
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its input signals, and for receiving the output signal of
the second linear combiner as another of its input
signals. The first multiplexer is conditioned to
reproduce as its output signal the first of its input
signals when and only when symbols used for data
synchronization are determined to appear in the stream of
2N-level symbols. Otherwise the first multiplexer is
conditioned, at least at selected times, to reproduce the
output signal of the second linear combiner as first
postcoded symbol decoding results.
HRIEF DESCRIPTION OF THE DRAWING
Figure 1 is a block schematic diagram of a digital
television signal receiver using an NTSC-rejection comb
filter before symbol decoding and a postcoding comb filter
after symbol decoding, in accordance with the invention,
and using a co-channel interference detector that compares
the energies of the baseband.
Figure 2 is a block schematic diagram of an NTSC
co-channel interference detector for use in the Figure 1
digital television signal receiver.
Figure 3 is a block schematic diagram of a portion
of digital television signal receiver using an
NTSC-rejection comb filter before symbol decoding and a
postcoding comb filter after symbol decoding, in
accordance with the invention, and using a co-channel
interference detector of a type described by the inventor
in U.S. patent application serial No. (Atty. Dkt. 1500-1)
filed 21 March 1997.
Figure 4 is a block schematic diagram of a portion
of a digital television signal receiver using an
NTSC-rejection comb filter before symbol decoding and a
postcoding comb filter after symbol decoding, in
accordance with the invention, and using a co-channel
interference detector of a type described by the inventor
in U.S. patent application serial No. (Atty. Dkt. 1501-1)
filed 21 March 1997.
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Figure 5 is a block schematic diagram showing
details of a portion of the Figure l, Figure 3 or Figure 4
digital television signal receiver concerning the
selection of final symbol decoding results, selected from
prescribed symbol decoding results during data
synchronization intervals and selected at other times from
data-slicer response to the received baseband symbol codes
or from postcoded data-slicer response to comb filter
response to the received baseband symbol codes, depending
on whether or not the received baseband symbol codes are
substantially free of NTSC co-channel interference.
Figure 6 is a block schematic diagram of circuitry
alternative to that of Figure 5.
Figure 7 is a block schematic diagram of other
circuitry alternative to that of Figure 5.
Figure 8 is a block schematic diagram showing
details of a portion of the Figure 1, Figure 3 or Figure 4
digital television signal receiver for generating
prescribed symbol decoding results during data
synchronization intervals.
Figure 9 is a block schematic diagram showing
details of a portion of the Figure 1, Figure 3 or Figure 4
digital television signal receiver when the NTSC-rejection
comb filter employs a 12-symbol delay.
Figure 10 is a block schematic diagram showing
details of a portion of the Figure 1, Figure 3 or Figure 4
digital television signal receiver when the NTSC-rejection
comb filter employs a 6-symbol delay.
Figure 11 is a block schematic diagram showing
details of a portion of the Figure 1, Figure 3 or Figure 4
digital television signal receiver when the NTSC-rejection
comb filter employs a 2-video-line delay.
Figure 12 is a block schematic diagram showing
details of a portion of the Figure 1, Figure 3 or Figure 4
digital television signal receiver when the NTSC-rejection
comb filter employs a 262-video-line delay.
Figure 13 is a block schematic diagram showing
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details of a portion of the Figure 1, Figure 3 or Figure 4
digital television signal receiver when the NTSC-rejection
comb filter employs a 2-video-frame delay.
Figure 14 is a block schematic diagram showing a
digital television signal receiver using a plurality of
NTSC-rejection comb filters for performing parallel symbol
decoding.
Figure 15 is an assembly diagram showing how
Figures 15A and 15B can be fitted together to form a
single figure referred to as Figure 15 in the detailed
description that follows, which Figure 15 shows details of
symbol code selection circuitry that can be used in a
digital television signal receiver of the type shown in
Figure 14.
Figure 15A is a block schematic diagram showing
details of circuitry in the Figure 14 digital television
signal receiver for generating prescribed symbol decoding
results during data synchronization intervals.
Figure 15B is a block schematic diagram showing
details of circuitry in the Figure 14 digital television
signal receiver for selecting among symbol decoding
results during time periods between data synchronization
intervals.
DETAILED DESCRIPTION
At various points in the circuits shown in the
Figures of the drawing, shimming delays have to be
inserted in order that the sequence of operation is
correct, as will be understood by those skilled in
electronic design. Unless there is something out of the
ordinary about a particular shimming delay requirement, it
will not be explicitly referred to in the specification
that follows.
Figure 1 shows a digital television signal receiver
used for recovering error-corrected data, which data are
suitable for recording by a digital video cassette
recorder or for MPEG-2 decoding and display in a
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television set. The Figure 1 DTV signal receiver is shown
as receiving television broadcast signals from a receiving
antenna 8, but can receive the signals from a cable
network instead. The television broadcast signals are
supplied as input signal to "front end" electronics 10.
The "front end" electronics 10 generally include a
radio-frequency amplifier and first detector for
converting radio-frequency television signals to
intermediate-frequency television signals, supplied as
input signal to an intermediate-frequency (I-F) amplifier
chain 12 for vestigial-sideband DTV signals. The DTV
receiver is preferably of plural-conversion type with the
IF amplifier chain 12 including an intermediate-frequency
amplifier for amplifying DTV signals as converted to an
ultra-high-frequency band by the first detector, a second
detector for converting the amplified DTV signals to a
very-high-frequency band, and a further
intermediate-frequency amplifier for amplifying DTV
signals as converted to the VHF band. If demodulation to
baseband is performed in the digital regime, the IF
amplifier chain 12 will further include a third detector
for converting the amplified DTV signals to a final
intermediate-frequency band closer to baseband.
Preferably, a surface-acoustic-wave (SAW) filter is
used in the IF amplifier for the UHF band, to shape
channel selection response and reject adjacent channels.
This SAW filter cuts off rapidly just beyond 5.38 MHz
remove from the suppressed carrier frequency of the VSB
DTV signal and the pilot carrier, which is of like
frequency and of fixed amplitude. This SAW filter
accordingly rejects much of the frequency-modulated sound
carrier of any co-channel interfering analog TV signal.
Removing the FM sound carrier of any co-channel
interfering analog TV signal in the IF amplifier chain 12
prevents artifacts of that carrier being generated when
the final I-F signal is detected to recover baseband
symbols and forestalls such artifacts interfering with
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data-slicing of those baseband symbols during symbol
decoding. The prevention of such artifacts interfering
with data-slicing of those baseband symbols during symbol
decoding is better than can be accomplished by relying on
comb-filtering before data-slicing.
The final I-F output signals from the IF amplifier
chain 12 are supplied to a complex demodulator 14, which
demodulates the vestigial-sideband amplitude-modulation
DTV signal in the final intermediate-frequency band to
recover a real baseband signal and an imaginary baseband
signal. Demodulation may be done in the digital regime
after analog-to-digital conversion of a final
intermediate-frequency band in the few megacycle range as
described for example by C.B. Patel et alii in U.S. patent
No. 5,479,449 issued 26 December 1995 and entitled
"DIGITAL VSB DETECTOR WITH PHASE TRACKER, AS FOR INCLUSION
IN AN HDTV RECEIVER". Alternatively, demodulation may be
done in the analog regime, in which case the results are
usually subjected to analog-to-digital conversion to
facilitate further processing. The complex demodulation
is preferably done by in-phase (I) synchronous
demodulation and quadrature-phase (Q) synchronous
demodulation. The digital results of the foregoing
demodulation procedures conventionally have 8-bit accuracy
or more and describe 2N-level symbols that encode N bits
of data. Currently, 2N is eight in the case where the
Figure 1 DTV signal receiver receives a through-the-air
broadcast via the antenna 12 and is sixteen in the case
where the Figure 1 DTV signal receiver receives
cablecast. The concern of the invention is with the
reception of terrestrial through-the-air broadcasts, and
Figure 1 does not show the portions of the DTV receiver
providing symbol decoding and error-correction decoding
for received cablecast transmissions.
Symbol synchronizer and equalizer circuitry 16
receives at least the digitized real samples of the
in-phase (I-channel) baseband signal from the complex
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demodulator 14; in the Figure 1 DTV receiver the circuitry
16 is shown also receiving the digitized imaginary samples
of the quadrature-phase (Q-channel) baseband signal. The
circuitry 16 includes a digital filter with adjustable
weighting coefficients that compensates for ghosts and
tilt in the received signal. The symbol synchronizer and
equalizer circuitry 16 provides symbol synchronization or
"de-rotation" as well as amplitude equalization and ghost
removal. Symbol synchronizer and equalizer circuitry in
which symbol synchronization is accomplished before
amplitude equalization is known from U.S. patent No.
5,479,449. In such designs the demodulator 14 will supply
oversampled demodulator response containing real and
imaginary baseband signals to the symbol synchronizer and
equalizer circuitry 16. After symbol synchronization, the
oversampled data are decimated to extract baseband
I-channel signal at normal symbol rate, to reduce sample
rate through the digital filtering used for amplitude
equalization and ghost removal. Symbol synchronizer and
equalizer circuitry in which amplitude equalization
precedes symbol synchronization, "de-rotation" or "phase
tracking" is also known to those skilled in the art of
digital signal receiver design.
Each sample of circuitry 16 output signal is
resolved to ten or more bits and is, in effect, a digital
description of an analog symbol exhibiting one of (2N=8)
levels. The circuitry 16 output signal is carefully
gain-controlled by any one of several known methods, so
the ideal step levels for symbols are known. One method_
of gain control, preferred because the speed of response
of such gain control is exceptionally rapid, regulates the
direct component of the real baseband signal supplied from
the complex demodulator 14 to a normalized level of
+1.25. This method of gain control is generally described
in U.S. patent No. 5,479,449 and is more specifically
described by C.B. Patel et al in allowed
U.S. patent No. 5,573,454
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CA 02241067 2000-07-24
entitled "AUTOMATIC GAIN CONTROL OF RADIO RECEIVER FOR
RECEIVING DIGITAL HIGH-DEFINITION TELEVISION SIGNALS".
The output signal from the circuitry 16 is supplied
as input signal to data sync detection circuitry 18, which
recovers data field synchronization information F and data
segment synchronization information S from the equalized
baseband I-channel signal. Alternatively, the input
signal to data sync detection circuitry 18 can be obtained
prior to equalization.
The equalized I-channel signal samples at normal
symbol rate supplied as output signal from the circuitry
16 are applied as the input signal to an NTSC-rejection
comb filter 20. The comb filter 20 includes a first delay
device 201 to generate a pair of differentially delayed
streams of the 2N-level symbols and a first linear
combiner 202 for linearly combining the differentially
delayed symbol streams to generate the comb filter 20
response. As described in U.S. patent No. 5,260,793, the
first delay device 201 can provide a delay equal to the
period of twelve 2N-level symbols, and the first linear
combiner 202 can be a subtractor. Each sample of the comb
filter 20 output signal is resolved to ten or more bits
and is, in effect, a digital description of an analog
symbol exhibiting one of (4N-1)=15 levels.
The symbol synchronizer and equalizer circuitry 16
is presumed to be designed to suppress the direct bias
component of its input signal (as expressed in digital
samples), which direct bias component has a normalized
level of +1.25 and appears in the real baseband signal
supplied from the complex demodulator 14 owing to
detection of the pilot carrier. Accordingly, each sample
of the circuitry 16 output signal applied as comb filter
20 input signal is, in effect, a digital description of an
analog symbol exhibiting one of the following normalized
levels: -7, -5, -3, -1, +1, +3, +5 and +7. These symbol
levels are denominated as "odd" symbol levels and are
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CA 02241067 1999-06-28
detected by an odd-level data-slicer 22 to generate
interim symbol decoding results of 000, 001, 010, 011,
100, 101, 110 and 111, respectively.
Each sample of the comb filter 20 output signal is,
in effect, a digital description of an analog symbol
exhibiting one of the following normalized levels: -14,
-12, -10, -8, -6, -4, -2, 0, +2, +4, +6, +8, +10, +12 and
+14. These symbol levels are denominated as "even" symbol
levels and are detected by an even-level data-slicer 24 to
generate precoded symbol decoding results of 001, 010,
011, 100, 101, 110, 111, 000, 001, 010, 011, 100, 101,
110, and 111, respectively.
The data-slicers 22 and 24 can be of the so-called
"hard decision" type, as presumed up to this point in the
description, or can be of the so-called "soft decision"
type used in implementing a Viterbi decoding scheme.
Arrangements are possible in which the odd-level
data-slicer 22 and the even-level data-slicer 24 are
replaced by a single data-sliver, using multiplexer
connections to shift its place in circuit and to provide
bias to modify its slicing ranges, but these arrangements
are not preferred because of the complexity of operation.
The symbol synchronizer and equalizer circuitry 16
is presumed in the foregoing description to be designed to
suppress the direct bias component of its input signal (as
expressed in digital samples), which direct bias component
has a normalized level of +1.25 and appears in the real
baseband signal supplied from the complex demodulator 14
owing to detection of the pilot carrier. Alternatively,
the symbol synchronizer and equalizer circuitry 16 is
designed to preserve the direct bias component of its
input signal, which simplifies the design of the
equalization filter in the circuitry 16 somewhat. In such
case the data-slicing levels in the odd-level data-sliver
22 are offset to take into account the direct bias
component accompanying the data steps in its input
signal. Providing that the first linear combiner 202 is a
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subtractor, whether the circuitry 16 is designed to
suppress or to preserve the direct bias component of its
input signal has no consequence in regard to the
data-slicing levels in the even-level data-slicer 24.
However, if the differential delay provided by the first
delay device 201 is chosen so that the first linear
combiner 202 is an adder, the data-slicing levels in the
even-level data-slicer 24 should be offset to take into
account the doubled direct bias component accompanying the
data steps in its input signal.
A comb filter 26 is used after the data-slicers 22
and 24 to generate a postcoding filter response to the
precoding filter response of the comb filter 20. The comb
filter 26 includes a 3-input multiplexer 261, a second
linear combiner 262, and a second delay device 263 with
delay equal to that of the first delay device 201 in the
comb filter 20. The second linear combiner 262 is a
modulo-8 adder if the first linear combiner 202 is a
subtractor and is a modulo-8 subtractor if the first
linear combiner 202 is an adder. The first linear
combiner 202 and the second linear combiner 262 may be
constructed as respective read-only memories (ROMs) to
speed up linear combination operations sufficiently to
support the sample rates involved. The output signal from
the multiplexer 261 furnishes the response from the
postcoding comb filter 26 and is delayed by the second
delay device 263. The second linear combiner 262 combines
precoded symbol decoding results from the even-level
data-slicer 24 with the output signal from the second
delay device 263.
The output signal of the multiplexer 261 reproduces
one of the three input signals applied to the multiplexer
261, as selected in response to first, second and third
states of a multiplexer control signal supplied to the
multiplexer 261 from a controller 28. The first input
port of the multiplexer 261 receives ideal symbol decoding
results supplied from memory within the controller 28
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during times when data field synchronization information F
and data segment synchronization information S from the
equalized baseband I-channel signal are recovered by the
data sync detection circuitry 18. The controller 28
supplies the first state of the multiplexer control signal
to the multiplexer 261 during these times, conditioning
the multiplexer 261 to furnish, as the final coding
results which are its output signal, the ideal symbol
decoding results supplied from memory within the
controller 28. The odd-level data-slicer 22 supplies
interim symbol decoding results as its output signal to
the second input port of the multiplexer 261. The
multiplexer 261 is conditioned by the second state of the
multiplexer control signal to reproduce the interim symbol
decoding results, as the final coding results which are
its output signal. The second linear combiner 262
supplies postcoded symbol decoding results as its output
signal to the third input port of the multiplexer 261.
The multiplexer 261 is conditioned by the third state of
the multiplexer control signal to reproduce the postcoded
symbol decoding results, as the final coding results which
are its output signal.
Running errors in the postcoded symbol decoding
results from the postcoding comb filter 26 are curtailed
by feeding back the ideal symbol decoding results supplied
from memory within the controller 28 during times data
sync detection circuitry 18 recovers data field
synchronization information F and data segment
synchronization information S. This is an important
aspect of the invention, which will be described in
greater detail further on in this specification.
The output signal from the multiplexer 261 in the
postcoding comb filter 26 comprises the final symbol
decoding results in 3-parallel-bit groups, assembled by a
data assembler 30 for application to a data interleaver
32. The data interleaver 32 commutates the assembled data
into parallel data streams for application to trellis
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decoder circuitry 34. Trellis decoder circuitry 34
conventionally uses twelve trellis decoders. The trellis
decoding results are supplied from the trellis decoder
circuitry 34 to data de-interleaver circuitry 36 for
de-commutation. Byte parsing circuitry 38 converts the
data interleaver 36 output signal into bytes of
Reed-Solomon error-correction coding for application to
Reed-Solomon decoder circuitry 40, which performs
Reed-Solomon decoding to generate an error-corrected byte
stream supplied to a data de-randomizer 42. The data
de-randomizer 42 supplies reproduced data to the remainder
of the receiver (not shown). The remainder of a complete
DTV receiver will include a packet sorter, an audio
decoder, an MPEG-2 decoder and so forth. The remainder of
a DTV receiver incorporated in a digital tape
recorder/reproducer will include circuitry for converting
the data to a form for recording.
An NTSC co-channel interference detector 44
supplies the controller 28 with an indication of whether
NTSC co-channel interference is of sufficient strength as
to cause uncorrectable error in the data-slicing performed
by the data-slicer 22. If detector 44 indicates the NTSC
co-channel interference is not of such strength, the
controller 28 will supply the second state of multiplexer
control signal to the multiplexer 261 at times other than
those times when data field synchronization information F
and data segment synchronization information S are
recovered by the data sync detection circuitry 18. This
conditions the multiplexer 261 to reproduce as its output
signal the interim symbol decoding results supplied from
the odd-level data-slicer 22. If detector 44 indicates
the NTSC co-channel interference is of sufficient strength
to cause uncorrectable error in the data-slicing performed
by the data-slicer 22, the controller 28 will supply the
third state of multiplexer control signal to the
multiplexer 261 at times other than those times when data
field synchronization information F and data segment
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synchronization information S are recovered by the data
sync detection circuitry 18. This conditions the
multiplexer 261 to reproduce as its output signal the
postcoded symbol decoding results provided as second
linear combining results from the second linear combiner
262.
Figure 2 shows a form the NTSC co-channel
interference detector 44 can take, which form is believed
to be novel in the art. A subtractor 441 differentially
combines the interim symbol decoding results supplied from
the odd-level data-slicer 22 and the postcoded symbol
decoding results provided as second linear combining
results from the second linear combiner 262. If the
amount of NTSC co-channel interference is negligible, and
if the random noise in the baseband I-channel signal is
negligible, these interim and postcoded symbol decoding
results should be similar, so the difference output signal
from the subtractor 441 should be low. If the amount of
NTSC co-channel interference is appreciable, however, the
difference output signal from the subtractor 441 will not
be generally low, but rather will often be high.
A measure of the energy in the difference output
signal from the subtractor 441 is developed by squaring
the difference output signal with a squarer 442 and
determining the mean average of the squarer response over
a prescribed short time interval with a mean averaging
circuit 443. The squarer 442 can be implemented using
read-only memory (ROM). The mean averaging circuit 443
can be implemented using a delay line memory for storing
several successive digital samples and an adder for
summing the digital samples currently stored in the delay
line memory. The short-term mean average of the energy in
the difference output signal from the subtractor 441, as
determined by the mean averaging circuit 443, is supplied
to a digital comparator connected to provide a threshold
detector 444. The threshold in the threshold detector 444
is sufficiently high not to be exceeded the short-term
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mean-average of differences in the random noise
accompanying the interim symbol decoding results and the
postcoded symbol decoding results applied to the
subtractor 441. The threshold is exceeded if the NTSC
co-channel interference is of sufficient strength as to
cause uncorrectable error in the data-slicing performed by
the data-slicer 22. The threshold detector 444 supplies
the controller 28 indication of whether or not the
threshold is exceeded.
Figure 3 shows a digital television receiver
differing from that of Figure 1 in that the circuitry for
determining whether or not NTSC co-channel interference is
of sufficient strength as to cause uncorrectable error in
the data-slicing performed by the data- dicer 22 is of the
type described by the inventor in U. S. patent
No. 5,801,790 and entitled "USING VIDEO SIGNALS FROM
AUXILIARY ANALOG TV RECEIVERS FOR DETECTING NTSC
INTERFERENCE IN DIGITAL TV RECEIVERS". The DTV signal, as
converted to IF by the "front end" electronics 10, is
supplied to an IF amplifier chain 46 for NTSC signals.
The IF amplifier chain 46 for NTSC signals differs from
the IF amplifier chain used in conventional NTSC signal
receivers. Insofar as midband gain characteristics are
concerned, amplifier stages in the IF amplifier chain 46
for NTSC signals correspond to the amplifier stages in the
IF amplifier chain 12 for DTV signals, having
substantially linear gain and having the same automatic
gain control as the corresponding amplifier stages in the
IF amplifier chain 12. The vestigial sideband of the NTSC
signal is not suppressed in the IF amplifier chain 46.
The portion of the full sideband of the NTSC signal that
is single-sideband in character is preferably suppressed
in the IF amplifier chain 46 to reduce the energy of
co-channel DTV signal. The reduces the dynamic range of
IF amplifier chain 46 response, facilitating additional
amplification of video carrier for locking the phase of a
local video carrier oscillator used
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in the complex demodulator 48. The filtering procedures
to establish the bandwidth of the IF amplifier chain 46
can be carried out by SAW filtering in a UHF IF amplifier
if plural-conversion receiver circuitry is used. The
amplified IF response of the IF amplifier chain 46 is
supplied to a complex demodulator 48 for NTSC video
signal, either directly or after some further
amplification. The complex demodulator 48 supplies an
in-phase I-channel response composed of samples of NTSC
signal and the real component of accompanying DTV
artifacts. The complex demodulator 48 also supplies a
quadrature-phase Q-channel response composed of samples of
the imaginary component of accompanying DTV artifacts,
which samples are applied to a Hilbert transformation
filter 50. The Hilbert transformation filter 50 response
is supplied to a linear combiner 52. The linear combiner
52 combines the Hilbert transformation filter 50 response
with suitably delayed in-phase I-channel response, to
recover samples of NTSC signal substantially free from
accompanying DTV artifacts. The linear combiner 52 is an
adder or a subtractor depending on relative video carrier
phasing during the synchronous demodulation procedures
used in the complex demodulator 48 to generate the
I-channel and Q-channel responses.
The NTSC signal substantially free from
accompanying DTV artifacts supplied from the linear
combiner 52 is applied to a lowpass filter 54 with a
cut-off frequency of 750 kHz or less. An estimate of
luminance signal energy in the co-channel interfering NTSC
signal is generated by squaring the lowpass filter 54
response with a squarer 56 and determining the mean
average of the squarer response over a prescribed short
time interval with a mean averaging circuit 58. This
estimate is supplied to a threshold detector 58. The
threshold in the threshold detector 58 is exceeded if the
NTSC co-channel interference is of sufficient strength as
to cause uncorrectable error in the data-slicing performed
- 27 -
CA 02241067 2000-07-24
by the data-slicer 22. The threshold detector 58 supplies
the controller 28 indication of whether or not the
threshold is exceeded.
Figure 4 shows a digital television receiver
differing from the Figure 1 and Figure 3 receivers in that
the circuitry for determining whether or not NTSC
co-channel interference is of sufficient strength as to
cause uncorrectable error in the data-slicing performed by
the data-slicer 22 is of the type described by the
inventor in U. S. patent No. 5,801,790 and entitled "USING
INTERCARRIER SIGNALS FOR DETECTING NTSC INTERFERENCE IN
DIGITAL TV RECEIVERS". The DTV signal, as converted to IF
by the "front end" electronics 10, is supplied to an IF
amplifier chain 62 of quasi-parallel type for NTSC sound
signals. The amplifier stages in the IF amplifier chain
62 for NTSC sound signals correspond to similar amplifier
stages in the IF amplifier chain 12 for DTV signals,
having substantially linear gain and having the same
automatic gain control as the corresponding amplifier
stages in the IF amplifier chain 12. The frequency
selectivity of the IF amplifier chain 62 is such as to
emphasize response within ~250 kHz of NTSC audio carrier
and within ~250 kHz or so of NTSC video carrier. The
filtering procedures to establish the frequency
selectivity of the IF amplifier chain 62 can be carried
out by SAW filtering in a UHF IF amplifier if
plural-conversion receiver circuitry is used. The
response of the IF amplifier chain 62 is supplied to an
intercarrier detector 64 which uses the modulated NTSC
video carrier as an exalted carrier for heterodyning the.
NTSC audio carrier to generate intercarrier sound
intermediate~frequency signal with a 4.5 MHz carrier
frequency. This intercarrier sound IF signal is amplified
by an intercarrier-sound intermediate-frequency amplifier
66, which 4.5 MHz IF amplifier 66 supplies amplified
intercarrier sound IF signal to an intercarrier amplitude
- 28 -
CA 02241067 1999-06-28
detector 68. The response of the amplitude detector 68 is
averaged over a prescribed short time interval with a mean
averaging circuit 70, and the resulting mean average is
supplied to a threshold detector 72. The threshold in the
threshold detector 72 is exceeded if the NTSC co-channel
interference is of sufficient strength as to cause
uncorrectable error in the data-slicing performed by the
data-slices 22. The threshold detector 72 supplies the
controller 28 indication of whether or not the threshold
is exceeded.
Figure 5 shows a preferred way in which the
multiplexes 261 in the postcoding comb filter 26 is
implemented. The 3-input multiplexes 261 is shown as
comprising two 2-input multiplexers 2611 and 2612. The
controller 28 applies the output signal from the NTSC
co-channel interference detector (e. g., 44) as control
signal to the 2-input multiplexes 2611.
If the NTSC co-channel interference is of
sufficient strength to cause uncorrectable error in the
data-slicing performed by the data-slices 22, the
resulting ONE output signal from the NTSC co-channel
interference detector conditions the multiplexes 2611 to
reproduce, for application to the second input port of the
multiplexes 2612, the postcoded symbol decoding results
the second linear combines 262 supplies to the first input
port of the multiplexes 2611.
If the NTSC co-channel interference is of
insufficient strength to cause uncorrectable error in the
data-slicing performed by the data-slices 22, the
resulting ZERO output signal from the NTSC co-channel
interference detector conditions the multiplexes 2611 to
reproduce the interim symbol decoding results the
data-slices 22 supplies to the second input port of the
multiplexes 2611. These reproduced interim symbol
decoding results are applied to the second input port of
the multiplexes 2612.
Figures 5, 6 and 7 each show an OR gate 281 being
- 29 -
CA 02241067 1999-06-28
included in the controller 28. The OR gate 281 supplies a
response that is a ONE, when the field segment sync
detector 181 supplies a ONE thereto in response to the
occurrence of a field sync segment being detected, and
when the data segment sync detector 182 supplies a ONE
thereto in response to the occurrence of a data sync code
being detected. At all other times the OR gate 281
supplies a response that is a ZERO.
In Figure 5 the OR gate 281 response is applied as
control signal to the multiplexer 2612. The OR gate 281
response being ZERO conditions the multiplexer 2612 to
reproduce, as final symbol decoding result for application
to the data assembler 30, the output signal of the
multiplexer 2611 supplied to the second input port of the
multiplexer 2612 as better estimate of symbol decoding
result. The OR gate 281 response being ONE conditions the
multiplexer 2612 to reproduce, as final symbol decoding
result for application to the data assembler, ideal
decoding results drawn from memory in the controller 28,
as will be described in detail further on in this
specification with reference to Figure 8 of the drawing.
Figure 6 shows an alternative construction 260 of
the post coding comb filter 26. The 3-input multiplexer
261 comprising two 2-input multiplexer 2611 and 2612 is
replaced by a 3-input multiplexer 2610 comprising three
2-input multiplexers 26101, 26102 and 26103.
Figure 7 shows a modification 2600 of the post
coding comb filter 26, in which the 3-input multiplexer
261 comprising two 2-input multiplexers 2611 and 2612 is
replaced by a 3-input multiplexer 26100 comprising two
2-input multiplexers 261001 and 261002 receiving their
respective control signals from the OR gate 281 and from
the NTSC co-channel interference detector. The post
coding comb filter 2600 provides somewhat different
operating result than the post coding comb filters 26 and
260. The multiplexer 261001 replaces postcoded symbol
decoding results with ideal symbol decoding results when
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CA 02241067 1999-06-28
the OR gate 281 response is ONE. When NTSC co-channel
interference detector supplies a ONE indicative that NTSC
co-channel interference is of sufficient strength to cause
uncorrectable error in the data-slicing performed by the
data-slicer 22, a multiplexer 261002 selects the resulting
modified postcoded symbol decoding results as final symbol
decoding results for application to the data assembler
30. When NTSC co-channel interference detector supplies a
ZERO indicative that NTSC co-channel interference is of
insufficient strength to cause uncorrectable error in the
data-slicing performed by the data-slicer 22, the
multiplexer 261002 selects the interim symbol decoding
results from the data-slicer 22 as final symbol decoding
results for application to the data assembler 30, without
any replacement of those interim symbol decoding results
by ideal symbol decoding results.
Figure 8 shows the multiplexer 2612 of Figure 5 in
greater detail, together with the circuitry for generating
the ideal symbol decoding results applied to the
multiplexer 2612. The multiplexer 2612 comprises the
output buffer registers of read-only memories (ROMs) 74,
76, 78 for selectively reading to a 3-bit-wide output bus
80 from the multiplexer 2612. The multiplexer 2612
further comprises a tri-state buffer 82 for selectively
forwarding the 3-bit-wide output of the multiplexer 2611
to the output bus 80.
The circuitry for generating the ideal symbol
decoding results applied to the multiplexer 2612 comprises
the ROMs 74, 76, 78; a symbol clock generator 84; an
address counter 86 for addressing the ROMs 74, 76, 78; jam
reset circuitry 88 for resetting the counter 86; the
address decoders 94, 96, 98 for generating read enable
signals for the ROMs 74, 76, 78; and a NOR gate 92 for
controlling the tri-state buffer 82. The address counter
86 counts input pulses received at symbol decoding rate
from the symbol clock generator 84, thereby to generate
successive addresses respectively descriptive of the
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CA 02241067 1999-06-28
symbols in one data frame. Suitable portions of these
addresses are applied to the ROMs 74, 76, 78 as their
input addresses. The jam reset circuitry 88 resets the
counter 86 to appropriate counts responsive to data field
synchronization information F and data segment
synchronization information S recovered by the data sync
detection circuitry 18 of Figure 1, 3 or 4.
It is preferable to configure the counter 86 so a
group of more significant bits counts the number of data
segments per data frame and so a group of less significant
bits counts the number of symbols per data segment. This
simplifies the design of the jam reset circuitry 88;
reduces the bit-widths of input signal to the address
decoders 94, 96, 98; and facilitates the ROMs 74, 76, 78
being addressed by partial addresses from the counter 86,
reducing the bit widths of ROM addressing.
The ROM 74 stores ideal symbol decoding results for
an odd field sync segment and is selectively enabled for
reading by receiving a ONE from the address decoder 94.
The ROM 74 is addressed by the group of less significant
bits of counter 86 output that counts the number of
symbols per data segment group; and the address decoder 94
responds to the group of more significant bits that counts
the number of data segments per data frame. The address
decoder 94 generates a ONE when and only when the data
segment portion of the address supplied by the address
counter 86 corresponds to the address of an odd field sync
segment.
The ROM 76 stores ideal symbol decoding results for
an even field sync segment and is selectively enabled for
reading by receiving a ONE from the address decoder 96.
The ROM 76 is addressed by the group of less significant
bits of counter 86 output that counts the number of
symbols per data segment group; and the address decoder 96
responds to the group of more significant bits that counts
the number of data segments per data frame. The address
decoder 96 generates a ONE when and only when the data
- 32 -
CA 02241067 1999-06-28
segment portion of the address supplied by the address
counter 86 corresponds to the address of an even field
sync segment.
The ROM 78 stores ideal symbol decoding results for
the start code group at the beginning of each sync segment
and is selectively enabled for reading by receiving a ONE
from the address decoder 98. The ROM 78 responds to the
two least significant bits of counter 86 output; and the
address decoder 98 responds to the group of less
significant bits of counter 86 output that counts the
number of symbols per data segment group. The address
decoder 98 generates a ONE when and only when the data
symbol per data segment count portion of the address
supplied by the address counter 86 corresponds to the
partial address of a start code group.
The NOR gate 92 receives the responses of the
address decoders 94, 96 and 98 at respective ones of its
three input connections. When ideal symbol decoding
results are available, one of the address decoders 94, 96
and 98 supplies a ONE as its output signal, conditioning
the NOR gate 92 to supply a ZERO response to the tri-state
data buffer 82. This conditions the tri-state data buffer
82 to exhibit a high source impedance to the data bus 80,
so the signal forwarded from the multiplexer 2611 will not
be asserted on the 3-bit-wide data bus 80 from the
multiplexer 2612. During those portions of data segments
for which ideal symbol decoding results are not
predictable, none of the address decoders 94, 96 and 98
supplies a ONE as its output signal, conditioning the NOR
gate 92 to supply a ONE response to the tri-state data
buffer 82. This conditions the tri-state data buffer 82
to exhibit a low source impedance to the data bus 80, so
the signal forwarded from the multiplexer 2611 will be
asserted on the 3-bit-wide data bus 80 from the
multiplexer 2612.
The Figure 8 circuitry for generating ideal symbol
decoding results applied to the multiplexer 2612 is
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CA 02241067 1999-06-28
readily adapted by one skilled in the art of digital
circuit design for use in the configurations shown in
Figures 6 and 7.
Figure 9 is a block schematic diagram showing
details of a portion of the Figure 1, Figure 3 or Figure 4
digital television signal receiver using a species 120 of
the NTSC-rejection comb filter 20 and a species 126 of the
postcoding comb filter 26. A subtractor 1202 serves as
the first linear combiner in the NTSC-rejection comb
filter 120, and a modulo-8 adder 1262 serves as the second
linear combiner in the postcoding comb filter 126. The
NTSC-rejection comb filter 120 uses a first delay device
1201 exhibiting a delay of twelve symbol epochs, and a
postcoding comb filter 126 uses a second delay device 1263
also exhibiting a delay of twelve symbol epochs. The
12-symbol delay exhibited by each of the delay devices
1201 and 1263 is close to one cycle delay of the artifact
of the analog TV video carrier at 59.75 times the analog
TV horizontal scan frequency fH. The 12-symbol delay is
close to five cycles of the artifact of the analog TV
chrominance subcarrier at 287.25 times fH. The 12-symbol
delay is close to six cycles of the artifact of the analog
TV sound carrier at 345.75 times fH. This is the reason
that the differentially combined response of the
subtractor 1202 to the audio carrier, to the video carrier
and to frequencies close to chrominance subcarrier
differentially delayed by the first delay device 1201
tends to have reduced co-channel interference. However,
in portions of a video signal in which edges cross a
horizontal scan line, the amount of correlation in the
analog TV video signal at such distances in the horizontal
spatial direction is quite low.
A species 1261 of the multiplexer 261 is controlled
by a multiplexer control signal that is in its second
state most of the time when it is determined there is
insufficient NTSC co-channel interference to cause
uncorrectable error in the output signal from the
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data-slicer 22 and that is in its third state most of the
time when it is determined there is sufficient NTSC
co-channel interference to cause uncorrectable error in
the output signal from the data-slicer 22. The
multiplexer 1261 is conditioned by its control signal
being in its third state to feed back the modulo-8 sum
results of the adder 1262, as delayed twelve symbol epochs
by the delay device 1263, to the adder 1262 as a summand.
This is a modular accumulation procedure in which a single
error propagates as a running error, with error recurring
every twelve symbol epochs. Running errors in the
postcoded symbol decoding results from the postcoding comb
filter 126 are curtailed by the multiplexer 1261 being
placed into its first state for four symbol epochs at the
beginning of each data segment, as well as during the
entirety of each data segment containing field sync. When
this control signal is in its first state, the multiplexer
1261 reproduces as its output signal ideal symbol decoding
results supplied from memory in the controller 28. The
introduction of ideal symbol decoding results into the
multiplexer 1261 output signal halts a running error.
Since there are 4 + 69(12) symbols per data segment, the
ideal symbol decoding results slip back four symbol epochs
in phase each data segment, so no running error can
persist for longer than three data segments.
Figure 10 is a block schematic diagram showing
details of a portion of the Figure 1, Figure 3 or Figure 4
digital television signal receiver using a species 220 of
the NTSC-rejection comb filter 20 and a species 226 of the
postcoding comb filter 26. The NTSC-rejection comb filter
220 uses a first delay device 2201 exhibiting a delay of
six symbol epochs, and the postcoding comb filter 226 uses
a second delay device 2263 also exhibiting a delay of six
symbol epochs. The 6-symbol delay exhibited by each of
the delay devices 2201 and 2263 is close to 0.5 cycle
delay of the artifact of the analog TV video carrier at
59.75 times the analog TV horizontal scan frequency fg,
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close to 2.5 cycles of the artifact of the analog TV
chrominance subcarrier at 287.25 times fH, and close to 3
cycles of any artifact of the analog TV audio carrier at
345.75 times fH. An adder 2202 serves as the first linear
combiner in the NTSC-rejection comb filter 220, and a
modulo-8 subtractor 2262 serves as the second linear
combiner in the postcoding comb filter 226. Since the
delay exhibited by the delay devices 2201 and 2263 is
shorter than the delay exhibited by the delay devices 1201
and 1263, although nulls near frequencies converted from
analog TV carrier frequencies are narrower band, there is
more likely to be good anti-correlation in the signals
additively combined by the adder 2202 than there is likely
to be good correlation in the signals differentially
combined by the subtractor 1202. The suppression of the
sound carrier is poorer in the NTSC-rejection comb filter
220 response than in the NTSC-rejection comb filter 120
response. However, if the sound carrier of a co-channel
interfering analog TV signal has been suppressed by SAW
filtering or a sound trap in the IF amplifier chain 12,
the poor sound rejection of the comb filter 220 is not a
problem. The responses to sync tips is reduced in
duration using the NTSC-rejection comb filter 220 of
Figure 10 rather than the NTSC-rejection comb filter 120
of Figure 9, so there is substantially reduced tendency to
overwhelm error-correction in the trellis decoding and
Reed-Solomon coding.
A species 2261 of the multiplexer 261 is controlled
by a multiplexer control signal that is in its second
state most of the time when it is determined there is
insufficient NTSC co-channel interference to cause
uncorrectable error in the output signal from the
data-slicer 22 and that is in its third state most of the
time when it is determined there is sufficient NTSC
co-channel interference to cause uncorrectable error in
the output signal from the data-slicer 22. The
multiplexer 2261 is conditioned by its control signal
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being in its third state to feed back the modulo-8 sum
results of the adder 2262, as delayed six symbol epochs by
the delay device 2263, to the adder 2262 as a summand.
This is a modular accumulation procedure in which a single
error propagates as a running error, with error recurring
every six symbol epochs. Running errors in the postcoded
symbol decoding results from the postcoding comb filter
226 are curtailed by the multiplexer 2261 being placed
into its first state for four symbol epochs at the
beginning of each data segment, as well as during the
entirety of each data segment containing field sync. When
this control signal is in its first state, the multiplexer
2261 reproduces .as its output signal ideal symbol decoding
results supplied from memory in the controller 28. The
introduction of ideal symbol decoding results into the
multiplexer 2261 output signal halts a running error.
Since there are 4 + 138(6) symbols per data segment, the
ideal symbol decoding results slip back four symbol epochs
in phase each data segment, so no running error can
persist for longer than two data segments. The likelihood
of a protracted period of running error in the postcoding
comb filter 226 is substantially less than in the
postcoding comb filter 126, although the running error
recurs more frequently and affects twice as many of the
twelve interleaved trellis codes.
FIGURE 11 is a block schematic diagram showing
details of a portion of the FIGURE 1, FIGURE 3 or FIGURE 4
digital television signal receiver using a species 320 of
the NTSC-rejection comb filter 20 and a species 326 of the
postcoding comb filter 26. The NTSC-rejection comb filter
320 uses a first delay device 3201 exhibiting a delay of
1368 symbol epochs, which delay is substantially equal to
the epoch of two horizontal scan lines of an analog TV
signal, and the postcoding comb filter 326 uses a second
delay device 3263 also exhibiting such delay. The first
linear combiner in the NTSC-rejection comb filter 320 is a
subtractor 3202, and the second linear combiner in the
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postcoding comb filter 326 is a modulo-8 adder 3262.
A species 3261 of the multiplexes 261 is controlled
by a multiplexes control signal that is in its second
state most of the time when it is determined there is
insufficient NTSC co-channel interference to cause
uncorrectable error in the output signal from the
data-slices 22 and that is in its third state most of the
time when it is determined there is sufficient NTSC
co-channel interference to cause uncorrectable error in
the output signal from the data-slices 22. The DTV
receiver preferably contains circuitry for detecting
change between alternate scan lines in the NTSC co-channel
interference, so that the controller 28 can withhold
supplying the third state of the multiplexes 3261 control
signal under such conditions.
The multiplexes 3261 is conditioned by its control
signal being in its third state to feed back the modulo-8
sum results of the adder 3262, as delayed 1368 symbol
epochs by the delay device 3263, to the adder 3262 as a
summand. This is a modular accumulation procedure in
which a single error propagates as a running error, with
error recurring every 1368 symbol epochs. This symbol
code span is longer than the span for a single block of
the Reed-Solomon code, so a single running error is
readily corrected during Reed-Solomon decoding. Running
errors in the postcoded symbol decoding results from the
postcoding comb filter 326 are curtailed by the
multiplexes 3261 being placed into its first state during
the entirety of each data segment containing field sync,
as well as for four symbol epochs at the beginning of each
data segment. When this control signal is in its first
state, the multiplexes 3261 reproduces as its output
signal ideal symbol decoding results supplied from memory
in the controller 28. The introduction of ideal symbol
decoding results into the multiplexes 3261 output signal
halts a running error. The 16.67 millisecond duration of
an NTSC video field exhibits phase slippage against the
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24.19 millisecond duration of a DTV data field, so the DTV
data segments containing field sync eventually scan the
entire NTSC frame raster. The 525 lines in the NTSC frame
raster each contain 684 symbol epochs, for a total of
359,100 symbol epochs. Since this is somewhat less than
432 times the 832 symbol epochs in a DTV data segment
containing field sync, one can guess with reasonable
confidence that running errors of duration longer than 432
data fields will be expunged by the multiplexer 3261
reproducing ideal symbol decoding results during DTV data
segments containing field sync. There is also phase
slippage between data segments, for the start code groups
of which ideal symbol decoding results are available, and
NTSC video scan lines. One can estimate 359,100 symbol
epochs, which is 89,775 times the four symbol epochs in a
code start group, are scanned during 89,775 consecutive
data segments. Since there are 313 data segments per DTV
data field, one can guess with reasonable confidence that
running errors of duration longer than 287 data fields
will be expunged by the multiplexer 3261 reproducing ideal
symbol decoding results during the code start groups. The
two sources of suppression of running errors are
reasonably independent of each other, so running errors of
duration longer than two hundred or so data fields are
quite unlikely. Furthermore, if NTSC co-channel
interference dips low at a time when the running error
recurs, to condition the multiplexer 3261 for reproducing
the response of the data-slicer 22 as its output signal,
the error may be corrected earlier than would otherwise be
the case.
The FIGURE 11 NTSC-rejection comb filter 320 is
quite good in suppressing demodulation artifacts generated
in response to analog TV horizontal synchronizing pulses,
as well as suppressing many of the demodulation artifacts
generated in response to analog TV vertical synchronizing
pulses and equalizing pulses. These artifacts are the
co-channel interference with highest energy. Except where
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there is scan-line-to-scan-line change in the video
content of the analog TV signal over the period of two
scan lines, the NTSC-rejection comb filter 320 provides
reasonably good suppression of that video content
regardless of its colour. The suppression of the FM audio
carrier of the analog TV signal is reasonably good, in
case it has not been suppressed by a tracking rejection
filter in the symbol synchronization and equalization
circuitry 16. Artifacts of most analog TV colour bursts
are suppressed in the NTSC-rejection comb filter 320
response, too. Furthermore, the filtering provided by the
NTSC-rejection comb filter 320 is "orthogonal" to the
NTSC-interference rejection built into the trellis
decoding procedures.
FIGURE 12 is a block schematic diagram showing
details of a portion of the FIGURE 1, FIGURE 3 or FIGURE 4
digital television signal receiver using a species 420 of
the NTSC-rejection comb filter 20 and a species 426 of the
postcoding comb filter 26. The NTSC-rejection comb filter
420 uses a first delay device 4201 exhibiting a delay of
179,208 symbol epochs, which delay is substantially equal
to the period of 262 horizontal scanning lines of an
analog TV signal, and the postcoding comb filter 426 uses
a second delay device 4261 also exhibiting such delay. A
subtractor 4202 serves as the first linear combiner in the
NTSC-rejection comb filter 420, and a modulo-8 adder 4262
serves as the second linear combiner in the postcoding
comb filter 426.
A species 4261 of the multiplexer 261 is controlled
by a multiplexer control signal that is in its second
state most of the time when it is determined there is
insufficient NTSC co-channel interference to cause
uncorrectable error in the output signal from the
data-slicer 22 and that is in its third state most of the
time when it is determined there is sufficient NTSC
co-channel interference to cause uncorrectable error in
the output signal from the data-slicer 22. The DTV
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receiver preferably contains circuitry for detecting
field-to-field change in the NTSC co-channel interference,
so that the controller 28 can withhold supplying the third
state of the multiplexes 4261 control signal under such
conditions.
The multiplexes 4261 is conditioned by its control
signal being in its third state to feed back the modulo-8
sum results of the adder 4262, as delayed 179,208 symbol
epochs by the delay device 4263, to the adder 4262 as a
summand. This is a modular accumulation procedure in
which a single error propagates as a running error, with
error recurring every 179,208 symbol epochs. This symbol
code span is longer than the span for a single block of
the Reed-Solomon code, so a single running error is
readily corrected during Reed-Solomon decoding. Running
errors in the postcoded symbol decoding results from the
postcoding comb filter 426 are curtailed by the
multiplexes 4261 being placed into its first state during
the entirety of each data segment containing field sync,
as well as for four symbol epochs at the beginning of each
data segment. When this control signal is in its first
state, the multiplexes 4261 reproduces as its output
signal ideal symbol decoding results supplied from memory
in the controller 28. The introduction of ideal symbol
decoding results into the multiplexes 4261 output signal
halts a running error. The maximum number of data fields
required to expunge running error in the multiplexes 4261
output signal is presumably substantially the same as
required to expunge running error in the multiplexes 3261
output signal. However, the number of times the error
recurs in that period is lower by a factor of 131.
The FIGURE 12 NTSC-rejection comb filter 420
suppresses most demodulation artifacts generated in
response to analog TV vertical synchronizing pulses and
equalizing pulses, as well as suppressing all the
demodulation artifacts generated in response to analog TV
horizontal synchronizing pulses. These artifacts are the
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co-channel interference with highest energy. Also, the
NTSC-rejection comb filter 420 suppresses artifacts
arising from the video content of the analog TV signal
that does not change from field to field or line-to-line,
getting rid of stationary patterns irrespective of their
horizontal spatial frequency or colour. Artifacts of most
analog TV colour bursts are suppressed in the
NTSC-rejection comb filter 420 response, too.
FIGURE 13 is a block schematic diagram showing
details of a portion of the FIGURE 1, FIGURE 3 or FIGURE 4
digital television signal receiver using a species 520 of
the NTSC-rejection comb filter 20 and a species 526 of the
postcoding comb filter 26. The NTSC-rejection comb filter
520 uses a first delay device 5201 exhibiting a delay of
718,200 symbol epochs, which delay is substantially equal
to the period of two frames of an analog TV signal, and
the postcoding comb filter 526 uses a second delay device
5261 also exhibiting such delay. A subtractor 5202 serves
as the first linear combiner in the NTSC-rejection comb
filter 520, and a modulo-8 adder 5262 serves as the second
linear combiner in the postcoding comb filter 526.
A species 5261 of the multiplexer 261 is controlled
by a multiplexer control signal that is in its second
state most of the time when it is determined there is
insufficient NTSC co-channel interference to cause
uncorrectable error in the output signal from the
data-slicer 22 and that is in its third state most of the
time when it is determined there is sufficient NTSC
co-channel interference to cause uncorrectable error in
the output signal from the data-slicer 22. The DTV
receiver preferably contains circuitry for detecting
change between alternate frames in the NTSC co-channel
interference, so that the controller 28 can withhold
supplying the third state of the multiplexer 5261 control
signal under such conditions.
The multiplexer 5261 is conditioned by its control
signal being in its third state to feed back the modulo-8
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sum results of the adder 5262, as delayed 718,200 symbol
epochs by the delay device 5263, to the adder 5262 as a
summand. This is a modular accumulation procedure in
which a single error propagates as a running error, with
error recurring every 718,200 symbol epochs. This symbol
code span is longer than the span for a single block of
the Reed-Solomon code, so a single running error is
readily corrected during Reed-Solomon decoding. Running
errors in the postcoded symbol decoding results from the
postcoding comb filter 526 are curtailed by the
multiplexer 5261 being placed into its first state during
the entirety of each data segment containing field sync,
as well as for four symbol epochs at the beginning of each
data segment. When this control signal is in its first
state, the multiplexer 5261 reproduces as its output
signal ideal symbol decoding results supplied from memory
in the controller 28. The introduction of ideal symbol
decoding results into the multiplexer 5261 output signal
halts a running error. The maximum number of data fields
required to expunge running error in the multiplexer 5261
output signal is presumably substantially the same as
required to expunge running error in the multiplexer 3261
output signal. However, the number of times the error
recurs in that period is lower by a factor of 525.
The FIGURE 13 NTSC-rejection comb filter 520
suppresses all demodulation artifacts generated in
response to analog TV vertical synchronizing pulses and
equalizing pulses, as well as suppressing all the
demodulation artifacts generated in response to analog TV
horizontal synchronizing pulses. These artifacts are the
co-channel interference with highest energy. Also, the
NTSC-rejection comb filter 520 suppresses artifacts
arising from the video content of the analog TV signal
that does not change over two frames, getting rid of such
very stationary patterns irrespective of their spatial
frequency or colour. Artifacts of all analog TV colour
bursts are suppressed in the NTSC-rejection comb filter
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520 response, too.
One skilled in the art of television system design
will discern other properties of correlation and
anti-correlation in analog TV signals that can be
exploited in the design of NTSC-rejection filters of still
other types than those shown in FIGURES 9-13. The use of
NTSC-rejection filters that cascade two NTSC-rejection
filters of the types already disclosed increases the 2N
levels of the baseband signals to (8N-1) data levels.
Such filters may be required to overcome particularly bad
co-channel interference problems despite their shortcoming
of reducing signal-to-noise for random noise interference
with symbol decoding.
FIGURE 14 shows a modification of a digital
television signal receiver as thusfar described,
constructed in accordance with a further aspect of the
invention so as to operate in parallel a plurality of
symbol decoders using respective even-level data-slivers,
each preceded by a different type of NTSC-rejection comb
filter and each succeeded by a respective postcoding comb
filter to compensate for the precoding introduced by the
preceding NTSC-rejection comb filter. An even-level
data-sliver A24 converts the response of an NTSC-rejection
filter A20 of a first type to first precoded symbol
decoding results for application to a postcoding comb
filter A26 of a first type. An even-level data-sliver B24
converts the response of an NTSC-rejection filter B20 of a
second type to second precoded symbol decoding results for
application to a postcoding comb filter B26 of a second
type. An even-level data-sliver C24 converts the response
of an NTSC-rejection filter C20 of a third type to third
precoded symbol decoding results for application to a
postcoding comb filter C26 of a third type. The odd-level
data-sliver 22 supplies interim symbol decoding results to
the postcoding comb filters A26, B26 and C26. The
prefixes A, B and C in the identification numbers for the
elements of Figure 14 are different integers which will
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correspond to respective ones of the integers 1, 2, 3, 4
and 5 when receiver portions as shown in ones of Figures
9-13 are employed.
Symbol decoding selection circuitry 90 in Figure 14
formulates a best estimate of correct symbol decoding for
application to the trellis decoding circuitry 34,
selecting from the interim symbol decoding results
received from the data-slicer 22 and the various postcoded
symbol coding results received from postcoding comb
filters A26, B26 and C26. The best estimate of symbol
decoding results are used to correct the summation
procedures in the postcoding comb filters A26, B26 and
C26.
Figure 15, comprising Figures 15A and 15B,
illustrates in greater detail a currently preferred way of
implementing the symbol decoding selection circuitry 90.
Figure 15A shows details of circuitry for generating
prescribed symbol decoding results for application, during
data synchronization intervals, to the 3-bit-wide output
data bus 800 of the symbol decoding selection circuitry
90. The 15A circuitry operates similarly to circuitry
described above with reference to Figure 8.
Figure 15B illustrates in greater detail circuitry
within the symbol decoding selection circuitry 90 for
selecting among the interim symbol decoding results and
the various postcoded symbol decoding results, for
generating final symbol decoding results during time
periods between data synchronization intervals. The
efficacies of the NTSC-rejection filters A20, B20 and C20
in removing NTSC co-channel interference from DTV signal
are determined by observing how well related
NTSC-rejection filters A100, B100 and C100 reduce the
energy of NTSC co-channel interference translated to
baseband and separated from DTV signal artifacts.
Separation of NTSC co-channel interference from DTV signal
proceeds as previously described with reference to Figure
3. The lowpass filter 54 response to baseband video that
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has been synchronously detected from the NTSC co-channel
interference is supplied as input signal to NTSC-rejection
filters A100, B100 and C100. The NTSC-rejection filter
A100 differs from the NTSC-rejection filter A20 of first
type insofar as the type of linear combiner that is used,
the linear combiner in one of the filters A20 and A100
being an adder and the linear combiner in the other of the
filters A20 and A100 being a subtractor. This is because
the filter A100 is supplied baseband video, but the
artifact of NTSC video carrier in the DTV signal supplied
to the filter A20 is not at baseband for video carrier.
For similar reasons, the NTSC-rejection filter B100
differs from the NTSC-rejection filter B20 of second type
insofar as the type of linear combiner that is used, and
the NTSC-rejection filter C100 differs from the
NTSC-rejection filter C20 of third type insofar as the
type of linear combiner that is used. The responses of
the NTSC-rejection filters A100, B100 and C100 are squared
by squarers A102, B102 and C102, respectively, for
determining the energies of these responses. The response
of the lowpass filter 54 is squared by a squarer 104 for
determining its energy.
Figure 15B modifies the Figure 8 circuitry to
replace the multiplexer 2611 and the tri-state data buffer
82 with four tri-state data buffers 082, A82, B82 and
C82. The tri-state data buffer 082 is used for
selectively asserting the interim symbol decoding results
from the data-slicer 22 onto the 3-bit-wide output data
bus 800 of the symbol decoding selection circuitry 90.
The three tri-state data buffers A82, B82 and C82 are used
for selectively asserting the postcoded symbol decoding
results from the postcoding comb filters A26, B26 and C26,
respectively, onto the data bus 800.
It is to be determined whether any of the responses
of the NTSC-rejection filters A100, B100 and C100 has
substantially less energy than the response of the lowpass
filter 54 to determine that one of the three tri-state
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data buffers A82, B82 and C82, rather than the tri-state
data buffer 082, is to be conditioned for providing low
source impedance when the NOR gate 92 response is ONE. If
such determination is made it is to be further determined
which of the responses of the NTSC-rejection filters A100,
B100 and C100 has the least remaining energy therein, to
govern which of the three tri-state data buffers 082, A82,
B82 and C82 is to be conditioned for providing low source
impedance when the NOR gate 92 response is ONE. Towards
these goals, the responses of squarers 104 and A102 are
compared by a comparator 106; the responses of squarers
104 and H102 are compared by a comparator 108; the
responses of squarers 104 and C102 are compared by a
comparator 110; the responses of squarers A102 and B102
are compared by a comparator 112; the responses of
squarers A102 and C102 are compared by a comparator 114;
and the responses of squarers B102 and C102 are compared
by a comparator 112.
A 3-input NOR gate 118 responds to none of the
comparators 106, 108 and 110 indicating that the response
of squarer 104 exceeds any of the responses of the
squarers A102, B102 and C102 to furnish a ONE as output
signal; otherwise the NOR gate 118 output signal is a
ZERO. A 2-input AND gate 120 supplies a ONE response that
conditions the three tri-state data buffer 082 for
providing low source impedance when and only when NOR gate
92 response is ONE at the same time that the NOR gate 118
response is a ONE.
A 3-input AND gate 122 furnishes a ONE output
signal responsive to the output of the comparator 106
being a ONE, indicative that the squarer A102 response has
less energy than the squarer 104 response, at the same
time both of the complemented outputs of the comparators
112 and 114 are ONES, indicative that the response of
squarer 104 has no more energy than the responses of the
squarers B102 and C102; otherwise the AND gate 122 output
signal is a ZERO. A 2-input AND gate 124 supplies a ONE
- 47 -
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response that conditions the three tri-state data buffer
A82 for providing low source impedance when and only when
NOR gate 92 response is ONE at the same time that the AND
gate 122 response is a ONE.
A 3-input AND gate 126 furnishes a ONE output
signal responsive to the complemented output of the
comparator 116 being a ONE, indicative that the squarer
B102 response has no more energy than the squarer C102
response, at the same both of the outputs of the
comparators 108 and 112 are ONES, indicative that the
response of squarer B102 has less energy than the
responses of the squarers 104 and A102; otherwise the AND
gate 126 output signal is a ZERO. A 2-input AND gate 128
supplies a ONE response that conditions the three
tri-state data buffer B82 for providing low source
impedance when and only when NOR gate 92 response is ONE
at the same time that the AND gate 126 response is ONE.
A 3-input AND gate 130 furnishes a ONE output
signal when the outputs of the comparators 110, 114 and
116 are all ONES, indicative that the response of squarer
C102 has less energy than the responses of the squarers
104, A102 and B102; otherwise the AND gate 130 output
signal is a ZERO. A 2-input AND gate 132 supplies a ONE
response that conditions the three tri-state data buffer
C82 for providing low source impedance when and only when
NOR gate 92 response is ONE at the same time that the AND
gate 130 response is a ONE.
Referring back to Figure 14, the NTSC-rejection
comb filter A20 and the postcoding comb filter A26
circuitry are advantageously chosen to be of types like
the NTSC-rejection comb filter 520 and the postcoding comb
filter 526 circuitry of Figure 13. This is so despite a
considerable cost in memory, since 718,200 symbols have to
be stored in each of the 2-video-frame delays 5201 and
5263. (However, the storage in the 2-video-frame delays
5201 provides the storage required for the Figure 15
co-channel interference detector A44. Furthermore, the
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same memory can be used for realizing the shorter delays
4201, 3201, 2201, 1201 and the shorter delays in the other
co-channel interference detectors of Figure 15. Also, the
storage in the 2-video-frame delay 5263 provides the
storage required for shorter delays 4263, 3263, 2263,
1263.)
The high-energy demodulation artifacts generated in
response to analog TV synchronizing pulses, equalizing
pulses, and colour bursts are all suppressed when the
NTSC-rejection comb filter A20 additively combines
alternate video frames. Also, artifacts arising from the
video content of the analog TV signal that does not change
over two frames are suppressed, getting rid of stationary
patterns irrespective of their spatial frequency or
colour. When NTSC-rejection comb filter A20 additively
combines alternate video frames, the NTSC-rejection comb
filter A100 differentially combines those alternate video
frames and together with the squarer A102 provides a
detector for change between alternate frames in the NTSC
co-channel interference.
The remaining problem of suppressing demodulation
artifacts primarily concerns suppressing those
demodulation artifacts arising from frame-to-frame
difference at certain pixel locations within the analog TV
signal raster. These demodulation artifacts can be
suppressed by intra-frame filtering techniques. The
NTSC-rejection comb filter B20 and the postcoding comb
filter B26 circuitry can be chosen to suppress remnant
demodulation artifacts by relying on correlation in the
horizontal direction, and the NTSC-rejection comb filter
C20 and the postcoding comb filter C26 circuitry can be
chosen to suppress remnant demodulation artifacts by
relying on correlation in the vertical direction.
Consider how such a design decision can be further
implemented.
If the sound carrier of a co-channel interfering
analog TV signal is not suppressed by SAW filtering or a
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sound trap in the IF amplifier chain 12, the
NTSC-rejection comb filter B20 and the postcoding comb
filter B26 circuitry are advantageously chosen to be of
types like the NTSC-rejection comb filter 120 and the
postcoding comb filter 126 circuitry of Figure 9. If the
sound carrier of a co-channel interfering analog TV signal
is suppressed by SAW filtering or a sound trap in the IF
amplifier chain 12, the NTSC-rejection comb filter B20 and
the postcoding comb filter B26 circuitry are
advantageously chosen to be of types like the
NTSC-rejection comb filter 220 and the postcoding comb
filter 226 circuitry of Figure 10. This is because the
anti-correlation between video components only six symbol
epochs away from each other is usually better than the
correlation between video components twelve symbol epochs
away from each other.
The optimal choice of the NTSC-rejection comb
filter C20 and the postcoding comb filter C26 circuitry is
less straightforward, because of the choice one must make
(in consideration of field interlace in the interfering
analog TV signal) whether to choose the temporally closer
scan line in the same field or the spatially closer line
in the preceding field to be combined with the current
scan line in the NTSC-rejection comb filter C20. Choosing
the temporally closer scan line in the same field is
generally the better choice, since jump cuts between
fields are less likely to ravage NTSC rejection by the
comb filter C20. With such choice, the NTSC-rejection
comb filter C20 and the postcoding comb filter C26
circuitry are of types like the NTSC-rejection comb filter
320 and the postcoding comb filter 326 circuitry of Figure
11. When NTSC-rejection comb filter C20 additively
combines alternate scan lines of video, the NTSC-rejection
comb filter C100 differentially combines those alternate
scan lines of video and together with the squarer C102
provides a detector for change between alternate scan
lines in the NTSC co-channel interference.
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With the other choice instead, the NTSC-rejection
comb filter C20 and the postcoding comb filter C26
circuitry are of types like the NTSC-rejection comb filter
420 and the postcoding comb filter 426 circuitry of Figure
12. The NTSC-rejection comb filter C100 and the squarer
C102 together then provide a detector for change between
fields in the NTSC co-channel interference.
The Figure 14 digital receiver apparatus is
modified in yet other embodiments of the invention to use
additional parallel data-slicing operations, each carried
out by a cascade connection of respective NTSC-rejection
filter followed by a respective even-level data-slicer
followed by a respective postcoding comb filter. While
two additional parallel data-slicing operations are shown
i.n Figure 14, modifications to use still further parallel
data-slicing operations can provide capability for
refining the best estimate of correct symbol decoding
result still further.
The trellis decoder circuitry 34 may be replicated
and the relative success of various symbol decoding
decisions can be compared to refine the best estimate of
symbol decoding result further. This involves
considerably more digital hardware, however.
In certain embodiments of the invention alternative
to those previously described, the symbol decoding
selection circuitry 90 includes voting circuitry for
polling the symbol codes supplied from the odd-level
data-slicer 22, the postcoding comb filter A26 of first
type, the postcoding comb filter B26 of second type, and
the postcoding comb filter C26 of third type. If all four
of the symbol decoding results concur, the symbol decoding
result concurred to is supplied to the data assembler 30.
If the symbol decoding results supplied from the odd-level
data-slicer 22, the postcoding comb filter A26 of first
type, the postcoding comb filter B26 of second type, and
the postcoding comb filter C26 of third type do not concur
a simple voting procedure is carried out by the voting
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circuitry to select the decoding result least likely to be
in error.
More accurate symbol decoding will be obtained more
of the time if a weighted voting procedure is followed in
the voting circuitry. The weights for voting can be
modified to take into account the variances of the
decoding results, reducing the weight accorded a decoding
result in the voting procedure if it departs from a
decoding result concurred in by a majority of the other
symbol decoding circuits. Using circuitry similar to some
of the circuitry shown in Figure 15B and some additional
circuitry, weights for voting can also be determined in
inverse relationships to the energy measurements generated
by the squarer 104, by the NTSC-rejection comb filter A100
and the squarer A102, by the NTSC-rejection comb filter
B100 and the squarer H102, and by the NTSC-rejection comb
filter C100 and the squarer C102.
Co-channel interference by analog television
signals of other standards than NTSC, such as the PAL
standard, may arise in digital television systems adapted
from the digital television system used for terrestrial
broadcasting in the United States of America. The
invention is readily modified as a mere matter of design
to accommodate such co-channel interference.
One skilled in the art of digital communications
receiver design and acquainted with the foregoing
specification and its drawing will be enabled to design
many embodiments of the invention other than the preferred
ones specifically described. This should be borne in mind
when construing the scope of the broader claims which
follow. In the claims which follow, the word "said" is
used whenever reference is made to an antecedent, and the
word "the" is used for grammatical purposes other than to
refer back to an antecedent.
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