Note: Descriptions are shown in the official language in which they were submitted.
CA 02241408 2000-09-25
USING SPECIAL NTSC RECEIVER TO DETECT WHEN CO-CHANNEL
INTERFERING NTSC SIGNAL ACCOMPANIES A DIGITAL TV SIGNAL
The invention relates to digital television as
transmitted by radio waves in a broadcast television band
and, more particularly, to a method for detecting in a
digital television receiver when a digital television
signal is accompanied by co-channel interfering NTSC
signal of substantial amplitude.
BACKGROUND OF THE INVENTION
A Digital Television Standard published 16
September 1995 by the Advanced Television Subcommittee
(ATSC) specifies the nature of vestigial sideband (VSB)
signals for transmitting digital television (DTV) signals
in 6-MHz-bandwidth television channels such as those
currently used in over-the-air broadcasting of National
Television Subcommittee (NTSC) analog television signals
within the United States. As long as NTSC analog
television signals continue to be broadcast, it will be
advantageous in a receiver for DTV signals to be able to
determine when an NTSC analog television signal causes
substantial co-channel interference with a DTV signal
being received. The DTV receiver can then be designed to
change its mode of operation responsive to a determination
that such co-channel interference is occurring, so that
the undesirable effects of such co-channel interference
can be mitigated, which is usually done by comb
filtering. The comb filtering employed in a DTV receiver
for suppressing NTSC co-channel interference is better
discontinued when such interference is not substantial,
since this can avoid the additional Johnson noise arising
from the plural paths through the comb filter. Generally,
co-channel interference from an NTSC analog television
signal is considered to be substantial if it has
sufficient energy as to cause frequent errors in the
data-slicing operations used during symbol decoding a DTV
signal as synchrodyned to baseband. (U.S. patent No.
5,801,790 issued Sept. 1, 1998 and entitled "USING VIDEO
SIGNALS FROM AUXILIARY ANALOG TV RECEIVERS FOR DETECTING
NTSC INTERFERENCE IN DIGITAL TV RECEIVERS" particularly
describes a
1
CA 02241408 1998-06-23
DTV receiver designed to change its mode of operation responsive to a
determination
that substantial co-channel interference is occurring, so that the undesirable
effects of
such co-channel interference can be mitigated. That application teaches that
the
detection of NTSC co-channel interference in digital TV receivers is more
readily
done after synchrodyning any such NTSC interference to baseband rather than
after
synchrodyning DTV signal to baseband.
U. S. patent No. x,122,879 issued 16 June 199? to Katsu Ito and entitled
"TELEVISION SYNCHRONOUS RECEIVER WITH PHASE SHIFTER FOR
REDUCING INTERFERENCE FROM A LOWER ADJACENT CHAUNEL"
describes an analog television receiver that synchronously detects recejved
NTSC
signal both in-phase and quadrature-phase. The Ito receiver synchrodynes the
radio-
frequency (RF) amplifier response directly to baseband, so an adjacent lower
channel
may appear as an image. The quadrature-phase synchronous detection response is
phase shifted 90° at all video frequencies above X00-7~0 kHz and
linearly combined
1 ~ with the in-phase synchronous detection response to suppress image
frequency
components translated to baseband during synchronous detection of the received
NTSC signal. In U. S. patent No. x,122,879 Ito does not disclose the fact that
this
procedure also cancels the video components above 7~0 kHz. The attendant loss
of
luminance high frequencies is acceptable in small-viewing-screen television
receivers,
however, such as those used in wrist watches.
Current DTV receiver designs use plural frequency conversion, with a first
conversion to an intermediate frequency in the ultrahigh frequency (UHF) band
above
the channels designated for television broadcasting, and with a second
conversion to
an intermediate frequency in the very high frequency (VHF) band below the
channels
2~ designated for television broadcasting. So image suppression is not a
problem.
Furthermore, the carrier of a VSB DTV signal is located only 310 kHz from
channel
edge so there is very little double sideband content as compared to an NTSC
signal.
The inventor points out that an NTSC receiver of the type linearly combining
in-phase synchronous video detection response with inverse-Hilbert-transformed
quadrature-phase synchronous video detection response is nevertheless of
interest in
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DTV reception, for use as an auxiliary receiver for detecting when a digital
television
signal is accompanied by co-channel interfering NTSC signal of substantial
amplitude. By arranging for the inverse Hilbert transformation of the
quadrature-
phase synchronous video detection response to frequencies well below 7~0 kHz,
such
an auxiliary receiver becomes substantially insensitive to artifacts of co-
channel DTV
signal. The suppression of the DTV artifacts simplifies measuring the
magnitude of
co-channel interfering NTSC signal.
SUMVIARY OF THE INVENTION
The invention in one of its aspects is embodied in a method for detecting in a
digital television receiver when a digital television signal is accompanied by
co-
channel interfering NTSC signal of substantial amplitude, which method
comprises
steps as follow. The video portion of any co-channel interfering NTSC signal
is
synchrodyned to baseband, for generating an in-phase demodulation result
including
first artifacts of the digital television signal, and for generating a
quadrature-phase
1 ~ demodulation result including second artifacts of the digital television
signal. The in-
phase and quadrature-phase demodulation results are subsequently
differentially
shifted in respective phase by 90° at frequencies above a few kilohertz
and then
linearly combined to generate a linear combining result substantially free of
the first
and second artifacts of the digital television signal. Then, whether or not
the
amplitude of said linear combining result exceeds a prescribed value is
detected, for
generating an indication of when the digital television signal is accompanied
by co-
channel interfering NTSC signal of substantial amplitude.
The invention in another of its aspects is embodied in a digital television
receiver including circuitry for detecting times when analog television signal
of
substantial amplitude occupies a television broadcast channel, which circuitry
is more
particularly described as follows. The receiver has input circuitry for
selecting from a
television broadcast channel a vestigial sideband amplitude-modulation signal
descriptive of a video signal portion of any analog television signal that
occupies the
television broadcast channel, converting the selected vestigial sideband
amplitude-
modulation signal to an intermediate frequency signal, and amplifying the
CA 02241408 1998-06-23
intermediate frequency signal to provide an amplified intermediate frequency
signal.
The vestigial sideband amplitude-modulation signal as originally received by
the
input circuitry includes a video carrier and full sideband in addition to a
vestigial
sideband. Video synchrodyning circuitry synchronously detects the amplified
intermediate frequency signal with respect to the video carrier signal and
with respect
to a carrier in quadrature phase with the video carrier signal, for generating
an
in-phase synchronous detection response and for generating a quadrature-phase
synchronous detection response. Phase shift circuitry referred to as an
inverse Hilbert
transform circuitry in this specification phase shifts by substantially
90° all frequency
components of the quadrature-phase synchronous detection response above a
prescribed frequency to generate phase shift circuitry response. Lineai
combining
circuitry linearly combines the in-phase synchronous detection response and
the phase
shift circuitry response for recovering linear combining circuitry response to
a portion
of the video signal described both in the full sideband and the vestigial
sideband of the
1 ~ vestigial sideband amplitude-modulation signal as originally received.
This linear
combining circuitry response is substantially free of response to any digital
television
signal that occupies the television broadcast channel being currently
received. A
threshold detector is included in the receiver for determining when the first
linear
combining circuitry response exceeds a prescribed threshold value, for
generating
indications that the co-channel analog television signal is of substantial
amplitude.
BRIEF DESCRIPTION OF THE DRAWING
FIGURES 1 and ? are each a schematic diagram of a television receiver that is
capable of receiving NTSC analog TV signals as well as DTV signals, which
receiver
employs the method of the invention for detecting the presence in DTV signals
of co-
channel interfering NTSC analog TV signals.
Figure 3 is a schematic diagram of a modification that can be made to either
of
the television receivers of FIGURES 1 and 2.
FIGURES 4, 5, 6 and 7 are flow charts showing steps of the methods for
detecting in a digital television receiver when a digital television signal is
CA 02241408 1998-06-23
accompanied by co-channel interfering NTSC signal of substantial amplitude,
which
methods embody the invention in its various aspects.
DETAILED DESCRIPTION
FIGURE 1 shows portions of a television receiver that is capable of receiving
NTSC analog TV signals as well as DTV signals. Over-the-air type television
broadcasting signals as received by an antenna 1 are amplified by an
adjustably tuned
radio-frequency amplifier 2 and supplied to a first detector 3. The RF
amplifier 2 and
the first detector 3 have adjustable tuning and together function as a tuner
for selecting
a digital television signal from one of channels at different locations in a
frequency
band. The first detector 3 includes a first local oscillator supplying first
local
oscillations tunable over a frequency range above the ultra-high-frequency
(UHF) TV
broadcast band and a first mixer for mixing the first local oscillations with
a TV signal
selected by the adjustably tuned RF amplifier 2 for upconverting the selected
TV
signal to generate a UHF intermediate-frequency signal in a 6-VIHz-wide UHF
1 ~ intermediate-frequency band located at frequencies above the assigned
channels in the
UHF TV broadcast band.
The first detector 3 supplies the high-IF-band signal to a UHF-band
intermediate-frequency amplifier 6 used in NTSC audio reception. The response
of
the UHF IF amplifier 6 is applied to a second detector 9 used in NTSC audio
reception. The second detector 9 includes a second local oscillator supplying
second
local oscillations of prescribed frequency above the ultrahigh frequency UHF
TV
broadcast band and a second mixer for mixing the second local oscillations
with the
response of the UHF IF amplifier 6 to generate a very-high-frequency (VHF)
intermediate frequency signal located at frequencies below the assigned
channels in
the VHF TV broadcast band. This VHF IF signal is supplied to a very-high-
frequency
intermediate-frequency amplifier 12.
The response of the VHF IF amplifier 12 is applied to an intercarrier sound
detector 34, which supplies 4.5 loll-iz intercarrier sound intermediate-
frequency signals
to an intercarrier sound intermediate-frequency amplifier 3~ which amplifies
and in
most designs symmetrically limits the amplified response for application to an
FVI
5
_3
' CA 02241408 1998-06-23
detector 36. The FM detector 36 reproduces baseband composite audio signal
supplied to the remaining portions of the analog TV receiver part of the DTV
receiver.
In regard to baseband composite audio signal, these remaining portions
typically
include stereophonic decoder circuitry. If the NTSC audio signals are selected
with
narrowband filtering in the IF amplifiers 6 and 12 that pass only the FM audio
carrier
as translated to intermediate frequencies, the intercarrier sound detector 3=l
can be
provided by a multiplier that multiplies the IF amplifier 12 response by video
carrier
selected to the multiplier by a narrowband filter responsive to the response
of the IF
amplifier 10 or 11. If the NTSC audio signals are selected with filtering in
the IF
amplifiers 6 and 12 that passes both the NTSC video and audio carriers as
translated
to intermediate frequencies, for implementing "quasi-parallel" sound, the
intercarrier
sound detector 34 can be a simple rectifier or can be a square-law device.
The first detector 3 also supplies the high-IF-band signal to a UHF-band
intermediate-frequency amplifier 37 used both in NTSC video reception and in
ATSC
1 ~ reception. A surface-acoustic-wave (SA'V) filter in the UHF IF amplifier
37, which
determines overall IF response for ATSC DTV signal and for NTSC video signal,
preferably rejects NTSC audio signal. Otherwise, the SAW filter has
substantially flat
amplitude response over the remainder of the 6-MHz-wide TV broadcast channel
as
translated to the UHF IF band and has substantially linear phase response
throughout
its passband. The SAW filter is preceded in the UHF IF amplifier 37 by a
transistor
amplifier designed to drive the SAW filter from a prescribed source impedance
that
minimizes multiple-reflection. Better to maintain this prescribed source
impedance,
the transistor amplifier gam 15 preferably fixed in value and suffices to
overcome the
insertion loss in the SAW filter. The response of the UHF IF amplifier 37 is
applied
2~ to a second detector 38 used in ATSC DTV reception and in NTSC video
reception.
The second detector 38 includes a second local oscillator supplying second
local
oscillations of prescribed frequency above the ultrahigh frequency UHF TV
broadcast
band and a second mixer for mixing the second local oscillations with the
response of
the UHF IF amplifier 37 to generate a very-high-frequency (VHF) intermediate
frequency signal located at frequencies below the assigned channels in the VHF
TV
broadcast band. The second detectors 9 and 38 preferably share the same second
local
oscillator.
CA 02241408 1998-06-23
The VHF IF signal from the second detector 38 is supplied to a very-high-
frequency intermediate-frequency amplifier 41, which includes controlled-gain
transistor amplifier stages that provide up to 60 dB or more amplification.
The VHF
IF amplifier 41 is provided with reverse automatic gain control derived in
response to
its output signal level, reverse AGC being preferred for the linearity of gain
it affords.
The RF amplifier 2 is provided with delayed reverse automatic gain control in
response to the output signal level of the IF amplifier 47.
Output signal from the VHF IF amplifier 47 is applied to an ATSC symbol
code detector 13, which detects baseband symbol codes therefrom. The symbol
code
detector 13 is one which uses an in-phase synchronous detector for detecting
the
vestigial-sideband amplitude-modulation of the data carrier and uses a
quadrature-
phase synchronous detector for developing automatic frequency and phase
control
(AFPC) signal for a controlled oscillator supplying synchrodyning signals to
the
synchronous detectors. The in-phase synchronous detector operates in the
analog
1 ~ regime and its output signal is digitized with L O-bit or so resolution by
an analog-to-
digital converter 14. Alternatively, the symbol code detector 13 and
succeeding ADC
14 can be replaced by a third detector for converting the VHF-band response of
the IF
amplifier 47 to a final intermediate-frequency band just above baseband, an
analog-to-
digital converter for digitizing the third detector response, and digital
synchrodyning
circuitry for synchrodyning the digitized third detector response to baseband.
Such
alternative circuitry is described by C. B. Patel et alii in U. S. patent No.
5,479,449,
issued 26 December 1995 and entitled "DIGITAL VSB DETECTOR WITH
BAU1DPASS PHASE TRACKER, AS FOR INCLUSION N AN HDTV
RECEIVER", and in U. S. patent No. 5,48,617, issued 20 August 199 and entitled
"DIGITAL VSB DETECTOR WITH BANDPASS PHASE TRACKER USING
RADER FILTERS, AS FOR USE N AuI HDTV RECEIVER", by way of examples.
When a DTV signal is being received, a direct signal resulting from the
synchronous
detection of the pilot signal accompanies the symbol codes as reproduced at
baseband
and is detected by a pilot carrier detector 1~ to generate a DTV ENABLE
signal,
which conditions the display portions of the DTV receiver to display DTV
images
rather than NTSC television images. The pilot carrier detector 1~ can, as
shown in
FIGURE 1, be of a type to respond to digital input signal or alternatively can
be of a
CA 02241408 2000-09-25
type to respond to analog input signal as supplied
directly from the symbol code detector 13.
FIGURE 1 shows the digitized baseband symbol codes
being supplied from the ADC 14 to a symbol decoder 20.
The symbol decoder 20 comprises a data slicer 21 for
data-slicing the symbol decoder 20 input signal to produce
a first symbol decoder response, an
NTSC-artifact-rejection comb filter 22 supplying a
response to the symbol decoder 20 input signal which
response suppresses any NTSC co-channel interfering
signal, a data slicer 23 for data-slicing the comb filter
22 response for generating an erroneous symbol decoder
response, a matching comb filter 24 for correcting that
erroneous symbol decoder response to produce a second
symbol decoder response, and a multiplexer 25 for
selecting one of the first and second symbol decoder
responses as the ultimate symbol decoder response supplied
by the symbol decoder 20 to a trellis decoder 16 typical
to a DTV receiver. In the absence of an indication of
substantial NTSC co-channel interfering signal being
received, the multiplexer 25 selects the first symbol
decoder response from the data slicer 21 to provide the
symbol decoder 20 output signal to the trellis decoder
16. In the presence of the indication of substantial NTSC
co-channel interfering signal being received, except
during symbol decoder initialization intervals, the
multiplexer 25 selects the second symbol decoder response
from the matching comb filter 24 to provide the symbol
decoder 20 output signal to the trellis decoder 16.
The symbol decoder 20 can be improved by modifying
the multiplexer 25 to supply ideal symbol decoding results
drawn from memory within the television receiver at times
data segment synchronization and field synchronization
code groups appear in a received DTV signal.
8
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Output signal from the VHF IF amplifier 47 is applied to circuitry 46 for
synchrodyning NTSC video carrier modulation to baseband. Both an in-phase
synchronous detector and a quadrature-phase synchronous detector are used in
the
circuitry :16 for synchrodyning NTSC video carrier modulation to baseband;
synchrodyning is presumed to be carried out in the digital regime after
converting to a
final intermediate-frequency band just above baseband; so the tlnal
intermediate-
frequency can be digitized. Alternatively, synchrodyning NTSC video carrier
modulation to baseband can be done in the analog regime, and the responses of
an in-
phase synchronous detector and a quadrature-phase synchronous detector used
for this
L O purpose can be digitized using respective analog-to-digital converters.
The response
Q of the quadrature-phase synchronous detector is the Hilbert transform of the
single
sideband components of the NTSC signal (i. e., those components above 760 kHz
in
frequency) plus the artifacts of the DTV signal as they appear in the response
I of the
in-phase synchronous detector. This Hilbert transform provided by the response
Q of
16 the quadrature-phase synchronous detector is phase shifted to provide
90° lag at all
frequencies (e:ccept possibly the lowest at which there should be little
response) by
inverse Hilbert transform circuitry 47.
Addition and subtraction are considered as being alternative forms of linearly
combining. One of linear combiners 47 and 48 is an adder and the other is a
20 subtractor. The inverse Hilbert transform response of the circuitry 47 is
linearly
combined with the response of-the-in-phase synchronous detector in the linear
combiner 48 to generate a composite video signal with high frequencies boosted
to
correct levels for application to the rest of the analog TV receiver
circuitry. In regard
to baseband composite video signal, these remaining portions typically include
sync
25 separation circuitry, color signal reproduction circuitry, and circuitry
for adapting the
4:3 aspect ratio ~1TSC image for presentation on a 16:9 viewscreen used for
displaying DTV images.
.. The inverse Hilbert transform response of the circuitry 47 is linearly
combined
in the linear combiner 49 with the in-phase baseband response I of the
synchrodyne
30 circuitry 46 to generate a luminance signal cutting off somewhat above 760
kHz,
which luminance signal is free of DTV artifacts. Whether the linear combiners
48 and
CA 02241408 1998-06-23
49 are respectively an adder and a subtractor, or whether the linear combiners
48 and
49 are respectively a subtractor and an adder depends on the whether the
operation of
the quadrature-phase synchronous detector is chosen to lead the operation of
the in-
phase synchronous detector or to lag it.
FIGURE 1 shows the band-limited luminance signal from the linear combines
49 being further filtered using a lowpass filter ~0 with a cut-off frequency
of 1 ivlHz
or so and then being squared by a squarer 31 for generating an indication of
the
energy of NTSC co-channel interfering signal during DTV reception. The squarer
31
could be constructed from a digital multiplier receiving that signal both as
multiplier
L O and as multiplicand, but is more practical to realize in read-only memory.
The squarer
31 output signal is an indication of the energy of NTSC co-channel interfering
signal
during DTV reception.
A digital threshold detector 32 determines when this indication is strong
enough to exceed a threshold value below which NTSC co-channel interfering
signal
1 ~ is considered not to be substantial enough to be likely to introduce
uncorrectable error
into the operation of the data slices 21. The threshold detector 32 response
is supplied
to multiplexes control circuitry 33. The multiplexes control circuitry 33
controls the
selection by the multiplexes 25 beriveen first and second symbol decoder
responses
that determines the ultimate symbol decoder response supplied as the symbol
decoder
20 20 output signal. The multiplexes control circuitry 33 conditions the
multiplexes 2~ to
select the first symbol decoder response as the symbol decoder 20 output
signal
during symbol decoder initialization intervals. At other times the multiplexes
control
circuitry 33 conditions the multiplexes 2~ to select the first symbol decoder
response
as the symbol decoder 20 output signal as long as the threshold detector 32
response
25 indicates that NTSC co-channel interfering signal is considered not to be
substantial
enough to be likely to introduce uncorrectable error into the operation of the
data
slices 21, otherwise conditioning the multiplexes 25 to select the second
symbol
decoder response as the symbol decoder 20 output signal.
FIGURE 2 shows the FIGURE 1 apparatus modified for supplying the
30 response of the linear combines 49 to the digital threshold detector 32
without
CA 02241408 1998-06-23
squaring by the squarer 31. The response of the linear combiner =t9 is
essentially
baseband luminance extending up to 750 kHz, so is always of the same polarity.
Accordingly, the squarer 31 can be dispensed with, and the digital threshold
detector
32 can be replaced with a digital threshold detector 032 with a prescribed
threshold
that is the square root of the prescribed threshold of the digital threshold
detector 32.
That is, the prescribed threshold of the digital threshold detector 32 is the
square of
the prescribed threshold of the digital threshold detector 032.
In the FIGURE 1 and FIGURE 2 television receivers the inclusion of the
lowpass filter ~0 eases the requirements on the inverse Hilbert transform
circuitry 47,
since exact 90° lag need not be provided at frequencies above the cut-
off frequency of
the lowpass filter ~0 in order to suppress DTV artifacts in that portion of
the
frequency spectrum. Where the inverse Hilbert transform circuitry =t7 provides
reasonably exact 90° lag up to 4.2 MHz or so, it is possible to replace
the lowpass
filter ~0 by a straight-through connection. Insofar as the combining of the
inverse
1 ~ Hilbert transform filter 47 response in linear combiner 48 with the in-
phase baseband
response I of the synchrodyne circuitry 46 to boost composite video signal
high
frequencies is concerned, the inverse Hilbert transform circuitry 47 need not
provide
exact 90° lag for frequencies clear up to 4.2 MHz, since video peaking
circuitry can be
employed to compensate for the roll-off of composite video signal high
frequencies
that attends incorrect lag if the error in lag is not too severe.
FIGURE 3 shows a modification that can be made to either of the television
receivers of FIGURES 1 and 2. In the FIGURE 3 modification, instead of the
inverse
Hilbert transform circuitry 47 shifting the quadrature-phase baseband response
Q of
the synchrodyne circuitry 46 for application to both the linear combiners 48
and 49,
inverse Hilbert transform circuitry ~1 shifts the quadrature-phase baseband
response
Q of the synchrodyne circuitry 46 for application just to the linear combiner
48; and
other inverse Hilbert transform circuitry ~2 shifts the quadrature-phase
baseband
response Q of the synchrodyne circuitry 46 for application just to the linear
combiner
49. The inverse Hilbert transform circuitry S1 provides reasonably exact
90° lag from
0.~ MHz to 4.2 MHz to optimize the spectral response of the composite video
signal,
but does not have to provide 90° la? at frequencies well below 0.~ MHz.
This avoids
11
CA 02241408 1998-06-23
the many, many tap finite-impulse-response (FIR) filter needed for providing
90° lag
at frequencies well below 0.5 MHz at the high digital sampling rates necessary
also to
provide 90° lag at frequencies up to 4.2 MHz. Owing to the use of the
lowpass filter
~0, the inverse Hilbert transform circuitry ~2 need provide reasonably exact
90° lad
only up to 1.0 MHz or so, but circuitry ~2 provides 90° lag at
frequencies well below
O.p NIHz, preferably down to a fraction of NTSC scan line rate. These
requirements
can be met at a decimated digital sampling rate, four times lower than the
digital
sampling rate used in the inverse Hilbert transform circuitry ~1,
substantially reducing
the temporary storage requirements to provide differentially delayed samples
for the
FIR filtering in the inverse Hilbert transform circuitry ~2. Indeed, the
lowpass filter
SO can be designed to have a still lower cut-off frequency below 0.5 ~IHz, so
the
decimated digital sampling rate used in the inverse Hilbert transform
circuitry 52 can
be eight times lower than the digital sampling rate used in the inverse
Hilbert
transform circuitry ~1. Or the cut-off frequency of the lowpass filter ~0 can
be further
1 ~ halved another time or few times, so the decimated digital sampling rate
used in the
inverse Hilbert transform circuitry ~2 can be further decimated from the
digital
sampling rate used in the inverse Hilbert transform circuitry ~1.
FIGURE 4 is a flow chart of the method of operation implemented by the
FIGURE 1 TV receiver. An initial step SO of receiving a digital television
signal apt
at times to be accompanied by a co-channel interfering analog television
signal having
a video portion is carried out by elements 1, 2, 3, 37, 38, and 41 of the
FIGURE 1 TV
receiver. The synchrodyne circuitry 46 carries out a subsequent step S1 of
synchrodyning the video portion of any co-channel interfering analog
television signal
to baseband, for generating an in-phase demodulation result including first
artifacts of
the DTV signal, and for generating a quadrature-phase demodulation result
including
second artifacts of the DTV signal.. The inverse Hilbert transformer circuitry
47
implements a subsequent step S2 of differentially shifting the in-phase and
quadrature-phase demodulation results in respective phase by 90° at
frequencies in a
prescribed frequency range extending well below 750 kilohertz. The linear
combiner
49 carries out a subsequent step S3 of linearly combining the in-phase and
quadrature-
phase demodulation results following their differential shifting in respective
phase by
90° at frequencies in the prescribed frequency range, to generate a
linear combining
12
CA 02241408 1998-06-23
result substantially free of the first and second artifacts of the digital
television signal
in the prescribed frequency range. (The cut-off of the lowpass filter ~0
determines the
upper boundary of this prescribed frequency range.) The squarer 31 performs a
step
S4 of squaring the linear combining result; and the digital threshold detector
32 then
carries out a final step S~ of detecting whether or not the resultin~ square
of said
linear combining result e;cceeds the square of said prescribed value, for
determining
whether the digital television signal is accompanied by co-channel interfering
analog
television signal of substantial amplitude.
FIGURE 5 is a flow chart of the method of operation implemented by the
FIGURE 1 TV receiver, differing from the FIGURE 4 flow chart in that the step
S2
implemented by the inverse Hilbert transformer circuitry 47 is more
particularly
shown as the step S2' of phase shifting the quadrature-phase demodulation
results by .
90° at frequencies in a prescribed frequency range e:ctending well
below 7~0 kilohertz.
FIGURE 6 is a flow chart of the method of operation implemented by the
1 ~ FIGURE 2 TV receiver. The method depicted in the FIGURE 6 flow chart uses
the
same steps S0, S1, S2 and S3 depicted in the FIGURE 4 flow chart. The step S4
of
squaring implemented by the squarer 31 in the FIGURE 1 TV receiver is omitted
in
the operation of the FIGURE 2 TV receiver, of course. The step S~ performed by
the
digital threshold detector 32 in the FIGURE 1 TV receiver is supplanted in the
operation of the FIGURE 2 TV receiver by a step S~' of detecting whether or
not the
amplitude of the linear combining result in the prescribed frequency range
e:cceeds a
prescribed value, for generating an indication of when the DTV signal is
accompanied
by co-channel interfering analog television signal of substantial amplitude.
This step
S~' is carried out by the digital threshold detector 032 in the FIGURE 2 TV
receiver.
FIGURE 7 is a flow chart of tl-~e method of operation implemented by the
FIGURE 6 TV receiver, differing from the FIGURE 4 flow chart in that the step
S2
implemented by the inverse Hilbert transformer circuitry 47 is more
particularly
shown as the step S2' of phase shifting the quadrature-phase demodulation
results by
90° at frequencies in a prescribed frequency range e:ctending well
below 7~0 kilohertz.
13
