Note: Descriptions are shown in the official language in which they were submitted.
CA 02241638 2000-07-31
NTSC VIDEO SIGNAL RECEIVERS WITH REDUCED SENSITIVITY
TO INTERFERENCE FROM CO-CHANNEL DIGITAL TELEVISION SIGNALS
The invention relates to NTSC television signal
receivers and, more particularly, to improvements in such
receivers for rendering them substantially less sensitive
to interference from co-channel digital television signals.
BACKGROUND OF THE INVENTION
U.S. patent No. 5,122,879 issued 16 June 1992 to
Katsu Ito and entitled "TELEVISION SYNCHRONOUS RECEIVER
WITH PHASE SHIFTER FOR REDUCING INTERFERENCE FROM A LOWER
ADJACENT CHANNEL" describes a receiver from the video
portion of an analog television signal that synchronously
detects received analog television signal both in-phase
and quadrature-phase. To improve noise figure by avoiding
amplifiers with varactor diode tuning, the Ito receiver
synchrodynes the radio-frequency (RF) amplifier response
directly to baseband, so an adjacent lower channel may
appear as an image. The quadrature-phase synchronous
detection response is phase shifted 90° at all video
frequencies above 750 kHz and linearly combined with the
in-phase synchronous detection response to suppress image
frequency components translated to baseband during
synchronous detection of the video portion of the received
NTSC signal. In U.S. patent No. 5,122,879 Ito does not
disclose the fact that this procedure also cancels the
video components above 750 kHz. The attendant loss of
luminance high frequencies is acceptable in
small-viewing-screen television receivers, however, such
as those used in wrist watches.
Hy modifying the band-limited video signal receiver
described by Ito so that the quadrature-phase synchronous
detection response is phase shifted 90° at all video
frequencies, artifacts of co-channel interfering digital
television signals are removed from the band-limited
baseband NTSC signal, it is here pointed out. Presuming
that the time constant of the automatic gain control
circuitry in the receiver is a few
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horizontal scan lines, the quadrature-phase synchronous detection response
need be
phase shifted 90° only for frequencies above a few kilohertz to keep
artifacts of
co-channel interfering digital television signals in the band-limited video
signal from
being visible on the television viewing screen or interfering with horizontal
synchronization.
SUIvIVIARY OF THE NVENTION
A video signal receiver with reduced sensitivity to interference from
co-channel digital television signals is constructed in accordance with a
principal
aspect of the invention to include input circuitry for selecting a vestigial
sideband
amplitude-modulation signal descriptive of a video signal, converting the
selected
vestigial sideband amplitude-modulation signal to an intermediate frequency
signal,
and amplifying the intermediate frequency signal to provide an amplified
intermediate
frequency signal. The vestigial sideband amplitude-modulation signal as
originally
received includes a video carrier and full sideband in addition to a vestigial
sideband.
The vestigial sideband amplitude-modulation signal is selected from any single
one of
a plurality of channels at least one of which is subject to containing at
times
co-channel interference from a digital television signal. Video synchrodyning
circuitry synchronously detects the amplified intermediate frequency signal
with
respect to video cazrier signal, for generating an in-phase synchronous
detection
response and for generating a quadrature-phase synchronous detection response.
All
frequency components of the quadrature-phase synchronous detection response
above
a prescribed frequency are phase shifted by substantially 90° by an
inverse Hilbert
transform filter and are linearly combined with suitably delayed in-phase
synchronous
detection response for recovering lower frequency portions of the video signal
described both in the full sideband and the vestigial sideband of the
vestigial sideband
amplitude-modulation signal, substantially free of artifacts from any co-
channel
interfering digital television signal. The term "linear combiner" as used in
this
specification and the claims appended thereto is a generic term for "additive
combiner", or adder, and for "differential combiner", or subtractor.
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CA 02241638 1998-06-23
In accordance with further aspects of the invention, the video signal receiver
includes circuitry for recovering higher frequency portions of the video
signal
described in the full sideband of the vestigial sideband amplitude-modulation
signal,
but not in its vestigial sideband. The vestigial sideband amplitude-modulation
signal
as supplied to this circuitry for recovering higher frequency portions of the
video
signal is selectively filtered to remove the pilot carrier signal component of
any
co-channel digital television signal. This is done to avoid any artifact of
such pilot
carrier signal being generated when recovering higher frequency portions of
the video
signal. The video signal receiver further includes circuitry for linearly
combining
those higher frequency portions of the video signal with the lower frequency
portions
of the video signal, which are described both in the full sideband and the
vestigial
sideband of the vestigial sideband amplitude-modulation signal, and which,,are
substantially free of artifacts from any co-channel interfering digital
television signal.
BRIEF DESCRIPTION OF THE DRAWING
FIGURES 1, 2, 3, 4, ~, 6 and 7 are each a schematic diagram of a respective
television receiver that is capable of receiving NTSC analog TV signals as
well as
DTV signals, which receiver employs the method of the invention for detecting
the
presence in DTV signals of co-channel interfering NTSC analog TV signals.
FIGURE 8 is a plot of preferred spectral responses for portions of the
television receivers of FIGURES 1, 2, 3, 4 and 5.
FIGURE 9 is a schematic diagram of synchrodyne circuitry as can be
employed in any of the television receivers of FIGURES l, 2, 3, 4, 5, 6 and 7.
FIGURE 10 is a plot of preferred spectral response for a portion of the
television receivers of FIGURES 6 and 7.
DETAILED DESCRIPTION
FIGURE 1 shows a television receiver that is capable of receiving NTSC
analog TV signals as well as DTV signals. Over-the-air type television
broadcasting
signals as received by an antenna 1 are amplified by an adjustably tuned
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radio-frequency amplifier 2 and supplied to a first detector 3. The RF
amplifier 2 and
the first detector 3 have adjustable tuning and together function as a tuner
for selecting
said digital television signal from one of channels at different locations in
a frequency
band. The first detector 3 includes a first local oscillator supplying first
local
S oscillations tunable over a frequency range above the ultra-high-frequency
(UHF) TV
broadcast band and a first mixer for mixing the first local oscillations with
a TV signal
selected by the adjustably tuned RF amplifier 2 for upconverting the selected
TV
signal to generate a UHF intermediate-frequency signal in a 6-MHz-wide UHF
intermediate-frequency band located at frequencies above the assigned channels
in the
UHF TV broadcast band.
The first detector 3 supplies the high-IF-band signal to a UHF-band
intermediate-frequency amplifier 4 for NTSC audio signal, to a UHF-band ~~
intermediate-frequency amplifier S for fullband NTSC video signal, and to a
UHF-band intermediate-frequency amplifier 6 for NTSC video highs signal. The
I S responses of the UHF-band IF amplifiers 4, S and 6 are supplied to
respective second
detectors 7, 8 and 9 to be downconverted to respective VHF-band
intermediate=frequency signals in a VHF band below the very high frequencies
assigned as TV broadcast channels. The second detectors 7, 8 and 9 share a
common
second local oscillator for generating second local oscillations and have
respective
second mixers for mixing those second local oscillations with the responses of
the
UHF-band IF amplifiers 4, 5 and 6, respectively. The VHF-band IF signals from
the
second detector detectors 7, 8 and 9 are respectively supplied to a VHF-band
intermediate-frequency amplifier 10 for NTSC audio signal, to a VHF-band
intermediate frequency amplifier 11 for fullband NTSC, video signal, and to a
VHF-band intermediate frequency amplifier 12 for NTSC video highs signal.
The UHF-band IF amplifiers 4, S and 6 include surface-acoustic-wave (SAW)
filters for UHF-IF-band NTSC audio signal, for UHF-IF-band fullband NTSC video
signal and for UHF-IF-band NTSC video highs signal, respectively. SAW filters
with
steep rejection skirts, but with pass bands having linear group delay and flat
amplitude
response, are more easily implemented at UHF than at VHF. This is the reason
for
preferring to determine overall IF response for UHF-IF-band NTSC audio signal,
for
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UHF-IF-band fullband NTSC video signal and for UHF-IF-band NTSC video highs
signal in the UHF IF band rather than in the VHF IF band.
The SAW filter in the IF amplifer 5 for determining overall IF response for
fullband NTSC video signal preferably has substantially flat amplitude
response for
those portions of the VSB AM signal ranging between frequencies X00 kHz to at
least
3.5 MHz above the lower limit of the 6-MHz-wide TV broadcast channel as that
VSB
AM signal is translated to the UHF IF band, that rejects in-channel and
adjacent-channel NTSC audio signals, and that has substantially linear phase
response
throughout its passband. The SAW filter in the IF amplifier 5 may suppress or
reject
the pilot carrier of any co-channel interfering ATSC DTV signal, so long as
linear
phase response is maintained to 750 kHz from NTSC video carrier frequency. The
self resonances of the IF filtering for fullband NTSC video signal, which are
stimulated by impulse noise, are near the middle of the IF passband. So,
ringing
effects caused by impulse noise are less apt to affect baseband video response
below
7~0 kHz if the IF filtering has at least 3 MHz bandwidth. '
The SAW filter in the IF amplifier 6 for determining overall IF response for
NTSC video highs signal rejects in-channel and adjacent-channel NTSC audio
signals, and preferably this SAW filter e;chibits a roll-off for the lower
1.75 MHz or so
of the 6-MHz-wide TV broadcast channel as translated to the UHF IF band and
has
substantially linear phase response throughout its passband. The roll-off for
the lower
1.75 NIHz or so of the 6-MHz-wide TV broadcast channel as translated to
intermediate frequencies rejects adjacent-channel NTSC audio signal, the pilot
carrier
of any co-channel interfering ATSC DTV signal and the in-channel NTSC video
carrier. The SAW filter in the IF amplifier 6 preferably e:chibits a roll-off
for the
upper 550 kHz or so of the 6-MHz-wide TV broadcast channel as translated to
the
UHF IF band and rejects in-channel sound signal.
FIGURE 8 shows the desired overall receiver responses, as referred to the
lower frequency of the original transmission channel, at the output ports of
the UHF
IF amplifiers 5 and 6.
CA 02241638 1998-06-23
r , ,
The UHF-band IF amplifiers 4, ~ and 6 can include wideband constant-gain
amplifiers for driving their component SAW filters from source impedances that
minimize multiple reflections and for overcoming the insertion losses of their
component SAW filters. The VHF-band IF amplifiers 10, 11 and 12 include
respective controlled-gain amplifiers that provide up to 60 dB or more
amplification.
The VHF-band IF amplifiers 10, 11 and 12 each include stages with forward
automatic gain control derived in response to the output signal level of the
IF
amplifier 11, forward AGC being preferred for the better noise figure it
affords. The
RF amplifier 2 is provided with delayed reverse automatic gain control in
response to
the output signal level of the IF amplifier 11.
The response of the VHF IF amplifier 10 is applied to an intercarrier sound
detector 13, which supplies 4.5 MHz intercarrier sound intermediate-frequency
signals
to an intercarrier sound intermediate-frequency amplifier 14 which amplifies
and in
most designs symmetrically limits the amplified response for application to an
FM
1~ detector 1~. The FM detector 1~ reproduces baseband composite audio signal
supplied to the sound reproduction portion 16 of the NTSC receiver per
conventional
practice. The sound reproduction portion 16 of the NTSC receiver typically
includes
stereophonic decoder circuitry. If the NTSC audio signals are selected with
narrowband filtering in the IF amplifiers 4 and 10 that pass only the FM audio
carrier
as translated to intermediate frequencies, the intercarrier sound detector 13
can be
provided by a multiplier that multiplies the IF amplifier 10 response by a
video carrier
supplied from a third local oscillator in circuitry 17 for synchrodyning the
fullband
NTSC video signal to baseband.
Alternatively, if the NTSC audio signals are selected with filtering in the IF
amplifiers 4 and 10 that passes both the NTSC video and audio carriers as
translated
to intermediate frequencies, for implementing "quasi-parallel" sound, the
intercarrier
sound detector 13 can be a simple rectifier or can be a square-law device. A
video
carrier is then no longer supplied from a third local oscillator in the
circuitry 17 for
synchrodyning the fullband NTSC video signal to baseband.
6 ..
CA 02241638 1998-06-23
Output signal from the VHF IF amplifier 11 is applied to the circuitry 17 for
synchrodyning NTSC video carrier modulation to baseband, which circuitry can
take
the form shown in FIGURE 9. Both an in-phase synchronous detector and a
quadrature-phase synchronous detector are used in the circuitry 17 for
synchrodyning
NTSC video carrier modulation to baseband. Synchrodyning is carried out in the
analog regime in the specific circuitry 17 for synchrodyning NTSC video
carrier
modulation to baseband shown in , and the responses of the in-phase
synchronous
detector 170 and the quadrature-phase synchronous detector 171 used for this
purpose
are digitized using respective analog-to-digital converters 172 and 173. The
third
local oscillator 174 in the circuitry 17 supplies oscillations in 0°
phasing to the
in-phase synchronous detector 170 and supplies oscillations in +90°
phasing or
in -90° phasing via a phase shift network 17~ to the quadrature-phase
synchronous
detector 171. The third local oscillator 174 is a controlled oscillator
provided
automatic frequency and phase control (AFPC) signal responsive to the unwanted
appearance of low frequency components in the quadrature-phase synchronous
detector 171 response. FIGURE 9 shows the AFPC signal being generated using
the
commonplace Costar loop arrangement in which the responses of the in-phase
synchronous detector 170 and the quadrature-phase synchronous detector 171 are
filtered by lowpass filters 176 and 177, the responses of the lowpass filters
176 and
177 are multiplicatively mi:ced in a mi:cer 178, and the.resulting product is
filtered by
a lowpass filter 179 to generate the AFPC signal for the third local
oscillator 174.
Alternatively, synchrodyning NTSC video carrier modulation to baseband can
be done in the digital regime after converting to a final intermediate-
frequency band
just above baseband, so the final intermediate-frequency can be digitized.
This avoids
any problems with the two analog-to-digital converters 172 and 173 differing
somewhat in conversion gain.
The digital response Q of the quadrature-phase synchronous detector 171 is
the Hilbert transform of the single sideband components of the NTSC signal (i.
e.,
those components above 750 kHz in frequency) plus the artifacts of the DTV
signal as
they appear in the response I of the in-phase synchronous detector 170. The
reader's
attention is now directed back to FIGURE 1. This Hilbert transform provided by
the
CA 02241638 1998-06-23
response Q of the quadrature-phase synchronous detector in the synchrodyne
circuitry
17 is phase shifted to provide 90° lag at all frequencies above a few
kilohertz by
inverse Hilbert transform circuitry 18. Finite-impulse-response digital
filters suitable
for the inverse Hilbert transform circuitry 18 are known in the digital
television
S receiver art.
The inverse Hilbert transform response of the circuitry 18 is linearly
combined
in a linear combiner 19 with the digital response I of the in-phase
synchronous
detector, to generate a luminance signal cutting off somewhat above 7~0 kHz.
This
luminance signal is generally free of DTV artifacts, owing to their single-
sideband
character as referred to NTSC video carrier frequency. Whether the linear
combiner
19 is an adder or a subtractor depends on the whether the operation of the
quadrature-phase synchronous detector is chosen to lead the operation of the
in-phase
synchronous detector or to lag it.
Output signal from the VHF IF amplifier 12 is applied to a quadrature-..phase
synchronous detector 20 for synchrodyning to baseband the NTSC video carrier
modulation that is descriptive of the higher-frequency portions of the
composite video
signal. The ~quadrature-phase synchronous detector 20 supplies a digital
response Q'.
By way of e:cample, if quadrature-phase synchronous detection is performed in
the
analog regime, an analog-to-digital converter is cascaded after the
synchronous
detector for digitizing its response. Synchronous carrier for the quadrature-
phase
synchronous detector 20 is supplied from the same source in the circuitry 17
for
synchrodyning NTSC video carrier modulation to baseband as supplies the
quadrature-phase synchronous detector within the synchrodyne circuitry 17 (e.
g.,
from the phase shift network 175). The response Q' of the quadrature-phase
synchronous detector 20 is the Hilbert transform of the single sideband
components of
the NTSC signal (i. e., those components above 750 kHz in frequency) plus the
artifacts of the portion of the DTV signal passed by the SAW filter in the IF
amplifier
6. This Hilbert transform provided by the response Q' of the quadrature-phase
synchronous detector 20 is phase shifted to provide 90° lag at least at
frequencies
above S00 kHz or so by inverse Hilbert transform circuitry 21. This procedure
generates a response that is the same at higher frequencies as the response of
an
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in-phase NTSC video detector, but which exhibits a low-frequency cut-off that
is
complementary to the high-frequency cut-off of the linear combiner 19.
A linear combiner 22 combines the responses of the linear combiner 19 and of
the quadrature-phase synchronous detector 20 to generate a fullband composite
video
signal for application to the portion 23 of the NTSC receiver used to
reproduce
pictures on a viewing screen. This portion 23 of the NTSC receiver typically
includes
sync separation circuitry and color signal reproduction circuitry; in a
combination
NTSC and HDTV receiver circuitry will also be included for adapting the 4:3
aspect
ratio NTSC image for presentation on a 16:9 viewscreen used for displaying DTV
images.
The inverse Hilbert transform circuitry 18 requires a substantial amount of
latency (or insertion delay) in order to provide 90° lag for
frequencies as low as a few
kilohertz. Providing 90° lag for frequencies that are a fraction of
horizontal scan line
rate means that uncancelled artifacts-of DTV signals will be of low enough
frequency
I ~ that receiver AGC will operate to suppress them. Shim delay is necessary
in the I
signal connection from synchrodyne circuitry 17 to linear combiner 19 for
equalizing
the latencies of the I and Q signals supplied to the linear combiner 19. Shim
delay
must be cascaded with the inverse Hilbert transform circuitry 21 to the extent
that its
latency is less than that of the inverse Hilbert transform circuitry 18.
Making the
inverse Hilbert transform circuitry 21 the same as the inverse Hilbert
transform
circuitry 18 is possible to avoid the need for such shimming. When such
modification
is made, the circuitry can be subjected to a reduction technique that
eliminates the
need for separate inverse Hilbert transform circuitry 18 and 21.
FIGURE 2 shows a television receiver that is capable of receiving NTSC
analog TV signals as well as DTV signals, which receiver results from such
reduction.
Elements 18-22 of the FIGURE 1 television receiver are replaced by an adder 24
for
combining the Q output signal of the synchrodyne circuitry 17 and the Q'
output
signal of the quadrature synchronous detector 20, inverse Hilbert transform
circuitry
25 responsive to the sum output signal from the adder 24, and linear combining
circuitry 26 for linearly combining the inverse Hilbert transform circuitry 25
response
CA 02241638 1998-06-23
with the I output signal of the synchrodyne circuitry 17 to generate a
luminance signal
cutting off somewhat above 750 kHz. This luminance signal is generally free of
DTV
artifacts, owing to their single-sideband character as referred to NTSC video
carrier
frequency. If the quadrature synchronous detector 20 and the quadrature
synchronous
detector within the synchrodyne circuitry 17 are operated out-of phase with
each
other, rather than in-phase with each other as presumed, the adder 2:! is
replaced by a
subtractor to achieve equivalent operation.
The synchronous detection of video high frequencies using quadrature-phase
video carrier is advantageous in that cross-over between video low frequencies
and
video high frequencies is automatically correct. Furthermore, cross-over
occurs at the
highest video frequencies possible so that DTV artifact cancellation extends
to as high
frequency as possible.
FIGURE 3 shows a modification of the FIGURE 1 television receiver, which
modification uses an in-phase synchronous detector 27 for synchronous
detection of
video high frequencies, rather than the quadrature-phase synchronous detector
20.
The quadrature-phase synchronous detector 20 is dispensed with, together with
the
inverse Hilbert transform circuitry 21 and the linear combiner 22. A cross-
over filter
28 lowpass filters the response of the linear combiner 19 and highpass filters
the I'
'output signal of the in-phase synchronous detector 27 before linearly
combining them
to generate a fullband NTSC composite video signal for application to the
portion 23
of the NTSC receiver used to reproduce pictures on a viewing screen. The cross-
over
frequency at which the lowpass filtering and highpass filtering cut off in the
cross-over filter 28 is preferably at least 500 kHz. The FIGURE 2 television
receiver
is more economical of hardware than the FIGURE 3 receiver, since the cross-
over
filter 28 is not required in the FIGURE 2 receiver.
In the FIGURE 1, FIGURE 2 and FIGURE 3 television receivers chroma
demodulation circuitry is presumed to be included in the portion 23 of the
NTSC
receiver used to reproduce pictures on a viewing screen, with chroma signal
being
separated from the fullband composite video signal applied to that portion 23
of the
NTSC receiver used to reproduce pictures on a viewing screen. However, it is
CA 02241638 1998-06-23
possible alternatively to separate chroma signal from the high frequency
component
of the composite video signal before its combination with the low frequency
component of the composite video signal.
FIGURE 4 shows a variant of the FIGURE 3 television receiver having
conventional chroma demodulation circuitry 29 connected to be directly
responsive to
the baseband video high frequencies as detected by the in-phase synchronous
detector
27. The chroma demodulation circuitry 29 is shown as being separate from a
portion
30 of the NTSC receiver used to reproduce pictures on a viewing screen. The
chroma
demodulation circuitry 29 supplies color difference signals to that portion 30
of the
NTSC receiver, which portion 30 receives fullband composite video signal from
the
cross-over filter 28.
FIGURE S shows a variant of the FIGURE 2 television receiver having
chroma demodulation circuitry 29 connected to be directly responsive to the
baseband
video high frequencies as detected by the quadrature-phase synchronous
detector 20.
Since color burst like other chroma signal components is phase shifted by
90°, the fact
of the Hilbert transform of color signal rather than actual color signal being
synchronously detected has no substantial effect on color difference signal
recovery.
The FIGURE 6 television receiver differs from the FIGURE 2 TV receiver in
that the UHF-band IF amplifier ~ is replaced by a UHF-band IF amplifier 31
with a
passband for the complete frequency spectrum of the VSB AM NTSC video carrier
modulation, the NTSC video lows second detector 8 is replaced by a second
detector
32 for the complete frequency spectrum of the VSB AM NTSC video carrier
modulation, and the VHF-band IF amplifier 11 is replaced by a VHF-band IF
amplifier 33 with a passband for the complete frequency spectrum of the VSB AM
NTSC video carrier modulation. FIGURE 10 shows the desired overall receiver
response, as referred to the lower frequency of the original transmission
channel, at
the output port of the UHF IF amplifier 33. This full bandwidth response
permits the
. UHF-band IF amplifier 6, the NTSC video highs second detector 9, the VHF-
band IF
amplifier 12, the NTSC video highs quadrature synchronous detector 20, and the
inverse Hilbert transform circuitry 21 to be dispensed with entirely. Instead,
a
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highpass filter 34 extracts video high frequencies from the response of the
inverse
Hilbert transform circuitry 18 for application to the linear combiner 22,
there to be
linearly combined with the video low frequencies supplied from the linear
combiner
19.
The FIGURE 7 television receiver differs from the FIGURE 3 TV receiver in
that the UHF-band IF amplifier S is replaced by a UHF-band IF amplifier 31
with a
passband for the complete frequency spectrum of the VSB AM NTSC video carrier
modulation, the NTSC video lows second detector 8 is replaced by a second
detector
32 for the complete frequency spectrum of the VSB AM NTSC video carrier
modulation, and the VHF-band IF amplifier 11 is replaced by a VHF-band IF
amplifier 33 with a passband for the complete frequency spectrum of the VSB
AiVI
NTSC video carrier modulation. FIGURE 10 shows the desired overall receiver
response, as referred to the lower frequency of the original transmission
channel, at
the output port of the UHF IF amplifier 33. This full bandwidth response
permits the
UHF-band IF amplifier 6, the NTSC video highs second detector 9, the VHF-band
IF
amplifier 12 and the NTSC video highs in-phase synchronous detector 27 to be
dispensed with entirely. Instead, the response I of the in-phase synchronous
detector
in the synchrodyne circuitry 17 is applied to the cross-over filter 28 to
supply it with
video high frequencies.
In variants of the FIGURE 1 TV receiver, chroma demodulation circuitry can
be arranged to directly respond to the response of the quadrature-phase
synchronous
detector 20 or of the inverse Hilbert transform circuitry 21. In variants of
the
FIGURE 6 and FIGURE 7 TV receivers, chroma demodulation circuitry can be
arranged to directly respond to either the I output signal or the Q output
signal of the
synchrodyne circuitry 17 or to directly respond to the response of the inverse
Hilbert
transform circuitry 18; these arrangements are possible in the FIGURE 1,
FIGURE 2
and FIGURE 3 TV receivers as well, if their video lows IF amplifiers are
modified to
have an overall fullband response as shown in FIGURE 10.
The effect of artifacts of co-channel interfering digital signal on chroma
demodulation results can (like other forms of random noise) be reduced by
transversal
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CA 02241638 2000-07-31
filtering, since digital television signals are random
from scan line to scan line while chrominance signals tend
to exhibit strong line-to-line anticorrelation before
demodulation and strong line-to-line correlation after
demodulation. Modifications of the television receivers
thus far disclosed to use PAL or SECAM signals rather than
NTSC signals are easily effected by one skilled in the art
of television receiver design when acquainted with the
foregoing disclosure. While the foregoing disclosure
describes NTSC television receivers that reproduce sound
and picture, the invention has application to NTSC
television receivers that do not reproduce sound and
picture, such as those incorporated into video tape
recorders or into NTSC signal cancellation filters for
digital television receivers. One skilled in the art of
television receiver design when acquainted with the
foregoing to disclosure will be enabled to design many
variants of the receivers as preferred embodiments, and
this should be borne in mind when determining the scope of
the claims which follow.
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