Note: Descriptions are shown in the official language in which they were submitted.
CA 02254370 1998-11-13
GROUP DELAY EQUALIZATION
OF RADIO FREQUENCY SIGNALS
BY PIECEWISE EQUALIZATION IN A LOWER FREQUENCY BAND
S
BACKGROUND OF THE INVENTION
1. Field of the Invention
The present invention relates to equalization of signals that have substantial
group
delay distortion that must be equalized. More specifically, the invention
relates to time-
delay equalization of digital television (DTV) signals that have significant
group delay
distortion that is introduced by a DTV channel band pass filter.
2. Related Art
The United States Federal Communications Commission (FCC) Sixth Report and
Order (April 1997), as modified by the Memorandum Opinion and Order on
Reconsideration of the Sixth Report and Order (M.O.&O., 47 C.F.R. ~ 73.622(h))
(February 23, 1998), has mandated that the out-of channel digital television
(DTV)
signals from DTV broadcast transmitters be suppressed much more than from
analog (for
example, NTSC) transmitters. Accordingly, DTV transmitters must use an output
band
pass filter (BPF) that has very steep skirts, and high levels of rejection,
just outside the
DTV channel in the surrounding channels.
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CA 02254370 1998-11-13
Unfortunately, the filtering needed for this skirt steepness and rejection
introduces
substantial group delay to the DTV signal components that are inside the DTV
channel,
near its edges. This group delay for the required DTV signal filtering (on the
order of
hundreds of nanoseconds) compares with much smaller group delay experienced
with
NTSC transmitters (on the order of tens of nanoseconds). FIG. 3(a)
schematically
illustrates the significant group delay distortion that a band pass filter
(BPF) introduces
into DTV signals near its channel edges. One must compensate for this group
delay
distortion.
Applicant has recognized that, in theory, it is possible to equalize (pre-
compensate for) the BPF-induced group delay distortion at intermediate
frequency (41-47
MHz), before amplification and frequency conversion to broadcast frequency.
However,
cascaded arrangements of inductors and capacitors that would comprise such IF
equalizers would demand extremely small-valued or large-valued components, and
the
Q of one or more stages of such equalizers could be as high as 100,000.
Accordingly,
equalizing DTV signals at the 41-47 MHz band at intermediate frequency (IF) is
not a
practical solution to compensating for group delay distortion of DTV signals
that is
introduced by RF channel band pass filters.
Thus, there is a need in the art for a practical implementation of an
equalizer that
can equalize group delay distortion, especially group delay distortion in DTV
signals that
is introduced by band pass filters. It is to meet this need that the present
invention is
directed.
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CA 02254370 1998-11-13
SLT1~IARY OF THE INVENTION
The present invention provides a system and method for equalizing undesirable
group delay characteristics caused by radio frequency (RF) band pass filters
(or other
sources) that cannot practically be equalized even at conventional
intermediate
S frequencies (IF). The invention provides frequency translation of the IF
signal to a
frequency range that allows the group delay to be equalized in piecewise
frequency
segments.
For example, one embodiment of the invention provides for two frequency
translations of a 41-47 MHz IF TV signal down to a secondary IF band (4-10
MHz) range
so that two equalizers that operate in the 4-7 MHz range can consecutively
equalize the
lower and upper halves of the 6 MHz broadband TV signal. Of course, the
invention
envisions that such consecutive piecewise equalization can involve translation
between
a variety of frequency bands, and can involve other than two piecewise
equalizations.
More generally, an arrangement is provided for compensating for group delay
distortion that includes: consecutively translating plural frequency segments
of a signal
in a first frequency band to one or more target frequency segments, and
equalizing, in the
target frequency segments, the consecutively-translated frequency segments of
the signal,
so as to compensate for the group delay distortion that is inherent in the
respective
frequency segments of the signal in the first frequency band.
Other objects, features and advantages of the present invention will be
apparent
to those skilled in the art upon a reading of this specification including the
accompanying
drawings.
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BRIEF DESCRIPTION OF THE DRAWINGS
The invention is better understood by reading the following Detailed
Description
of the Preferred Embodiments with reference to the accompanying drawing
figures, in
which like reference numerals refer to like elements throughout, and in which:
FIG. 1(a) illustrates the application of the inventive group delay
equalization
arrangement in a television transmitter, in this example, a transmitter that
is a combined
analog (NTSC) and digital television (DTV) broadcast transmitter.
FIGS. 1 (b) and 1 (c) illustrate application of the inventive group delay
equalization
arrangement in alternative television transmitters, namely, in an analog
(NTSC) broadcast
transmitter and a digital television (DTV) broadcast transmitter,
respectively.
FIG. 2 is a block diagram of a preferred embodiment of the group delay
equalization circuit according to the present invention.
FIGS. 3(a) through 3(e) illustrate group delay as a function of frequency at
corresponding points in the embodiment of FIG. 2.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
In describing preferred embodiments of the present invention illustrated in
the
drawings, specific terminology is employed for the sake of clarity. However,
the
invention is not intended to be limited to the specific terminology so
selected, and it is
to be understood that each specific element includes all technical equivalents
that operate
in a similar manner to accomplish a similar purpose. Moreover, choice of
components
CA 02254370 1998-11-13
and design procedures that are well understood to those skilled in the art
(such as
equalizer filter design, choice of mixers, use of signal splitters and
amplifiers, and the
like) are omitted as not being essential to the invention that is claimed.
FIG. 1(a) illustrates an example of a transmitter in which the group delay
equalization scheme according to the present invention may be employed.
Briefly, the
transmitter transmits analog (e.g., NTSC) and DTV signals using a common
antenna
1150 after amplifying them with a single wideband (12 MHz) amplifier 1145. The
top
portion of FIG. 1 (a) shows the NTSC portion, and the bottom portion of FIG. 1
(a) shows
the DTV portion. Such a circuit was disclosed in co-pending U.S. Application
No.
09/050,109, which is incorporated herein by reference.
Referring to FIG. 1 (a), a video processor 210 manipulates an input video
signal
to compensate, in advance, for distortion that is expected to occur at
intermediate
frequency (If) and broadcast radio frequency (RF) to the modulated visual
signal. Video
processor 210 provides such compensation functions as differential gain
compensation
and differential phase compensation and luminance non-linearity compensation,
to
achieve linearity of response.
An NTSC modulator 220 receives the compensated video signal from the video
processor 210, along with an associated audio signal. Essentially, the
modulator
modulates the video and audio signals onto an intermediate frequency carrier
from a
phase lock loop (PLL) 225 that is phase locked to common frequency reference
100. The
modulator may comprise, for example, a vestigial side band (VSB) filter
implemented
with surface acoustic wave (SAW) technology. NTSC modulator 220 outputs an
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intermediate-frequency signal with NTSC-standard visual and aural carriers at
45.75
MHz and 41.25 MHz carrier frequencies, respectively.
Frequency reference 100 may include, for example, an oscillator 101 that
provides
a stable-frequency reference signal (such as 10 MHz) to various circuit
components via
a signal splitter 102. Oscillator 101 may be implemented as a temperature-
controlled
crystal oscillator, an oven-controlled crystal oscillator, a GPS (global
positioning system)
reference signal, and the~like. Signal splitter 102 may be any circuit that
fans out the
reference signal, ensuring a constant phase relationship throughout the
circuits it drives.
The NTSC IF signal from modulator 220 is pre-distorted by an NTSC IF
processor 230 to compensate for distortion that is expected to occur at radio
frequency
(RF) to the modulated NTSC signal. NTSC IF processor 230 may compensate for
such
undesirable phenomena as intermodulation distortion, cross-modulation
distortion, and
incidental carrier phase modulation distortion, and the like, resulting in a
compensated,
purely amplitude-modulated signal that is desired. The inventive group delay
equalization circuit may be included in IF processor 230. In this manner, the
invention
can, at IF and iow power, pre-correct group delay distortion caused by a band
pass filter
(BPF) 1146 acting on an RF signal at high power. The pre-compensated NTSC IF
signal
is provided to IF-to-broadcast-frequency converter 240.
IF-to-broadcast-frequency converter 240 also receives a sinusoidal carrier of
a
frequency determined by the desired broadcast frequency of the particular
broadcast
channel "N", such as in the IJHF range, that is allocated to the broadcast
site involved.
The carrier is provided by a phase lock loop (PLL) 250, which is phase-locked
to an
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output from the frequency reference 100. IF-to-broadcast-frequency converter
240
includes a mixer that provides a low-power (for example, one watt) modulated
NTSC
signal at Channel N's broadcast frequency. Converter 240 reverses the
frequency order
of the aural and visual components of this low-power, broadcast-frequency,
modulated
NTSC signal, so that the visual component carrier is now below the aural
component
carrier, in accordance with broadcast standards.
A series of amplifiers, shown by exemplary internlediate power amplifier (IPA)
260 and driver amplifier 270, amplify the low-power, broadcast-frequency,
modulated
NTSC signal from converter 240 to a power level closer to broadcast power
levels. For
example, driver amplifier 240 may output a signal of 2.5 kW peak average power
at sync,
with 125 W aural power. This signal is provided to the first input of the
combiner 1140.
Preferably, the signal output to the combiner is subject to automatic gain
control
(AGC). For this purpose, one example (not shown) of an AGC feedback path is
provided
from the output of driver amplifier 270 to the IF-to-broadcast-frequency
converter 240.
Gain control circuitry that may be of conventional design, and located within
converter
240, ensures that an NTSC signal of substantially constant power level is
provided to the
combiner.
Like the description of the NTSC signal modulator, the following description
omits conventional elements known to those skilled in the art, with the
understanding that
commercially-available products perform the same overall function. Further,
the present
description is abbreviated because the functions performed by elements 320,
325, 330,
340, 350, 360, 370, and 377 perform functions that are analogous to the
functions
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CA 02254370 1998-11-13
performed by elements 220, 225, 230, 240, 250, 260, 270, and 277,
respectively.
Modulator 320 receives a carrier frequency signal that is phase locked by PLL
325 to a reference carrier from frequency reference element 100. Modulator 320
further
encodes a 19.39 MHz, SMPTE 310M-compliant MPEG bit stream, and modulates a
pilot carrier at 46.69 MHz in accordance with (for example) the 8-VSB standard
accepted
by the Federal Communications Commission for terrestrial broadcast. Modulator
320
outputs an intermediate-frequency analog signal with a pilot carrier at 46.69
MHz, at the
upper edge of the 41-47 MHz band allocated for television signals at IF.
A DTV IF processor 330 processes the IF signal from the modulator, performing
pre-compensation and pre-conditioning functions generally analogous to that
performed
by processor 230 for NTSC signals. However, DTV IF processor 330 is preferably
implemented as a digital signal processor (DSP) to perform the pre-
compensation and
pre-conditioning functions on a digital-content signal, using techniques (such
as finite
impulse response filters) that are better suited to processing of such
signals.
The inventive group delay distortion device of FIG. 2 may be part of processor
330 (FIG. 1 (a)). The FIG. 2 circuit is used at IF, before the corrected DTV
signal is RF-
converted, and amplified by the main DTV amplifier 1145. Group delay
pre-compensation is made at low power and at IF, with the goal of achieving
equalization
of the group delay at the high-power, RF output of BPF 1146.
The DTV IF processor 330 provides a pre-compensated and pre-conditioned
signal to an IF-to-broadcast-frequency converter 340. Converter 340 converts
the analog
IF signal from the DTV IF processor 330 to a broadcast-frequency signal. In
the
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CA 02254370 1998-11-13
preferred application of the invention, in which the DTV channel is
immediately adjacent
the corresponding NTSC channel in the frequency spectrum in accordance with
FCC
channel assignments, two situations are encountered. The "N-1" situation
involves a
DTV channel that is immediately below the NTSC channel, and the "N+1"
situation
S involves a DTV channel that is immediately above the NTSC channel. The IF-to-
broadcast-frequency converter 340 is therefore illustrated as providing a
signal on
Channel N-1 or Channel N+1, where "N" is the channel assigned the
corresponding
NTSC channel. Essentially including a mixer, converter 340 receives a
sinusoidal carrier
signal from a phase lock loop 350 that is driven by frequency reference
element 100 to
be modulated by the DTV IF signal.
The converter's operation results in a reversal of the frequency order of the
DTV
signal from the upper end of the channel (46.69 MHz is near 47 MHz) to the
lower end
ofthe channel at broadcast frequencies. It is noteworthy that, in the N+1
situation, this
placement of the DTV signal at the lower end of the DTV channel places it only
510 kHz
away from the deviated NTSC aural carrier.
Converter 340 provides a low-power, broadcast-frequency signal to a series ~of
amplifiers, shown as including an intermediate power amplifier (IPA) 360 and a
driver
amplifier 370. Driver amplifier 370 provides to the combiner, an 8-VSB-
compliant DTV
signal, in Channel N-1 for the "N-1" channel allocation situation or in
Channel N+1 for
the "N+1 " channel allocation situation. The signal from driver amplifier 370
is of a
power level sufficient to drive the high power amplifier 1145 to provide the
desired
broadcast output power. An AGC feedback path (not shown) may be provided from
the
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CA 02254370 1998-11-13
drivers's output back to the converter 340, which ensures that the signal
provided to the
combiner is of substantially constant power level.
Combiner 1140 is preferably implemented as a conventional quadrature hybrid
combiner of the type discussed in detail in commonly-assigned U.S. Patent No.
4,804,931. As is readily appreciated by those skilled in the art, hybrid
combiners are
four-port devices that have two outputs, each one of which receives half the
signal power
from each of the combiner's two inputs. Thus, an undesirable characteristic of
hybrid
combiners is that they halve the power of the sum signal. In the present use
of hybrid
combiners, half the power from each input signal is provided to the high-power
amplifier
1145, while the other half of the power from each input signal is wasted
through
dissipation in a resistance to ground. Despite the power loss to resistance,
the desirable
linearity of the hybrid combiner, and the isolation of the input signals from
each other to
thereby avoid undesirable mixing of the two inputs, make it a preferred
implementation
for combiner 1140.
High-power amplifier 1145 fulfills the demand of flatness of response (less
than
1 dB) across a two-channel-wide bandwidth (12 MHz in the United States, 16 MHz
in
most countries outside the U.S.), and the requirement for meaningful power to
the
NTSC+DTV signals with minimal inter-channel interference. A tetrode-class
device, and
especially a diacrode amplifying device such as a Thomson TH-680, provide
optimum
performance for this application. Tetrode and diacrode implementations are
preferred
because of their ability to operate with cavity sections tunable to wide (two-
channel wide)
bandwidths, to exhibit sufficient linearity so that cross modulation and
intermodulation
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distortion may be corrected with established methods, and to provide
meaningful
broadcast power levels. Of course, the scope of the invention should not be
limited to
tetrode and diacrode solutions; alternative implementations, such as those
involving solid
state amplifiers, also lie within the contemplation of the invention. The
diacrode or
tetrode power amplifier may be replaced by a suitable broadband solid state
amplifier
using an appropriate number of power RF transistors to get to the required
power, and
advantageously can operate in both the UHF and VHF bands.
In an exemplary embodiment, the TH-680 can provide 104 kW of peak envelope
power, which may (as a non-limiting example) include the following allocation
of power
levels. To reduce interference with channels outside the adjacent-channel
pair, a suitable
two-channel-wide (12 MHz in the U.S.) band pass filter (BPF) 1146 is provided
at the
output to the amplifier 1145. To comply with broadcast power standards, the
amplifier
1145 must amplify the combined NTSC+DTV signal so that the BPF provides a
signal
25 kW average peak-of sync power (NTSC), 1.25 kW NTSC average aural power, and
2.5 kW average DTV power. Of course, variation of the above particulars in
accordance
with commonly-known principles lies within the ability of those skilled in the
art.
As is readily appreciated by those skilled in the art, such an amplifier
involves a
tube that performs the power amplification, as well as a resonator cavity that
limits the
frequency range in which the tube amplifies signals. For any assigned adjacent-
channel
pair (either Channel N-1 through N, or Channel N through N+1), one skilled in
the art,
upon reading this specification, is readily capable, without undue
experimentation, of
implementing a properly-tuned amplifier using a suitable tetrode-class device
and
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resonator cavity. The implementation is different for each adjacent-channel
pair, but the
design principles remain the same regardless of the particular assignment, and
further
details need not be provided here to illustrate the implementation and
operation of the
invention.
For many applications, high-power amplifier 1145 comprises a tetrode-class
device, especially a Thomson TH-680 diacrode, available from Thomson Tubes
Electronique. It is to be understood that the scope of the invention should
not be limited
to a particular component or to a specific set of signal types.
FIG. 1(a) emphasizes an implementation of the power level AGC feedback
arrangements that ensure that output power levels are maintained substantially
constant.
In FIG. 1(a), feedback paths 276 and 376 are shown leading from the broadcast
signal
output by the band pass filter 1146, back to respective IF-to-broadcast-
frequency
converters 240 and 340. Paths 276 and 376 are provided in lieu of paths (not
shown)
from amplifiers 270, 370, respectively. NTSC channel bandpass filter 277, and
DTV
channel band pass filter 377, are provided in feedback paths 276, 376,
respectively, so
that only in-channel frequency components are returned to converters 240, 340.
The feedback arrangements operate on similar principles of feedback control,
known to those skilled in the art. When average power varies from a desired
steady-state
power level, either at the outputs of driver amplifiers 270, 370 or at the
output of BPF
1146, feedback arrangements within converters 240, 340 act to correct the
variation to
return the power level at the sensed point back to the desired steady-state
power level.
The gain factor that converters 240, 340 apply to the feedback signals on
paths 275, 375
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or 276, 376 are different, and are determined by the differences in magnitude
of power
between the outputs of driver amplifiers 270, 370 and of BPF 1146. However,
the
principles remain the same.
Gain correction achieved locally (within the respective NTSC and DTV paths)
compensates only for variations that occur through amplification paths 260,
270 and 360,
370. However, the implementation shown in FIG. 1 (a) achieves a more
comprehensive
gain correction over the entire path between the IF-to-broadcast-frequency
converters
240, 340 and the ultimate output of BPF 1146.
Those skilled in the art will recognize that, although the disclosed
embodiment
is designed especially for pre-compensating group delay distortion in
broadband,
noise-like, DTV signals, the invention is equally applicable to pre-
compensating group
delay distortion in analog (e.g., NTSC) signals that essentially constitute a
set of carrier
signals at discrete frequencies. Accordingly, an embodiment with the structure
of FIG. 2
may be used to implement part of element 230 (FIG.1(a)) as well as element 330
(FIG. 1 (a)).
Moreover, the embodiment with the structure of FIG. 2 may be used in
transmitters that transmit only analog (e.g., NTSC) or only digital television
(DTV)
signals, and not both. In particular, FIG. 1 (b) shows that the inventive
equalization
scheme may be used in element 230 of a purely analog-format transmitter, and
FIG. 1 (c)
shows that the inventive equalization scheme may be used in element 330 of a
purely
digital-format transmitter. The elements in FIGS. 1 (b) and 1 (c) generally
perform the
same functions as like-numbered elements in FIG. 1(a). However, the pass band
of the
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band pass filters 1146 are only one channel wide in FIGS. 1 (b) and 1 (c), in
contrast with
the two-channel-wide BPF in FIG. 1 (a).
Thus, it is to be understood that the invention has utility in pre-
compensating
group delay distortion in either or both analog (e.g., NTSC) format signals
and digital
format (DTV) signals.
Referring now to FIG. 2, a preferred embodiment of the group delay
equalization
circuit according to the present invention is illustrated.
An intermediate-frequency (IF) television signal is input to a first mixer
10A,
which is also driven by a 51 MHz oscillator signal from an oscillator 60A.
First mixer
l0A drives a series combination of a low pass filter (LPF) 20A and an
equalizer 30A.
Equalizer 30A drives a second mixer 40A, which is also driven by the 51 MHz
oscillator
signal from oscillator 60A. Second mixer 40A drives a band pass filter (BPF)
SOA.
Elements 10A, 20A, 30A, 40A, SOA and 60A may be considered to be a first
stage of the equalization arrangement. A topologically identical and
functionally similar
second stage, including elements IOB, 20B, 30B, 40B, SOB and 60B, receives the
output
of BPF SOA. BPF SOB provides the output of the equalization circuit.
In operation, the unequalized IF TV signal situated in the 41-47 MHz band is
input to mixer 10A. The unequalized signal does not have the desired pre-
compensation
for the group delay distortion that will be introduced by RF BPF 1146 (FIG. 1
(a)).
Accordingly, the undesirable group delay characteristics inherent in the
unequalized IF
signal may be represented by the delay-versus-frequency graph of FIG. 3(a).
The "Filter Passband" in FIG. 3(a) relates to the bandwidth of the RF BPF 1146
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(FIG. 1 (a)). In the FIG. 1 (a) embodiment, the pass band is 12 MHz wide, to
compensate
for the side-by-side NTSC and DTV channel signals. However, in transmitters in
which
only a single channel is transmitted (such as FIGS. I (b) and 1 (c)), the pass
band is 6 MHz
wide. (In many countries outside the U.S., these figures would be 14 MHz and 7
MHz,
or 16 MHz and 8 MHz, respectively.) In either event, a single channel at a
time (in this
example, 6 MHz wide) is equalized, based on the IF (41-47 MHz) signal.
If more than one signal is transmitted at a time (as is the case in the FIG. 1
(a)
transmitter), then the two pre-compensations that occur in IF processors 230,
330 are
reflected in the "combined" signal that passes through BPF 1146. (The
"combined"
signal has an analog and a digital signal that are side-by-side in the RF
frequency
spectrum.)
Referring again to FIG. 2, mixer l0A translates the frequency range of the
41-47 MHz IF signal down to the 4-10 MHz range because of the mixing that
occurs with
the 51 MHz oscillator signal from oscillator 60A. LPF 20A, which is preferably
an
approximately 15 MHz LPF, passes the 4-10 MHz signal to equalizer 30A without
introducing additional group delay. LPF 20A also rejects the oscillator signal
and the
mixing "sum" signal that occupies the 92-98 MHz band.
Equalizer 30A may be implemented as a S to 10-section, all-pole equalizer with
a designed operating range of 4-7 MHz. This is purposely chosen as the lower
half of the
4-10 MHz range occupied by the signal that is input to the equalizer. The
lower 4-7 MHz
range is preferred because it only requires equalization stages with Qs of
less than 10,
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which are practically implementable with capacitors and inductors having
unloaded Qs
of less than 100.
In contrast, for signal components between 7 MHz and 10 MHz (the higher half
of the 4-10 MHz range), section Qs approach 50,000, rendering equalization in
the 7-10
S MHz band impractical. In any event, the equalizer 30A outputs a signal whose
pre-compensation for group delay is represented in FIG. 3(b). FIG. 3(b)
illustrates how
the group delay distortion in the lower (4-7 MHz) half of the band has been
pre-compensated.
Second mixer 40A (FIG. 2) translates the signal represented in FIG. 3(b) back
up
to IF (41-47 MHz). Using the same oscillator for both down-conversion and
up-conversion ensures that precise frequency and phase information is not
changed.
A band pass filter SOA (which may pass, for example, 39-50 MHz) passes the
re-translated signal that is in the 41-47 MHz range without introducing
additional group
delay. FIG. 3(c) illustrates how the group delay distortion in the 44-47 MHz
band (which
corresponds to the 4-7 MHz band before re-translation) has been pre-
compensated in the
signal output by BPF SOA.
To summarize, elements 10A, 20A, 30A, 40A, SOA, and 60A have
pre-compensated for group delay distortion to be expected in the upper half
(44-47 MHz)
of the 41-47 MHz signal. The group delay distortion to be expected in the
lower half
(41-44 MHz) of the 41-47 Hz signal must still be pre-compensated. This
pre-compensation is achieved by elements IOB, 20B, 30B, 40B, SOB and 60B.
Elements IOB, 20B, 30B, 40B and SOB function in substantially the same way as
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corresponding elements 10A, 20A, 30A, 40A, and 50A, respectively. However, the
frequency of oscillator 60B is 37 MHz, rather than the 51 MHz frequency of
oscillator
60A. Significantly, the oscillator signals are located the same frequency
difference away
from the closest edge of the 41-47 MHz band of the original IF signal. The
choice of a
37 MHz oscillator frequency causes a translation of the un-pre-compensated 41-
44 MHz
band down to the 4-7 MHz range. This frequency translation is achieved by
third mixer
10B.
Equalizer 30B equalizes the signal in the 4-7 MHz range that is received from
LPF 20B. FIG. 3(d) shows the pre-compensation that is inherent in the signal
output by
equalizer 30B. At this point, both the upper and lower halves of the original
IF signal
have been pre-compensated by equalization below 7 MHz.
The equalized signal provided by equalizer 30B is re-translated up to the 41-
47
MHz IF signal band by a fourth mixer 40B, which is driven by the 37 MHz signal
from
second oscillator 60B. A suitable band pass filter 50B passes the equalized 41-
47 MHz
signal without adding group delay.
The signal provided by BPF 50B may be considered the output of the
equalization
arrangement as a whole. The group delay pre-compensation inherent in this
signal is
illustrated in FIG. 3(e). When this signal is frequency-translated and
amplified in the
transmitter (such as FIG. 1(a)), it retains the pre-compensation needed to
compensate for
the group delay distortion that is introduced by the transmitter's output band
pass filter
(such as element 147 in FIG. 1 (a)).
Although the invention has been described with reference to a 6 MHz DTV
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television signal occupying, at IF, a 41-47 MHz band, that is twice down-
converted to
the 4-10 MHz range for purposes of pre-compensating for group delay distortion
introduced at RF by a transmitter's output band pass filter, it is understood
that the
invention is not to be limited to these types of signals, these particular
frequency ranges,
this number of consecutive down-conversions, or this source of group delay
distortion.
Rather, the invention may be adapted to a variety of signal types, frequency
ranges,
number of consecutive down-conversions, and distortion sources. Thus,
modifications
and variations of the above-described embodiments of the present invention are
possible,
as appreciated by those skilled in the art in light of the above teachings. It
is therefore
to be understood that, within the scope of the appended claims and their
equivalents, the
invention may be practiced otherwise than as specifically described.
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