Note: Descriptions are shown in the official language in which they were submitted.
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RECEIVER IMPROVEMENT FOR
EXPANDED INFORMATION CAPACITY FOR
EXISTING COMMUNICATION
TRANSMISSIONS SYSTEMS
BACKGROUND OF THE INVENTION
This application claims priority to a U.S. Serial No. 60/171,384 filed
December 22, 1999, entitled, "Receiver Improvement for Expanded Data
Capacity,"
which is incorporated herein by this reference.
~o Field of the Invention
The present invention relates to communications systems and methods for
transmitting additional information via television transmissions and other
transmissions. Although the invention applies to a wide range of communication
transmissions, this disclosure focuses, although not in a limiting way,
largely on the
applicability to television.
Background
The standard method for over-the-air transmission of television signals in the
United States in called NTSC. This is an analog system in which the picture is
transmitted in a vestigial sideband modulation format on the visual carrier
and the
2o sound component transmitted as frequency modulation on a separate sound
carrier.
Relatively recent developments in the television industries have focused upon
the
transmission of High Definition Television (HDTV) that requires a substantial
increase in transmitted information and hence greatly expands the required
baseband
video signal bandwidth. Great progress has been made in the area of digital TV
bandwidth compression so that one or more HDTV signals can be conveyed in the
standard television bandwidth of 6 MHz. These HDTV developments have, by a
combination of techniques, substantially reduced the bandwidth required for
fully
digital transmission of the video information. Subsequently, it was determined
that
the same compression techniques can be used to put multiple "Standard
Definition
3o Television", SDTV, signals in the same bandwidth. Both of these techniques,
HDTV
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and SDTV, are generically called Digital Television, DTV. A further move from
the
original HDTV goal is the allowance by the FCC of data transmission instead of
television transmission.
The development of DTV and its acceptance as a future broadcast standard has
also led to the requirement for a transition period between broadcasting the
present
analog TV to that of compressed DTV. Since the transmission of standard analog
NTSC will remain for many years before the complete transition to DTV, the
availability of a technique allowing simultaneous, non-interfering
transmission of
digital signals within the same channel as an analog NTSC signal would result
in a
to two (or more) channel capacity increase in the existing broadcast frequency
assignments. Alternatively, data transmission can be accommodated in addition
to or
instead of DTV along with the analog NTSC signal. As more efficient means of
bandwidth compression emerge, the transmission of HDTV si~rultaneously with
analog NTSC in the same 6 MHz bandwidth is an attractive possibility.
While the focus of this work has been on NTSC television signals, it applies
equally well to any vestigial sideband signal transmission system.
TECHNICAL CONSIDERATIONS
The standard NTSC format allocates 6 MHz of spectrum to the transmission of
the combined video and audio signals. The visual carrier is placed 1.25 MHz
above
2o the lower band edge and the aural carrier 5.75 MHz above the lower band
edge. The
visual information is impressed on the visual carrier using a vestigial
sideband
amplitude modulation (AM) technique so that the frequency components below the
visual carrier occupy no more than the 1.25 MHz of the available spectral
assignment,
while the frequency range allocated to the visual information extends for
approximately 4.2 MHz above the visual carrier.
The color information is carried on the color subcarrier of the main visual
carrier at approximately 3.58 MHz above the visual carrier (4.83 MHz above the
lower band edge). The modulation of the color information is both in-phase and
quadrature and contains more lower sideband components than upper.
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The aural information is carried on the separate aural carrier at 4.5 MHz
above
the visual carrier (5.75 MHz above the lower band edge) and is frequency
modulated
(FM) with a peak deviation of 25 KHz over the range of audio frequencies
extending
to somewhat above 15 KHz.
It should be noted that I) the amplitude modulation of the visual luminance
information is solely an in-phase variation with no quadrature component prior
to the
vestigial filter while, 2) the aural subearrier is solely frequency modulated.
Subcarriers are sometimes added to the aural carrier but they too are
frequency
modulated on the main aural carrier. It can be seen then that not all of the
information
carrying capacity in the 6 MHz analog channel is occupied. There are no
quadrature
components in the region close to the visual carrier and no amplitude
modulation
components in the aural carrier region.
Other opportunities exist which make possible the transmission of additional
information within the normal TV spectral assignments. One company called
WavePhore, Inc. utilizes a signal which puts single sideband phase shift data
in the
area of approximately 3.9 to 4.2 MHz above the visual carrier (5.15 MHz to
5.45 MHz
above the lower band edge) and is capable of transmitting in the order of 500
Kbits /
second while causing only minor interference to the analog television signal.
This
system is covered under U.S. patent(s).
2o Another approach of transmitting data within the NTSC broadcast format has
been developed by Digideck, Inc. This technique is called the D-Channel and
operates at a reduced level in the lower frequency portion of the video
vestigial
sideband is capable of transmission of something in the order of 750 Kbits per
second
with only minor interference to the analog television signal and is covered
under U.S.
z5 patent(s).
In order to transmit a digitally compressed NTSC signal, a data transmission
rate in the order of 1 to 5 megabits per second will be required when
employing
present day compression techniques. The above mentioned prior art systems are
incapable of these high data rates without causing significant and unacceptab~
3o interference to the analog NTSC signal.
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The Federal Communications Commission (FCC) has authorized the use of
the prior art systems for transmitting digital data in analog television
systems and
invited other inventors to come forth with improved systems for this type of
data
transmission in Report & Order (R & O), "Digital Data Transmission Within the
Video Portion of Television Broadcast Station Transmissions", in MM docket No.
95-
42. Consistent with that invitation, the FCC has granted the inventors an
experimental license on September 23, 1999 to transmit its signal on Channel
62 in
Scottsdale Arizona.
Further background information is found in U.S. Patent application 09/062225
to filed April 17, 1998 Still further background information is found in the
book:
"Modern Cable Television Technology: Video, Voice, and Data Communications" by
Walter Ciciora, James Farmer, David Large, published December 1998, Morgan
Kaufmann Publishers, (Web Page: http://www.mkp.com) 912 pages; ISBN 1-55860-
416-2, which is incorporated herein by this reference.
is The Need for an Improvement in the Receiver for Expanded Information
Capacity for Existing Communication Transmission Systems.
The interference visible on non-ideal television receivers due to data
transmission using the invention of U.S. Patent application 09/062225 filed
April 17,
1998 are governed by several factors including:
20 1. The desired coverage area in broadcast applications.
2. The number of data levels.
3. The performance of the data receiver.
4. The injection level of the data signal.
5. The amount of abatement signal applied.
25 6. The acceptable level of video errors.
7. The acceptable level of data errors.
All of these factors interact. Most importantly, the better the performance of
the data receiver, the less data signal injection is required for a given data
rate and a
given coverage area. This minimizes the visibility of the interference of data
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transmission as experienced by non-ideal television receivers. Conversely,
other
trade-offs can be made. For example, for a given amount of visible
interference, a
better performing data receiver will allow either greater coverage area or
other
desirable benefits.
While U.S. Patent application 09/062225 filed April 17, 1998 focuses on, but
is not necessarily limited to, the double sideband region of the NTSC signal,
the
present invention facilitates improved performance when going beyond the
double
sideband region of the NTSC signal. This allows greater data capacity.
Increased
data capacity is a clear benefit satisfying an important need.
to There are certain requirements for the filters in the receivers disclosed
in U.S.
Patent application 09/062,225 filed April 17, 1998. The improvements of the
present
invention relax the need for higher performance filters in the data receiver
of the
invention of U.S. Patent application 09/062.225 filed April 17. 1998.
There is always a need for more cost-effective implementation of the data
receivers such as in the invention of U.S. Patent application 09/062.225 filed
April 17,
1998. Modern mixed signal techniques for designing integrated circuits such as
those
developed by Microtune Incorporated (http://www.microtune.com) and others are
especially suitable for cost-effective implementations. The present invention
facilitates using those techniques for the implementation of a data receiver
for the
2o invention of U.S. Patent application 09/062,225 filed April 17, 1998.
While the focus of this invention is on the data receiver, the invention is
applicable to the improved reception of analog television signals as well. The
reception of VSB signals for television display involves significant
compromises.
This invention overcomes many of those compromises.
2s SUMMARY OF THE INVENTION
Systems according to U.S. Patent application 09/062,225 filed April 17, 1998
enable transmission of data (including compressed digital video), in the range
of
several megabits / second by techniques involving amplitude modulation of the
aural
carrier and quadrature or phase modulation of the visual carrier. Although the
3o discussion of the specification of U.S. Patent application 09/062.225 filed
April 17,
1998 and the disclosure of this invention focuses on the NTSC-M television
standard,
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similar concepts apply to analog television systems including but not limi~d
to the
multiple versions of PAL, other versions of NTSC, and SECAM and other
vestigial
sideband systems and frequency modulation systems.
The present invention improves the television receiver and more specifically
the data receiver of the visual portion of the signal of U.S. Patent
application
09/062,225 filed April 17, 1998. While the reception of the aural portion of
the signal
of U.S. Patent application 09/062,225 filed April 17, 1998 is not further
discussed
here, it remains an important part of the total system according to a
preferred
embodiment. It is not further discussed here because the present invention
does not
to change its implementations.
The invention of U.S. Patent application 09/062,225 filed April 17, 1998
includes a Compensator Subsystem which adjusts the transmitted data spectrum
so
that when passed through the Nyquist filter of an existing television
receiver, the data
spectrum will become symmetrical about the television signal's visual carrier.
In the
~5 double sideband region of the spectrum after the television receiver's
Nyquist filter,
the data signal will be in quadrature with the visual signal to the extent
that the
Compensator Subsystem accurately compensates for the effects of the Nyquist
filter.
The visual signal will have both in-phase and quadrature components because
the
television receiver's Nyquist filter has made the visual spectrum
unsymmetrical about
2o its carrier. An ideal synchronous detector that is phase locked to the
visual carrier in
the television receiver will only respond to the in-phase components of the
visual
signal and will thus ignore the quadrature data signal as well as the
quadrature
components of the visual signal.
The data receiver of the invention of U.S. Patent application 09/062,225 filed
25 April 17, 1998 should ideally perform in a complementary manner. That is,
the data
receiver will not have a Nyquist filter. Thus the visual spectrum in the
double
sideband region will remain symmetrical about its carrier and thus will not
have any
quadrature components. The data spectrum, however, will have both in-phase and
quadrature components due to the action of the Compensator Subsystem in the
data
3o transmitter. An ideal synchronous detector that is phase locked to ninety
degrees
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relative to the visual carrier would respond only to quadrature data
components and
ignore the visual signal in the double sideband region which is completely in-
phase
with the visual carrier as well as the in-phase components of the data signal.
There are at least two practical problems in implementing the data receiver.
First, the visual signal has quadrature components outside the double sideband
region.
Consequently, the data signal demodulated by the data detector will be
contaminated
with the quadrature components of the visual signal. In one approach to
minimize
this, the quadrature components must be strongly attenuated by a filter to
prevent their
detection by the data synchronous detector. Ideally, this filtering would
eliminate the
to visual quadrature components. But practical filters will not accomplish
this. Such
detection of the quadrature components of the visual signal by the data
detector would
result in ''eye closure" of the data signal. As is appreciated by those of
ordinary skill
in the data detection arts, this will reduce the overall system performance
margin and
lead to data detection difficulties. This filter must remove as much of the
visual
t5 quadrature components as possible while not unduly damaging the amplitude
or phase
of the data components. This is a severe constraint. The invention of U.S.
Patent
application 09/062,225 filed April 17, 1998 proposes an alternative
feedforward
approach which has some benefits, but still leaves a need for better
techniques.
Secondly, if the network transfer function experienced by the data signal or
the visual
zo signal has phase and amplitude characteristics that upset the quadrature
relationships,
further contamination of the data signal by the visual signal and distortion
of the
detected data signal will result. This reduces the performance of the data
receiver and
forces undesirable trade-offs in total system outcomes.
The present invention alleviates these problems by synthesizing a double side
25 band signal in such a manner that both the visual signal and the d~a signal
are each
symmetrical with respect to their carriers in both amplitude and phase. This
allows
synchronous detection techniques to be applied to cleanly separate the data
signal
from the visual signal.
The double sideband synthesis is accomplished by reversing the received
3o spectrum and adding it back to itself. Reversing the received spectrum
interchanges
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the upper sideband and the lower sideband in a manner that causes the lowest
frequency of the lower sideband in the received spectrum to become the highest
frequency of the upper sideband in the reversed spectrum. Likewise, the
highest
frequency of the upper sideband in the received spectrum will become the
lowest
frequency of the lower sideband in the reversed spectrum. One method of
implementing spectrum reversal is presented in the preferred embodiment. Other
methods of spectrum reversal can be implanted by those of ordinary skill in
these arts.
While the main focus of this invention is the improved recovery of data
signals, the invention has benefits for television receiver design even for
signals that
to do not include data. Also, this invention will improve the reception of
data signals
that are phase modulated onto the visual carrier.
The invention will be described with the aid of the attached figures.
BRIEF DESCRIPTION OF THE DRAWINGS
FIGURE 1 a is a graph of a television signal spectrum normalized to 0.0 Hz.
FIGURE 1b is a graph of the output of a flat response television receiver.
FIGURE 1 c is a graph of an idealized and typical TV receiver response curve.
Figure 2a is a graph of Data & Video Modulator Signals.
Figure 2b is a graph of Data Receiver Signals.
Figure 2c is a graph of television Receiver Signals.
2o Figure 2d is a block diagram of Data Detection receiver using a Filter. The
receiver is subject to "Rude Video" interference.
Figure 3a is a graph of the Data Spectrum subjected to Data RF Tx Filter.
Figure 3b is a graph of the Data Spectrum subjected to TV Receiver's Nyquist
Filter. This yields a Q component only (note: in the receiver the spectrum is
reversed
at IF).
Figure 4a is a block diagram of the Data Demodulator and an optional NTSC
demodulator using the present Synthesis of a Double Sideband Signal invention.
Figure 4b is a graph of the NTSC spectrum and the Data Spectrum after the
tuner and before Nyquist filter in a conventional television receiver. (The
spectra is
3o shown "as is'' in the receiver; i.e. opposite to at RF). The two spectra
are shown
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separately for purposes of illustration. In the actual system, the two spectra
are
combined in a manner that does not allow them to be viewed separately.
Figure 4c is a graph of the TV Receiver's Nyquist filter characteristic.
Figure 4d is a graph of the NTSC spectrum and the Data Spectrum after the
tuner, Nyquist filter, and Precision Phase~Correct Delay.
Figure 4e is a graph of the NTSC spectrum and the Data Spectrum at output of
Spectrum Reverser.
Figure 4f is a graph of the NTSC spectrum and the Data Spectrum at output of
Summer.
Figure 5a block diagram of the Spectrum Reverser.
Figure Sb is a graph of the IF spectrum as seen at IF frequencies (the reverse
is
seen at RF frequencies.)
Figure Sc is a graph of the spectrum of Figure Sb heterodyned by a cosine
wave with N = 3 and band pass filtered to select the desired sideband.
t5 Figure 5d is a graph of the spectrum of Figure Sc heterodyned by a cosine
wave with N = 3 {(N + 2) = 5} and band pass filtered to select the desired
sideband,
Figure 6a is a block diagram of a Precision Phase-Correct Delay Example
using exactly the same filters as used in the Spectrum Reverser.
Figure 6b is a graph of the IF spectrum as seen at IF frequencies (the reverse
is
2o seen at RF frequencies.)
Figure 6c is a graph of the spectrum of Figure 6b heterodyned by a cosine
wave with M = 3 and band pass filtered to select the desired sideband.
Figure 6d is a graph of the spectrum of Figure 6c heterodyned by a cosine
wave with M = 5 and band pass filtered to select the desired sideband.
25 Figure Se is a block diagram of the Spectrum Reverser Without filters.
Figure 5f is a graph of the IF spectrum as seen at IF frequencies (the reverse
is
seen at RF frequencies.)
Figure Sg is a graph of the spectrum of Figure Sf heterodyned by a cosine
wave with N = 3
3o Figure Sh is a graph of the spectrum of Figure Sg heterodyned by a cosine
wavewithN=3 {(N+2)=5}.
Figure 6e Precision Phase-Correct Delay Without Filters.
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Figure 6f is a graph of the IF spectrum as seen at IF frequencies (the reverse
is
seen at RF frequencies.)
Figure 6g is a graph of the spectrum of Figure 6f heterodyned by a cosine
wave with M = 5
Figure 6h is a graph of the spectrum of Figure 6g heterodyned by a cosine
wave with M = S.
Figure 6i is a graph of the spectrum resulting from adding the spectra of Figs
~h and 6h.
Figure 6j is a graph of the spectrum of a cosine wave at the IF frequency.
to Figure 6k is a graph of the spectrum of Figure 6i heterodyned by a cosine
wave at the IF frequency and a low pass filter to receive the baseband signal.
Figure 7a is a graph of the television spectrum and the data spectrum at the
output of a Nyquist Filter and a Precision Phase-Correct Delay as see at IF
(at RF,
spectrum is reversed).
Figure 7b is a graph of the television spectrum and the data spectrum at the
output of the Spectrum Reverser.
Figure 7c is a graph of the sum of the television spectrum and the data
spectrum found at the output of Precision Phase-Correct Delay and at the
output of the
Spectrum Reverser.
2o Figure 8a is a block diagram of the Data Demodulator and an optional NTSC
demodulator using the present Double Sideband Synthesis invention but without
a
Nyquist Filter. This Figure is similar to Figure 4a.
Figure 8b is a graph of the television spectrum and the data spectrum at the
output of a Precision Phase-Correct Delay as see at IF (at RF, spectrum is
reversed)
but without first passing through a Nyquist filter.
Figure 8c is a graph of the television spectrum and the data spectrum at the
output of the Spectrum Reverser, but without first passing through a Nyquist
filter.
Figure 8d is a graph of the sum of the television spectrum and the data
spectrum found at the output of Precision Phas~Correct Delay and at the output
of the
3o Spectrum Reverser, but without first passing through a Nyquist filter.
Figure 9a is a graph of the Data Spectrum after the tuner and Precision Phase-
Correct Delay but without first passing through a Nyquist filter. Note: This
spectrum
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has both Q components and I because it is not symmetrical about the carrier.
(Spectra
shown ''as is'' in the receiver; i.e. opposite to at RF).
Figure 9b is a graph of the Data Spectrum at output of Spectrum Reverser, but
without first passing through a Nyquist filter.
Figure 9c is a graph of the Data Spectrum at output of the Summer. This
spectrum has only a Q component because it has been made symmetrical about the
carrier frequency. The delayed and reversed data spectra prior to summation
are
shown to illustrate that they add to the summer output spectra.
Figure I Oa is a block diagram of the Data Demodulator of the receiver using
to the present Double Sideband Synthesis invention. Note that this figure is
identical to
Figure 8a.
Figure lOb is a graph of the television spectrum and an expanded bandwidth
data spectrum at the output of a Precision Phase-Correct Delay as see at IF
(at RF,
spectrum is reversed) but without first passing through a Nyquist filter.
t5 Figure lOc is a graph of the television spectrum and the expanded bandwidth
data spectrum at the output of the Spectrum Reverser, but without first
passing
through a Nyquist filter.
Figure I Od is a graph of the sum of the television spectrum and the expanded
bandwidth data spectrum found at the output of Precision Phas~Correct Delay
and at
2o the output of the Spectrum Reverser, but without first passing through a
Nyquist filter.
Figure 11 a is a graph of the amplitude transfer function of the Data RF Tx
Filter found at the transmitter.
Figure 11 b is a graph of the amplitude transfer function of the NTSC
Vestigial
Sideband Filter found at the transmitter.
25 Figure l lc is a graph of the amplitude transfer function of the Composite
Filtering applied to Data at Transmitter found by combining the amplitude
transfer
functions of Figure 1 la and Figure I Ib.
Figure 11 d is a graph of an expanded bandwidth Data Spectrum extending to
1.25 MHz and the Composite Filter function of Figure 11 c along with the
resulting
3o non symmetric Filtered Data Spectrum.
Figure 12a is a graph of the application of the present Double Sideband
Synthesis invention to receive the expanded bandwidth Data Spectrum of Figure
I 1d.
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The present invention yields perfect reconstruction of the data spectrum
between +
0.75 MHz and modest distortion between + 0.75 MHz and +1.25 MHz. This modest
distortion could be pre-distorted at the transmitter.
Figure 12b is a graph of the expanded data spectrum after the television
receiver's Nyquist Filter. In the television receiver, the expanded bandwidth
Data
spectrum has been made symmetrical and has been reduced in bandwidth to the
double sideband bandwidth of the NTSC signal. The data spectrum is all in
quadrature with the visual signal and a Synchronous detector in the television
receiver
will be blind to this data signal.
~o DETAILED DESCRIPTION
Reference will now be made in detail to embodiments of the invention,
examples of which are illustrated in the accompanying drawings. Wherever
possible,
the same reference numbers or numbers with the same trailing digits will be
used
throughout the drawings to refer to the same or like parts.
t 5 Figure 1 a describes the standard analog NTSC television signal comprising
a
Video Carrier 102 that is Amplitude Modulated (AM), a Sound Carrier 103 that
is
Frequency Modulated (FM) and a Color Carrier 105, which is modulated with an
in-
phase and a quadrature phase components. The lower sideband of the television
signal is unattenuated to a frequency 750 KHz below the visual carrier. The
signal is
2o severely attenuated at frequencies more than 1.25 MHz below the visual
carrier. The
signal is rolled off between 750 KHz and 1.25 MHz below the visual carrier in
a
manner that can be realized with practical filters. This filtering results in
a vestige of
the lower sideband leading to the term "Vestigial Side Band" (VSB) modulation.
The
VSB technique was motivated by the desire to provide increased picture
resolution
25 while remaining within the allocated 6 MHz frequency band for a television
channel.
The details of the VSB implementation were governed by practical consideration
for
the economic realization of the required filters.
Figure 1b illustrates the difficulty this VSB technique causes in the
demodulation process. Because the region 104 of Figure 1 a between 750 KHz
below
3o the visual carrier and 750 KHz above the visual carrier is ordinary double
sideband
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modulation and the region 106 in Figure la between 1.25 above the visual
carrier and
4.0 MHz above the visual carrier is single sideband modulation, the
demodulated
signal will have twice the strength in the baseband frequencies which are less
than 750
KHz compared to the signals in the baseband frequencies between 1.25 MHz and
4.0
MHz.
Figure lc illustrates the solution adopted in analog television receivers to
deal
with this problem. Before demodulation, the signal is weighted with a filter
that is
antisymmetric about the visual carrier and results in half strength passed
through the
filter at the visual carrier frequency 102. This normalizes the strength of
the
to demodulated signal from zero baseband frequency to its highest frequency.
This type
of filter, with an antisymmetric characteristic, is called a Nyquist filter.
Figure 2a describes the data and video modulator signals at the signal
origination point. Figure 2b Illustrates the processing of the signals in a
data receiver
of the invention of U.S. Patent application 09/062225 filed April 17, 1998
while
Figure 2c shows the processing of these signals in an ordinary television
receiver.
It is important to recognize at this point that a conventional double sideband
signal has only in-phase components. It has no quadrature components. Also, a
signal which is not a conventional double sideband structure has both in-phase
and
quadrature components. As a consequence, the visual signal of Figure la
between
750 KHz below the visual carrier and 750 KHz above the visual carrier is a
conventional double sideband signal and has only in-phase components. The
signal
with frequencies higher than 750 KHz above the visual carrier is not a
conventional
double sideband signal and thus has both in-phase and quadrature components.
This
is illustrated in the center section of Figure 2a. The left part of the top
section of
Figure 2a shows the VSB filter characteristic 202 that converts a conventional
double
sideband television signal into the VSB signal 204 of the center section of
Figure 2a.
The well-known process of synchronous demodulation wherein a signal is
multiplied by a cosine wave of the same phase and frequency as the signal's
carrier
responds only to the in-phase components of the signal. If that same signal
were
3o multiplied by a ninety degree phase-shifted cosine wave (which becomes a
sine
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wave), then only the quadrature components would be demodulated. In this
manner,
in-phase and quadrature components can be completely separated from each
other.
The top section of Figure 2c shows the television receiver's Nyquist filter
characteristic 206 and the center section of Figure 2c shows the resulting
television
signal spectrum 208 after passing through the television receiver's Nyquist
filter 206.
All portions of this spectrum have in-phase and quadrature components because
no
portion of this spectrum is symmetric about the can ier.
A more detailed discussion of conventional analog and digital television
signals is contained Chapters 2 and 3 in one of the inventor's texts "Modern
Cable
to Television Technology, Video, Voice, and Data Communications", Ciciora,
Farmer &
Large, 1999 Morgan Kaufmann Publishers, ISBN 1-55860-416 2.
The left portion of the bottom section of Figure 2a shows a data signal 210
double sideband modulated onto a quadrature phase shifted visual carrier 212
and the
quadrature phase shifted carrier then completely suppressed. This data signal
210 has
is only quadrature components; no components ofthis data signal exist in-phase
with the
television signal's visual carrier 214. If this signal was to pass through the
Nyquist
filter 208 of the center section of Figure 2c, its double sideband nature
would be
destroyed resulting in the creation of components in-phase with the visual
carrier 214
of the television signal. These components would become visible as
interference on
2o the television screen. This is undesirable and is likely to be unacceptable
unless the
interference is reduced to such a low level as to become invisible to viewers
under
normal circumstances. To prevent this difficulty, the data signal is
predistorted in a
manner such that passage through the receiver's Nyquist filter 206 will
convert it back
into a conventional double sideband signal 216 with only quadrature
components.
25 This signal will then be ignored by the television receiver's synchronous
demodulator.
This is accomplished with the Data Radio Frequency Transmit Filter 218 (Data
RF Tx
Filter) with characteristic described in the right hand side of the top
segment of Figure
2a. This filter characteristic 218 has the shape of the Nyquist filter used in
the
receiver 206 but with the frequencies reversed. This causes the transmitted
data signal
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spectrum 220 to appear as in the bottom section of Figure 2a. That spectrum
has both
in-phase and quadrature components.
The data demodulation process of the prior art is shown in Figure 2b.
Synchronous demodulation is employed to separate the quadrature data signal
from
the in-phase visual signal. However, the visual signal is conventional double
sideband only in the range between 750 KHz below the visual carrier and 750
KHz
above the visual carrier. Outside that range, the signal will have components
222 in
quadrature which will be demodulated by the data signal synchronous
demodulator
224 and will interfere with the detection of data; i.e. the discrimination of
the discrete
to levels of the data signal at baseband after demodulation.
Even if the data demodulation filter 226 has perfect symmetry, the visual
signal will have undergone the VSB filtering 202 process at the transmitter on
one
side of its spectrum but not the other. Consequently, the two sides of the
data
spectrum are not perfectly symmetrical and therefore a quadrature component
has
is been created. This quadrature component has been called "rude video" 228
because it
interferes with the detection of the data signal after demodulation. Since the
visual
signal 204 is much stronger than the data signal 210, relatively small
asymmetries in
the composite filter characteristics will result in "rude video" and lead to
problems in
detecting the data.
2o Note that if the filtering process were to be made completely symmetrical
by
the use of a reversed VSB filter in the data receiver, the video would have
only
components that are in-phase with the visual carrier and would not be
detected. Since
the VSB filter 202 is not precisely specified, there would be some difficulty
in
choosing a proper reversed VSB filter design. The relative strength of the
visual
25 signal relative to the data signal makes the precision of this choice
important. Note
also that since in the television receiver, the data signal 216 is of much
lower strength
than the visual signal 208, the degree of precision in matching the
transmitting data
filter 218 (which ideally is the reverse of the receiver's Nyquist filter 206)
is much
less critical. The situation further benefits from the fact that the data
signal 210 is
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uncorrelated to the visual signal 204 and imperfections will appear as noise
rather
than as some annoying pattern or image.
Figure 3a shows a data double sideband spectrum 302 (of raised cosine shape,
chosen for illustrative purposes only and not as a limitation). A data
transmit filter
characteristic 304 shape is shown (as a linear filter, chosen for illustrative
purposes
only and not as a limitation). The resulting transmitted data signal 310 has
been made
unsymmetrical and therefore will have both in-phase and quadrature components.
Figure 3b shows the transmitted data signal 308 passing through the television
receiver's Nyquist filter 306 and being converted once again into a
symmetrical
to spectrum shape 312 that will have only spectral components that are in
quadrature to
the visual signal. Thus a synchronous demodulator in the television receiver
will not
respond to the data signal. Since the data signal is of much lower strength,
any
inaccuracies in converting it into a symmetrical spectrum will result in
relatively
minor interference, not visible under ordinary viewing condition.
t5 Figure 4 shows one embodiment of the principle of the current invention
which is called "Data Separation from Video by Synthesis of a Double Sideband
Signal" and synchronous demodulation. In Figure 4a, the invention is installed
in a
device that is to optionally receive television signals. The TV receiver has a
Nyquist
Filter 406 whose output is split into three paths. One path 402 feeds a
mechanism for
2o recovering the carrier signal. A phase locked loop 404 is shown here for
illustrative
purposes but not as a limitation. The recovered carrier signal is used in a
mixer 408
for synchronous demodulation of the visual signal. It is also phase shifted
410 for use
in another mixer 412 for synchronous demodulation of the data signal. The
second
path 414 feeds a Spectrum Reverser block 416. This reverses the entire
received
25 spectrum, after the Nyquist filter 406, interchanging high frequencies with
low
frequencies and low frequencies with high frequencies symmetrically about the
visual
carrier. The third path 418 feeds a precision phase-correct delay 420 to
ensure the
correct timing of the otherwise unprocessed signal. The Spectrum Reversed
signal
and the precision delayed signal are added to form a double sideband signal.
Figure
30 4b shows the received television 422 and data signals 424 prior to Nyquist
filtering
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426. The television signal 422 is shown as an NTSC signal for illustrative
purposes
but not as a limitation. The television signal 422 and the data signal 424 are
shown as
two separate signals for purposes of illustration. Their algebraic sum would
be seen
on a spectrum analyzer instrument in the frequency domain or an oscilloscope
instrument in the time domain. Figure 4c shows the television receiver's
Nyquist
filter 426. Figure 4d shows the television signal 428 and the data signal 430
after the
television receiver's Nyquist filter 426. The data signal 430 has become
symmetrical
in its frequency range and thus has only quadrature components. The television
signal
448 in this same frequency range has become unsymmetrical having both in-phase
1o and quadrature components. Figure 4e shows the output 432, 434 of the
Spectrum
Reverser 416 that flips the spectrum around the visual carrier frequency 436.
Figure
4f shows the sum of the precision delayed spectrum and the reversed spectrum
438,
440. At this point, both the visual signal 438 and the data signal 440 have
become
double sideband. The visual signal 438 has only in-phase components throughout
and
is the data signal 440 has only quadrature components throughout. The two
signals can
be completely separated with synchronous demodulation without the need for
precision filters. The data path no longer requires a well designed filter to
remove
quadrature components of the visual signal 438 from the data path. There are
no
quadrature video components in the video signal 438 in the range of
frequencies
20 occupied by the data signal 440 or in the regions of frequencies anywhere
near the
frequencies occupied by the data signal 440. There is no "rude video" to
contend
with.
Figure 5 describes an implementation of a Spectrum Reverser 502. The
Spectrum Reverser 502 is not the invention itself. It will be appreciated that
other
25 methods of implementation of spectrum reversal would also be effective in
implementing this invention. Other methods of implementation will be
understood by
those of ordinary skill in these arts. The first Local Oscillator 504 operates
at a
frequency N times the receiver's intermediate frequency, IF. Mixer # 1 506
multiplies
this oscillator's 504 cosine wave output with the combined video and data
signal.
3o Figure Sb shows the combined video and data signal 508. Figure Sc shows the
result
when N is set equal to 3. This choice is for illustrative purposes only and is
not a
limitation. The mixer 506 behaves as a doubly balance mixer yielding and
output
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which is comprised of the sum frequencies and the difference frequencies. The
input
components are balanced out and do not appear at the output. The local
oscillator
signal 510 is depicted in Figure 5c as a dashed vector to indicate its
location at the
input to the mixer 506, but to also indicate that it is not present in the
output signal.
The sum frequencies 512 which form the upper sideband in Figure 5c are
retained
with the Band Pass Filter # 1 514 and the lower sideband 516 is rejected by
that filter
514. Note that the spectrum 512 is not reversed at this point, but merely
shifted to
another frequency. Next Local Oscillator 518 # 2 operates at a frequency (N +
2)
times the IF frequency. Since N has been set equal to 3, Local Oscillator # 2
518 in
to this illustration operates at five times the IF frequency. This choice is
for illustrative
purposes only and is not a limitation. Mixer # 2 520 multiples this
oscillator's cosine
wave output with the output of Band Pass Filter # 1 514. Figure 5 d shows
output of
Mixer # 2 520. The mixer behaves as a doubly balance mixer yielding and output
which is comprised of the sum frequencies and the difference frequencies. The
input
t5 components are balanced out and do not appear at the output. The local
oscillator 518
signal is depicted in Figure 5d as a dashed vector 520 to indicate its
location at the
input to the mixer, but to also indicate that it is not present in the output
signal. The
sum frequencies that form the upper sideband 522 in Figure 5d are rejected
with Band
Pass Filter # 2 524 and the lower sideband 526 is passed by that filter. Note
that the
2o passed spectrum 526 is now reversed and at the same IF frequency 528 as the
original
spectrum 508 of Figure 5b. The misers 506, 520 and especially the Band Pass
Filters
514, 524 of Figure 5a have propagation times that result in a delay of the
signal. This
gives rise to the need for the Precision Phase-Correct Delay 420 element of
Figure 4a.
Figure 6 describes an implementation of a Precision Phasa~Correct Delav. The
25 Precision Phase-Correct Delay is not the invention itself. It will be
appreciated that
other methods of implementation of a Precision Phas~Correct Delay would also
be
effective in implementing this invention. Other methods of implementation will
be
understood by those of ordinary skill in these arts. Comparison of Figure 5a,
Figure
5c, and Figure 5d respectively with Figure 6a, Figure 6c, and Figure 6d
illustrates that
3o the respective filters 514 and 614, 524 and 624 are identical and the
mixers 406 and
606, 420 and 620 are identical. Thus the propagation time through this circuit
will be
identical to that of the Spectrum Reverser 502 to the precision of the
matching of the
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components. Optional Phase Adjusters 630, 632 have been added to compensate
for
mismatches in the implementation. The Local Oscillators 504, 518, 604 operate
at
different frequencies, but that does not impact the propagation delay through
the
system. The explanation of Figure 5a through Figure 5d provides an
understanding of
the operation of Figure 6a through Figure 6d.
It is noted that the information signals in the vestigial sideband portion of
the
received spectra are correlated with the information signals in the
unattenuated other
side. As a result, the voltages add when spectrum reversal and addition is
done. The
noise in these two sidebands is uncorrelated and the noise powers add. This
results in
to an advantageous signal to noise ratio improvement.
Now that the basic principle of the invention has been presented,
simplifications will be described.
The filters of Figure 5a and Figure 6a 514, 524. 614, 624 aid in understanding
the operation. In the implementation itself, the primary function of the
filters is to
is prevent overloading the mixers 506, 606, 520, 620. Ideal mixers would not
require
these filters. That is, mixers with sufficient dynamic range would not require
these
filters. The filters 514, 524, 614, 624 are a source of expense and complexity
and
delay that can be avoided with adequate mixer design. Figure 5e shows the
Spectrum
Reverser 502 of Figure 5a, but without the band pass filters 514, 524. Figure
6e
2o shows the Precision Phase-Correct Delay 634 of Figure 6a, but without the
band pass
filters 614. 624. Spectrum Reverser Figures 5f, 5g, and5h correspond to and
perform
the same functions as Figures 5b, 5c, and Se. In the former set of Figures,
the
undesired spectra are not filtered out; they are just left in place.
Similarly, the
Precision Phase-Correct Delay Figures 6f, 6g, and 6h correspond to and perform
the
25 same functions as Figures 6b, 6c, and 6e. In the former set of Figures, the
undesired
spectra are not filtered out; they are just left in place. Figure 5h and
Figure 6h
describe the input to the Summer 450 of Figure 4a and Figure 6i describes the
output
of that Summer 450 . Note that the shapes of the spectra in these figures is
schematic,
intending to easily illustrate when spectra are reversed. These are not
intended tc
3o show actual spectra or even the actual summation of spectra. Figure 6j is
the output
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of the Phase Shifter 410 of Figure 4a that feeds the Mixer 412 connected to
the Data
Output. Figure 6k shows the output of that Mixer 412. Note that it is the sum
and
difference of the IF frequency and the spectra of Figure 6i. If the result is
then low
pass filtered with a relatively simple low pass filter 652, only the data
baseband
spectra 654 remains. The low pass filter 652 is simple and inexpensive because
the
closest interfering spectrum 656 is at twice the IF frequency, some ninety MHz
away
from the baseband data spectra which consists of less than a few MHz. In fact,
normal parasitic reactances will attenuate the higher frequencies. If these
higher
frequency components were not completely removed, they would only have the
effect
to of slightly closing the eye pattern when the data signal is detected and
converted into
a digital stream. Depending on the system signal to noise ratio, this may be
quite
acceptable.
Note that expense, complexity, and delay of any band pass filters 514, 524,
614, 624 has been avoided. This is especially of importance in an integrated
circuit
~5 implementation where filters are a significant challenge. Even in a
discrete
realization, filters are a labor intensive and expensive part of the design
that will not
be missed. The signal to noise ration is significantly improved with this
invention.
These are just some of the advantages of the innovation of this invention.
Figure 7 shows the spectra in a larger, easy to see form. Figure 7a is the
20 output of the Nyquist Filter 406 and the Precision Phase Correct Delay 420.
Figure
7b shows the output of the Spectrum Reverser 416. Figure 7c sums the spectra
at the
outputs of the Precision Phase-Correct Delay 420 and the Spectrum Reverser
416.
Both the television 738 and the data spectra 740 are double sideband after the
Summer. The television signal 738 has just in-phase components and the data
signal
25 740 has only quadrature components. These components are easily separated
with
synchronous demodulation techniques.
Figure 8 shows a further simplification in cases where high quality video is
not
required or where just a data receiver is implemented. In that case, the
Nyquist filter
406 is eliminated. There is no Nyquist filter 406 at all. As the Figures show,
the
3o result is again a symmetrical double sideband spectrum 838, 840. Both the
television
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838 and the data spectra 840 are double sideband. The television signal 838
has just
in-phase components and the data signal 840 has only quadrature components.
These
components are easily separated with synchronous demodulation techniques. The
television signal 838 is modestly distorted in that it has more strength in
its lower
frequencies than is expected. This will not be objectionable on a small screen
television receiver such as used in portable applications.
In the case where an adaptive equalizer is used to combat multipath in the
signal, the increased signal strength in the visual signal will be of
advantage when the
visual signal is used as a "training signal" for the data. In the special case
where the
to antenna is adjusted by observing the NTSC picture, the adjustment will be
aided by
the increase signal strength in the range of frequencies occupied by the data.
The elimination of the Nyquist filter 406 saves expense and complexity. This
is especially of importance in an integrated circuit implementation where
filters are a
challenge. The absence of Nyquist filter 406 facilitates implementation with a
just a
few or even just one integrated circuit.
Figure 9a displays the transmitted data spectrum 924. Figure 9b shows the
reversed data spectrum 934. Figure 9c shows the addition of the two spectra
924, 934
yielding a symmetrical spectrum 940 with only quadrature components.
There is a further significant advantage in the implementation of Figure 8
2o compared to the implementation of Figure 4. In Figure 4, the Nyquist Filter
406
attenuates data signal 424 frequencies that are closer to the band edge while
simultaneously attenuating the NTSC signal 422. Since the NTSC signal 422 and
the
data signal 424 are combined, it is not possible to attenuate portions of the
NTSC
signal 422 and not attenuate portions of the data signal 424 where they both
occupy
the same frequencies. This attenuation of the data frequencies in the receiver
reduces
the signal to noise ratio and that lowers the margin of data recover
performance. The
implementation of Figure 8 avoids this reduction in signal to noise ratio by
utilizing
the entire received data spectrum rather than rejecting a significant portion
of it. In
both implementations, the signals in the sidebands are correlated so their
voltages add
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while the noise is uncorrelated and so the noise powers add. The result is an
important improvement in signal to noise ratio.
Figure 10 demonstrates the results of extending the data spectrum beyond the
normal double sideband region of the NTSC signal 1060 (which consists of those
frequencies between 750 KHz above the visual carrier and 750 KHz below the
visual
carrier). Figure 11 shows the spectra details. Figure 1 I a is the Data RF
Transmit
filter 1118. Note that it severely attenuates the data signal between 750 KHz
above
the visual carrier and 1.25 MHz above the visual carrier. Figure llb is the
NTSC
VSB filter 1102 that strongly attenuates the data (and visual) signal between
750 KHz
to below the visual carrier and 1.25 MHz below the visual carrier. At
frequencies more
than 1.25 MHz below the visual carrier, the VSB filter 1102 severely
attenuates the
data and visual signals. Figure l lc is the composite filter function 1166 of
these two
filters at the transmitter. Figure 1 1d repeats this composite filter function
I 166, shows
a raised cosine data spectra 1168 (for illustrative purposes only and not as a
limitation) and the result I 170 of the composite filter function operating on
the data
spectra 1168. It can be seen that the transmitted data spectra 1170 is highly
unsymmetrical. It slumps I 172 between 750 KHz below the visual carrier and
1.25
MHz below the visual carrier. It is severely attenuated between 750 KHz above
the
visual carrier and 1.25 MHz above the visual carrier. However, the information
2o carrying bandwidth of this signal extends to 1.25 mHz. This is 50% more
bandwidth
than the previous implementations discussed.
Figure 12a shows the received data spectra 1270, the same as in Figure 11 d.
It
also shows a spectrum-reversed data spectra 1272 and the sum 1274 of the
received
1270 and reversed spectra 1272. For comparison, the original data spectra 1276
is
also shown in Figure 12a. Within the NTSC double sideband region, + 750 kHz
around the carrier, the shape of the sum 1274 of the received 1270 and the
reversed
spectra 1272 is identical to the original data spectrum 1276. It will be noted
that there
is some minor distortion in the reconstructed data spectra outside of this
region.
While it is symmetrical and therefore only has quadrature components which can
be
3o separated from the video by synchronous demodulation, the spectra slumps in
the
region between 1.25 MHz below the visual carrier and 750 KHz below the visual
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carrier and in the region between 1.25 MHz above the visual carrier and 750
KHz
above the visual carrier. This distortion is likely not serious and will
result in only a
slight closing of the eye pattern of the data. It is noted that this could be
compensated
with a predistortion at the point of origination and this effect eliminated.
Figure 12b displays the data signal 1216 in existing television receivers. The
television receiver's Nyquist filter 1206 severely attenuates the data in the
region
between 750 KHz below the visual carrier and 1.25 MHz below the visual
carrier.
This causes the data spectra 1216 in the television receiver to become
symmetrical
and have only quadrature components. It also limits the data spectrum 1216 in
the
television receiver to the NTSC double sideband region. Synchronous
demodulation
will separate the desired visual components from the data components 1216.
Since
the visual signal is much stronger than the data signal 1216, small
asymmetries in the
data spectra will result in only a small impact on the video. Since the data
is
uncorrelated with the video the small in-phase contribution only adds a
trivial amount
of noise to the video. This will not be objectionable under nearly all
practical
circumstances.
The foregoing is provided for purposes of disclosure of preferred embodiments
of the invention. Modifications, deletions, additions and alternative
techniques for
creating systems according to the present invention and for carrying out
processes
according to the present invention may be accomplished without departing from
the
scope or spirit of the invention.
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