Note: Descriptions are shown in the official language in which they were submitted.
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THAN ONE VOLUME.
THIS IS VOLUME 1 OF 2
NOTE: For additional volumes please contact the Canadian Patent Office.
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WO 97127550 PCT/US97 144
COMMUNICATION SYSTEM
WITH MMULTICARRIER TELEPHONY TRANSPORT
Field of the Invention
The present invention relates generally to the field of communication
systems. More particularly, the present invention relates to communication
systems with multicarrier telephony transport.
a.c o ir, d o the lnvfr tion
Two information services found in households and businesses today
include television, or video, services and telephone services. Another
information service involves digital data transfer which is most frequently
accomplished using a. modem connected to a telephone service, All fiirther
references to telephony herein shall include both telephone services and
digital
data transfer services.
Characteristics of telephony and video signals are different and therefore
telephony and video networks are designed differently as well. For example,
telephony information occupies a relatively narrow band when compared to the
bandwidth for video signals. In addition, telephony signals are low frequency
whereas NTSC standard video signals are transmitted at carrier frequencies
greater than 50 NlHz. Accordingly, telephone transmission networks are
relatively narrow band systems which operate at audio frequencies and which
typically serve the customer by twisted wire drops from a curb-side junction
box.
On the other hand, cable television services are broad band and incorporate
various frequency carrier mixing methods to achieve signals compatible with
conventional very high frequency television receivers. Cable television
systems
or video services are typically provided by cable television companies through
a
shielded cable service connection to each individual home or business.
One attempt to combine telephony and video services into a single
network is described in U.S. Patent No. 4,977,593 to Balance entitled
''Optical
Communications Network." Balance describes a passive optical
communications network with an optical source located in a central station.
The
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optical source transmits time division multiplexed optical signals along an
optical fiber and which signals are later split by a series of splitters
between
several individual fibers servicing outstations. The network allows for
digital
speech data to be transmitted from the outstations to the central station via
the
same optical path. In addition, Balance indicates that additional wavelengths
could be utilized to add services, such as cable televisionõ via digital
multiplex to
the network.
A 1988 NCT'A technical paper, entitled "Fiber Backbone: A Proposal For
an Evolutionary Cable TV network Architecture," by James A. Chiddix and
David M. P grac, describes a hybrid optical fiber/coaxial cable television
(CATV) system architecture. The architecture builds upon existing coaxial
CA V networks. The architecture includes the use of a direct optical fiber
path
from a head end to a number of feed points in already existing CATV
distribution system.
U.S. Patent No. 5,153,763 to Pidgeon, entitled "CATV Distribution
Networks Using Light Wave Transmission Lines," describes a CATV network
for distribution of broad band, -multichannel CATV signals from a head end to
a
plurality of subscribers. Electrical to optical transm. at the head end and
optical to electrical receivers at a fiber node launch and receive optical
signals
corresponding to broad band CATV electrical signals. Distribution from the
fiber node is obtained by transmitting electrical signals along coaxial cable
transmission lines. The system reduces distortion of the transmitted byroad
band
CATV signals by block conversion of all or part of the broad band of CATV
signals to a frequency range which is less than an octave. Related U.S. Patent
No. 5,262,883 to Pidgeon, entitled "CATV Distribution Networks Using Light
Wave Transmission Lines," . er describes the distortion reducing system.
Although the above-mentioned networks describe various concepts for
transmitting broad band video signals over various architectures, which may
include hybrid optical fiber/coax architectures, none of these references
describe
a cost effective, flexible, communications system for telephony co unications.
Several problems are in erent in such a co uniication system.
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One such problem is the need to optimize the bandwidth used for
transporting data so that the bandwidth used does not exceed the allotted
bandwidth. Bandwidth requirements are particularly critical in multi. point to
point communication where multiple transmitters at remote units must be
accommodated such that allotted bandwidth is not exceeded.
A second problem involves power consumption of the system. The
communication system should minimize the power used at the remote units for
the transport of data, as the equipment utilized at the remote units for
transmission and reception may be supplied by power distributed over the
transmission medium of the system.
Another problem arises from a fault in the system preventing
communication between a head end and multiple remote units of a multi-point to
point system. For example, a cut transmission line from a head end to many
remote units may leave many users without service. After the fault is
corrected,
it is important bring as many remote units back into service as quickly as
possible.
Data integrity must also be addressed. Both internal and external
interference can degrade the communication. Internal interference exists
between data signals being transported over the system. That is, transported
data signals over a common communication link may experience interference
therebetween, decreasing the integrity of the data. Ingress from external
sources
can also effect the integrity of data transmissions. A telephony communication
network is susceptible to "noise" generated by external sources, such as HAM
radio. Because such noise can be intermittent and vary in intensity, a method
of
transporting data over the system should correct or avoid the presence of such
ingress.
Iese problems and others as will become apparent from the description
to follow, present a need for an enhanced corn mu cation system. Moreover,
once the enhanced system is described, a number of practical problems in its
physical realization are presented and overcome.
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Another embodiment provides a method and apparatus for a fast Fourier
transform. This invention relates to the field of electronic communication
systems, and more specifically to an improved method and apparatus for
providing a fast Fourier transform ("FF1 ").
There are many advanced digital signal-processing applications requiring
analysis of large quantities of data in short time periods, especially where
there is
interest in providing "real time" results. Such applications include signal
processing in modems which use OF DM (orthogonal frequency division
multiplexing). In order to be useful in these and other applications, Discrete
Fourier Transform (;DFT) or Fast Fourier Transform (FF'T) signal processors
must accommodate large numbers of transforms, or amounts of data, in very
short processing times, often called high data throughput.
In addition to the speed and data-throughput requirements, power
consumption is a major concern for many applications. In some
signal-processing applications, power is supplied by portable generation or
storage equipment, such as batteries, where the ultimate power available is
limited by many enviro ent. In such applications, processor power
consumption must be as low as possible. One useful measure of utility or merit
for FFT processors is the energy dissipation per transform point. Ultimately,
one
key problem with any FFT processor is the amount of power consumed per
transform. Generally, high-performance, efficient FFT processors exhibit
energy
dissipations per transform in the range of 100 to 1000 times log2N nanojoules,
where X. is the number of points in a given trans-form. As a consequence,
reasonably large transforms required to process large arrays of data, result
in
large power consumption.
Machine-implemented computation of an FFT is often simplified by
cascading together a series of simple multiply-and-add stages. When a
recursive
process is used, data circulates through a single stage and the computational
structure of the stage is made variable for each circulation. Each circulation
through the stage is referred to as a ' pass14.
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A plurality of computational elements, each known as a radix-r butterfly,
may be assembled to define a single stage for carrying out a particular pass.
A
radix-r butterfly receives r input signals and produces a corresponding number
of
r output signals, where each output signal is the weighted sum of the r input
5 signals. The radix number, r, in essence, defines the number of input
components which contribute to each output component.
By way of example, a radix-2 butterfly receives two input signals and
produces two output signals. Each output signal is the weighted sum of the two
input signals- A radix-4 butterfly receives four input signals and produces
four
corresponding output signals. Each output signal of the radix-4 butterfly
constitutes a weighted sum of the four input signals.
Completion of an N-point Fast Fourier Transform (FFT) requires that the
product of the butterfly radix values, takers over the total number of stages
or
passes, equals the total point count, N. Thus, a 64-point FFT can be performed
by one radix-64 butterfly, or three cascaded stages where each stage has
sixteen
radix-4 butterflies (the product of the radix values for stage-I and stage-2
and
stage-3 is 4 x 4 x 4 = 64), or six cascaded stages where each of the six
stages
comprises 32 radix-2 butterflies (the product of the radix values for stage- t
through stage-6 is 2 x 2 x 2 x 2 x 2 x 2_ 64).
A multi-stage or multi-pass FFT process can be correctly carried out
under conditions where the number of butterfly elements changes from one pass
(or stage) to the next and the radix value, r, of the butterfly elements also
changes from one pass (or stage) to the next. A paper by Gordon DeMuth,
"ALGORITHMS FOR DEFINING MIXED RADIX FFT FLOW GRAPHS",
IEEE Transactions on Acoustics, Speech, and Signal Processing, Vol 37, No. 9,
September 1989, Pages 1349-1358, describes a generalized method for
performing an FFT with a mixed-radix system. A mixed-radix system is one
where the radix va sue, r, in one stage or pass is different from that of at
least one
other stage or pass.
Am advantage of a mixed-radix computing system. is that it can be
"tuned" to optimize the signal-to-noise ratio of the. ,lansforrxn (or more
correctly
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speaking, to minimize the accumulated round-off error of the total transform)
for
each particular set of circumstances. By way of example, it is advantageous in
one environment to perform a 512-point FFT using the mixed-radix sequence:
4, 4, 4, 4, 2_ In a different environment, it may be more advantageous to use
the
mixed-radix sequence: 4, 2, 4, 4, 4.:ound-off error varies within a machine of
finite precision as a function of radix value and the peak signal magnitudes
that
develop in each stage or pass.
In addition, it may be advantageous to scale intermediate results between
each stage or pass, in order to minimize round-off errors and the problem of
overflow. Further, it may be advantageous to vary the amount of scaling
performed between each pass, e.g., either to scale by 1/4 between each radix-4
stage or to scale by 1/2 for some stages and 1/8 for other stages.
Heretofore, FFT processors generally fetched data values from their
working storage in a serial manner, thus limiting the speed which could be
obtained. Further, current FFT processors generally were limited in speed by
loading the working storage with input values, then processing the data in the
working storage, then unloading the result values.
There are many advanced digital signal-processing applications requiring
analysis of large quantities of data in short time periods, especially where
there is
interest in providing; "real time" results. Such applications include signal
processing in modems which use OFDM (orthogonal frequency division
multiplexing).
One need in the art is for an accurate analog-to-digital conversion (ADC)
at moderate frequencies having limited bandwidth. One technology known in
the art is the " Sigma-Delta"' C w_ iich provides very good resolution (high
number of bits in the digital result), but only for signals whose converted
signal
bandwidth is low.
Another need is for an ADC which provides bandwidth-limited digital I
and Q signals (representing amplitude and quadrature) fur a 200 kl-lz
bandwidth
received analog modem signal, wherein the digital result has very high
resolution
and accuracy.
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What is needed is a method and apparatus which addresses the above
problems in the art.
urrunaryf tF~e Invenizorn
The present invention describes a multi-point to point communication
system including multicarrier telephony transport. The multi-point to point
communication system includes a hybrid fiber/coax distribution network..
head end terminal provides for downstream transmission of downstream control
data and downstream telephony information in a fist frequency bandwidth over
the hybrid fiber/coax distribution network and reception of upstream telephony
information and upstream control data in a second frequency bandwidth over the
hybrid fiber/coax distribution network. The head end terminal includes a head
end multicarrier modem for modulating at least downstream telephony
information on a plurality of orthogonal carriers in the first frequency
bandwidth
and demodulating at least upstream telephony information modulated on a
plurality of orthogonal carriers in the second frequency bandwidth. The head
end terminal finther includes a head end controller operatively connected to
the
head end multicarrier modem for controlling transmission of the downstream
telephony information and downstream control data and for controlling receipt
of
the upstream control data and upstream telephony information. The system
further includes at least one service unit, each service unit associated with
at
least one remote unit and operatively connected to the hybrid fiber/coax
distribution network for upstream transmission of upstream telephony
information and upstream control data in the second frequency bandwidth and
for receipt of the downstream control data and downstream telephony
information in the first frequency bandwidth.. Each service unit includes a
service unit multicarrier modem for modulating at least the upstream telephony
information on at least one carrier orthogonal at the head end to at least one
other
-carrier in the second frequency bandwidth and for demodulating at least the
downstream telephony information modulated on at least a band of a plurality
of
orthogonal carriers in the first frequency bandwidth. Each service unit also
includes a service unit controller operatively connected to the service unit
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g
multicarrier modem for controlling the modulation of and demodulation
performed by the service unit multicarrier modem..
Another embodiment prevents untoward spectral effects in the
multicarrier signal from variations in channel activity and from highly
repetitive
data patterns in the payload channels, Data is the payload channels can be
scrambled with pseudorandom sequences, and different sequences can be applied
to different channels in order to produce a more balanced multicarrier
spectrum.
In another embodiment, the plurality of orthogonal carriers in the first
frequency bandwidth include at least one control channel for transmission of
downstream contro:. data and a plurality of telephony information channels for
transmission of downstream telephony information. Further, the plurality of
orthogonal carriers in the second frequency bandwidth include at least one
control channel for transmission of upstream control data and a plurality of
telephony information channels for transmission. of upstream telephony
information.
In other embodiments, a plurality of control channels are interspersed
among the telephony information channels in the first frequency bandwidth and
a plurality of control channels are interspersed acnong the telephony channels
of
the second frequency bandwidth. The telephony channels may be divided into
subbands each having multiple data or payload channels and a control channel;
this allows the remote moderns to be realized as less expensive and/or better
performing narrow--band modems.
The clock signals for generating the carriers and the symbols
representing the transmitted data may be locked to each other or generated
from
the same source, to reduce intersymbol interference significantly.
Another technique for reducing intersyrnbol interference is the
transmission of each symbol with more than 360 of phase in one cycle of its
carrier, in order to allow some leeway in tracking the phase of a channel
carrier
in a receiving modem.
Some applications demand more or different error detection and
correction capability than, others. An embodiment is shown which handles both
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unencoded parity-type detection/correction and more multiple types of more
powerful methods, such as Reed-Solomon encodiacg, in a transparent, real-time
fashion, by packing the data words differently for each case. Moreover, the
processor loading involved in these error-correction techniques can be spread
out
in time, so that not all channels need to be handled at the same time. This is
accomplished by staggering the beginning times of different data messages.
In another embodiment, the at least one service unit includes a service
modem for upstream. transmission of upstream telephony information and
upstream control data within a channel band of the second frequency bandwidth
corresponding to one of the channel bands of the first frequency bandwidth in
which the service modem receives downstream telephony information and
downstream control information. Alternatively, the at least one service unit
includes a multi-service modem for upstream transmission of upstream
telephony information and upstream control data witldn EL plurality of channel
bands of the second frequency bandwidth corresponding to a plurality of the
channel bands of the first frequency bandwidth in which the multi-service
modem receives downns. earn telephony information and downstreann. control
information.
In still another embodiment, the plurality of control channels of the first
frequency bandwidth and the plurality of control channels of the second
frequency bandwidth each include at least one synchronization channel.
In other embodiments, different modulation techniques are utilized for
different carriers. For example, different modulation techniques are utilized
for
different telephony channels. As another example, the aforementioned IOC
channels may be modulated as differential binary phase-shift keyed (BPSK)
signals, while the payload data channels are modulated as 5-bit quadrature
amplitude modulated (QAM32) signals, in order to enhance the use of IOC
channels for subband tracking, and for other purposes. The constellation
defining the modulated signals can be constructed to achieve a minimal number
of bit errors for small errors in amplitude or phase of the received signal;
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broadly, the constellation points are mapped to bit combinations in a scheme
analogous to a Gray code.
A communication system which addresses the problems inherent in the
system, in particular,, ingress problems is also described. The communication
5 system includes a distribution network between a head end terminal and at
least ffi
one remote unit. The head end terminal receives upstream telephony
information and upstream control data in a frequency bandwidth over the
distribution network. The head end terminal includes a head end multic er
demodulator for demodulating at least upstream telephony information
10 modulated on a plurality of orthogonal carriers in. the frequency
bandwidth. The
demodulator includes at least one polyphase filter for filtering the at least
upstrearn telephony information modulated on the plurality of orthogonal
carriers to provide ingress protection, for the modulated orthogonal carriers.
The
head end terminal also includes a head end controller operatively connected to
the head end multicarrier demodulator for controlling receipt of the upstream
control data and upstream telephony information. The system further includes
at
least one service unit modulator, each service unit modulator associated with
at
least one remote unit and operatively connected to the distribution network
for
modulating at least upstre telephony information on at least one carrier
orthogonal at the head end terminal to at least one other carrier in the
frequency
bandwidth. The system also includes a service unit controller operatively
connected to the service unit multicarrier modulator for controlling the
modulation performed by the service unit rnultica-rier modulator.
in another embodiment, the plurality of orthogonal carriers in the
frequency bandwidth include a plurality of telephony information channels for
transmission of upstream telephony information after modulation of telephony
information thereon and at least one control channel associated with the
plurality
of telephony channels for transmission of upstream control data thereon. Here
also, the IOC may be placed in the midpoint of the, subbands.
In another embodiment, the at least one polyphase filter includes a first
and second polyphase filter. T he first polyphase falter filters a first
plurality of
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channel sets and passes a first plurality of at least telephony channels
within each
channel set of the first plurality of channel sets. The second polyphase
filter
filters a second plurality of channel sets and passes a second plurality of at
least
telephony channels within each channel set of the second plurality of channel
sets. The first and second polyphase filter are offset froro one another such
that
all at least telephony channels of the first and second plurality of channel
sets are
passed. In another em,rnbodiment, the polyphase filters include at least two
overlapping polyphase filters.
In another alternate embodiment, the demodulator includes a tunable
notch filter for filtering the at least upstream telephony information
modulated
on a plurality of orthogonal carriers to prevent passage of corrupted
modulated
orthogonal carriers.
In addition, a method of polyphase filtering in a communication system
is also described. The method includes receiving a plurality of orthogonal
carriers having modulated telephony information thereon. The plurality of
orthogonal carriers include a first and second plurality of noncontiguous
channel
sets. The first plurality of noncontiguous channel sets are filtered and a
first
plurality of channels of each channel set of the first plurality of
noncontiguous
channel sets are passed. The second plurality of noncontiguous channel sets
are
filtered and a second plurality of channels of each channel set of the second
plurality of noncontiguous channel sets are also passed. The second plurality
of
channels passed include channels of the first plurality of noncontiguous
channel
sets not passed when filtering the first plurality of noncontiguous channel
sets.
A receiver apparatus is also described which receives a frequency
bandwidth having a plurality of modulated orthogonal carriers. At least one
polyphase filter provides ingress protection for the frequency bandwidth by
filtering a plurality of channel sets of the modulated orthogonal carriers.
The use of channel monitoring to address some of the problems inherent
in a multi-point to point communication system, in particular, with respect to
ingress, is also described. The monitoring method of the present invention
monitors a telephony communication n-bit channel wherein one of the bits is a
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parity bit. The parity bit of the n-bit channel is sampled and a probable bit
error
rate is derived from the sampling of the parity bit.
In one embodiment, the probable bit error rate over a time period is
compared to a predetermined bit error rate value representing a minimum bit
error rate to determine if the n-bit channel is corrupted. A corrupted channel
can
then either be reallocated or, in another embodiment, the transmission power
of
the channel can be increased to overcome the core option.
In an alternate method embodiment, the method comprises the steps of
sampling the parity bit of the n-bit channel over a first time period,
deriving a
probable bit error rate from the sampling of the parity bit over the first
time
period, comparing the probable bit error rate over the first time period to a
pre-
determined bit error rate value to determine if the a-bit channel is
corrupted, and
accumulating a probable bit error rate over a plurality of successive time
periods
if the n-bit channel is not corrupted.
In another alternate method embodiment, the method comprises the steps
of sampling the parity bit of the n-bit channel and deriving a probable bit
error
rate from the sampling of the parity bit over a first time period. The
probable bit
error rate over the first time period is compared to a first predetermined bit
error
rate value to determine if the n-bit channel is corrupted. A probable bit
error rate
from the sampling of the parity bit over a second time period is derived. The
second time period is longer than the first time period and runs concurrently
therewith. The probable bit error rate over the second time period is compared
to
a second predetermined bit error rate value to determine if the n-bit channel
is
corrupted.
In yet another alternate embodiment, a method for monitoring at least
one telephony communication channel includes equalizing a signal on the
channel and monitoring the equalization of the signal to produce a probable
bit
error rate as a function of the equalization.
In still yet another alternate embodiment, a method for monitoring at
least one unallocated telephony coax munication channel includes periodically
monitoring the at least one unallocated telephony communication channel. Error
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data for the at least one unallocated telephony communication channel
accumulated and the at least one unallocated telephony communication channel
is allocated based on the error data,
A multi-point to point communication system utilizing a distributed loop
method is also described. The communication system in accordance with the
present invention includes a distribution network and a head end terminal for
downstream transmission of downstream control data and downstream telephony
information in a first frequency bandwidth over the distribution network. The
head end terminal receives upstream telephony information and upstream control
data in a second frequency bandwidth over the distribution network. The head
end terminal further includes a head end multicarrier modem for modulating at
least downstream telephony information on a plurality of orthogonal carriers
in
the first frequency bandwidth and demodulating at least upstream telephony
information modulated on a plurality of orthogonal carriers in the second
frequency bandwidth. A head end controller is operatively connected to the
head
end multicarrier modem for controlling transmission of the downstream
telephony information and downstream control data and for controlling receipt
of
the upstream control data and upstream telephony information. The system
includes a plurality of service units. Each service wnit is associated with at
least
one remote unit and operatively connected to the distribution network for
upstream transmission of upstream telephony information and upstream control
data in the second frequency bandwidth and for receipt of the downstream
control data and downstream telephony information in the first frequency
bandwidth. Each service unit includes a service unit multicarrier modem for
modulating at least the upstream telephony information on at least one carrier
orthogonal to at least one other carrier in the second frequency bandwidth and
for demodulating at least the downstream telephony information modulated on at
least a band of a plurality of orthogonal carriers in the first frequency
bandwidth.
Each service unit also includes a service unit controller operatively
connected to
the service unit multicarrier modem for controlling the modulation of and
demodulation performed by the service unit multicarrier modem. The service
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unit controller adjusts at least one local transmission characteristic in
response to
an adjustment command from the head end controller transmitted in the
downstream control data to the at least one remote unit. The head end
controller
further includes a detector for detecting the at least one local transmission
characteristic of the service unit modem associated with the at least one
remote
unit and for generating the adjustment command as a function of the detected
at
least one transmission characteristic for transmittal to the service unit
associated
with the at least one remote unit in the downstream control data.
Distributed system control is also employed for acquiring and tracking
remote service units newly connected to (or activated within) the system. An
acquisition process makes rough estimates of the frequency, phase, and data-
symbol timing of the head-end transmitter, calculates the round-trip delay of
data
to and from the head end, and then tracks changes in the frequency, phase, and
timing of the head end, all with minimal overhead to the transmission of
payload
data within the system. A special non-valid data signal is used to signal the
start
of a training pattern for acquisition purposes. Maintaining accurate power
balancing or leveling among the remote units transmitting upstream to the head
end is both necessary and difficult in a multipoint-to-point multicarrier
system.
One embodiment of the invention uses both an automatic gain control or
equalizer at the head end and a transmitter output or power control at the
remote
end to achieve the conflicting goals of wide dynamic range and high resolution
amplitude control.
Furthermore, communication system having a distribution network
between a head end and a plurality of remote units using a scanning method is
described. The system includes the transmission, from. the head end, of a
plurality of modulated orthogonal carriers having telephony information
modulated thereon in a plurality of regions of a first frequency bandwidth.
Each
of the regions has at least one control channel associated therewith having
control information modulated thereon. A scanner at the remote units, scans
each of the plurality of regions in the first frequency bandwidth and locks
onto
the at least one control channel associated with each of the plurality of
regions to
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detect a unique identifier to determine which region of the first frequency
bandwidth the remote unit is to tune to and which region in a second frequency
bandwidth the remote unit is to transmit within.
In another embodiment, the communication system includes a
5 distribution network between a head end and a plurality of remote units. The
head end includes a head end terminal for downstream transmission of
downstream control data and downstream telephony information in a first
frequency bandwidth- over the distribution network and for receipt of upstream
telephony information and upstream control data in a second frequency
10 bandwidth over the distribution network. The head end terminal includes a
head
end rnulticarrier modem for modulating at least downstrearn telephony
information on a plurality of orthogonal carriers in a plurality of regions of
the
first frequency bandwidth. The head end multicarrier modem also demodulates
at least upstream telephony information modulated on a plurality of orthogonal
15 carriers of a plurality of regions in the second frequency bandwidth. The
plurality of orthogonal carriers in each of the regions includes a plurality
of
telephony information channels for transmission of telephony information
thereon with each of the regions having at least one control channel
associated
therewith for transmission of control data. The head end terminal also
includes a
head end controller operatively connected to the head end multicarrier modem
for controlling transmission of the downstream telephony information and
downstream control data and for controlling receipt of the upstream control
data
and upstream telephony information. The system further includes a plurality of
service unit modems with each service unit modem associated with at least one
remote unit and operatively connected to the distribution network for upstream
transmission of upstrearn telephony information and upst:re control data in
one
of the plurality of regions of the second frequency bandwidth and for receipt
of
the downstream control data and downstream telephony information in one of
the plurality of regions in the first frequency bandwidth. Each service unit
modem includes a scanner for scanning each of the plurality of regions in the
first frequency bandwidth and for locking onto the at least one control
channel in
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each of the plurality of regions to detect a unique identifier for each
service unit
modem to determine which region of the first frequency bandwidth the service
unit modem is to tune to and which region in the second frequency bandwidth
the service unit modem is to transmit within.
This invention further provides a three-part RAM structure, the functions
of which can be permuted between input, conversion, and output functions. In
one embodiment, the conversion RAM section is configured to offer four values
to be accessed simultaneously in order to speed operations.
In another embodiment, this invention rellates to the field of electronic
communication systems, and more specifically to an improved method and
apparatus for providing a Sigma-Delta analog-to-digital conversion and
decimation for a modem.
According to another aspect of the invention there is provided a data
delivery system having a head end connected to P. plurality of remote
subscribers
over a network, the data originating from general purpose digital computers,
wherein the data is transmitted to the remote subscribers in a plurality of
data
channels, and at the start of a data transmission session, one or more
channels are
selectively assigned to carry the data between the head end and the subscriber
premises with each of the data channels maintaining a substantially constant
rate
of data transfer between the head end and the destination premise. The system
and method further allows that the number of assigned channels assigned to a
particular subscriber can be changed from one connection to another to
accommodate changes in overall system loading, but at all times maintaining a
minimum number of assigned channels so that a minimum rate of data transfer
can be maintained between the head and and a subscriber premise. The system
also provides asymmetrical operation so that the number of data channels
assigned in the do tnstream path from the head end to the subscribers is much
greater in number than the number of upstream data channels.
According to another aspect of the invention there is provided a system
of computer data and telephony data transmission over a telecommunications
network having a head end connected to a plurality of remote subscribers, the
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computer data originating from general purpose digital computers, and the
telephony data received from or conveyed to the public telephone network. The
system comprises transmitting the computer data and telephony data to the
remote subscribers in a plurality of data channels, and establishing a
computer
data or telephony data connection between the head end and a subscriber
premise
independently of one another. Each of the computer data or telephony data
connections are established by assigning one or more of the data channels to
carry the computer data and one or more of the data channels to carry the
telephony data with at least some of the data channels being available to
carry
either computer data or telephony data. The system further allows that the
number assigned data channels can be changed from one connection to another
to the same subscriber so that the overall bandwidth of the network can be
reallocated.
The system further allows that the data channels are transmitted using
Radio Frequency (RF) signaling, and that the RF is scrambled, thereby
providing
data security.
According to Yet another aspect of the invention, the system allows that a
computer data received at the head end can initiate a connection to a remote
subscriber, wherein the connection comprises the assignment of one or more
data
channels to carry the computer data to the subscriber.
According to yet another embodiment, the system 10 or 500 of the
present invention includes an ATM modulator which can receive ATM data and
modulate it onto the IHIFC network. In one preferred embodiment, digital video
data is delivered over an ATM network, multiplexed and modulated onto the
HFC in. RF digital OFDM format on data connections established between the
head end and a subscriber, as for example described above with respect to
system 500. A digital set top box receives the digital video, for example in
4.0
Mbps MPEG or equivalent, and converts it to video for display on a television.
A return path over a telephony or data channel allows for interactive digital
video.
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In another embodiment, a method for transmitting data over a
telecommunication system from a head end to a service unit is provided. The
service unit is assigned to subband of a transmission channel of the
telecommunication system. The subband includes a number of payload channels
that transmit data at a first rate and a control channel that transmits data
at a
second rate. The second rate is slower than the first rate. The system
receives a
request to transmit data to a service unit at the second, slower rate. The
system
further determines whether to transmit the data at the first, faster, rate
based on
the size of the data. When a payload channel in the subband is available to
transmit the data at the first rate, the system allocates the payload channel
to
transmit the data to the service unit over the payload channel at the first
rate.
When the payload channels are allocated to service units and at least one of
the
allocated payload channels is idle, the system allocates the idle payload
channel
to transmit the data to the service unit over the payload channel at the first
rate.
This method can be used to download software and transmission protocols as
well as other data that is not time sensitive. Further, the method
advantageously
provides flexibility in providing bandwidth for transmission of data in the
telecommunications system.
In another embodiment, a telecommunications system dynamically
allocates bandwidth among a plurality of service units. The system comprises a
head end that transmits data over a transmissionmedium to the service units.
The head end includes a modem circuit for narrow band transmission in at least
one transmission channel. Each transmission charnel includes a number of
subbands having a number of payload channels and a control channel in each
subband. Further, a control circuit in the head end assigns each service unit
to a
subband for transmission and receipt of data. The control circuit also
allocates a
payload channel to a service unit in response to a request for bandwidth for a
service unit.
In another embodiment, the control Circuit assigns a number of service
units to each subband. The control circuit dynamically allocates bandwidth to
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the service units for selective use of the payload channels in the subband.
This
increases the number of service units that can be coupled to the system.
In another embodiment, a method for dynamically allocating bandwidth
to a service unit in a telecommunications system is provided. The system uses
a
multi-channel transmission scheme with transmission channels that include a
number of subbands. Each subband further includes a number of payload
channels. The meth.od.. begins by receiving a request for a payload channel
for a
service unit that is assigned to a first subband. The method selects an
available
payload channel in the first subband and determines if the payload channel is
acceptable for providing service to the service unit, e.g., acceptable
transmission
quality. When the payload channel is acceptable, the method allocates the
payload channel to the service unit. When, however, the payload channel is
unacceptable, the method selects other channels to find an acceptable payload
channel.
In another embodiment, a telecommunications system implements a
method for allocating payload channels for a service that use multiple payload
channels to communicate with a service unit. The system assigns an identifier
for each payload channel that indicates the relative order of the multiple
payload
channels for the service. The system further monitors the quality of the
payload
channels of the system. When the quality of one of the multiple payload
channels drops below a threshold, the system allocates a different payload
channel to replace the original payload channel for the service. Once
reallocated,
the system uses the identifier for the original payload channel so that the
proper
order for the allocated payload channels is maintained by the service
irrespective
of the order that the payload channels are received at the service unit.
In another embodiment, a telecommunications system provides a method
for using an upstream payload channel to inform the head end of errors that
occur in downstream payload channels. The system, monitors a downstream
-tr ansmission channel at a service unit for transmission errors. Further, the
system generates a signal at the service unit that indicates transmission
errors in
the downstream payload channel. The system also transmits the signal to the
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head end on an associated upstream payload channel thus allowing the head end
to monitor and respond to the performance of the service unit and associated
payload channels.
In another embodiment, a method for controlling a plurality of service
5 units in a telecommunication system is provided. The method first assigns an
identifier to each service unit. The method further assigns each service unit
to a
subband of a transmission channel of a narrow band transmission scheme. In the
transmission channel, each subband includes a control charnel for receiving
and
transmitting control signals. The method broadcasts the control signals for
the
10 service units over the control channels. The method identifies the terminal
to use
the control signal with the identifier.
In another embodiment, a service unit for use with. a communication
system transmits signals with a narrow band transrrnission scheme. The
transmission channel is divided into a number of suxbbands with each subband
15 including a number of payload channels and a control channel. The service
unit
includes a modem that is tunable to receive telephony and control signals on a
subband of a transmission channel. The service unit further includes a
controller
circuit coupled to the modeni to receive control signals over the control
channel
and to determine which control signals to use to control the operation of the
20 modem. The service unit also includes interface circuits coupled to the
controller for providhag signals to a channel unit.
In another embodiment, a method for controlling power usage at a
service unit of a telecommunications system is provided. The method comprises
determining the type of service supported by each :line of a service unit.
When
the service unit supports analog telephony serviceõ the method determines the
hook status of all of the lines of the service unit. When the lines are ors-
hook, the
method powers down the service units to conserve power usage until a request
is
received to use a line of the service unit.
The present invention describes a method +ox establishing communication
between a head end and a plurality of remote units in a multi-point to point
communication system, such as when a fault as described above has left many
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users of the system without service. The method includes transmitting
information from the :,":lead end to the plurality of rep Mote units in a
plurality of
regions of a first frequency bandwidth. Each of the regions has at least one
control channel associated therewith. The in-formation transmitted includes
identification information corresponding to each of n remote units of the
plurality of remote units. Such information is periodically transmitted for
the n
remote units from the head end on the at least one control channel of one of
the
plurality of regions of the first frequency bandwidth during a first
predetermined
time period. The identification information for each of the plurality of n
remote
units is transmitted out of phase with respect to the identification
information for
the other of the n remote units. At each of the n remote units, the at least
one
control channel of each of the plurality of regions in the first frequency
bandwidth is scanned to detect identification information corresponding to
each
of the n remote units to identify a particular region. of the plurality of
regions that
each of the n remote units is to use for receiving information from the head
end.
in one embodiment, a region is identified in a second frequency
bandwidth in which each of the n remote units is to transmit within. The
method
further includes serially performing synchronization for each of the n remote
units for communication with the head end, during a second predetermined time
period after the first predetermined time period.
A multi npoint to point communication system having a distribution
network between a head end and a plurality of remote units for accomplishing
the above method includes means for transmitting information from the head end
to the plurality of remote units in a plurality of regions of a first
frequency
bandwidth. Each of the regions has at least one control channel associated
therewith. The transmitting means further periodically transmits
identification
information corresponding to each of a set of n remote units of the plurality
of
remote units on. at least one control channel of one of the plurality of
regions of
the first frequency bandPidth during a first predetermined time period of an
identification and synchronization time period. The identification information
for
each of the plurality of n remote units is transmitted out of phase with
respect to
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the identification information for the other of the n remote units. The system
further includes at each of the n remote units, means for scanning the at
least one
control channel of each of the plurality of regions in the first frequency
bandwidth to detect identification information during the first predetermined
time period corresponding to each of the n remote units to identify a
particular
region of the plurality of regions that each of the n remote units is to use
for
receiving information from the head end. Further, at each of the n remote
units,
the system includes .means for modulating at least upstream telephony
information on at least one carrier in a second frequency bandwidth orthogonal
at the head end terminal to at least one other carrier in the second frequency
bandwidth and for adjusting at least one local transmission characteristic in
response to an adjustment command from the head end. Means at the head end
for detecting the at least one local transmission characteristic of each of
the n
remote units and for generating the adjustment commands as a function of the
detected at least one transmission characteristic for transmittal to the n
remote
units to serially perform synchronization for each of the n remote units
during a
second predetermined time period of the identification and synchronization
time
period is also included in the system.
The present invention is a hybrid fiber/coax video and telephony
communication network which integrates bi-directional telephony and
interactive video services into one network including optical fiber and
coaxial
cable distribution systems. The present invention utilizes optical fiber as
the
transmission medium. for feeding a plurality of optical distribution nodes
with
video and telephony information from a head end. Coaxial cable distribution
systems are utilized for connection of the distribution nodes to a plurality
of
remote units. J -he head end optically transmits the video information
downstream to the nodes where it is converted to electrical signals for
distribution to the remote units, Telephony information is also optically
transmitted to the nodes in frequency bandwidths unused by the video
information. The do- stream telephony and video optical signals are converted
to electrical telephony and video signals for distribution to the plurality of
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remote units. The network provides for transmission of upstream electrical
data
signals, for example telephony signals, to the head end by transmitting from
the
remote units upstream electrical data signals to the distribution nodes where
such
upstream electrical data signals are converted to upstream optical signals for
transmission to the head end.
In one embodirne:nt, the head end includes a first distribution terminal
having at least one optical transmitter for transmitting optical downstream
telephony signals on at least one optical fiber. in addition, the head end
includes
a second distribution terminal having a separate optical transmitter for
transmitting an optical downstream video signal on an optical fiber line.
in another embodiment, the video and telephony signal distribution
network transmits optical downstream video and telephony signals on at least
one optical fiber in a first frequency bandwidth. In this embodi.anent, a
second
frequency bandwidth is reused for transmission of upstream electrical data
signa; s generated at the remote units. The second frequency bandwidth is
reused
for transmission by each remote emit.
In another embodiment of the invention, a filter is utilized at service
units which interface the coaxial distribution systems to user equipment. The
ingress filter allows for passage of downstream video signals to video
equipment
units and blocks downstream telephony signals transmitted in a different
frequency bandwidth,
escri tion of `the Drawing
Figure '1 shows a block diagram of a communication system in accordance
with the present invention utilizing a hybrid fiber/coax
distribution network;
Figure 2 is an alternate embodiment of the system of Figure 1;
Figure 3 is a detailed block diagram of a host digital terminal DT) with
associated transmitters and receivers of the system of Figure 1;
Figure 4 is a block diagram of the associated transmitters and receivers of
Figure 3"
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Figure 5 is a block diagram of an optical distribution mode of the system of
Figure 1,
Figures 6, 7 are embodiments of frequency shifters for use in the optical
distribution node of Figure 5 and the telephony upstrearrn receiver
of Figure 4, respectively;
Figure 8 is a general block diagram of an integrated service unit (ISU) such
as a home integrated service unit (IRSU) or a multiple integrated
service u it (MISU) of Figure 1;
Figures 9, 10, 11 show data Exame structures and Ex a signaling utilized in
the 1' of Figure 3;
Figure 12 is a general block diagram of a coax master card (CXMC) of a
coax master unit (C J) of Figure :3
Figure 13 shows a spectral allocation for a first transport embodiment for
telephony transport in the system of Figure Ã;
Figure 14 shows a mapping diagram for QAM :r iodula.tion;
Figure 15 shows a mapping diagram for BPSK odulation;
Figure 16 shows a subband diagram for the spectral allocation of Figure 13;
Figures 17, 18 show alternative mapping diagrams or constellations for Q AM
modulation;
Figure 19 shows a timing diagram, of an identification and synchronization
process;
Figure 20 shows a timing diagrarn of a burst identification and
synchronization process.;
Figure 21 is a block diagram of a raster coax card (MCC) downstream
transmission architecture of the C U for the first transport
embodiment of the system of Figure I ;
Figure 22 is a block diagram of a coax transpoft unit (CXTU) downstream
receiver architecture of an MISU for the first transport
embodiment of the system of Figure 1;
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Figure 23 is a block, diagram of a coax home module (CXHM) downstream
receiver architecture of an IHSTJ for the first transport
embodiment of the of the system of Figure
Figure 24 is a block; diagram of a CX1M upstream transmission
5 architecture associated with the CXHM downstream receiver
architecture of Figure 23;
Figure 25 is a block diagram of a CX T(J upstream transmission architecture
associated with the CXTU downstream receiver architecture of
Figure 22;
10 Figure 26 is a block diagram of an MCC upstream receiver architecture
associated with the MCC downstream transmission architecture
of Figure 21;
Figure 27 is a flow diagram of a acquisition distributed loop routine for use
with the system of Figure 1;
15 Figure 28 is a flow diagram of a tracking distributed loop architecture
routine fir use with the system of Figure 1;
Figure 29 shows a magnitude response of a polyphase alter bank of the
MCC upstream receiver architecture o.f: Fire 26;
Figure 30 is an enlarged view of part of the magnitude response of Figure
20 29;
Figure 31 is a block diagram of an ingress filter structure and FFT of the
MCC upstream receiver architecture of Figure 26;
Figure 32 is a block diagram of a polyphase filter structure of the ingress
filter stricture and. F FT of Figure 31;
25 Figure 33 is a block diagram of a carrier, amplitude, timing recovery block
of the downstream receiver architectures of the first transport
embodiment;
Figure 34 is a block diagram of a carrier, amplitude, timing recovery block
of the MCC upstream receiver architecture of the first transport
embodiment;
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Figure 35 is a block diagram of internal equalizer operation for the receiver
architectures of the first transport emiodiment;
Figure 36 is a spectral allocation of a second transport embodiment for
transport in the system of Figure 1;
Figure 37 is a block diagram of an MCC modem architecture of the CXMU
for the second transport embodiment of the system of Figure 1;
Figure 38 is a block diagram of a subscriber modem architecture of the
HISU for the second transport embodiment of the system of
Figure 1;
Figure 39 is a block diagram of a modem of the subscriber modem
architecture of Figure 38;
Figure 40 is a block diagram for channel monitoring used in the system of
Figure 1;
Figures 41, 42,43 are flow diagrams for error monitor portions of channel
monitor routines of Figure 400;
Figure 44 is an alternate flow diagram for the diagram of Figure 42;
Figure 45 is a flow diagram for a background monitor portion of the channel
monitor routines of Figure 40,
Figure 46 is a flow diagram for a backup portion of the channel monitor
routines of Figure- 40;
Figures 47, 48 are a flow diagram of an acquisition distributed loop routine
for
use with another embodiment of the system of Figure 1;
Figure 49 is a. flow diagram of a downstream tracking loop for use with the
embodiment of Figures 47 and 48.
Figure 50 is a flow diagram of an upstream tracking loop for use with the
embodiment of Figures 47 and 48.
Figure 51 is a block diagram showing the locking of all clocks within a
system.
Figures 52, 53 depict phase diagrams of symbol waveforms in an embodiment of
the invention-
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Figures 54, 55, 56, 57 describe error rates and message-encoding methods for
use in a system according to the invention,
Figure 58 is a block diagram of a scrambler for use in the invention.
Figure 59 is a block diagram of a control circuit for a CXMU of an HD T in
a telecommunications system;
Figures 60, 61, 62 aru flow charts that illustrate methods for assigning
subbands and allocating payload channels in a
telecommunications system that uses a multi-carrier
communication sclaeme;
Figures 63, 64, 65, 66, 67 are frequency spectrum diagrams that illustrate
examples of assigning service units to subbands;
Figure 68 is a flowchart that illustrates error monitoring by the channel
manager;
Figure 69 is a flowchart that illustrates a method for allocating an ISU data-
link (IDL) channel in a telecommunications system;
Figure 70 is a block diagram of FFT system; 210 0;
Figure 71 is a block diagram of modem 2400 which includes a FFT system
2100 configured to perform an IFFT in transmitter section 2401
and another FFT system 2,100 configured to perform an FFT in
receiver section 2402;
Figure 72 is a block diagram of three logical baths of R : an input RAIN4
2251, an output RAM 2253, and a corcversion RAM 2252;
Figure 73 is a block diagram of one embodiment: of a physical
implementation which provides the function of input RAM 2241,
conversion RAM 2242, and output RAM 224.3;
Figure 74 is a block diagram of one embodiment of a dual radix core 2600;
Figures 75, 76, 77, 78, 79, 80, 81, 82 together form a table showing the order
of
calculations for a "norrmal butterfly sub-
operation";
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Figures 83, 84, 85, 86, 87, 88, 89, 90 together form et table showing the
order of
calculations for a "transposed butterfly sub-operation.";
Figure 91 is a block diagram of one embodiment of dual- ix core 2600
showing the nomenclature used for the products output by
multipliers 2620 through 2627 and for adder-subtractor-
accumulators 2633;
Figure 92 is a block diagram of one embodiment of an adder-subtractor-
accumulator 2633;
Figure 93 is a block diagram of modern 2400 which includes a Sigma-Delta
ADC and decimator system to drive FFT system 2100;
Figure 94 is a more detailed block diagram of modem receiver 2402;
Figure 95 is a detailed block diagram of one embodiment of a Sigma-Delta
converter 2840;
Figure 96 is an overall schematic diagram of the data delivery transport
system according to the present invention;
Figure 97 is a simplified block diagram of the head-end terminal 12 of the
system :500 according to the present invention;
Figure 98 illustrates a Personal Cable Data Modern (PCDM) 540 and a Data
Modem. Service Module (DMSM) 550;
Figure 99 illustrates in greater detail a PCDM 540;
Figure 100 illustrates a Data Modern Ch el U'rat DMCU) 560;
Figure 101 shows a graph of average bandwidth, per user as a function of the
number of users for the system 500 according to the present
invention;
Figure 102 is a simplified block diagram of the data transport and framing of
the system;. 500 according to the present invention;
Figure 103 illustrates a Local Area Network Unit (LANIJ) 580 according to
the present invention;
Figure 104 illustrates in more detail a DMSM 550 according to the present
invention;
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Figure 105 illustrates in more detail a. DMCU 560 according to the present
invention;
Figures 106, 107, 108, 109 illustrate the call setup for a data connection on
the
system 500 according to the present invention-,
Figure 110 illustrates a call termination sequence on the system 500
according to the present invention;
Figure 1 I l illustrates the software of a LANIJ 580 according to the present
invention;
Figure 112 illustrates a PCDM 620 adapted for asymmetrical data delivery;
Figure 113 illustrates the head-end configuration for asymmetrical data
delivery according to the present invention;
Figures 114, 115 illustrate another alternate embodiment of the invention
wherein digital video is received over an ATM network
and transmitted over a modified form of system 10/500;
Figure 116 shows a block diagram of a hybrid fiber/coax network in
accordance with the present invention;
Figure 117 is a block diagram of a head end host distribution terminal of the
network of Figure 116;
Figure 118 is a block diagram of an optical distribution node of the network
of Figure 116;
Figure 119 is a block diagram of a home coaxial line unit of the network of
Figure 13.6;
Figure 120 is a block diagram of an alternative embodiment for transmission
from the head end to the optical distribution nodes in accordance
with the present invention;
Figure 121 is a block diagram of an impulse shaping technique utilized in
accordance with the present invention;
Figure 122 is a block diagram of an alternative enim bodiment of the optical
to
electrical converter of the head end host distribution terminal of
Figure 1 i 7;
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Figure 123 is a block diagrarn. of an alternative embodiment of the head end
host distribution terminal of Figure 117.
gjaUed_Description of the Preferred' nabodiment
The ca unication system 10, as shown in Figure 1, of the present
5 invention is an access platform primarily designed to deliver residential
and
business telecommunication services over a hybrid fiber-coaxial (HFC)
distribution network 11. The system 10 is a cost-effective platform for
delivery
of telephony and video services. Telephony services rnmay include standard.
telephony, computer data and/or telemetry. In addition, the present system is
a
10 flexible platform for accommodating existing and emerging services for
residential subscribers.
The hybrid fiber-coaxial distribution network 11 utilizes optical fiber
feeder lines to deliver telephony and video service to a distribution node 18
(referred to hereinafter as the optical distribution node (ODNN)) remotely
located
15 from a central office or a head end. 32. From the ODNs 18, service is
distributed
to subscribers via a coaxial network. Several advantages exist by utilizing
the
HFC-based communication system 10. By utilizing fiber installed in the feeder,
the system 10 spreads the cost of optoelectronics across hundreds of
subscribers.
Instead of having a separate copper loop which runs from a distribution point
to
20 each subscriber ("star" distribution approach), the system 10 implements a
bused
approach where a distribution coaxial leg 30 passes each home and subscribers
"tap" the distribution coaxial leg 30 for service. The system 10 also allows
non-
video services to be modulated for transmission using more cost-effective R.F
modem devices in dedicated portions of the sped . Finally, the system 10
25 allows video services to be carried on existing coaxial facilities with no
additional subscriber equipment because the coaxial distribution links can
directly drive existing cable-ready television sets.
It should be apparent to one skilled in the art t nat the modem transport
architecture described herein and the functionality of the architecture and
30 operations surrounding such architecture could be utilized with
distribution
networks other than hybrid fiber coax networks. For example, the functionality
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may be performed with respect to wireless systems. Therefore, the present
invention contemplates use of such systemss in accordance with the
accompanying claims.
The system 10 includes host digital terminals 12 (}lI)Ts) which
implement all common equipment functions for telephony transport, such as
network interface, synchronization, DSO grooming, and operations,
administration, maintenance and provisioning (OAM&P) interfaces, and which
include the interface between the switching network and a transport system
which carries information to and from customer interface equipment such as
integrated service units 100 (ISUs). Integrated services units (ISUs) 100,
such as
home integrated service units (HISUs) 68 or multiple user integrated service
units (MISUs) 66, whici may include a business integrated service unit as
opposed to a multiple dwelling integrated service unit, implement all customer
interface functions and '.interface to the transport system which carries
information to and frorr, the switched network. In the present system, the HDT
12 is normally located in a central office and the ISUs 100 are remotely
located
in the field and distributed in various locations. The HDT 12 and ISUs 100 are
connected via the hybrid fiber-coax distribution network I 1 in a multi-point
to
point configuration. In the present system, the modem ctionality required to
transport information over the HFC distribution network I 1 is performed by
interface equipment in both the ITT 12 and the ISUs 100. Such modem
functionality is performed utilizing orthogonal frequency division
multiplexing.
The communication system shall now be generally described with
reference to Figures 1, s and 8. The primary components of system 10 are host
digital terminals (HD T s) 12, video host distribution terminal O /MT) 34,
telephony downstream transmitter 14, telephony upstrean receiver 16, the
hybrid fiber coax (HFC) distribution net--work l including optical
distribution
node 18, and integrated service units 66,68 (shown generally as ISU 100 in
Figure 8) associated with remote units 46. The HDT 12 provides telephony
interface between the switching network (noted generally by trunk line 20) and
the modem interface to the HFC distribution network for transport of
telephony.
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information. The telephony downstream transmitter 14 performs electrical to
optical conversion of coaxial R-F downstre telephony information outputs 22
of an. HDT 12, shown is Figure 3, and transmits onto redundant downstream
optical feeder lines 24. The telephony upstream receiver 16 performs optical
to
electrical conversion of optical signals on redundant upstream optical feeder
lines 26 and applies electrical signals on coaxial upstream telephony
information inputs 28 of T 12. The optical distribution node (ODN) 18
provides interface between the optical feeder lines 24 and 26 and coaxial
distribution legs 30. Die ODN 18 combines do stream video and telephony
onto coaxial distribution legs 30. The integrated services units provide modem
interface to the coaxial distribution network and service interface to
customers.
The T 12 and ISUs 100 implement the telephony transport system
modulator-demodulator (modem) functionality. The )T 12 includes at least
one MCC modem. 82, shown in Figure 3 and each ISU 100 includes an RF
IS1r.J modem 101, shown in Figure 8. The MCC modems 82 and ISU modems
101 use a multi-carrier ' transmission technique to transport telephony
information, such as DSO+ channels, between the F3.DT 12 and ISUs 100. This
multi-carrier technique is based on oogonal frequency division multiplexing
(OFDM) where a bandwidth of the system is divided up into multiple carriers,
each of which may represent an information channel. Multi-carrier modulation
can be viewed as a technique which takes time-division multiplexed information
data and transforms it to frequency-division multiplexed data. The generation
and modulation of data on multiple carriers is accomplished digitally, using
an
orthogonal transformation on each data channel, The receiver performs the
inverse sformation on segments of the sampled wavefor m to demodulate the
data. The multiple carriers overlap spectrally. However, as a consequence of
the
orthogonality of the transformation, the data in each carries can be
demodulated
with negligible interference from the other carriers, thus reducing
interference
between data signals transported. Multi--c ier transmission obtains efficient
utilization of the transmission bandwidth, particularly necessary in the
upstream
corn munication of a multi-point to point system. Multi-c er modulation also
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provides an efficient means to access multiple multiplexed data streams and
allows any portion of the band to be accessed to extract such multiplexed
information, provides superior noise immunity to impulse noise as a
consequence of having relatively long symbol times, and also provides an.
effective means for eliminating narrowband interference by identifying
carriers
which are degraded and inhibiting the use of these carriers for data
transmission
(such channel monitoring and protection is described. in detail below).
Essentially, the telephony transport system can disable use of carriers which
have interference and poor performance and only use carriers which meet
transmission quality targets.
Further, the DNNs 18 combine downstream video with the telephony
information for transmission onto coaxial distribution legs 30. The video
information from existing video services, generally shown by trunk line 20, is
received by and processed by head end 32. Head end 32 or the central office,
includes a video host distribution terminal 34 (VHDT) for video data
interface.
The V14DT 34 has optical transmitters associated therewith for communicating
the video information to the remote units 46 via the ODN.s 1.8 of the C
distribution networkf 1.
The telephony transmitter 14 of the HDTs 12, shown in Figure 3 and 4,
includes two transmitters for downstream telephony transmission to protect the
telephony data transmitted. These transmitters are conventional and relatively
inexpensive narrow band laser transmitters. One transmitter is in standby if
the
other is functioning properly. Upon detection of a fault in the operating
transmitter, the transmission is switched to the standby transmitter. In
contrast,
the transmitter of the VI-IDT 34 is relatively expensive as compared to the
transmit-ters of HDT 12 as it is a broad band analog D9B laser transmitter.
Wherefore, protection of the video information, a non-essential service unlike
telephony data, is left unprotected. By splitting the telephony data
transmission
from the video data transmission, protection for the telephony data alone can
be
achieved. If the video data information and the telephony data were
transmitted
over one optical fiber line by an expensive broad band analog laser, economies
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:34
may dictate that protection for telephony services may not be possible.
Therefore, separation of such transmission is of importance.
F " er with reference to Figure 1, the video information is optically
transmitted downstream via optical fiber line 40 to splitter 38 which splits
the
optical video signals for transmission on a plurality of optical fiber lines
42 to a
plurality of optical distribution nodes 18. The telephony transmitter 14
associated with the T 12 transmits optical telephony signals via optical fiber
feeder line 42 to the optical distribution nodes 18. The optical distribution
nodes
18 convert the optical video signals and optical telephony signals for
transmission as electrical outputs via he coaxial distribution portion of the
hybrid fiber coax (HF ) distribution network 11 to a plurality of remote w its
46. The electrical downstream video and telephony signals are distributed to
ISUs via a plurality of coaxial legs 30 and coaxial taps 44 of the coaxial
distribution portion of the C distribution network 11.
The remote units 46 have associated therewith an ISU 100, shown
generally in Figure 8, that includes means for transmitting upstream
electrical
data signals including telephony information, such as from telephones and data
terminals, and in addition may include means for tr nsmit .ng set top box
information from set top boxes 45 as described further below. The upstream
electrical data signals are provided by a plurality of:IIUs 100 to an optical
distribution node 18 connected thereto via the coaxial portion of the C
distribution network 11. The optical distribution node 18 converts the
upstream
electrical data signals to an upstream optical data signal for transmission
over an
optical fiber feeder lire 26 to the head end 32.
Figure 2 generally shows an alternate embodiment. for providing
transmission of optical video and optical telephony signals to the optical
distribution nodes 18 from head and 32, the HDT 112 and "H-DT 34 in this
embodiment utilize the same optical transmitter and the same optical fiber
feeder
line 36. The signals from HDT 12 and V T 34 are combined and transmitted
optically from headend 32 to splitter 38. The combined signal is then split by
splitter 3 g and four split signals are provided to the -optical distribution
nodes 18
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for distribution to the remote units by the coaxial distribution legs 30 and
coaxial
taps 44. Return optical telephony signals from the ODNs 18 would be combined
at splitter 38 for provision to the headend. However, as described above, the
optical transmitter utilized would be relatively expensive due to its broad
band
5 capabilities, lessening the probabilities of being able to afford protection
for
essential telephony services.
As one skilled in the art will recognize, the fiber feeder lines 24, 26, as
shown in Figure 1, may include four fibers, two for transmission downstream
from downstream telephony transmitter 14 and two for transmission upstream to
10 upstream telephony receiver 16. With the use of directional couplers, the
number of such fibers may be cut in half. In addition, the number of
protection
transmitters and fibers utilized may vary as known to one skilled in the art
and
any listed number is not limiting to the present invention as described in the
accompanying claims.
15 The present invention stall now be described in fiather detail. The first
part of the description shall primarily deal with video i ransport. The
remainder
of the description shall primarily be with. regard to telephony transport.
VIDEO TRANSPORT
The communication system 10 includes the head end. 32 which receives
20 video and telephony information from video and telephony service providers
via
trunk line 20. Head end 32 includes a plurality of HDTs 12 and a VHDT 34.
The HDT 12 includes a network interface for communicating telephony
information, such as TI, ISDN, or other data services irafbrmation, to and
from.
telephony service providers, such communication also shown generally by trunk
25 line 20. The VHDT 34 includes a video network interface for communicating
video information, such as cable TV video information and interactive data of
subscribers to and from video service providers, such. communication also
shown
generally by trunk line 20.
The VHDT 34 transmits downstream optical signals to a splitter 38 via
30 video optical fiber feeder line 40. The passive optical sputter 38
effectively
makes four copies of the downstream high bandwidth optical video signals. The
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duplicated downstream optical video signals are distributed to the
correspondingly connected optical distribution nodes 18. One skilled in the
art
will readily recognize that although four copies of the downstream video
signals
are created, any number of copies may be made by an appropriate splitter and
that the present invention is not limited to any specific number.
The splitter is a passive means for splitting broad band optical signals
without the need to employ expensive broad band optical to electrical
conversion
hardware. Optical signal splitters are commonly known to one skilled in the
art
and available from numerous fiber optic component manufacturers such as
Gould, Inc. In the alternative, active splatters may also be utilized. In
addition, a
cascaded chain of passive or active splitters would further multiply the
number
of duplicated optical signals for application to an additional number of
optical
distribution nodes and t-herefore increase further the remote: units
serviceable by
a single head end. Such alternatives are contemplated in accordance with the
present invention as described by the accompanying claims.
The VHDT 34 can be located in a central office, cable TV head end, or a
remote site and broadcast up to about 112 NTSC channels. The VHDT 34
includes a transmission system like that of a LiteAMpT system available from
American Lightwave Systems, Inc., currently a subsidiary of the assignee
hereof.
Video signals are transmitted optically by amplitude modulation of a 1300
nanometer laser source at the same frequency at which the signals are received
(i.e. the optical transmission is a teraher=tz optical carrier which is
modulated
with the video signals). The downstream video transmission bandwidth is
about 54-725 MHz. One advantage in using the same frequency for optical
transmission of the video signal as the frequency of the video signals when
received is to provide high bandwidth transmission with reduced conversion
expense. This same-frequency transmission approach means that the modulation
downstream requires optical to electrical conversion or proportional
conversion
with a photodiode and perhaps amplification, but no frequency conversion. In
addition, there is no sample data bandwidth reduction and little loss of
resolution.
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An optical distribution node 18, shown in further detail in Figure 5,
receives the split downstream optical video signal from the splitter 38 on
optical
fiber feeder line 42. The downstream optical video signal is applied to a
downstream video receiver 400 of the optical distribution node 18. The optical
video receiver 400 utilized is like that available in the Lite AMp' product
line
available from American Lightwave Systems, Inc. The converted signal from
video receiver 400, proportionally converted utilizing photodiodes, is applied
to
bridger amplifier 403 along with converted telephony signals from downstream
telephony receiver 402. The bridger amplifier 403 simultaneously applies four
downstream electrical telephony and video sign-gals to diplex filters 406
which
allow for full duplex operation by separating the transmit and receive
functions
when signals of two different frequency bandwidths are utilized for upstream
and
downstream transmission. There is no frequency conversion performed at the
ODN 18 with respect to the video or the downstream telephony signals as the
signals are passed through the ODNs to the remote units via the coaxial
portion
of the 1C distribution network 11 in the same frequency bandwidth as they are
received at the ODNs 18,
After the ODN 18 has received the downstream optical video signals and
such signals are converted to downstrearn electrical video signals, the four
outputs of the ODN 18 are applied to four coaxial legs 30 of the coaxial
portion
of the HFC distribution network 11 for transmission of the downstream
electrical
video signals to the remote units 46. Such transmission for the electrical
video
signals occurs in about the 54-725 X4Hz bandwidth. Each ODN 18 provides for
the transmission on a plurality of coaxial legs 30 and any number of outputs
is
contemplated in accordance with the present invention as described in the
accompanying claims.
As shown in Figure 1, each coaxial cable leg 30 can provide a significant
number of remote units 46 with downstream electrical video and telephony
signals through a plurality of coaxial taps 44. Coaxial taps are commonly
known
to one skilled in the art and act as passive bidirectional pickoffs of
electrical
signals. Each coaxial cable leg 30 may have a number of coaxial taps 44
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connected in series. In addition, the coaxial portion of the HFC distribution
network 11 may use any number of amplifiers to extend the distance data can be
sent over the coaxial poition of such C distribution network 11.
Downstream video signals are provided from the coaxial taps 44 to the
remote units 46. The video signal from the coaxial tap 44 is provided to an
HISU 68 which is generally shown by the block diagram of ISU 100 in Figure 8.
The ISU 100 is provided with the downstream electrical video and telephony
signal from tap 44 and it is applied to diplex filter 104. The downstream
electrical video and telephony signal is passed through the diplex filter 104
to
both an ingress filter 105 and ISU modem 101. The do Cream video signal is
passed by the ingress filter 105 to video equipment via an optional set top
box
45. The downstream electrical telephony signal applied from the diplex. filter
104 to the ISU modem 101 is processed as described in ftirther detail below.
Ingress filter 105 provides the remote unit 46 with protection against
interference of signals applied to the video equipment as opposed to those
provided to other user equipment such as telephones or computer terminals.
Ingress filter 105 passes the video signals; however, is blocks those
frequencies
not utilized by the video equipment. By blocking those frequencies not used by
the video equipment, stray signals are eliminated that may interfere with the
other services by the network to at least the same remote unit.
The set top box 45 is an optional element at the remote unit 46.
Interactive video data from. set top box 45 would be tr smitted by an
additional
separate modem provided by the video service provider at a relatively low
frequency in the bandwidth of about 5 to 40 MHz, Such frequency must not be
one used for the transport of upstream and downstream telephony data and
downstream video.
For an MISU 6,651, a separate coaxial line from coaxial tap 44 is utilized to
provide transmission of video signals from the coaxial tap 44 to the set top
box
45 and thus for providing downstream tream video signals to video equipment 4-
11. The
ingress filter 105 as shown in Figure 8 is not a part of the MISU 66 as
indicated
by its dashed representation.
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Alternative embodiments of the V T 34 may employ other modulation
and mixing schemes or techniques to shift the video signals in frequency, and
other encoding methods to transmit the information in a coded format. Such
techniques and schemes for transmitting analog video data, in addition to
those
transmitting digital video data, are known to one skilled in the art and are
contemplated in accordance with the spirit and scope of the present invention
as
described in the accompanying claims.
TELEPHONY TRANSPORT
With reference to Figure 3, telephony information and ISU operations
and control data (hereinafter referred to as control data) modulated on
carriers by
MCC modem 82 is transmitted between the I.1DT 12 and the telephony
downstream transmitter 14 via coaxial lines 22. Telephony information and
control data modulated on carriers by ISUs 100 is received at telephony
upstream receiver 16 and communicated to the MCC modem 82 via coaxial
cable lines 28. The telephony downstream transmitter 14 and the telephony
upstream receiver 16 transmit and receive, respectively, telephony information
and control data via optical fiber feeder lines 24 and 26 to and from a
corresponding optical distribution node 18. The control data may include all
operations, administration, maintenance & provisioning (OAM&P) for providing
the telephony services of the system 10 and any other control data necessary
for
providing transport of telephony information between the T 12 and the ISUs
100.
A block diagram of the HDT 12 is shown in Figure :i. The HDT 12
includes the following n.iodules: Eight DSI Units (DS1U) (seven quad-DS I
units
48 plus one protection unit 50), one protection switch. test conversion unit
52
(PSTU), two clock & time slot interchange units 54 (CTSUs) (one active and one
standby/protection unit), six coax master units 56 (CXI\6Us) (three active and
three standby/protection units), two shelf control units 58 (SCNUs) (one
active
and one standby/protection unit), and two power supply units 60 (P Us) (two
load-sharing units which provide the appropriate T voltages from a central
office supply).
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The DS1U units can also be adapted to transfer data in the standard E1U
format, if desired.
The T 12 comprises all the common equipment functions of the
telephony transport of the communication system 10. The FIST 12 is normally
5 located in a central office and directly interfaces to a local digital
switch or
digital network element equipment. The HDT provides the network interface 62
for all telephony information. Each pi' accommodates from 2 to 28 DS -1
inputs at the network interface 62, representing a maximum of 672 DSO
channels. The HDT 12 also provides all synchronization for telephony transport
10 in the system 10. The HIDT 12 may operate in any one oftlar ee
synchronization
modes: external timing, line timing or internal timing. External timing refers
to
synchronization to a building integrated timing supply reference which is
sourced from a central office in which the HDT 12 is iocated. Line timing is
synchronized to the recovered clock from a DS -1 signal normally derived from
15 the local digital switch. Internal timing is a free- Wing or hold-over
operation
where the HDT maintains its own synchronization in the absence of any valid
reference inputs.
The HDT 12 also provides quarter-DSO grooming capabilities and
implements a 4096 x 4096 full-access, rton-blocking quarter-DSO (16 kbps)
20 cross-connect capability. This allows DSOs and quarter-DSOs (ISDN "D"
channels) to be routed from any timeslot at the DSO.-sI network interface 62
to
any customer serviced by any ISU 100.
The T 12 further provides the modern functionality required for
telephony transport over the HFC distribution network i 1 including the MCC
25 modern 82. The HDT :i 2 accommodates up to three active CXMUs 56 for
providing the modern interface to the HFC distribution network I 1 and also
provides one-for-one protection for each active C vl.U 56.
The T 12 coordinates the telephony transport system including
control and communication of many IS'Us of the mini-point to point
30 communication system 10. Each ITT 12 module performs a function. The
DS1U module 48 provides the interface to the digital network and DS: -1
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termination. The PSTU 52 provides DSIU equipment protection by switching
the protection DS I U 50 for a failed DS I U module 48. The CCTSU 54 provides
the quarter-DSO timesiot grooming capability and all. system synchronization
functions. The CTSU 54 also coordinates all call processing in the system. The
CXMU 56, described in further detail below, provides the modem functionality
and interface for the OFDM telephony transport over the HFC distribution
network I I and the SCNU 58 supervises the operation of the entire
communication system. providing all OAM&P ctions for telephony transport.
Most processing of requests for provisioning is performed by the SCNU 58.
Downstre Telelzorn rnsmitter
The downstream telephony transmitter 14, shown in Figure 4, takes the
coaxial outputs 22 from the active CXMUs 56 of the HDT 12 which carry
telephony information and control data and combines the outputs 22 into a
downstream telephony transmission signal. -the electrical-to-optical
conversion
logic required for the optical transmission is implemented in a stand-alone
downstream telephony transmitter 14 rather than in the HD` 12 to provide a
more cost effective transport solution. By placing this function in a separate
component, the expense of this function does not need to be replicated in each
CXMU 56 of the 1 T 12, This reduces the cost of the C. 56 finiction and
allows the CXMU 56 to transmit and receive over coax instead of fiber. 'Me
downstream telephony transmitter 14 also provides 1. )r transmission on
redundant downstream fiber feeder lines 24 to an ODN 18.
The downstream telephony transmitter 14 is co-located with the HD T 12
preferably within a distance of 100 feet or less. The downstream telephony
transmitter 14 receives the coaxial RF outputs from the active CXMUs 56, each
within a 6 MHz frequency band, and combines them at combiner 25 into a single
R.F signal. Each 6 MHz frequency band is separated by a guard band as is
known to one skilled in the art. Downstream telephony information is then
transmitted in about the 725-800 MHz frequency band. The telephony
transmitter 14 passes he combined signal through a I-to-2 splitter (not
shown),
thereby producing redundant downstream electrical si s.. The two redundant
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signals are each delivered to redundant laser transmitters 501 for electrical-
to-op-
tical conversion and 'die, redundant signals modulate an optical output such
that
the output of the downstream telephony transmitter 14 is on two optical feeder
lines 24, each having an identical signar modulated thereon. This provides
protection for the downstream telephony portion of the present system. Both
Fabry-Perot lasers in the telephony transmitter 14 are active at all times.
All pro-
tection functions are provided at the receive end of the optical transmissior.
(located at the ODN 18) where one of two receivers is selected as 91 active;
therefore, the telephony transmitter 14 requires no protection switching
capabilities.
Upstreafl2 Telep ony Deceiver
The upstream telephony receiver 16 performs the optical-to-electrical
conversion on the upstream optical telephony signals on the upstream optical
feeder lines 26 from the ODN 18. The -upstream telephony receiver 16 is
normally co-located in the central office with the HDT 12, and provides an
electrical coaxial output to the ]E DT 12, and a coaxial output 23 to be
provided to
a video set-top controller (not shown). Upstream telephony information is
routed via coax lines 28 from the upstream telephony receiver 16 to active
C 1bfUs 56 of the T 12. The coaxial link 28 between the T 12 and the
upstream telephony receiver 16 is preferably limited to a distance of 100 feet
or
less and is an intra-office link- Video set-top controller information, as
described
in the Video Transport section hereof, is located in a bandwidth of the RF
spectrum of 5-40 MHz which is not utilized for upstream telephony transport
such that it is transmitted along with the upstream telephony information.
The upstream telephony receiver 16 has dual receivers 502 for the dual
upstream optical fiber feeders lines 26. These feeder lines 26 carry redundant
signals from the ODN 18 which contain both telephony information and control
data and also video set top box information. The upstream telephony receiver
16
performs automatic protection switching on the upstream feeder lines 26 from
the ODN. The receiver 502 selected as "active" by protection logic is split to
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feed the coaxial outputs 28 which drive the HDT 12 and output 23 is provided
to
the set-top controller (not shown).
Optical Distribution No
Referring to Figure 5, the ODN 1 provides the interface between the
optical feeder lines 24 and 26 from the IT 12 and the coaxial portion of the
HFC distribution network 11 to the remote units 46. As such., the ODN 18 is
essentially an optical-to-electrical and electrical-to-optical converter. The
maximum distance over coax of any ISU 100 from an ODN 18 is preferably
about 6 km and the maximum length of the combined optical feeder line/coaxial
drop is preferably about 20 km. The optical feeder line side of the ODN 18
terminates six fibers although such number may vary. They include: a
downstream video feeder line 42 (single fiber from video splitter 38), a
downstream telephony feeder line 24 (from downstream telephony transmitter
14), a downstream telephony protection feeder line 24 (from downstream
telephony transmitter 14), an upstream telephony feeder line 26 (to upstream
telephony receiver 16), an upstream protection feeder line 26 (to upstream
telephony receiver 16), and a spare fiber (not shown). The t1DN 18 provides
protection switching functionality on the receive optical feeder lines 24 from
the
downstream telephony transmitter. The ODDN provides redundant transmission
on the upstream optical :feeder lines 26 to the upstream telephony receiver,
Protection on the upstream optical feeder lines is controlled at the upstream
telephony receiver 16. On the coaxial distribution side of DN 18, the ODNN
1.8
terminates up to four coaxial legs 30-
In the downstream direction, the ODN 18 includes downstream telephony
receiver 402 for converting the optical downstream telephony signal into an
electrical signal and a bridger amplifier 403 that combines it with the
converted
downstream video signal from downstream video receiver 400 terminated at the
ODN 18 from the VHDT 34. This combined wide-band electrical
telephony/video signal is then transported in the spectrum allocated for
downstream transmission, for example, the 725-800 MHz band, on each of the
four coaxial legs of the coaxial portion of the HFC distribution network 11.
As
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such, this electrical telephony and video signal is carried over the coaxial.
legs 30
to the ISUs 100; the bridger amplifier 403 simultaneously applying four
downstream electrical telephony and video signals to diplex filters 406. The
diplex filters 406 allow for full duplex operation by separating the transmit
and
receive functions when signals at two different frequency bandwidths are
utilized
for upstream and downstream transmission. There is no frequency conversion
available at the ODN 18 for downstream transport as the telephony and video
signals are passed through the ODN 18 to the remote units 46 via the coaxial
portion of HFC distribution network i 1 in the same frequency bandwidth as
they
are received at the ODN 18. As shown in Figure 1, each coaxial leg 30 can
provide a significant number of remote units 46 with downstream electrical
video and telephony signals through a plurality of coaxial taps 44. Coaxial
taps
44 commonly known to one skilled in the art act as passive bidirectional
pickoffs
of electrical signals. Each coaxial leg 30 may have a number of coaxial taps
connected in a series. In addition, the coaxial portion of the C distribution
network 11 may use any number of amplifiers to extend the distance data can be
sent over the coaxial portions of the system 10. The downstream electrical
video
and telephony signals are them provided to an 1SU 100 (Figure 8), which, more
specifically, may be an HISU 68 or an MISU 66 as shown in Figure 1.
In the upstream direction, telephony and set top box: information is
received by the ODN 19 at diplex filters 406 over the, four coaxial legs 30 in
the
.lip spectrum region from 5 to 40 MHz. The ODN 18 may include optional
frequency shifters 64 equipped on up to three of four coaxial legs 30. These
frequency shifters 64, if utilized, mix the upstream spectrum on a coaxial leg
to a
higher frequency prior to combining with the other three coaxial legs.
Frequency
shifters 64 are designed to shift the upstream spectrum in multiples of 50
Mliz.
For example, the frequency shifters 64 may be provisioned to mix the upstream
information in the 5-40 MHz portion of the spectrum to any of the following
ranges: 50 to 100 MHz., 100 to 150 MHz, or 150 to 200 MHz. This allows any
coaxial leg 30 to use the same portion of the upstream lip spectrum as another
leg without any spectrum contention when the upstream information is combined
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at the ODN 18. Provisioning of frequency shifters is optional on a coaxial leg
30. The ODN 18 includes combiner 408 which combines the electrical upstream
telephony and set top box information from all the coaxial legs 30 (which may
or
may not be frequency shifted) to form one composite upstream signal having all
5 upstream information present on each of the four coaxial legs 30. The
composite
electrical upstream signal is passively 1:2 split and each signal feeds an
upstream
Fabry-Perot laser transmitter which drives a corresponding upstream fiber
feeder
line 26 for transmission to the upstream telephony receiver 16.
Figure 6 illustrates an embodiment of a frequency shifter, indicated
10 generally at 64`, for use in ODN 18 of Figure 5. Frequency shifter 64'
comprises
a mixer 700 that is coupled to receive and shift the frequency band of RF
signals
in the upstream direction from diplex filter 406 for a coaxial leg 30. An
output
of mixer 700 is coupled through a bandpass filter 704 to combiner 408. Local
oscillator 702 is coupled to provide a signal to control the operation of
mixer
15 700.
In operation, frequency shifter 64' shifts a block of RF signals from a first
frequency range to a second frequency range. For example, as mentioned above,
the RF signals provided to frequency shifter may comprise RF signals in the
range from 5 to 40 In one embodiment, ODN 1.8 comprises three
20 frequency shifters 64. In this embodiment, the local oscillators 702 of the
three
frequency shifters provide signals of 76 MHZ, 149 Ml , and 222 Z,
respectively. Thus, frequency shifters 64' respectively shift the upstream RF
signals approximately to the 50 to 100 MI, 125 to 175 and 200 to 250
M-HZ ranges.
25 If the upstream telephony and set top box signals are upshifted at the
ODN 18, the upstream telephony receiver 16 includes frequency shifters 31 to
downshift the signals according to the upshifting done at the DN 18. A
combiner 33 then combines the downshifted signals for application of a
combined signal to the HDT 12. Such downshifting and combining is only
30 utilized if the signals are. upshifted at the ODN 18.
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Figure 7 illustrates an embodiment of a frequency shifter, indicated
generally at 31', for use in telephony upstream receiver 16 of Figure 8.
Frequency shifter 31' returns a block of RF signals shifted by frequency
shifter
64' to original frequency range of the block. For example, frequency shifter
31'
may return a block of RF signals to 5 to 40 MHZ from 50 to 100 MHZ .
As discussed in more detail below, the upstream telephony signals
processed by frequency shifters 31' and 64' are typically OFDM signals. Thus,
frequency shifters 64' must return the RF signals to the original frequency
range
without introducing adverse phase and frequency errors. To reduce the
likelihood of this corruption of the OFDM signals, frequency shifter 31' locks
its
local oscillator to the local oscillator of a corresponding frequency shifter
64'
using a pilot tone transmitted from ODN 18 to telephony upstream receiver 16.
Frequency shifter 31' includes a bandpass filter 706 that is coupled to
receive an signal from ODN 18. Bandpass filter 706 is coupled to a splitter
708. Splitter 708 is coupled to provide the RF signal to an input of mixer
718.
Further, splitter 708 provides a second output that is used to generate a
local
oscillator signal for mixer 718. This local oscillator signal is phase locked
with a
corresponding local oscillator 702 of frequency converter 64'. This second
output of splitter 708 is coupled to phase detector 712 through bandpass
filter
710. Phase detector 712 is coupled to provide a control signal to voltage
controlled oscillator 714. Voltage controlled oscillator 714 is coupled
through
splitter 716 to provide the local oscillator signal to mixer 718. Splitter 716
further provides a feedbback signal to phase detector '712..
In operation, phase detector 712 phase locks local oscillator signal of
frequency shifter 31' with local oscillator 702 of a corresponding frequency
shifter 64'. Phase detector 712 compares the pilot tone from ODN 18 with the
feedback signal from voltage controlled oscillator 714 to generate the control
signal for voltage controlled oscillator 714. Consequently, the local
oscillator
signal provided to mixer 718 is phase locked with the corresponding local
oscillator 702 of frequency shifter 64'. Mixer 718 uses the local oscillator
signal
from splitter 716 and voltage controlled oscillator 714 to shift the block of
RF
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signals received by frequency shifter 31 to the original frequency range of
the
block of RF signals. Advantageously, unacceptable modifications of the OFDM
upstream signal by frequency shifters 64' and 31 B are thus avoided.
13ni
Integrated Services
Referring to Figure 1, the ISUs 100, such as HISU 68 and IISU 66,
provide the interface between the HFC distribution network 11 and the customer
services for remote units 46. Two basic types of ISUs are shown, which provide
service to specific customers. Multiple user integrated service unit 66
(MISUs)
may be a multiple dwelling integrated service unit or a business integrated
service unit. The multiple dwelling integrated service unit may be used for
mixed residential and business environments, such as multi-tenant buildings,
small businesses and clusters of homes. These customers require services such
as plain old telephone service (POTS), data services, DS I services, and
standard
TR-57 services. Business integrated service units are designed to service
business environments. They may require more services, for example, data
services, ISDN, DS 1 services, higher bandwidth services, such as video confer-
encing, etc. Home integrated services units 68 SUs) are used for residential
environments such as single-tenant buildings and duplexes, where the intended
services are POTS and basic rate integrated digital services network (ISDN).
Description for ISUs shall be limited to the HISUs and MISUs for simplicity
purposes as multiple dwelling and business integrated service units have
similar
functionality as far as the, present invention is concerned.
All ISUs 100 implement RF modem functionality and can be generically
shown by ISU 100 of Figure 8. ISU 100 includes ISU modem 101, coax slave
controller unit (CXSU) 1. 02, channel units 103 for providing customer service
interface, and diplex filter/tap 104. In the downstream direction, the
electrical
downstream telephony and video signal is applied to diplex filter/tap 104
which
passes telephony information to ISU rn6dem 101 and video information to video
equipment via an ingress filter 1 05 in the case of a HISU. When the ISU 1 00
is
a I SU 66, the video information is rejected by the diplex filter. The ISU
modem. 10 1 demodulates the downstream telephony information utilizing a
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modem corresponding to the MCC modem 82 used for modulating such
information on orthogonal multicarriers at HDT 12. ISU 100 demodulates
downstream telephony :information from a coaxial distribution leg 30 in a
provisionable 6 MHz frequency band. Timing generation 107 of the ISU modem
101 provides clocking for CXSU 102 which provides processing and controls
reception and transmission by ISU modem 101. The demodulated data from ISU
modem 101 is passed to the applicable channel units 103 via CXSU 102
depending upon the service provided. For example, the channel units 103 may
include line cards for POTS, DS I services, ISDN, other data services, etc.
Each
ISU 100 provides access to a fixed subset of all channels available in a 6 MHz
frequency band corresponding to one of the C s of . T 12. This subset of
channels varies depending upon the type of ISU 100. An MISU 66 may provide
access to many DSO channels in a 6 z frequency band, 'le an HISU 68
may only provide access to a few DSO channels.
The channel units 103 provide telephony information and control data to
the CXSU 102, which provides such data to ISU modem 101 and controls ISU
modem 101 for modulation of such telephony data and control data in a
provisional 6 MHz frequency band for transmission onto the coaxial
distribution
leg 30 connected thereto. The upstream: 6 MHz frequency band provisionable
for transmission by the ISU 100 to the HOT 12 corresponds to one of the
downstream 6 MHz bands utilized for transmission by the CXIMIUs 56 of HDT
12.
The CXSI_J 102 which applies demodulated data from the ISU modem 101
to the applicable channel units, performs data integrity checking on the
downstream 10 bit DSO+ packets received from the ISU modem 101. Each ten
bit 13SO+ packet as described below includes a parity or data integrity bit.
The
CXSU 102 will check the parity of each downstream 10 bit DS 0+ channel it
receives. Further, the parity of each upstream DSO+ received from the channel
units 103 is calculated and a parity bit inserted as th;.-; tenth bit of the
upstream
DSO+ for decoding and identification by the HDT 12 of an error in the upstream
data. If an error is detected by CXSU 102 when checking the parity of a
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downstream 10 bit DSO+ channel it receives, the parity bit of the
corresponding
upstream channel will be intentionally inverted to inform the HDT 12 of a
parity
error in the downstream, direction. Therefore, the upstream parity bit is
indicative of errors in the downstream DSO+ channel and the corresponding
upstream DSO+ channel. An example of such a parity bit generation process is
described in U.S. Patent No. 5,519,830 entitled Point to Multipoint
Performance Monitoring and Failure Isolation System". This upstream parity bit
is utilized in channel monitoring as described further below., As would be
apparent to one skilled in the part, the parity checking and generation may be
performed, at least in part, in other elements of the IS U or associated
therewith
such as the channel units.
Each ISU 100 recovers synchronization from downstream transmission,
generates all clocks required for ISU data transport and locks these clocks to
the
associated HDT timing. The ISUs 100 also provide call processing functionality
necessary to detect customer line seizure and line idle conditions and
transmit
these indications to the HDT 12. ISUs 100 terminate and receive control data
from the HDT 12 and process the control data received therefrom. Included in
this processing are messages to coordinate dynamic channel allocation in the
communication system 10. Finally, ISUs 100 generate ISU operating voltages
from a power signal received over the HFC distribution network 11 as shown by
the power signal 109 taken from diplex filter/tap 104.
Data Path in T
The following is a detailed discussion of the data path in the host digital.
terminal (HDT) 12. Referring to Figure 3, the data path between the network
facility at the network interface 62 and the downstream telephony transmitter
14
proceeds through the DS I U'48, CTSU 54, and C Cl 56 modules of the HDT
12, respectively, in the downstream direction. Each DS 1 U 48 in ties T 12
takes f o u r DS 1 s from the, network and formats this inibrrnati'.on into
four 24-
channel, 2.56 Mbps data streams of modified DSO signals referred to as CTSU
inputs 76. Each DSO in the CTSU input has been modified by appending a ninth
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bit which can carry rnultifrarne timing, signaling information and
control/status
messages (Figure 9). This modified DSO is referred to as a "DSO+." The ninth
bit signal (NBS) carries a pattern which is updated each frame and repeats
every
24 frames. This maps each 64 kbps DSO from the network into a 72 kbps DSO+.
5 Thus, the twenty-four DSO channels available on each OSI are formatted along
with overhead information into twenty-four DSO+ channels on each of four
CTSU input streams.
The ninth bit signaling (.SIBS) is a mechanism developed to carry the
multiframe timing, out-of-band signaling bits and miscellaneous status and
10 control information associated with each DSO between the DS1TJ and the
channel units. Its main functions are to carry the signaling bits to channel
units
103 and to provide a muitiframe clock to the channel units 103 so that they
can
insert upstream bit signaling into the DSO in the correct frame of the
multiframe.
Because downstream DSOs may be coming from DS is which do not share the
15 same multiframe phase each DSO must carry a multiframe clock or marker
which
indicates the signaling frames associated with the origination DS1. The NBS
provides this capability. Ninth bit signaling is transparent to the OFDM modem
transport of the communication system 10.
Up to eight DSIUs 48 may be equipped in a single HDT 12; including
20 seven active DS I Us 48 and a protection DS I U module 50. Thus, 32 CTSU
inputs are connected between the DS IUs and the CTSUs 54 but a maximum of
28 can be enabled to carry traffic at any one time. The four remaining CTSU
inputs are from either the protection DS I U or a failed DS I U. The PSTU
includes switch control for switching the protection DS IU 50 for a failed DS
1 U.
25 Each CTSU input is capable of carrying up to 32, 10-bit channels, the first
24 channels carry DSO+s and the remaining bandwidth is unused. Each CTSU
input 76 is clocked at 2.56 Mbps and is synchronized to the 8 kHz internal
frame
signal (Figure 11). 'T'his corresponds to 320 bits per 125 p sec frame period.
These 320 bits are framed as shown in Figure 9. The fourteen gap bits 72 at
the
30 beginning of the frame carry only a single activity pulse in the 2nd bit
position,
the remaining 13 bits are not used. Of the following 288 bits, the first 216
bits
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normally carry twenty-four DSO+ channels where each DSO+ corresponds to a
standard 64 kbps DSO channel plus the additional 8 kbps signaling bit. `T'hus,
each DSO+ has a bandwidth of 72 kbps (nine bits every 8 kHz frame). The
remaining 72 bits are reserved for additional DSO+ payload channels. The final
eighteen bits 74 of the frame are unused gap bits.
The clock and time slot interchange unit 54 (CTSU) of the I-IDT 12 takes
information from up to 28 active CTSU input data streams 76 and cross-connects
them to up to twenty-.four 32-cx el., 2.56 Mbps output data streams 78 which
are input to the coax master units (C s) 56 of the HDT 12. The format of
the data streams between the CTSU 54 and the CXMUs 56 is referred to as a
CTSU output. Each OTSU output can also carry up to 32, 10-bit channels like
the CTSU input. The first 28 carry traffic and the remaining bandwidth is
unused. Each CTSU output is clocked at 2.56 Mbps and is synchronized to the 8
kHz internal framing signal of the HDT 12 (Figure I1), This corresponds to 320
bits per 125 tisec frame period. The frame structure for the 320 bits are as
described above for the CTSU input structure.
The HDT 12 has the capability of time and space manipulation of quarter-
DSO packets (16 kbps), This function is implemented with the time slot
interchange logic that is part of CTSU 54. The CTSU implements a 4096 x 4096
quarter-DSO cross-connect function, although not all time slots are utilized.
In
normal operation, the CTSU 54 combines and relocates up to 672 downstream
DSO+ packets (or up to 2688 quarter-DSO packets) arranged as 28 CTSU inputs
of 24 DSO+s each, into 720 DSO+ packets (or 2880 quarter-DSO packets)
arranged as 24 CTSU outputs of 32 DSOs each.
The system has a maximum throughput of 672 DSO+ packets at the
network interface so not all of the CTSU output bandwidth is usable. If more
than the 672 channels are assigned on the "CTSU output" side of the CTTSU,
this
implies concentration is being utilized. Concentration is discussed father
below.
Each CXMU 56 is connected to receive eight active CTSU outputs 78
from the active CTSU 54. The eight CTSU outputs are clocked by a 2.56 MHz
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system clock and each carries up to 32 DSOs as described above. The DSO+s
are further processed by the CXNIU 56 and a tenth parity bit is appended to
each
DSO- resulting in a 10 bit DSO+. These 10 bit packets contain the DSO, the
NBS (ninth bit signal) and the parity or data integrity bit (Figure 10). The
10 bit
packets are the data transmitted on the HFC distribution network 11 to the
ISUs
100, The 10th bit or data integrity bit inserted in the downstream channels is
decoded and checked at the ISU and utilized to calculate and generate a parity
bit
for corresponding channels in the upstream as described above. This upstream
parity bit which may be representative of an error in the downstream or
upstream
channel is utilized to provide channel protection or monitoring as further
described herein.
CTSU 54 is also capable of applying a conventional Reed-Soloman code
to transmitted payload data for running error correction. Such codes carry an
overhead which must be borne by the inclusion of error-correcting symbols in
each block or message transmitted. Generally, two check symbols are required
to correct one corrupted data symbol in a message. ;The incorrect symbol may
contain any number of errors in its 5 bits, as long as all bit errors are
confined to
the same symbol. But even a single incorrect bit in each of two symbols counts
as two errors.) Shoat messages impose less computational overhead on a system,
but can correct fewer errors in the message. Conversely, long messages require
more computation and more latency before the corrections can be applied, but
their error-correction ability is greater. Figure 54 represents, for an
example
system, the probability of an uncorrectable error in a :'L'ame for various
error
probabilities in one individual symbol. The solid curve shows the error
performance for a 21-frame message having 19 fi-ames of data symbols and two
frames of error-correction code; the dashed curve represents a 41-frame
message
having 37 data and four code frames; the dotted curve gives the best
performance, with 73 data frames and eight code frames in an 81-frame message.
The present system allows a choice of different error-correction abilities
for different types of data. For example, voice data is highly redundant, and
needs little defense against errors. Financial transaction data, on the other
hand,
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wants a large degree of data integrity. In addition, it may be desirable to
allow a
user to select -- and pay for -- whatever degree of error correction that he
desires.
CTSU 54, Figure 3, includes a conventional "provisioning table", which
specifies a number of parameters relating to particular payload channels.
Figure 55 shows a provisioning table 4411 having an added column containing
indications for several, different amounts of error protection. In method
4410,
step 4412 reads the entry for a particular channel to, be set up. In this
implementation, the entry may specify message lengths of 21, 41, or 3 1 bits,
respectively having the ability to correct 1, 2, or 4 symbols; the entry may
also
specify no correction, in which case message blocks do not apply. Step 4413
encodes the table entry in an IOC message and sends it to the ISU whose
address
appears in that row of table 4111. .1 general-purpose processor in CXSU 102 of
the ISU stores the frame length in step 4414. As the CXSU receives data from
modem 101, Figure 9, it decodes the frames of an entire message, 4415, then
decodes the check symbols for the message, 4416, and signals an error, 4417,
if
one exists in the message. Steps 4415-4417 repeat. for subsequent messages.
The ISU employs the same process to send frames upstream to the head end,
using the frame length setting specified in step 4414.
Within both CXS .3 102 at the ISU and the C IU, Figure 12 at the HDT,
a 21-frame message or block requires 19 symbol or frame times to decode the
message, then has two frames of latency while its two check symbols are
decoded. A 41-frame message uses four frames of time for computation of any
errors from the four check symbols following its 37 data symbols. An 81-frame
message presents any error indication 8 frame times after the end of its 73
data
frames. (One extra frame of delay is imposed in the downstream direction due
to
remapping at the HDT.) If all messages were to start at the same time for all
channels in an entire band, the computational load in the 1{DT would peak
during the check-symbol frames, and would be lower at other times. Since the
processor must be capable of handling the peak loads, its power is
underutilized
at other times.
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The present system allows a lower-capacity processor to handle error
correction by staggering the beginning times of different messages in
different
subbands of channels, so that not all of them conic due at the same time. That
is,
the start of a message in any channel of a subband is offset from the start
'of a
message or "multiframe" signal, to be described, by a predetermined number of
frame times of 125 sec. each. The table below shows how the 24 subbands of
Figure 16 are offset, for each message length which can be selected.
Table I
Subband 21 fimmes 41 frames 81 frames
0 0 0 0
1 0 0
2 0 2
3 0 3
4 2 4
5 2 5
6 3 6
7 3 i
8 2 4 8
9 2 4 9
10 2 5 10
11 2 5 1,-
12 3 12
Only 13 subband settings are required, since no unit tunes more than 130
channels. Giving all 10 channels of each subband he same offset does not
overload the processors of the remote units. However, the head end ( T),
which receives and transmits all chamnnels, can enjoy significant relief from
not
having to encode or decode the check symbols for all channels at the same
time.
Figure 56 shows steps 4120 for performing fame staggering. Step 4421
repeats method 4420 for all active payload channels. Step 4422 accesses the
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current messages for the channels in one subband. Step 4423 calculates the 1,
2,
or 4 Reed-Soloman check words for the 21, 41, or 81 message data words. Step
4424 waits N frames past the start of a nxultiframeõ whereupon step 4425 sends
the message to modem 82, Figure 3 for transmission.
5 At a remote ISU, CXSU performs the same steps 4420 for upstream
messages. Step 4422 is required only in an MISU, because all channels in an
HISU reside in the same subband. Method 4420 may be performed at both ends
of the system, as described herein; it may also be performed only at one end,
either T or ISU. Staggering from the ISU to the T is preferable if only
10 one end is staggered, because the most critical processing load is the
error-
correction of all 240 channels in the upstream receiving modem, Figure 26.
The use of error-correcting codes along with unencoded data raises
problems in a real-time transport system. Data arrives from the trunk line 20,
Figure 1, at a constant rate. This data must be transmitted downstream in the
15 same time duration, whether it is encoded along the way, or sent unencoded.
Likewise, upstream data usually must be transmitted at the same rate whether
or
not it is encoded. That is, the use of error-correcting codes must be time-
transparent at both ends of the system. But error-correcting codes require the
transmission of check digits or symbols along with the data. The present
system.
20 resolves this difficulty by packing the data words differently if they are
encoded.
As explained above, the basic unencoded word length for a DSO + channel is ten
bits: eight data bits, a. signaling (NBS) bit, and a parity bit. When encoding
is
used, however, this format is changed to nine-bit words, with a single parity
bit
for the entire message. This is the reason for the choice of frame sizes for
the
25 encoded modes. A 21-frame message contains 19 data frames, which would
ordinarily be transmitted as 10 x 19--190 bits. Those same data frames,
packaged
as nine-bit words along with two nine-bit check words, require (19+2)X9=189
bits; adding one more parity bit covering the entire message lock gives 190
bits,
the same number as that required for the unencoded version of the same data.
30 The 41-frame message.- has 37 frames of data, or 370 bits in unencoded 10-
bit
format. Encoded as 37 nine-bit words along with four check words, the same
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message requires (37+4)x9=369 bits; again, a single additional parity bit
yield
the 370 bits of the same data in unencoded form. The 81-frame format has 73
data words, 8 check words, and a parity bit, yielding the same number of bits
as
73 data words in 10-bit form.
There are many other combinations of numbers which yield similar
results. These can be found heuristically without a great deal of
experimentation.. The first step is to estimate rough numbers of large (parity-
bearing) words in one or more message sizes, and the number of errors desired
to
be correctable for each size. The next step is to determine a number of
smaller
(non-parity) words that carry the same amount of data, but which form a total
message the same size or slightly smaller than the total number of bits in the
large-word format. Any excess bits then are assigned to parity over the block
or to any other function, for that matter. For example, if two bits are left
over
instead of one for each message, they could represent two parity bits over the
message, two control or format-designation bits, etc. The use of check
symbols,
of course, greatly reduces the need for parity or other forms of error
detection.
In fact, while the present system uses the message-parity bit as parity in the
downstream direction, the ISti deliberately sets the parity bit to an
incorrect
value in an upstream message if it was incorrect in the downstream message.
This serves to signal the HDT that a bit error was encountered, when the HDT
would not otherwise be aware of it; this in turn allows the HDT to keep more
accurate statistics on channel quality for reallocating channels, or for other
purposes.
Figure 57 shows a method 4430 for adding the "code packing" feature to
the method 4420 of Figure 56. Step 4431 repeats the steps for all channels.
Step
4432 determines whither the data for the channel is to be encoded or not. If
not,
step 4433 merely transmits it word-by-word to t e modem. If it is to be
encoded, step 4434 strips the parity for other) bit(s) from each word. After
step
4435 has formed the check words, step 4436 calculates the message-wide parity,
or other desired function. Thereafter, step 4437 waits the proper number of
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frame times (as specified by method 4420, and step 4338 sends the message to
the modem as before.
In the upstream direction, the reverse path through the HDT is
substantially a mirror of the forward path through the HDT 12. For example,
the
tenth parity bit is processed at the CXMU 56 and the signal from the CXMU 56
to the CTSU 54 is in the format of Figure 9.
The round trip delay of a DSO is the same for every data path. The time
delay over the path from the downstream CTSU output, through CXMU 56, over
the HFC distribution network to the ISU 100 and then from the ISU 100, back
over the 14FC distribution network 11, through CXMU 56 and to CTSU 54 is
controlled by upstream synchronization, as described in detail below.
Generally,
path delay is measured for each ISU and if it is not the correct number of
frames
long, the delay length is adjusted by adding delay to the path at the ISU 100.
Coax Master Unit (CXMU
The coax master unit 56 (CXM', shown in Figure 3, includes the coax
master card logic 80 (CXMC) and the master coax card (MCC) modem 82. As
previously described, up to six CXMUs may be equipped in an HDT 12. The 6
CXMUs 56 include three pairs of CXMUs 56 with each pair providing for
transmit in a 6 z bandwidth. Each pair of CXI%4Us 56 includes one active
CXMU and a standby CXMU. Thus, one to one protection for each CXJVRJ is
provided. As shown in Figure 3, both C s of the pair are provided with
upstream telephony data from the upstream telephony receiver 16 and are
capable of transmitting via the coaxial line 22 to the downstream telephony
transmitter 14. As such, only a control signal is required to provide for the
one-
to-one protection indicating which CXMU 56 of the pair is to be used for
transmission or reception.
Coax .stet' Card Logic (C. C)
The coax master card logic 80 (C C) of the CX U 56 (Figure 12),
provides the interface between the data signals of the HDT 12, in particular
of
the CTSU 54, and the.. modem interface for transport of data over the HFC
distribution network 11. The C C 80 interfaces directly to the MCC modem
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82. The CXMC 80 also implements an ISU operations channel transceiver for
multi-point to point operation between the HDT 12 and all ISUs 100 serviced in
the 6 MHz bandwidth in which the CXMU 56 controls transport of data within.
Referring to Figure is 2, the CXMC includes controller and logic 84,
downstream
data conversion 88, upstream data conversion 90, data integrity 92, IOC
transceiver 96, and timing generator 94.
Downstream data conversion 88 performs the conversion from the nine-bit
channel format from CTSU 54 (Figure 9) to the ten-bit channel format (Figure
10) and generates the data integrity bit in each downstream channel
transported
over the HFC distribution network 11. The data integrity bit represents odd
parity. Downstream data conversion 88 is comprised of at least a FIFO buffer
used to remove the 32 gap bits 72, 74 (Figure 9) present in the downstream
CTSU outputs and insert the tenth, data integrity bit, on each channel under
control of controller and logic 84.
The upstream data conversion 90 includes a least a FIFO buffer which
evaluates the tenth bit (data integrity) appended to each of the upstream
channels
and passes this information to the data integrity circuitry 92. The upstream
data
conversion 90 converts the data stream of ten-bit channels (Figure 10) back to
the nine-bit channel format (Figure 9) for application to CTSU 54. Such
conversion is performed under control of controller and logic 84.
The controller and logic 84 also manages call processing and channel
allocation for the telephony transport over the HFC distribution network I 1
and
maintains traffic statistics over the HFC distribution network I i in modes
where
dynamic time-slot allocation is utilized, such as for providing TR-303
services,
concentration services commonly known to those skilled in the art. In
addition,
the controller 84 maintains error statistics for the channels in the 6 MHz
band in
which the CXMU transports data, provides software protocol for all ISU
operations channel communications, and provides control for the corresponding
MCC modem 82.
The data integrity 92 circuitry processes the output of the tenth bit
evaluation of each upstream channel by the upstream conversion circuit 90. In
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the present system, parity is only guaranteed to be valid on a provisioned
channel
which has a call in progress. Because initialized and activated ISU
transmitters
may be powered down when the ISUs are idle, the parity evaluation performed
by the CXMC is not always valid. A parity error detected indicates either a
transmission error in an upstream channel or a transmission error in a
downstream channel corresponding to the upstream channel.
'Me ISU operations channel (IOC) transceiver 96 of the CXMC 80
contains transmit buffers to hold messages or control data from the controller
and logic 84 and loads these IOC control messages which are a fixed total of 8
bytes in length into a 64 kbps channel to be provided to the MCC modem 82 for
transport on the 14FC distribution network 11.
In this implementation, all IOC channels carry the same information at all
times. That is, the IOC messages are broadcast simultaneously over all the
channels. This allows the use of inexpensive and rugged narrow-band modems
in the ISUs, reserving the more expensive and critical wideband models for the
HDT, which uses only one modem for an entire 6TMHz band, and which can be
located centrally in a controlled environment. In the upstream direction, the
IOC
transceiver receives the 64 kbps channel via the MCC modem 82 which provides
the controller and logic 84 with such messages.
The timing generator circuit 94 receives redundant system clock inputs
from both the active and protection CTSUs 54 of the HD T 12. Such clocks
include a 2 kHz HFC multiframe signal, which is generated by the CTSU 54 to
synchronize the round trip delay on all the coaxial legs of the HFC
distribution
network. This signal indicates multiframe alignment on tl e ISU operations
channel and is used to synchronize symbol timing and data reconstruction for
the
transport system. A 8 k m e signal is provided for indicating the first "gap"
bit of a 2.56 Mldz, 32 channel signal from the CTSU 54 to the CXMU 56.A
2.048 MHz clock is generated by the CTSU 54 to the SCNU 58 and the CXMU
56. The CXMU 56 uses this clock for ISU operations channel and modem
communication between the C xC 80 and the MCC modem 82. A 2.56 MHz
bit clock is used for transfer of data signals between the USIUs 48 and CTSUs
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54 and the CTSUs 54 and. C Cs 56. A 20.48 MHz bit'. clock is utilized for
transfer of the 10-bit data channels between the CXMC and the MCC.
r Coate Card (MCC) Modem
The master coax card (MCC) modem 82 of the CXMU 56 interfaces on
5 one side to the CXM.C 80 and on the other side to the telephony transmitter
14
and receiver 16 for transmission on and reception from the f-IFC distribution
network 11. The MCC modem 82 implements the modem functionality for
OFDM transport of telephony data and control data. The block diagram of
Figure 3 identifies the associated interconnects of the MCC modem 82 for both
10 upstream and downstream communication. The MCC modem 82 is not an
independent module in the HDT 12, as it has no interface to the HDT 12 other
than through the CXMC 80 of the CXMU 56. The MCC modem 82 represents
the transport system logic of the HDT 12. As such, it is responsible for
implementing all requirements associated with information transport over the
15 HFC distribution network 11. Each MCC modem 82 of the CXMUs 56 of RDT
12 is allocated a maximum bandwidth of 6 MHz in the downstream spectrum for
telephony data and control data transport. The exact location of the 6 MHz
band
is provisionabie by the CXMC 80 over the communication interface via the IOC
transceiver 96 between the CXMC 80 and MCC modem 82. The downstream
20 transmission of telephony and control data is in the :F spectrum of about
725 to
800 MHz.
Each MCC modern 82 is allocated a maximum of 6 MHz in the upstream
spectrum for receipt of control data and telephony data from the ISUs within
the
RF spectrum of about 5 to 40 MHz. Again, the exact location of the 6 MHz
25 band is provisionable by the CXMC 80 over the communication interface
between the CXMC 80 and the MCC modem 82.
The MCC modem 82 receives 256 DSO+ channels from the CXMC 80 in
the form of a 20.48 MHz signal as described previously above. The MCC
modem 82 transmits this information to all the ISUs 1 00 using the
multicarrier
30 modulation technique based on OFDM as previously discussed herein. The
MCC modem 82 also recovers 256 DSO+ multicarner channels in the upstream
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transmission over the IFC distribution network and converts this information
into a 20.48 Mbps stream which is passed to CXN4C 80. As described
previously, the multicarrier modulation technique involves encoding the
telephony and control data, such as by quadrature amplitude modulation, into
symbols, and then performing an inverse fast Fourier transform technique to
modulate the telephony and control data on a set of orthogonal multicarriers.
Symbol alignment is a necessary requirement for the multicarrier
modulation technique implemented by the MCC modern 82 and the ISU modems
101 in the ISUs 100. In the downstream direction of transmission, all
information at an IST 100 is generated by a single C J 56, so the symbols
modulated on each multicarrier are automatically phase aligned. However,
upstream symbol alignment at a receiver of the MCC modem 82 varies due to
the multi-point to point nature of the HFC distribution network 11 and the
unequal delay paths of the ISUs 100. In order to maximize receiver efficiency
at
the MCC modem 82, all upstream symbols must be aligned within a narrow
phase margin. This is done by utilizing an adjustable delay parameter in each
ISU 100 such that the symbol periods of all channels received upstream from
the
different ISUs 100 are aligned at the point they reach the HDT 12. This is
part
of the upstream synchronization process and shall be described further below.
In
addition, to maintain orthogonality of the multicarriers, the carrier
frequencies
used for the upstream transmission by the ISUs 100 must be frequency locked to
the HDT 12.
Incoming downstream information from the CXMC 80 to the MCC
modem 82 is frame aligned to the 2 kHz and 8 kHz clocks provided to the MCC
modem 82. The 2 kFiz multi-frame signal is used by the MCC modem 82 to
convey downstream symbol timing to the ISUs as described in further detail
below. This multifraane clock conveys the channel correspondence and indicates
the multi-carrier frame structure so that the telephony data may be correctly
reassembled at the ISU 100. Two kHz represents the greatest common factor
between 10 kHz (the modem symbol rate) and 8 kHz (the data frame rate).
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All ISUs 100 will use the synchronization information inserted by the
associated MCC modem 82 to recover all downstream timing required by the
ISUs 100. This synchronization allows the ISUs 100 to demodulate the
downstream information and modulate the upstream transmission in such a way
that all ISU 100 transmissions received at the HDT 12 are synchronized to the
same reference. Thus, the carrier frequencies used for all ISU 100 upstream
transmission will be frequency locked to the HDT 12.
The symbol alignment is performed over synchronization channels in the
downstreaam and upstream 6 MHz bandwidths under the responsibility of the
MCC modem 82, in addition to providing path delay adjustment, initialization
and activation, and provisioning over such synchronization channels until
initialization and activation is complete as ft-ther described herein. These
parameters are then tracked by use of the IOC channels. Because of their
importance in the system, the IOC channel and synchronization channels may
use a different modulation scheme for transport of control data between the
MCC modem 82 and ISUs 100 which is more robust or oflesser order (less
bits/sec/Hz or bits/symbol) than used for transport of telephony data. For
example, the telephony data may be modulated using quadrature amplitude
modulation, while the IOC channel and synchronization channel may be
modulated utilizing BPSK modulation techniques.
The MCC modem 82 also demodulates telephony and control data
modulated on multicarriers by the IS Us 100_ Suck). demodulation is described
further below with respect to the various embodiments of the telephony
transport
system.
Functions with respect to the OFDM transport system for which the MCC
modem 82 is responsible, include at least the following, which are further
described with respect to the various embodiments in further detail. The MCC
modem 82 detects a, :received amplitude/level of a. synchronization
pulse/pattern
from an ISU 100 within a synchronization c 4el and passes an indication of
this level to the CXMC 80 over the communication interface therebetween. The
CXMC 80 then provides a command to the MCC rr odem 82 for transmission to
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the ISU 100 being leveled for adjustment of the amplitude level thereof. The
MCC modem 82 also provides for symbol alignment of all the upstream
multicarriers by correlating an upstream pattern modulated on a
synchronization
channel with respect to a known symbol boundary and passing a required symbol
delay correction to the CXMC 80 over the communication interface
therebetween. The CX:MC 80 then transmits via the MCC modem 82 a message
downstream to the ISU 100 to adjust the symbol delay of the ISU 100.
Likewise, with regard to synchronizing an ISU 100 for overall path delay
adjustment, the MCC modern 82 correlates an upstream multiframe pattern
modulated in the proper bandwidth by the ISU 100 on the IOC channel with.
respect to a known reference boundary, and passes a required path delay
correction to the CX1VJ'C 80 over the modem interface therebetween. The CXMC
80 then transmits via the MCC modem 82 over the IOC channel a message
downstream to adjust the overall path delay of an ISU 100.
S arv of Bidirectional Multi-Point to Point Te . honer s rt
The following summarizes the transport of telephony and control
information over the FIFC distribution network 11. Each CXMU 56 of HUT 12
is provisioned with respect to its specific upstream and downstream operating
frequencies. The bandwidth of both upstream and downstream transmission by
the CXMU 56 are a maximum of 6 MHz, with the downstream transmission in a
6 MHz band of the spectrum of about 725-800 .
In the downstream direction, each. MCC modem 82 of the CXM`U 56
provides electrical telephony and control data signals to the downstream
telephony transmitter 14 via coaxial lire 22 in its provisional 6 Nfflz
bandwidth.
The RF electrical telephony and control data signals from the MCC modems 82
of the HDT 12 are corribined into a composite signal. The downstream
telephony transmitter then passes the combined electrical signal to redundant
electrical-to-optical converters for modulation onto a pair of protected.
downstream optical feeder lines 24.
The downstream optical feeder lines 24 carry the telephony information
and control data to an ODN 18. At the ODN 18, the optical signal is converted
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back to electrical and combined with the downstream video information (from
the video head-end feeder line 42) into an electrical downstream RF output
signal. The electrical RF output signal including the telephony information
and
control data is then fed to the four coaxial distribution legs 30 by ODN 18.
All
telephony information and control data downstream is broadcast on each coaxial
leg 30 and carried over the coaxial portion of the IIFC distribution network
11.
The electrical downstream output RF signal is tapped from the coax and
terminated on the receiver modem 101 of an ISU 100 through diplex filter 104,
shown in Figure 8.
The RF electrical output signals include telephony information and control
data modulated on orthogonal multicaniers by MCC modem 82 utilizing
orthogonal frequency division multiplexing techniques; the telephony
information and con xol data being mapped into symbol data and the symbols
being modulated on a plurality of orthogonal carriers using fast Fourier
transform techniques. As the symbols are all modulated on carriers at a single
point to be transmitted to multiple points in the system 10, orthogonality of
the
multicarriers and symbol alignment of the symbols modulated on the orthogonal
multicarriers are automatically aligned for transport over the HFC
distribution
network 11 and the teelephony information and conrtrol data is demodulated at
the
ISUs 100 by the modern l0"a.
The ISU 100 receives the RF signal tapped from the coax of the coaxial
portion of the HFC network 11. The RF modem 101 of the ISU 100
demodulates the signal and passes the telephony information and control data
extracted to the CXSU controller 102 for provision to channel units 103 as
appropriate. The ISU 100 represents the interface where the telephony
information is converted for use by a subscriber or customer.
The C IUs 56 of the HDT 12 and the ISUs 100 implement the
bidirectional multi-point to point telephony transport system of the
communication system. 10. 'Me C IUs 56 and the'1S~Js, therefore, carry out
the modem functionality. The transport system in. accordance with the present
invention may utilize three different modems to implement the modem
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functionality for the transport system. The first modem is the MCC modem 82
which is located in each CXMU 56 of the HDT 12. The HDT 12, for example,
includes three active -MCC modems 82 (Figure 3) and is capable of supporting
many ISUs 100, representing a multi. point to point transport network. The
5 MCC modem 82 coordinates telephony information transport as well as control
data transport for controlling the ISUs 100 by the " T 12. For example, the
control data may include call processing messages, dynamic allocation and
assignment messages, ISU synchronization control messages, ISU modem
control messages, channel unit provisioning, and any other ISU operation,
10 administration, maintenance and provisioning (DAM P) information.
The second modem is a single family subscriber or HISU modern
optimized to support a single dwelling residential unit. Therefore, it must be
low
in cost and low in power consumption. The third modem is the multiple
subscriber or MTSU modem, which is required to generally support both
15 residential and business services.
The HISU modern and the MISU modem may take several forms. For
example, the HISU modem and the MISU modern may, as described further in
detail below with regard to the various embodiments of the present invention,
extract only a small portion of the multicarriers transmitted from the UDT 12
or
20 a larger portion of the multicarriers transmitted from the HDT 12. For
example,
the HISU may extract 20 multicarriers or 10 payload channels of telephony
information transported from the HDT 12 and the MISU may extract information
from 260 multicarriers or 130 payload channels transported from the HDT 12.
Each of these modems may use a separate receiver portion for extracting the
25 control data from the signal transported by the HDT 12 and an additional
receiver portion of the HISU modem to extract the telephony information
modulated on the multicarriers transported from the :T 12. This shall be
referred to hereinafter as an out of band ISU modem. The MCC modem 82 for
use with an out of band ISU modem may modulate control information within
30 the orthogonal carrier waveforms or on carriers somewhat offset from such
orthogonal carriers. In contrast to the out of band ISU modem, the I ISU and
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MISU modems may utilize a single receiver for the ISU modem and extract both
the telephony information and control data utilizing the single receiver of
the
modem!. This shall be referred to hereinafter as an in-band ISU modem. In such
a case, the control data is modulated on carriers within the orthogonal
carrier
waveform but may utilize different cwrier modulation techniques. For example,
BPSK for modulation of control data on the carriers as opposed to modulation
of
telephony data on payload carriers by QAM techniques. In addition, different
modulation techniques may be used for upstream or downstream transmission
for both control data and telephony data. For example, downstream telephony
data may be modulated on the carriers utilizing 256 QA.:i and upstream
telephony data may be modulated on the carriers utilizing 32 QAM. Whatever
modulation technique is utilized for transmission dictates what demodulation
approach would be used at the receiving end of the transport system.
Demodulation of the downstream telephony information and control data
transported by the N DT 12 shall be explained in further detail below with
reference to block diagrams of different modem embodiments.
In the upstream direction, each ISU modem 101 at an ISU 100 transmits
upstream on at least one orthogonal rnulticarrier in a 6 MHz bandwidth in the
RF
spectrum of about 5 to 40 MHz; the upstream 6 WLHZ band corresponding to the
downstream 6 MHz band in which transmissions are received. The upstream
electrical telephony and control data signals are transported by the ISU
modems
101 to the respectively connected optical distribution node 18 as shown in
Figure
1 via the individual coaxial cable legs 30. At the ODN 18, the upstream
signals
from the various IS-L-Ts are combined and transmitted optically to the T 12
via
optical feeder lines 26. As previously discussed, the upstream electrical
signals
from the various ISL s may, in part, be frequency shifted prior to being
combined
into a composite upstream optical signal. In such a caseõ the telephony
receiver
16 would include corresponding dov rish.ifting circ .iitay.
Due to the multi-point to point nature of transport over the HFFC
distribution network 11 frorn multiple ISUs 100 to a single I-HD T 12, in
order to
utilize orthogonal frequency division multiplexing techniques, symbols
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modulated on each carrier by the ISUs 100 must be aligned within a certain
phase margin. In addition, as discussed in further detail below, the round
trip
path delay from the. network interface 62 of the HDT 12 to all ISUs 100 and
back
from the ISUs 100 to the network interface 62 in the communication system 10
must be equal. This is required so that signaling multiframe integrity is
preserved throughout the system. In addition, a signal of'proper amplitude
must
be received at the H DT 12 to perform. any control :unctions with respect to
the
ISU 100. Likewise, with regard to OFDM transport from the ISUs 100, the ISUs
100 must be frequency locked to the 14DT 12 such that the multicarriers
transported over the H'-1FC distribution network 11 are orthogonally aligned.
The
transport system implements a distributed loop technique for implementing this
multi-point to point transport utilizing orthogonal frequency division
multiplexing as further described below. When the HDT 12 receives the
plurality of multicarriers which are orthogonally aligned and which have
telephony and control data modulated thereon with symbols aligned, the MCC
modems 82 of the CXMUs 56 demodulate the telephony information and control
data from the plurality of multicarriers in their corresponding 6 MHz
bandwidth
and provide such telephony data to the CTSU 54 for delivery to the network
interface 62 and the control data to the CXMC 80 for control of the telephony
transport.
As one skilled in the art will recognize, the spectrum allocations,
frequency assignments, data rates, channel numbers, types of services provided
and any other parameters or characteristics of the system which may be a
choice
of design are to be taken as examples only. The invention as described in the
accompanying claims contemplates such design choices and they therefore :fall
within the scope of such claims. in addition, many functions may be
implemented by software or hardware and either implementation is contemplated
in accordance with the scope of the claims even though reference may only be
made to implementation by one or the other.
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Fir ibo it em of'Iel_enlxonv Transnort System
The first embodiment of the telephony transport system in accordance
with the present invention shall be described with, particular reference to
Figures
13-35 which include block diagrams of MCC modems 82, and HISU modems
and MISU modems shown generally as ISU modem 101. in Figure 4. Such
modems implement the upstream and downstream modern transport
functionality. Following this description is a discussion on the theory of
operation utilizing such modems.
Referring to Figure 13, the spectrum allocation for one 6 MHz band for
upstream and downstream transport of telephony information and control data
utilizing OFDM techniques is shown. The waveform preferably has 240 payload
channels or DSO+ channels which include 480 carriers or tones for
accommodating a net data rate of 19.2 Mbps, 24 IOC channels including 46
carriers or tones, and 2 synchronization channels. Each synchronization
channel
includes two carriers or tones and is each offset from 24 IOC channels and 240
payload channels by. 10 unused carriers or tones, utilized as guard tones. The
total carriers or tones is 552. The synchronization tones utilized for
synchronization fun tonns as described further below are located at the ends
of
the 6 MHz spectrum, and the plurality of orthogonal carriers in the 6 MHz band
are separated fro.,,n carriers of adjacent 6 MHz bands by guard bands (516.0
kHz)
at each end of the 6 z spectrum. The guard bands are provided at each end of
the 6 MHz band to allow for .filter selectivity at the transmitter and
receivers of
the system. The synchronization carriers are offset from the telephony data or
payload carriers such that if the synchronization carrier utilized for
synchronization during initialization and activation is not orthogonal with
the
other tones or carriers within the 6 MHz band, the synchronization signal is
prevented from destroying the structure of the orthogonally aligned waveform.
The synchronization. tones are, therefore, outside of the main body of payload
carriers of the band and interspersed IOC channels, although the
synchronization
channel could be considered a special IOC channel.
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To minimize die power requirement of the ISUs, the amount of bandwidth
that an ISU processes is minimized. As such, the telephony payload channels
and IOC channels of the 6 MHz band are interspersed in the telephony payload
channels with an IOC channel located every 10 payload channels. With such a
distributed technique, wherein subbands of payload channels greater than 10
include an IOC chana :nel, the amount of bandwidth an ISM "sees" can be
limited
such that an IOC channel is available for the HDT 12 to communicate with the
ISU 100. Such subband distribution for the spectral allocation shown in Figure
13 is shown in Figure 16. There are 24 subbands in the 6 z bandwidth with
each subband including 10 payload channels with an IOC: channel between the
5th and 6th payload channels. A benefit of distributing the IOC channels
throughout the 6 M z band is protection from narrow band ingress. If ingress
destroys an IOC channnel, there are other IOC channels available and the HDT
12
can re-tune an ISU 100 to a different portion of the 6 MHz band, where an IOC
channel that is not corrupted is located.
Preferably, the MISU 66 sees approximately 3 MHz of the 6 MHz
bandwidth to receive up to 1.30 payload channels which bandwidth also includes
numerous IOC channels for communication from the T 12 to the MISU 66.
The HISU 68 sees about 100 kHz of the 6 u z bandwidth to receive 11
channels including at least one IOC channel for communication with the HDT
12.,
The primary difference between the downstream and upstream paths are
the support of downstream synchronization and upstream synchronization. In
the downstream direction, all ISUs lock to information from the HDT (point to
multi-point). The initialization and activation of ISUs are based on signals
supplied in the upstream synchronization charnel. During operation, ISUs track
the synchronization -tria the IOC channels. In the upstream, the upstream
synchronization process involves the distributed (inulti-point to point)
control of
amplitude, frequency, and timing; although frequency control can also be
provided utilizing only the downstream synchronization channel as described
further below. The process of upstream synchronization occurs in one of the
two
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upstream synchronization chanrnels, the primary or the secondary
synchronization channel.
Referring to Figure 21, the downstream transmission architecture of the
MCC modem 82 is shown. Two serial data inputs, approximately 10 Mbps each,
5 comprise the payload data from the CXMC 56 wl.-ich is clocked by the 8 kHz
frame clock input. The IOC control data input from the CXMC 56 is clocked by
the IOC clock input, which is preferably a 2.0 kHz clock. The telephony
payload
data and the IOC control data enter through serial. ports 132 and the data is
scrambled as known to one skilled in the art by scrambler 134 to provide
10 randomness in the waveform to be transmitted over the IIFC distribution
network 11. Without scrambling, very high peaks in the waveform may occur;
however, if the waveform is scrambled the symbols generated by the MCC
modem 82 become sufficiently random and such peaks are sufficiently limited.
Figure 58 details the operation of a typical scrambler, such as 134,
15 Figure 21. Symbol clock 4501 clocks a seed pattern through a linear-
feedback
shift register 4510 having nine stages, 4510-0 through 4510-8.With XOR gate
4511 positioned as shown, the generator polynomial is binary "100 010 000".
The seed initially loaded into register 4510 at input 4502 is "111 001 110".
Two
identical translation tables 4520 and 4521 receive two-bit inputs from
register
20 4510 at every symbol time. The high- and low-order bits of table 4520
proceed
from the outputs of stages 4510-7 and 4510-6, respectively. High-order bit
4523
of table 4521 also receives output 4510-6, but as its high-order bit; stage
output
4510-5 provides its low-order bit. Logic gates 4530 peri?orris an XOR between
,he five-bit output of table 4520 and the upper five bits of a 10-bit DSO
word,
25 while gates 4531 do the same for the lower five bits of the same DSO word.
Outputs 4505 and 4506 carry the two 5-bit scrambled symbols for the DSO word.
Each descrambler such as 176, Figure 22 or 23, is identical to its
corresponding
scrambler. It recovers the original bit pattern of each symbol by decoding it
with
the same polynomial. and seed.
30 The polynomial and seed for register 4510 of the scramblers and
descrambiers selected by known techniques to yield a maximal-length pseudo-
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random sequence. Inversion of the order of the input bits as between table
4520
and table 4521 increases the scrambling of the two symbols of the DSO word.
To increase the randomness among different sequences even more, different
scramblers in the system have different polynomials and seeds. Randomness
could be further increased by using more than four different table entries-,
however, the added complexity overrode the gain, for this particular
embodiment. Only the payload channels are scrambled; the IOC channels are
not scrambled.
The scrambled signals are applied to a symbol snapping function 136. The
symbol mapping function 136 takes tl'le input bits and maps them into a
complex
constellation point. For example, if the input bit., are mapped into a symbol
for
output of a BPSK signal, every bit would be mapped to a single symbol in the
constellation as in the mapping diagram for BPSK of Figure IS. Such mapping
results in in-phase and quadrature values (1/Q values) for the data. BPSK is
the
modulation technique preferably used for the upstream and downstream IOC
channels and the synchronization channels. BPSK encoding is preferred for the
IOC control data so as to provide robustness in the system as previously
discussed. For QPSK modulation, every two bits -would map into one of four
complex values that represent a constellation point. In the preferred
embodiment, 32 QAM is utilized for telephony payload data, wherein every five
bits of payload data is mapped into one of 32 constellation points as shown in
Figure 14. Such mapping also results in I/Q values. As such, one DSO+ signal
(10 bits) is represented by two symbols and the two symbols are transmitted
using two carriers. Thus, one DSO+ channel is transported over two carriers or
tones of 6 MHz spectrum.
One skilled in the art will recognize that various mapping or encoding
techniques may be utilized with different carriers. For example, telephony
channels carrying ISDN may be encoded using QPSK as opposed to telephony
channels carrying POTS data being encoded using 32 QAM. Therefore,
different telephony channels carrying different services m.aay be modulated
differently to provide for more robust telephony channels for those services
that
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require such quality. The architecture in accordance with the present
invention
provides the flexibility to encode and modulate any of the channels
differently
from the modulation technique used for a different charnel.
Within the framework of QAM32 modulation, Figure 17 shows a
constellation which has improved characteristics. Here, the in-phase and
quadrature values are shown encoded by three bits each instead of the four
shown in Figure 13; their analog values, however, they remain in the ranges -5
to
+5. The constellation of Figure 17 approaches as closely as possible to an
analogy to a Gray code scheme, in which a transition from one row to the next
and from one column to the next result in only a single bit change in the 5 -
bit
symbol code. (The exceptions are four transitions from the first column to the
second, and from the fifth to the sixth, which have two transitions each. The
corner cells have zero transitions between these columns, which do not detract
from the advantages of the scheme.) If a symbol is received incorrectly after
transmission, the most likely error is a slight change in either amplitude or
phase.
If the bit strings represented by the symbols have as few bit transitions as
possible for single-value phase and amplitude changes, then a reception error
will create fewer bit errors on the final digital output. That is, small
(symbol)
errors in produce small (bit) errors out.
The constellations shown in Figures 14 and 17 use all points of a six-cell
square except the four corners. Hence, they have two axes of symmetry, and
appear identical when rotated by 90'. 180 , and 270 . If a phase error ever
exceeds 45 , an attempted correction may pull the phase to an incorrect
orientation. This is called four-fold phase ambiguity. However, deliberately
using one and only one of the corner points as a valid symbol provides a key
for
identifying the correct phase for errors as great as a full 180 . For example,
designating the symbol for "16" as 1=010 (+5) and Q=010 (+5) instead of the
1=001, Q=010 (+3, +5) in Figure 17 introduces a symbol at this comer point
whenever a "16" is sent upstream or downstream. Because only one corner is
used, any received value having both I and Q values +5 requires phase rotation
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until I=+5 and Q-t'-5, This assignment also preserves the nearly Gray-code
structure of the constellation.
Any other symbol assignment which breaks the symmetry of the
constellation would produce the same effect. Even a constellation retaining
only
one axis of symmetry would allow twice the phase-correction range of the
constellation of Figure 17. For examele, using both the upper left and lower
right corners as valid symbols allows correction of phase errors up to 90 .
Each symbol that gets represented by the 1/Q values is mapped into a fast
Fourier transform (FFT) bin of symbol buffer 138. For example, for a DSO+,
running at 8 kHz frame rate, five bits are mapped into one FFT bin and five
bits
into another bin. Each bin or memory location of the symbol buffer 138
represents the payload data and control data in the frequency domain as 1/Q
values. One set of F-1, T bins gets mapped into the time domain through the
inverse FFT 140, as is known to one skilled in the art. The inverse FFT 140
maps the complex I/Q values into time domain samples corresponding to the
number of points in the FFT. Both the payload data and JOG data are mapped
into the buffer 138 and transformed into time domain samples by the inverse
FFT 140. The number of points in the inverse FFT 140 may vary, but in the
preferred embodiment the number of points is 256. The output of the inverse
FFT 140, for a 256 point FFT, is 256 time domain samples of the waveform.
In conventional practice, buffer 138 clocks symbols into inverse FFT 140
at exactly the same rate that inverse FFT 140 clocks out the in-phase and
quadrature values FFT I and FFT Q in Figure 21. To put the matter another way,
the 256 digital waveform samples from buffer 138 represent 360 , or 2t'
radians,
of a Q.AlN4 32 waveform having the amplitude and. phase of the 5 bits of its
symbol, as determined by mapping unit 136. The F F T I and Q outputs
represent
256 samples of a frequency spectrum corresponding to the same time period. At
the receiving end, however, any misalignment at all in the phase
synchronization
causes FFT 170, Figure 22, or 180, Figure 23, to decode a portion of a
previous
or subsequent symbol's waveform along with somewhat less than the full cycle
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of the desired symbol; this intersymbol interference can cause misreading the
symbol as a different valid symbol, resulting in as many as five bit errors.
In a presently preferred embodiment, the 256 samples clocked into inverse
FFT 140 represent an extra 45 (it!4 radians) above a complete cycle. Another
ti
way to think of this is that the symbols are clocked into the F'A'T at an
effective 9
kHz rate, and clocked out at the nominal 8 kHz symbol rate. Figure 52 shows an
unrnodulated sine wave, i.e., one having 1=0, Q=0 in the units used herein.
The
upper portion shows one cycle, 0-360 , at the nominal 8 kHz frame rate. The
lower portion shows the same wave at a 9 k_1Hz rate, so that the amount of
time
previously occupied by 360 now takes up 405 of phase -- from -22.5 to
+382.5 . Obviously, there are phase discontinuities between successive cycles
of the wave. Figure 53 shows a typical QAM 32 wave modulated at a different
amplitude and a slightly different phase from those of Figure 52. These might
correspond to, say, :=-1, Q=+1 in the scheme used herein. The small portions
at
the ends of this wave represent unmodulated cycles, as in Figure 52. The phase
of this wave is advanced from the corresponding wave of the lower portion of
Figure 52; it does not cross the zero axis at 0 and 180 of its proper
cycle. It
does, however, include the extra 22.5 of excess phase at each end, for 45
extra
over an 8 kHz cycle. Again, a phase discontinuities exist at the ends of the
total
405 phase degrees of this wave,
In fact, this characteristic gives the excess-phase improvement an
advantage over its primary function of providing a guard band for the symbol
decoder, for reducing intersymbol interference. In Figure 52, successive
cycles
of a wave modulated with the same symbol (or with no symbol), produce a
continuous waveform with no breaks or other features to distinguish the
beginnings and endings of individual cycles. The lower part of this diagram
demonstrates that even an unmodulated excess-phase waveform contains
discontinuity features serving as markers at the ends of each cycle..
repeating
string of idle symbols, or any other symbols, likewise produces these markers.
In the frequency- and phase-acquisition and tracking aspects discussed below,
such markers therefore provide definite waveform features for synchronizing
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purposes, without having to guarantee the transmission of any special string
of
varying characters strictly for synchronization. This saves the overhead of
interrupting the payload and/or 10C channels to provide such a string, and the
complexity of storing or diverting payload information while the sync string
is
5 present. It also allows, sync to take place at times = ,hen, because of the
above
factors, it would not be feasible otherwise.
At the receiving end, FFT 170 (in an MISU) or ISO SU) decodes the
256 time slots for one frame time as 405 of a cycle to symbol decoder 174,
which matches the cycle to the nearest 5-bit string of bits. Because any phase
10 difference up to 22.5 will never conflate the proper wave with that for
another
syrxtbol, no intersymbol interference at all occurs within this margin of
error in
phase tracking. This provides a form of guard band for each symbol In the
upstream direction, units 186,, 188, and 190 or 191 provide excess phase in
the
transmitting MISU and HISU modems of Figures 24 and 25; and the head-end
15 receiving modem of Figure 26 decodes and tracks this phase as described
above.
The inverse FFT 140 has separate serial outputs for in-phase and
quadrature (IIQ) components, FFTI and FFTO. Digital to analog converters 142
take the in-phase and quadrature components, which is a numeric representation
of baseband modulated signal and convert it to a discrete waveform. The signal
20 then passes through reconstruction filters 144 to remove harmonic content.
This
reconstruction is needed to avoid problems arising from multiple mixing
schemes and other filtering problems. The signal is summed in a signal
conversion transmitter 146 for up-converting the IIQ components utilizing a
synthesized waveform that is digitally tunable with the in-phase and
quadrature
25 components for mixing to the applicable transmit frequency. For example, if
the
synthesizer is at 600 MHz, the output frequency will be at 600 MHz. The
components are summed by the signal conversion transmitter 146 and the
waveform including a plurality of orthogonal carriers is then amplified by
transmitter amplifier 148 and filtered by transmitter filter 150 before being
30 coupled onto the optical fiber by way of telephony transmitter 14. Such
functions are performed under control of general purpose processor 149 and
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other processing circuitry of block 147 necessary to perform such modulation.
The general purpose processor also receives ISU adjustir0ent parameters from
carrier, amplitude, timing recovery block 222 (Figure 26) for carrying out
distributed loop symbol alignment, frequency locking, amplitude adjustment,
and path delay functions as described further belo~ w.AT.
In conventional practice, the relationship between the frequency of a
carrier and the frequency and timing of data symbols modulated onto that
carrier
is arbitrary and unimportant. In the present system, however, it has been
found
that even very small frequency drifts between the 8 z symbol or frame clock
and the frequencies of the tones upon which they ride can produce significant
intersymbol interference and distortion at the receiving end. Such drifts tend
to
destroy the orthogonality of the channel signals produced by inverse FFT 140
in
Figure 21. The present system also,',.however, provides a simple, inexpensive
way to overcome this problem. Figure 51 shows a portion 4200 of the HDT
clock/sync logic in CTSU 54, Figure 3. Timing recovery loop 4210 produces a
single master reference clock output at 10.24 :z. Although loop 4210 could
be a free-running oscillator, it is in fact slaved to the network 10, Figure
1. With
which the entire system communicates. This connection. is convenient in
eliminating gross or unpredictable differences between the data speeds of the
network and the system.
Smoothing loop 4220 evens out shorn-term. variations in the signal from
loop 4210. Phase comparator 4221 controls a voltage-controlled crystal
oscillator at 40.96 MBz divider 4223 provides feedback at the proper
frequency.
Comparator 4221 includes a low-pass integrator which gives phase-lock loop
4220 a bandwidth of about 130Hz. Divider 4230 reduces the frequency of
VCXO 4222 to 2.56Mhz. A second phase-lock loop 4240 has a phase
comparator 4241, again with low-pass characteristics, feeding a voltage-
controlled oscillator running at 1267..2MHz; divider provides feedback at the
proper frequency. Divider 4250 produces the final clock frequency, IRP 30
9.9MHz,at output 4251. The network clock is sufficiently accurate over long
periods of time, but it is subject to significant amounts of short-period.
jitter. The
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large amount of smoothing provided by loops 4220 and 4240 overcome the
intolerance of analog RF components for short-ten.: variations.
Meanwhile, digital divider 4260 divides the master 10.24MHz clock by a
tl
factor of 80 to produce an 8kHz symbol or frame clock. output 4261. Output
4261 does not require the smoothing, because it clocks only digital circuits,
which are relatively insensitive to short-term frequency changes.
RF master clock 4261 proceeds to RF synthesizer 143 in HDT
transmitting modern- 82, as shown in Figure 21, where it directly controls the
frequency of the tunable 500-850MHHz RF carrier far the entire band carrying
all
of the channels shown in Figures 13 and 16. Symbol clock 4261 proceeds to the
frame-clock inputs in Figure 21, where it controls the symbol timing, and,
because it also contra is the FFT speed, the frequencies of the channels in
the
entire band. Clock lock 4200 thus provides a solid link which inherently
preserves the orthogonality of the band signals in a multicarrier system, by
deriving the RF carrier clock and the symbol or frame clock from the same
source. At the same time, it provides a small amount of gradual variation for
satisfying the demands of the analog RF components.
The overall purpose of locking the two clocks together at the 14DT is to
lock the carrier clocks, and the symbol (frame) clocks throughout the system;
and
the purpose of this In turn is to preserve the orthogonality of the signals it
a
multicarrier system which is capable of bidirectional operation- that is, as a
muitipoint-to-point-configuration as well as in the usual point-to-multipoint
"broadcast" direction. Clock generator 166, Figures 22 and 23, of timing
generator 107, Figure 6 locks to the frequencies of the incoming signals to
provide the clocks used in the remote ISU modules. Th erefore, the carrier and
frame clocks in each upstream transmitter portion, .Figure :24, of remote -
modem
108, Figure 8, are also locked to each other, by virtue of being locked to the
incoming signal from the ITT.
At the downstream receiving end, either an MISI.7 or an MS'U provides
for extracting telephony information and control data from. the downstream
transmission in one of the 61, z bandwidths. With respect to the MIS1166, the
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MISU downstream receiver architecture is shown in Figure 22. It includes a 100
MHz bandpass filter 152 to reduce the frequency band of the received 600 .. to
850
MHz total band broadcast downstrew- n. The filtered signal then passes through
voltage tuned filters 154 to remove out of band interference and further
reduce
the bandwidth. The signal is down converted to baseband frequency via
quadrature and in-phase down converter 158 where the signal is mixed at
complex mixers 156 utilizing synthesizer 157 which is controlled from an
output of serial ports 178. The down converted LIQ components are passed
through filters 159 and converted to digital format at analog to digital
converters
160. The time domain samples of the I/Q components are placed in a sample
buffer 162 and a set of samples are input to down. converter compensation unit
164. The compensation unit 164 attempts to mitigate errors such as DC offsets
from the mixers and differential phase delays that occur in the down
conversion.
Carrier, amplitude and timing signaling are extracted from the
compensated signal, by the carrier, amplitude, and timing recovery block 166
by
extracting control data from the synchronization channels during
initialization
and activation of the ISU and the IOC channels during tracking as further
described below with reference to Figure 33. The compensated signal in
parallel
form is provided to fast Fourier transform (FFT) 170 to 'be converted into a
vector of frequency domain elements which are essentially the complex
constellation points witxhi I/Q components originally created upstream at the
MCC modem 82 for the DSO+ channels which the Si> sees. Due to
inaccuracies in channel filtering, an equalizer 172 removes dynamic errors
that
occur during transmission and reception. Equalization in the upstream receiver
and the downstream receiver architectures shall be explained in further detail
below with reference to Figure 35. From the equalizer 1.72, the complex
constellation points are converted to bits by symbol to bit converter 174,
descrambled at descrambler 1 76 which is a mirror element of scrambler 134,
and
the payload telephony information and IOC control data, are output by the
serial
ports 178 to the CXSU 102 as shown in Figure 8. Block 153 includes the
processing capabilities for carrying out the various functions as shown
therein.
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Referring to Figure 23, the HISU 68 downstream receiver architecture is
shown. The primary difference between the HISU downstream receiver
architecture (Figure 2y) and the MISU downstream receiver architecture (Figure
22) is the amount of bandwidth being processed. The front ends of the
receivers
up to the FFT processing are substantially the same, except during the down
conversion, the analog to digital converters 1 60 can be operated at a much
slower rate. For instance, if the bandwidth of the signal being processed is
100
kHz, the sample rate can be approximately 200 kHz. In an MISU processing a 3
MHz signal, the sample rate is about 6 MHz. Since the HISU is limited to
receiving a maximum of 10 DSO+s, the FFT 180 can be of'-" a smaller size. A 32
point FFT 1 80 is preferably used in the HISU and can be implemented more
efficiently, compared to a 128 or 256 point FFT utilized in the MISU,
Therefore, the major difference between these architectures is that the HISU
receiver architecture requires substantially less signal processing capability
than
the MISU receiver and as such has less power consumption. Thus, to provide a
system wherein power consumption at the remote units is minimized, the smaller
band of frequencies seen by the HISU allows for such low consumption. One
reason the HISU is allowed to see such a small band of carriers is that the
IOC
channels are interspersed throughout the 6 Nfflz spec .
Referring to Figure 24, the upstream transmission architecture for the
I-IISU 68 is shown. The IOC control data and the telephony payload data from
the CXSU 102 (Figure 8) is provided to serial ports 182 at a much slower rate
in
the KISU than in the 1, .I.SU or HDT transmission architectures, because the
I4-ISU supports only 10 DSO+ channels. The HISU upstream transmission
architecture implements three important operations. It adjusts the amplitude
of
the signal transmitted, the timing delay (both symbol and path delay) of the
signal transmitted, and the carrier frequency of the signal transmitted. The
telephony data and IOC control data enters through the serial ports 182 under
control of clocking signals generated by the clock generator 173 of the HISU
downstream receiver architecture, and is scrambled by scrambler 184 for the
reasons stated above with regard to the MCC downstream transmission
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architecture. The incoming bits are mapped into symbols, or complex
constellation points, including I/Q components in the frequency domain, by
bits
to symbol converter 186. The constellation points are then placed in symbol
buffer 188. Following the buffer 188, an inverse FFT 190 is applied to the
5 symbols to create time domain samples; 32 samples corresponding to the 32
point FFT. A delay buffer 192 is placed on the output of the inverse FFT 190
to
provide multi-frame alignment at MCC modem upstream receiver architecture as
a function of the upstream synchronization process controlled by the HDT 12.
The delay buffer 192. therefore, provides a path delay adjustment prior to
digital
10 to analog conversion by the digital to analog converters 194 of the in-
phase and
quadrature components of the output of the inverse FFT 1.90. Clock delay It 96
provides a fine tune adjustment for the symbol alignment at the request of IOC
control data output obtained by extracting control data from the serial stream
of
data prior to being scrambled. After conversion to analog components by
digital
15 to analog converters :194, the analog components therefrom are
reconstructed
into a smooth analog waveform by the reconstruction filters 198. The upstream
signal is then directly up converted by direct converter 197 to the
appropriate
transmit frequency under control of sytnthesizer block 195. Synthesizer block
195 is operated under control of commands -from an IOC control channel which
20 provides carrier frequency adjustment. co ands thereto as extracted in the
MSU downstream receiver architecture. The up converted signal is then
amplified by transmitter amplifier 200, filtered by transmitter filter 202 and
transmitted upstream to be combined with other signals transmitted by other
ISUs 100. The block 181 includes processing circuitry for carrying out the
25 functions thereof.
Referring to Figure 27, the upstrearn transmitter architecture for the MISU
66 is shown and is substantially the same as the upstream transmitter
architecture
of HISU 68. However, the MISU 66 handles more channels and cannot perform
the operation on a single processor as can the IIISU68. Therefore, both a
30 processor of block 181 providing the functions of block 181 including the
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inverse FFT 190 and a general purpose processor 206 to support the
architecture
are needed to handle the increased channel capacity.
Referring to Figure 26, the MCC' upstream receiver architecture of each
CXMU 56 at the T 12 is shown. A 5 to 40 band pass filter 208 filters
the upstream signal which is then subjected to a direct down conversion to
baseband by mixer and synthesizer circuitry 211. 'I he outputs of the down
conversion is applied to anti-alias filters 210 for conditioning thereof and
the
output signal is converted to digital format by analog to digital converters
212 to
provide a time domain sampling of the in-phase and quadrature components of
the signal to narrow band ingress filter and FFT 112. The narrow band ingress
filter and FFT 112, as described below, provides protection against narrow
band
interference that may affect the upstream transmission.
The ingress filter and FFT 112 protects ten channels at a time, therefore, if
ingress affects one of the available 240 DSO+s in the 6 MHz spectrum received
by MCC modem 82, a maximum of tea channels will be destroyed from the
ingress. The ingress filter and FFT 112 includes a polyphase structure, as
will
be recognized by one skilled in the art as a common filter technique. It will
be
further recognized by one skilled in the art that the nu .ber of channels
protected
by the polyphase filter can be varied. The output of the ingress filter and
FFT
1.12 is coupled to an equalizer 214 which provides correction for inaccuracies
that occur in the channel, such as those due to noise from reference
oscillators or
synthesizers. The output symbols of the equalizer 214, are applied to a
symbols
to bits converter 216 Ywvhere the symbols are mapped into bits. The bits are
provided to descramblers 218, which are a mirror of the scramblers of the ISUs
100 and the output of the descramblers are provided to serial ports 220. The
output of the serial ports is broken into two payload! streams and one IOC
control
data stream just as is provided to the MCC downstream transmitter architecture
in the downstream direction. Block 217 includes the necessary processing
circuitry for carrying out the functions -therein.
In order to detect the downstream information, the ampplitude, frequency,
and timing of the arriving signal must be acquired using the downstream
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synchronization process. Since the downstream signal constitutes a point to
multi-point node topology, the OFDM waveform arrives via a single path in an
inherently synchronous manner, in contrast to the upstream signal. Acquisition
of the waveform parameters is initially performed on the downstream
synchronization channels in the downstream synchronization bands located at
the ends of the 6 MT-1z spectrum. These synchronization bands include a single
synchronization carrier or tone which is BPSK modulated by a 2 kHz framing
clock. This tone is used to derive initial amplitude, frequency, and timing at
the
ISU. The synchronization carrier may be located. in the center of the receive
band and could be considered a special case of an TOC. After the signal is
received and the receiver architecture is tuned to a typical IOC channel, the
same
circuitry is used to track the synchronization parameters using the TOC
channel.
The process used to acquire the necessary signal parameters utilizes earlier,
amplitude and timing recovery block 166 of the ISU receiver architecture,
which
is shown in more detail in block diagram form in. Figure 33. The carrier,
amplitude and timing recovery block 166 includes a Costas loop 330 which is
used to acquire the frequency lock for the received waveform. After the signal
is
received from the compensation unit 164, a sample and hold 334 and analog to
digital conversion 332 is applied to the signal with the resulting samples
from
the converters 332 applied to the Costas loop 330. The s pling is performed
under control of voltage controlled oscillator 340 as divided by divider 333
which divides by he number of points of the FFT utilized in the receiver
architecture, M. The mixers 331 of the Costas loop 330 are fed by the arriving
signal and the feedback path, and serve as the loop phase detectors. The
output
of the mixers 331 are filtered and decimated to reduce the processing
requirements of subsequent hardware. Given that the received signal is band-
limited, less samples are required to represent the synchronization signal, If
orthogonality is not preserved in the receiver, the filter will eliminate
undesired
signal components from the recovery process. Under conditions of
orthogonality, the LPF 337 will completely remove effects from adjacent OFDM
carriers. When carrier fiequency lock is achieved, the process will reveal the
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desired BPSK waveforrn in the in-phase arm of the loop. The output of the
decimators are fed through another mixer, then processed through the loop
filter
with filter function H(s) and numerically controlled oscillator (NCO),
completing the feedback path to correct for frequency error. When the error is
at
a "small" level, the loop is locked. In order to achieve fast acquisition and
minimal jitter during tracking, it will be necessary to employ dual loop
bandwidths. System operation will require that frequency- lock is achieved and
maintained within about 4% of the OFDM channel spacing (360 Hz).
The amplitude of the signal is measured at the output of the frequency
recovery loop at BPSK power detector.- 336. The total signal power will be
measured, and can be used to adjust a numerically controllable analog gain
circuit (not shown). The gain circuit is intended to normalize the signal so
that
the analog to digital converters are used in an optimal operating region.
Timing recovery is performed using an early-late gate type algori' of
early-late gate phase detector 338 to derive timing error, and by adjusting
the
sample clock or oscillator 340 in response to the error signal. The early-late
gate
detector results in an advance/retard command during an update interval. This
command will be applied to thesample clock or oscillator 340 through filter-
341.
This loop is held off until frequency lock and amplitude lock have been
achieved. endue timing loop is loked, it generates a lock indicator signal.
The same clocks are -!so used for the upstream transmission. The carrier,
timing
and amplitude recovery block 166 provides a reference for the clock generator
168. The clock generator 168 provides all of the clocks needed by the : ISJ,
for
example, the 8 kHz frame clock and the sample clock.
Carrier, aniplitude, and timing recovery block 222 of the MCC modem
upstream receiver architecture (Figure 26), is shown by the synchronization
loop
diagram of Figure 34, It performs detection for upstream synchronization on
signals on the upstream synchronization channel. For initialization and
activation of an !S U, upstream synchronization is performed by the HDT
commanding one of the ISUs via the downstream IOC control channels to send a
reference signal upstream on a synchronization channel. The carrier,
amplitude,
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and timing recovery block 222 measures the parameters of data from the ISU
100 that responds on the synchronization channel and estimates the frequency
error, the amplitude error, and the timing error compared to references at the
HDT 12. The output of the carrier, amplitude, and timing recovery block 222 is
turned into adjustment commands by the H DT 12 and sent to the ISU being
initialized and activated in the downstream direction on an IOC control
channel
by the MCC downstream transmitter architecture.
The purpose of the upstream synchronization process is to initialize and
activate ISUs such that the waveform from distinct ISUs combine to a unified
waveform at the T 12. Thep meters that are estimated at the HDT 12 by
carrier, amplitude, and timing recovery block 222 and adjusted by the ISUs are
amplitude, timing, and frequency. The amplitude of an ISDs signal is
normalized so that DSO+s are apportioned an equal mount of power, and
achieves a desired signal to noise ratio at the HDT 12. In addition, adjacent
ISUs must be received at the correct relative level nor else weaker DSO+
channels
will be adversely impacted by the transient behavior ofthe stronger DSO+
channels. If a payload channel is transmitted adjacent to another payload
channel with sufficient frequency error, orthogonality in the OFDM waveform
deteriorates and error rate performance is compror raised. Therefore, the
frequency of the ISU must be adjusteet . to close tole ces Timing of the
recovered signal also impacts orthogonality. A. symbol which is not aligned in
time with adjacent symbols can produce transitions within the part of the
symbol
that is subjected to the FFT process. If the transitions of all symbols don't
fall
within the guard interval at the HDT, approximatel 16 tones (8 DSO+s)
relative to the non-orthogonal channel will be unrecoverable,
During upstream synchronization, the ISUs will be commanded to send a
signal, for example a square wave signal, to establish amplitude and frequency
accuracy and to align symbols. The pattern signal may be any signal which.
allows for detection of the parameters by carrier, amplitude and timing
recovery
block 222 and such signal may be different for detecting different parameters.
For example, the signal may be a continuous sinusoid fbr amplitude and
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frequency detection and correction and a square wave fbr symbol timing. The
carrier, amplitude and timing recovery block 222 estimates the three
distributed
loop parameters. In all three loops, the resulting error signal will be
converted to
a command by the CXMC 80 and sert via the MCCI modem 82 over an I0C
5 channel and the CXSUJ will receive the command and control the adjustment
made by the ISU.
As shown in Figure 34, the upstream synchronization from the ISU is
sampled and held 434. and analog to digital converted 432 under control of
voltage controlled oscillator 440. Voltage controlled oscillator is a local
10 reference oscillator which is divided by M, the points of the FFT in the
receiver
architecture, for control of sample and hold 434 and analog to digital
converter
432 and divided by k to apply an 8 kHz signal to phase detector 438.
Frequency error may be estimated utilizing the Costas loop 430. The
Costas loop 430 attempts to establish phase lock with the locally generated
15 frequency reference. After some period of time, loop adaptation will be
disabled
and phase difference with respect to the time will be used to estimate the
frequency error. The frequency error is generated by filter function H(s) 444
and
provided to the CXMC 80 for processing to send a equency adjustment
command to the ISU via an IOC control channel. The frequency error is also
20 applied to the numerically controlled oscillator CO) to complete the
frequency
loop to correct for frequency error.
The amplitude error is computed based on the magnitude of the carrier
during the upstream synchronization by detecting the carrier amplitude of the
in-
phase arm of the Costas loop 430 by power detector 436. The amplitude is
25 compared with a desired reference value at reference comparator 443 and the
error will be sent to the CXMC 80 for processing to send an amplitude
adjustment con and to the ISU via an 1OC control channel.
When the local reference in the i has achieved phase lock, the BPPISK
signal on the synchronization channel. arriving from the ISU is available for
30 processing. The square wave is obtained on the in-phase arm of the Costas
loop
430 and applied to early-late gate phase detector 4.38. fur comparison to the
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locally generated S z signal from divider 435. 'l.-he phase detector 438
generates a phase or symbol timing error applied to loop filter 441 and output
via
line 439. The phase or symbol timing error is then provided to the CXMC 80 for
processing to send a symbol timing adjustment command to the ISM" via an IOC
control channel.
The mechanisms in the ISU which adjust the parameters for upstream
synchronization include implementing an amplitude change with a scalar
multiplication of the time domain waveform as it is being collected from the
digital processing algorithm, such as inverse FFT 190, be% the digital to
analog
converters 194 (Figure 24). Similarly, a complex. mixing signal could be
created
and implemented as a complex multiply applied to the input to the digital to
analog converters 1914.
Frequency accuracy of both, the downstream sample clock and upstream
sample clock, in the ISU, is established by phase locking an oscillator to the
downstream synchronization and IO information. Upstream transmission
frequency is adjusted, for example, at synthesizer block 1 95 as commanded by
the '1 12.
Symbol timing corrections are implemented as a delay function. Symbol
timing alignment in the ISU upstrearn direction is therefore established as a
delay in the sample d g accomplished by either blanking a sample interval
(two of the same samples to go out simultaneously) or by putting in an extra
clock edge (one sample is clocked out and lost) via clock delay 196 (Figure
24).
In. this manner, a delay function can be cort olled witho-at data storage
overhead
beyond that already required.
After the ISU is initialized and activated into the system, ready for
transmission, the IS'J will maintain required upstream synchronization system
parameters using the carrier, amplitude, frequency recovery block 222. An
unused but initialized and activated ISU will be commanded to transmit on an
IOC and the block 222 will estimate the parameters therefrom as explained
above.
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In both the upstream transmitter architectures for the MISU 66 (Figure 24)
and the HISU 68 (Figure 25), frequency offset or correction to achieve
orthogonality of the carriers at ITT 12 can be determined on the ISU as
opposed
to the frequency offset being determined at the HD T during synchronization by
carrier, amplitude and timing recovery block 222 (]Figure 26) and then
frequency
offset adjustment commands being transmitted to the ISU for adjustment of
carrier frequency via the synthesizer biocks 1.95 and 199 of the HISU 68 and
If1SU 66, respectively. Thus, frequency error would no longer be detected by
carrier, amplitude and timing recovery block 222 as described above. Rather,
in
such a direct ISU implementation, the ISU, whether an HISU 68 or MISU 66,
estimates a frequency error digitally from the downstream signal and a
correction
is applied to the upstream data being transmitted.
The HDT 12 derives all transmit and receive frequencies from the same
fundamental oscillator. Therefore, all mixing signals are frequency locked in
the
UDT. Similarly, the ISU, whether an ITTSU 68 or MISU 66, derives all transmit
and receive frequencies from the same fundamental oscillator; therefore, all
the
mixing signals on the ISU are also frequency locked. There is, however, a
frequency offset present in the ISU oscillators relative to the HDT
oscillators.
The amount of frequency error (viewed from the ISU) will be a fixed percentage
of the mixing frequenccy. For example, if the ISU oscillator is I OPPM off in
frequency from the HDT oscillators, and the downstream ISU receiver mix
frrequency was 100 MHz and the ISU upstream transmit mixing frequency were
101 , the ISU would have to correct for 1 kHz on the downstream receiver
and create a signal with a 100 Hz offset on the upstream transmitter. As such,
with the ISU direct implementation, the frequency offset is estimated from the
downstream signal.
The estimation is performed with digital circuitry performing numeric
calculations, i.e. a processor. Samples of the synchronization channel or I t
channel are collected in hardware during operation of the system. A tracking
loop drives a digital numeric oscillator which is digitally mixed against the
received signal. This process derives a signal internally that is essentially
locked
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to the HDT. The internal numerical mix accounts for the frequency offset.
During the process of locking to the downstream signal in the ISU, the
estimate
of frequency error is derived and with the downstream frequency being known, a
fractional frequency error can be computed. Based on the knowledge of the
mixing frequency at the HDT that will be used to down convert the upstream
receive signal, an offset to the ISU transmit frequency is computed. This
frequency offset is digitally applied to the ISU transmitted signal prior to
converting the signal to the analog domain, such as by converters 194 of
Figure
24. Therefore, the frequency correction can be performed directly on the ISU.
Referring to Figures 31 and 32, the narrow band ingress filter and FFT
112 of the MCC upstream receiver architecture, including a polyphase filter
structure, will be described in further detail. Generally, the polyphase
filter
structure includes polyphase filters 122 and 124 and provides protection
against
ingress. The 6 MHz band of upstream OFDM carriers from the ISUs 100 is
broken into subbands through the polyphase filters which provide filtering for
small groups of carriers or tones, and if an ingress affects carriers within a
group
of carriers, only that group of carriers is affected and the other groups of
carriers
are protected by such filtering characteristics.
The ingress filter structure has two parallel banks 122, 124 of polyphase
filters. One bank has approximately 17 different non-overlapping bands with
channel spaces between the bands. A. magnitude response of a single polyphase
filter bank is shown in Figure 29. The second bank is offset from the first
bank
by an amount so that the channels that are not filte:r ed by the first bank
are
filtered by the second bank. Therefore, as shown in the closeup magnitude
response of a single polyphase filter bank in Figure 30, one band of channels
filtered may include those in frequency bins 38-68 with the center carriers
corresponding to bins 45-61 being passed by the falter. The overlapping filter
provides for filtering carriers in the spaces bet=ween the bands and the
carriers not
passed by the other filter bank. For example, the overlapping filter may pass
28-
44. The two channel banks are offset by 16 frequency bins so that the
combination of the two filter banks receives every one of the 544 channels.
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Referring to Figure 31, the ingress filter structure receives the sampled
waveform x(k) from the analog to digital converters 212 and then complex
mixers 118 and 120 provide the stagger for application to the polyphase
filters
122, 1.24. The mixer 118 uses a constant value and the mixer 120 uses a value
to
achieve such offset. The outputs of each mixer enters one of the polyphase
filters 122, 124. The output of each polyphase filter bank comprises 18 bands,
each of which contain 16 usable FFT bins or each band supports sixteen
carriers
at the 8 kHz rate, or 8 DSO+s. One band is not utilized.
Each band output of the polyphase filters 122, 124 has 36 samples per 8
kHz frame including =4 guard samples and enters a Fast Fourier Transform (FFT)
block 126, 128. The first operation performed by the FF1 blocks 126, 128 is to
remove the four guard samples, thereby leaving 32 time domain points. The
output of the each FFT in the blocks is 32 frequency bins, 16 of which are
used
with the other bins providing filtering. The output of the FFTs are staggered
to
provide overlap. As seen in Figure 31, carriers 0 - 15 are output by FFT #1 of
the top bank, carriers 16 - 31 are output by FFT #1 of the bottom bank,
carriers
32 - 48 are output by FFT #2 of the top bank and so on.
The polyphase filters 122, 124 are each standard polyphase filter
construction as is known to one skilled in the art and each. is shown by the
structure of Figure 32. The input signal is sampled at a 5.184 mega-- sample
per
second rate, or 648 samples per frame. The input is then decimated by a factor
of 18 (1 of 18 samples are kept) to give an effective sample rate of 288 kHz.
This signal is subjected to the finite impulse response (FIR) filters, labeled
H0.0(Z) through Hd.;6(Z), which include a number of taps, ,preferably 5 taps
per
filter. As one skilled in the art will recognize the number of taps can vary
and is
not intended to limit the scope of the invention. The outputs from the filters
enter an 18 point inverse FFT 130. The output of the transform is 36 samples
for
an 8 kHz e including 4 guard samples and is provided' to FFT blocks 126
and 128 for processing as described above. The FF'T tones are preferably
spaced
at 9 kHz, and the information rate is 8 kilosymbols per second with four guard
samples per symbol allotted. The 17 bands from each polyphase filter are
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applied to the FFT blocks 126, 128 for processing and output of the 544
carriers
as indicated above. One band, the 18th band, as indicated above, is not used.
The equalizer 214 (Figure 26) and 172 (Figure 22), in both upstream and
downstream receiver architectures, is supplied to account for changes in group
5 delay across the cable plant. The equalizer tracks out phase and gain or
amplitude variations due to environmental changes and can therefore adapt
slowly while maintaining sufficiently accurate tracking. The coefficients 360
of
the equalizer 172, 214, for which the internal equalizer operation is
generally
shown in Figure 35, represent the inverse of the channel frequency response to
10 the resolution of the FFT 112, 170. The downstream coefficients will be
highly
correlated since every channel will progress through the same signal path as
opposed to the upstream coefficients which may be uncorrelated due to the
variant channels that individual DSO+s may encounter in. the multi-point to
point
topology. While the channel characteristics are diverse, the equalizer will
15 operate the same for either upstream or downstream receiver.
The downstream equalizer will track on only the 1OC channels, thus
reducing the computational requirements at the ISUs and removing the
requirement for a preamble in the payload channels, described further below,
since the IOC channels are always transmitted. The upstream, however, will
20 require equalization on a per DSO+ and IOC channel basis.
The algorithm used to update the equalizer coefficients contains several
loca`, minima when operating on a 32: QAM constellation and suffers from a
foar -fold phase ambiguity. Furthermore, each DS0+ in the upstream can
emanate from a separate ISU, and can therefore have an independent phase
shift.
25 To mitigate this problem, each communication onset will be required to post
a
fixed symbol preamble prior to data transmission. Note that the IOC channels
are excluded from this requirement since they are not equalized and that the
preamble cannot be scrambled. It is known that at the time of transmission,
the
HDT 12 will still have accurate frequency lock and symbol timing as
established
30 during initialization and activation o: the ISU and will maintain
synchronization
or., the continuously available downstream IOC channel.
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The introduction of the preamble requires that the equalizer have
knowledge of its process state. Three states are introduced which include:
search, acquisition, and tracking mode. Search mode is based on the amount of
power present on a cT annel. Transmitter algorithms will place a zero value in
unused FFT bins, resulting in no power being transmitted on that particular
frequency. At the receiver, the equalizer will determine that it is in search
mode
based on the absence of power in the FFT bin.
When transmission begins for an initialized and activated ISU, the
equalizer detects the presence of signal and enters the acquisition mode. The
length of the preamble may be about 15 symbols. The equalizer will vary the
equalization process based on the preamble- The initial phase and amplitude
correction will be large but subsequent updates of the coefficients will be
less
significant.
In order to differentiate the training pattern from any other data sequence,
when the HDT informs an ISU to connect a new payload channel, the ISU
transmits 16 consecutive symbols having 1=0 and Q=0, which is not a valid data
symbol in the constellations of Figures 14 or 17. The ISU then transmits 8
valid
data symbols, allowing the equalizer for that channel to set its coefficient
properly to adjust for amplitude and phase of the incoming signal.
After acquisition, tutee equalizer will enter a tracking= mode with the update
rate being reduced to a minimal level. The tracking mode will continue until a
loss of power is detected or, the channel for a period of time. The channel is
then
in the unused but initialized and activated state, The equalizer will not
train or
track when the receiver is being tuned and the coefficients will not be
updated.
The coefficients may be accessed and used such as by signal to noise detector
305 (Figure 26) for channel monitoring as discussed further below.
For the equalization process, the I/Q components are loaded into a buffer
at the output of the FFT, such as FFT 112, 180. As will be apparent to one
skilled in the art, the following description of the equalizer structure is
with
regard to the upstream receiver equalizer 214 but is equally applicable to the
downstream receiver equalizer 172. The equalizer 214 extracts time domain
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samples from the buffer and processes one complex sample at a time. The
processed information is then output therefrom. Figure 35 shows the basic
structure of the equalizer algorithm less the state control algorithm which
should
be apparent to one skilled in the art. The primary equalization path performs
a
complex multiply at multiplier 370 with the value from the selected FFT bin.
The output is then quantized at symbol quantize block 366 to the nearest
symbol
value from a storage table. The quantized value (hard decision) is passed out
to
be decoded into bits by symbols to bits converter 216. The remainder of the
circuitry is used to update the equalizer coefficients. An error is calculated
between the quantized symbol value and the equalized sample at summer 364.
This complex error is multiplied with the received sample at multiplier 363
and
the result is scaled by the adaptation coefficient by multiplier 362 to form
an
update value. The update value is summed at summer 368 with the original
coefficient to result in a new coefficient value.
Operation of First_E,mbodiment
In the preferred embodiment, the 6 MHz frequency band for each MCC
modern 82 of HDT 12 is allocated as shown in Figure 13. Although the MCC
modem 82 transmits and receives the entire 6 MHz band, the ISU modems 100
(Figure 8) are optimized for the specific application for which they are
designed
and may terminate/generate fewer than the total number of carriers or tones
allocated in the 6 MHz band- The upstream and downstream band allocations
are preferably symmetric. The upstream. 6 MHz bands from the MCC modems
82 lie in the 5-40. z spectrum and the downstream 6 MHz bands lie in the
725-760 MHz spectrum. One skilled in the art will recognize that if different
transmission media are utilized for upstream and downstream transmission, the
frequencies for transmission may be the same or overlap but still be non-
interfering.
There are three regions in each 6 MHz frequency band to support specific
operations, such as transport of telephony payload' data, transport of ISU
system
operations and control data (IOC control data), and upstream and downstream
synchronization. Each carrier or tone in the OFDMV? frequency band consists of
a
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sinusoid which is modulated in amplitude and phase to form a complex
constellation point as previously described. The fundamental symbol rate of
the
OFDM waveform is 8 Uiz, and there are a total of 552 tones in the 6 MHz band.
The following Table 2 summarizes the preferable modulation type and
bandwidth allocation for the various tone classifications.
Table 2 ~ -----
Band Number f Tones Modulation Capacity I Bandwidth
Allocation Carriers
Synch 24 ton--s(2 synch tones BPSK. n/a 216 kHz
Band at each end and 10
guard tones at each
said;
Payload 480 (240 DSO+ 32 QAM 19.2 4.32 Nffiz
Data channels) PS
48 (a every 20 data BPSK 384 kbps 432 kHz
channels or 24 IOC
channels)
lntra-band Remainder on each n/a n/a 1.0:12 MHz
guard end (516 kHz at
each end)
Composite 552 n/a n/a 6.0 MHz
Signal
Guard bands are provided at each end of the spectrum to allow for
selectivity filtering after transmission and prior to reception. A total of
240
telephony data channels are included throughout the band, which accommodates
a net data rate of 19.2 Mbps. This capacity was designed to account for
additive
ingress, thereby retaining enough support to achieve concentration of users to
the
central office. The IOC channels are interspersed throughout the band to
provide
redundancy and communication suppoa, to narrowband receivers located in the
HESUs. The IOC data rate is 16 kbps (two 13PSK tones at the symbol rate of 8
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kHz frames per second). Effectively, an IOC is provided for every 10 payload
data channels. An ISU, such as an HISU, that can only see a single IOC channel
would be forced to retune if the IOC channel is corrupted. However, an ISU
which can see multiple IOC channels can select an alternate IOC channel in the
event that the primary choice is corrupt, such as for an MISU.
The synchronization channels are duplicated at the ends of the band for
redundancy, and are offset from the main body of usable carriers to guarantee
that the synchronization channels do not interfere with the other used
channels.
The synchronization channels were previously described and will be further
described below. The synchronization channels are operated at a lower power
level than the telephony payload channels to also reduce the effect of any
interference to such channels. This power reduction also allows for a smaller
guard band to be used between the synchronization c eels and the payload
telephony channels.
l5 One synchronization or redundant synchronization channels may also be
implemented within the telephony channels as opposed to being offset
therefrom. In order to keep them from interfering with the telephony channels,
the synchronization channels may be implemented using a lower symbol rate.
For example, if the telephony channels are implemented at an 8 kHz symbol
rate,
the synchronization channels could be implemented at a 2 kHz symbol rate and
also may be at a lower power level.
The ISUs 1 00 are designed to receive a subband, as shown in Figure 16, of
the total aggregate 61 z spectrum. As an example, the HISU 68 will
preferably detect only 22 of the available 552 channels. 'this implementation
is
primarily a cost/power savings technique. By reducing the number of channels
being received, the sample rate and associated processing, requirements are
dramatically reduced and can be achievable with common conversion parts on
the market today.
A given HISU 68 is limited to receiving a maximum of 10 DSOs out of the
payload data channels in the HISU receiver's frequency view. The remaining
channels will be used as a guard interval. Furthermore, in order to reduce the
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power/cost requirements, synthesizing frequency steps will be limited to 198
kHz. An IOC channel is provided for as shown in Figure 16 so that every HISU
68 will always see an IOC channel for control of the HISU 68 via HILT 12.
The MISU 66 is designed to receive 13 subbands, as shown in Figure 16,
5 or 130 of the 240 available DSOs. Again, the tuning steps will be limited to
198
kHz to realize an efficient synthesizer implementation. These are preferred
values for the HISU 63 and MISU 66, and it will be noted by one skilled in the
art that many of the values specified herein can be varied without changing
the
scope or spirit of the invention as defined by the accompanying claims.
10 As known to one skilled in the art, there may be need to support operation
over channels in a bandwidth of less than 6 MFIz. With appropriate software
and
hardware modifications of the system, such reconfiguration is possible as
would
be apparent to one skilled in the art. For example, for a 21 system, in the
downstream, the T" 12 would generate the channels over a subset of the total
15 band. The HISUs are inherently narrowband and would be able to tune into
the
2 MHz band. The MISUs supporting 130 channels, would receive signals
beyond the 2 MHz band. They would require reduction in filter selectivity by
way of a hardware modification. An eighty (80) channel MISU would be able to
operate with the constraints of the 2 MHz. system. In the upstream, the HISUs
20 would generate signals within the 2 MHz band and the MISUs transmit section
would restrict the information generated to the narrower band. At the HDT, the
ingress filtering would. provide sufficient selectivity against out of band
signal
energy. The narrowband system would require synchronization bands at the
edges of the 2 MHz band.
25 As previously described, acquisition of signal parameters for initializing
the system for detection of the downstream information is ,performed using the
downstream synchronization channels. The ISUs use the carrier, amplitude,
timing recovery block 166 to establish the downstream synchronization of
frequency, amplitude and timing for such detection of downstream information.
30 The downstream signal constitutes a point to multi-point topology and the
OFDM waveform arrives at the ISUs via a single path in an inherently
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synchronous manner. In the upstream direction, each ISU 100 must be
initialized and activated through a process of upstream synchronization before
an
HDT 12 can enable the ISU 100 for transmission. The process of upstream
synchronization for the ISUs is utilized so that the waveform from distinct
ISUs
combine to a unified waveform at the HDT. The upstream synchronization
process, portions of which were previously described, involves various steps.
They include- ISU transmission level adjustment, upstream multicarrier symbol
alignment, carrier frequency adjustment, and round trip path delay adjustment.
Such synchronization is performed after acquisition of a 6 MHz band of
operation.
Generally, with respect to level: adjustment,, the HDT 12 calibrates the
measured signal strength of the upstream transmission received from an ISU 100
and adjusts the ISU 100 transmit level so that all ISUs are within acceptable
threshold. Level adjustment is performed prior to symbol alignment and path
delay adjustment to maximize the accuracy of these measurements.
Generally, symbol alignment is a necessary requirement for the
multicarrier modulation approach implemented by the MCC modems 82 and the
ISU modems 101. In the downstream direction of transmission, all information
received at the ISU 100 is generated by a single G MI3 56, so the symbols
modulated on each multicarrier are automatically phase aligned. However,
upstream symbol alignment at the MCC modem 82 receiver architecture varies
due to the multi-point to point nature of the IIFC distribution network 11 and
the
unequal delay paths of the ISUs 100. In order to have maximum receiver
efficiency, all upstream symbols must be aligned within a narrow phase margin.
This is done by providing an adjustable delay path parameter in each ISU 100
such that the symbol periods of all channels received upstream from the
different
ISUs are aligned at the point they reach the HDT 12.
Generally, round trip path delay adjustment is performed such that the
round trip delay from the HDT network interface 62 to all ISUs 100 and back to
the network interface 62 from all the ISUs 100 in a system must be equal. This
is required so that signaling m.ultifranze integrity is preserved throughout
the
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system. All round trip processing for the telephony transport section has a
predictable delay with the exception of the physical delay associated with
signal
propagation across the HFC distribution network 11 itself. ISUs 100 located at
close physical distance from the T 12 will have., less round trip delay than
ISUs located at the maximum distance from the T 12. Path delay adjustment
is implemented to force the transport system of all ISUs to have equal round
trip
propagation delay. Mu .s also maintains DS 1 multiframe alignment for DS I
channels transported through the system, maintaining in-band channel signaling
or robbed-bit signaling with the same alignment for voice services associated
with the same DS 1.
Generally, carrier frequency adjustment must be performed such that the
spacing between carrier frequencies is such as to maintain orthogonality of
the
carriers. If the multicarriers are not received at the MCC modem 82 in
orthogonal alignment., interference between the multicarriers may occur. Such
carrier frequency adjustment can be performed in a manner like that of symbol
timing or amplitude adjustment or may be implemented on the ISU as described
previously above.
In the initialization process, when the ISU has just been powered up, the
ISU 100 has no knowledge of which downstream 6i MHz frequency band it
should be receiving in. This results in the need for the acquisition of 6 MHz
band in the initialization process. Until an ISU 100 has successfully acquired
a 6
MHz band for operation, it implements a "sc gb approach to locate its
downstream frequency band. Generally, a local processor of the CXSU
controller 102 of ISU 100 starts with a default 6 MHz receive frequency band
somewhere in the range from 625 to 850 MHz. The ISU 100 waits for a period
of time, for example 100 milliseconds, in each 6 MHz band to look for a valid
6
MHz acquisition command which matches a unique identification number for the
ISU 100 after obtaining a valid synchronization signal; which unique
identifier
may take the form of or be based on a serial number of the ISU equipment. If a
valid 6 MHz acquisition command or valid synchronization command is not
found in that 6 MHz band, the CXSU controller 102 looks at the next 6 MHz
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band and the process is repeated. In this manner, as explained further below,
the
T 12 can tell the ISU 100 which 6 MHz band it should use for frequency
reception and later Waich band for frequency transmission upstream.
The process of initialization and activation of ISUs, as generally described
above, and tracking or follow-up synchronization is further described below.
This description is written using an MISU 66 in conjunction with a CXSU
controller 102 but is equally applicable to any ISU 100 implemented with an
equivalent controller logic. The coax master card logic (C C) 80 is instructed
by the shelf controller unit (SCNU) 53 to initialize and activate a particular
ISU
100. The SCNU uses an ISU designation number to address the ISU 100. The
CXMC 80 correlates the ISU designation number with are. equipment serial
number, or unique identifier, for the equipment. No two ISU equipments
shipped from the factory possess the same unique identifier. If the ISU 100
has
never before been initialized and activated in the current system database,
the
CXMC 80 chooses a personal identification number (PIN) code for the ISU 100
being initialized and activated. This PIN code is then stored in the CXMC 80
and
effectively represents the "address" for all communications with that ISU 100
which will follow. The CXMC 80 maintains a lookup table between each ISU
designation number, t:he unique identifier for the ISU equipment, and the PIN
code. Each ISU 100 associated with the CXMU 56 has a unique PIN address
code assignment. One PIN address code will be reserved for a broadcast feature
to oil ISUs, which allows for the HDT to send messages to all initialized and
activated ISUs 100.
The CXMC 80 sends an initialization and activation enabling message to
the MCC modem 82 which notifies the MCC modem 82 that the process is
beginning and the associated detection functionality in the MCC modem 82
should be enabled. Such functionality is performed at least in part by
carrier,
amplitude, timing recovery block 222 as shown in the MCC upstream receiver
architecture of Figure 26 and as previously discussed.
tr
The CXMC 80 transmits an identification message by the MCC modem
82 over all I C channels of the 6 MHz band in which it transmits. The message
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includes a PIN address code to be assigned to the ISM" being initialized and
activated, a command indicating that ISU initialization and activation should
be
enabled at the ISU 100, the unique identifier for the ISU equipment, such as
the
equipment serial number, and cyclical redundancy checksum (CRC). The
messages are sent periodically for a certain period of time, TsCAN, which is
shown
as 6.16 seconds in Figure 20 and which is also shown in Figure 19. This period
of time is the maximum time which an ISU can scan all downstream 6 Nffiz
bands, synchronize, and listen for a valid identification message. The
periodic
rate, for example 50 cosec, affects how quickly the ISU learns its identity.
The
CXMC 80 will never attempt to synchronize more than one ISU at a time, but
will attempt to identify several ISUs during burst identification as described
further below. A software timeout is implemented if an ISU does not respond
after some maximum time limit is exceeded. This timeout must be beyond the
maximum time limit required for an ISU to obtain synchronization functions.
During periodic transmission by CXMC 80, the ISU implements the
scanning approach to locate its downstream frequency band. The local processor
of the CXSU starts with a default 6 MHz receive frequency band somewhere in
the range from 625 to 850 MHz. The ISU 100 selects the primary
synchronization channel of the 6 MHz band and then tests for loss of
synchronization after a period of time, If loss of synchronization is still
present,
the secondary synchronization channel is selected and tested for loss of
synchronization after a period of time, If loss of synchronization is still
present,
then the ISV restarts selection of the synchronization channels on the next 6
band which may + be 1 MHz away but still 6 MHz in width. When loss of
synchronization is not present on a synchronization channel then the ISU
selects
the first subband including the IOC and listens for a correct identification
message. If a correct identification message is found which matches its unique
identifier then the PIN address code is latched into an appropriate register.
If a
correct identification message is not found in the first subband on that IOC
then
a middle subband and IOC is selected, such as the I Ith subband, and the ISU
again listens for the correct identification message. If the message again is
not
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correctly detected, then the ISU restarts on another 6 MHz band. The ISU
listens
for the correct identification message in a subband for a period of time equal
to
at least two times the CXMU transmission time, for example 100 msec when
trarnasÃnission time is 50 msec as described above. The identification command
is
a unique command in the ISU 100, as the 1SU 100 will not require a PIN address
code match to respond to such commands, but only a vali.d unique identifier
and
CRC match. If an un-initialized and tin-activated ISU 100 receives an
identification command from the CXMC 80 via the MCC modem 82 on an IOC
channel, data which matches the unique identifier and a valid CRC, the CXSU
102 of the ISU 100 will store the PIN address code transmitted with the
command and unique identifier. From this point on, the ISU 100 will only
respond to commands which address it by its correct PIN address code, or a
broadcast address code; unless, of course, the ISU is re-activated again and
given
a new PIN address code.
After the ISU 1 00 has received a match to its unique identifier, the ISU
will receive the upstraatmn frequency band command with a valid PIN address
code that tells the ISU 100 which 6 MHz band to use for upstream transmission
and the carrier or tone designations for the upstream IOC channel to be used
by
the ISU 100. The CC.SU controller 102 interprets the command and correctly
activates the ISU modem 101 of the ISU 100 for the correct upstream frequency
band to respond in. Once the ISU modem 101 has acquired the correct 6 z
band, the CXSU conFtroller 102 sends a message command to the ISU modem
101 to enable upstream transmission. 'Distributed loops utilizing the carrier,
amplitude, and timing recovery block 222 of the MCC modem upstream receiver
architecture of the h'DT 12 is used to lock the various ISU parameters for
upstream transmission, including amplitude, carrier frequency, symbol
ale ent, and path delay.
The HIT is then given information on the new ISU and provides
downstream commands for the various parameters to the subscriber ISU unit.
The ISU begins transmission in the upstream and he HDT 12 locks to the
upstream signal. The HDT 12 derives an error indicator with regard to the
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parameter being adjusted and commands the subscriber ISU to adjust such
parameter. The adjustment of error is repeated in the process until the
parameter
for ISU transmission is locked to the HDT 12.
More specifically, after the ISU 100 has acquired the 6 MHz band for
operation, the CXSU 102 sends a message command to the ISU modem 101 and
the ISU modem 101 transmits a synchronization pattern on a synchronization
channel in the primary synchronization band of the spectral allocation as
shown
in Figures 13-15. The upstream synchronization channels which are offset from
the payload data channels as allocated in Figures 13-18 include both a primary
and a redundant synchronization channel such that upstream synchronization can
still be accomplished if one of the synchronization channels is corrupted. The
HD T monitors one channel for every ISU.
The MCC modem 82 detects a valid signal and performs an amplitude
level measurement on a received signal from the ISU. The synchronization
pattern indicates to the CXMC 80 that the ISU 100 has received the activation
and initialization and frequency band commands and is ready to proceed with
upstream synchronizaation. The amplitude level is compared to a desired
reference level, The C C 80 determines whether or not the transmit level of
the ISU 100 should be adjusted and the amount of such adjustment. If level
adjustment is required, the CXMC 90 transmits a message on the downstream
IOC channel instructing the CXSU 102 of the ISU '100 to adjust the power level
of the transmitter of the ISU modem 1.01. The CX AC 80 continues to check the
receive power level from the ISU 100 and provides adjustment commands to the
ISU 100 until the level transmitted by the ISU 100 is acceptable. The
amplitude
is adjusted at the ISU as previously discussed. If amplitude equilibrium is
not
reached within a certain number of iterations of amplitude adjustment or if a
signal presence is never detected utilizing the primary synchronization
channel
then the same process is used on the redundant synchronization channel. If
amplitude equilibrium is not reached within a certain number of iterations of
amplitude adjustment or if a signal presence is never detected utilizing the
prhuary or redundant synchronization channels then the ISU is reset.
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Once transmission level adjustment of time ISU 100 is completed and has
been stabilized, the CXMC 80 and MCC modem 82 perform carrier frequency
locking. The MCC modem 82 detects the carrier frequency as transmitted by the
ISU 100 and performs a correlation on the received transmission from the ISU
to
calculate a carrier frequency error correction necessary to orthogonally align
the
multicarriers of all the upstream transmissions from the ISUs. The MCC modem.
82 returns a message. to the CS C 80 indicating the amount of carrier
frequency
error adjustment required to perform, frequency alignment for the ISU. The
CXMC 80 transmits a message on a downstream IOC chancel via the MCC
modem. 82 instructing the CXSU 102 to adjust the transmit frequency of the ISU
modern 101 and the process is repeated until the frequency has been
established
to within a certain tolerance for the OFDM channel spacing. Such adjustment
would be made via at least the synthesizer block 195 (Figures 24 and 25). If
frequency locking and adjustment is accomplished on the ISU as previously
described, then this frequency adjustment method is not utilized.
To establish orthogonality, the CXMC 80 and MCC modem 82 perform
symbol alignment. The MCC modern 82 detects the synchronization channel
modulated at a 8 kI4z frame rate transmitted by the ISU modem 101 and
performs a hardware correlation on the receive signal to calculate the delay
correction necessary to symbol align the upstream ISU transmission from all
the
distinct ISUs 100. The MCC modern 82 returns a message to the CXMC 80
indicating the amount of delay adjustment required to symbol align the ISU 100
such that all the symbols are received at the HDT 12 simultaneously. The
CXMC 80 transmits a message in a downstream IOC channel by the MCC
modem 82 instructing the CXSU controller 102 to adjust the delay of the ISt
modem 101 transmission and the process repeats until ISU symbol alignment is
achieved. Such symbol alignment would be adjusted via at least the clock delay
196 (Figures 24 and 25)_ Nurnerous iterations may be necessary to reach symbol
alignment equilibrium and if it is not reached within a predetermined number
of
iterations, then the ISU may again be reset.
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Simultaneously with symbol alignment, the CXMC 80 transmits a
message to the MCC modem 82 to perform path delay adjustment. The CXMC
80 sends a message on a downstream IOC channel via the MCC modem 82
instructing the CXSU controller 102 to enable the ISU modem 101 to transmit a
another signal on a synchronization channel which indicates the multiframe (2
kHz) alignment of the ISU 100_ The MCC modern 82 detects this multiframe
alignment pattern and performs a hardware correlation on the pattern. From
this
correlation, the modem 82 calculates the additional symbol periods required to
meet the round trip path delay of the communication system. The MCC modem
82 then returns a message to the C?IC 80 indicating the additional amount of
delay which must be added to meet the overall path delay requirements and the
CXMC then transmits a message on a downstream. IOC channel via the MCC
modem 82 instructing the CXSU controller 102 to relay a message to the ISU
modem 101 containing the path delay adjustment value. 'Numerous iterations
may be necessary to reach path delay equilibrium and if it is not reached
within a
predetermined number of iterations, then the ISU may again be reset. Such
adjustment is made in the ISU transmitter as can be seen in the display delay
buffer "n" samples 1 92 of the upstream transmitter architectures of Figures
24
and 25. Patti delay and symbol alignment may be performed at-the same time,
separately or together using the same or different signals sent on the
synchronization channel.
Until the ISU is initialized and activated, the ISU 100 has no capability of
transmitting telephony data information on any of the 480 tones or carers.
After such initialization and activation has been completed, the ISUs are
within
tolerance required for transmission within the OFUM waveform and the ISU is
informed that transmission is possible and upstream synchronization is
complete.
After an ISU 100 is initialized and activated for the system, follow-up
synchronization or tracking may be performed periodically to keep the ISTJs,
calibrated within the required tolerance of the OFDM transport requirements.
The follow-up process is implemented to account for, drift of component values
with temperature. Him ISU 100 is inactive for extreme periods of time, the ISU
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can be tuned to the synchronization channels and requested to update upstream
synchronization parameters in accordance with the upstream synchronization
process described above. Alternatively, if an ISU has been used recently, the
follow-up synchronization or tracking can proceed over an IOC channel. Under
this scenario, as generally shown in Figure 28, the ISU 100 is requested to
provide a signal over an IOC channel by the T 12. The HDT 12 then
acquires and verifies that the signal is within the tolerance required for a
channel
within the OFDM waveform. If not, then the ISI5 is requested to adjust such
errored parameters. In addition, during long periods of use, ISUs can also be
requested by the 11DT 12 to send a signal on an IOC channel or a
synchronization channel for the purpose of updating the upstream
synchronization parameters.
In the downstream direction, the IOC channels transport control
information to the ISUs 100, The modulation format is preferably
differentially
encoded BPSK, although the differential aspect of the downstream modulation is
not required. In the upstream direction, the IOC channels transport control
information to the I-IDT 12. The IOC channels are differentially BPSK
modulated to mitigate the transient time associated with. the equalizer when
sending data in the upstream direction. Control data is slotted on a byte
boundary (500 lts frame). Data from any ISU can be e mitted on an IOC
channel asynchronously; therefore, there is the potential for collisions to
occur.
As there is potential for collisions, detection of collisions on the upstream
IOC channels is accomplished at a data protocol level. The protocol for
handling
such collisions may,, for example, include exponential backoff by the ISUs. As
such, when the T 12 detects an error in transmission;, a retransmission
command is broadcast to all the ISUs such that the ISUs retransmit the
upstream
signal on the IOC channel after waiting a particular time; the wait time
period
being based on an exponential function.
One skilled in the art will recognize that upstream synchronization can be
implemented allowing for multi-point to point transmission using only the
symbol timing loop for adjustment of symbol timing by the ISUs as commanded
SU ` L T SHEET (RULE 26)
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