Note: Descriptions are shown in the official language in which they were submitted.
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NETWORK-ASSISTED GLOBAL POSITIONING SYSTEMS, METHODS
AND TERMINALS INCLUDING DOPPLER SHIFT AND CODE PHASE
ESTIMATES
Field of the Invention
This invention relates to cellular wireless communications systems,
methods and mobile terminals, and more particularly to cellular wireless
communications systems, methods and mobile terminals that include Global
Positioning System (GPS) capabilities.
Backuound of the Invention
Cellular wireless communications systems, methods and mobile terminals
are widely used for voice and/or data communications. As is well known to
those
having skill in the art, cellular wireless communications systems, methods and
mobile terminals include terrestrial cellular wireless communications systems,
methods and mobile terminals, and/or satellite cellular wireless
communications
systems, methods and mobile terminals. As used herein, the term "mobile
terminal" includes cellular
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and/or satellite radiotelephones with or without a multi-line display;
Personal
Communications System (PCS) terminals that may combine a radiotelephone with
data processing, facsimile and/or data communications capabilities; Personal
Digital
Assistants (PDA) that can include a radio frequency transceiver and a pager,
Internet/intranet access, Web browser, organizer, calendar and/or a global
positioning
system (GPS) receiver; and/or conventional laptop and/or palmtop computers or
other
appliances, which include a radio frequency transceiver.
It may be desirable, and may be mandatory in the future, that mobile terminals
be equipped to determine the geographical location thereof, for example, to
support
emergency position reporting, often referred to as "E911" position reporting.
One
way to accomplish this result is to add a GPS receiver to a mobile terminal.
As is
well known to those having skill in the art, GPS is a satellite navigation
system that is
funded by and controlled by the U.S. Department of Defense, that provides
specially
coded satellite signals that can be processed in a GPS receiver, enabling the
receiver
to compute position, velocity and/or time. A description of the GPS system may
be
found in the publication entitled Global Positioning System Overview by Peter
H.
Dana, 1999, the disclosure of which is hereby incorporated herein by reference
in its
entirety as if set forth fully herein. As used herein, the term "GPS" also
includes other
satellite-based systems that can be used to measure positions on the earth,
such as
GLONASS.
GPS receivers may be expensive, increase mobile terminal size and/or
consume the limited amount of battery power that is available to the mobile
terminal.
Accordingly, techniques have been proposed to integrate some or all of a GPS
receiver into a mobile terminal. See, for example, U.S. Patents 6,424,826 to
Horton et
al. entitled Systems and Methods for Sharing Reference Frequency Signals
Within a =
Wireless Mobile Terminal Between a Wireless Transceiver and a Global
Positioning
System Receiver; and 6,097,974 to Camp, Jr. et al. entitled Combined GPS and
Wide
Bandwidth Radiotelephone Terminals and Methods.
Moreover, it is known that GPS receivers may suffer from latency in
achieving a time to first fix, due to the time that it may take for the GPS
receiver to
download the necessary GPS satellite ephemeris data. Accordingly, systems have
been proposed to shorten the time to first fix. See, for example, U.S. Patents
6,415,154 to Wang et al. entitled Method and Apparatus for Communicating
Auxilliary Information and Location Information Between a Cellular Telephone
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Network and a Global Positioning System Receiver for Reducing Code Shift
Search
Time of the Receiver; 6,295,023 to Bloebaum entitled Methods, Mobile Stations
and
Systems for Acquiring Global Positioning System Timing Information; 6,169,514
to
Sullivan entitled Low-Power Satellite-Based Geopositioning System; and
5,663,734 to
Krasner entitled GPS Receiver and Method for Processing GPS Signals.
In-building GPS operation may be particularly challenging, because the GPS
receiver may need to overcome an additional 20dB-25dB degradation in received
signal-to-noise ratio due to in-building attenuation. This may produce a
hundredfold
or more increase in the GPS signal processing latency time, compared to an
outdoor
use.
Summary of the Invention
Wireless communications systems according to some embodiments of the
present invention include a terrestrial wireless network that is configured to
transmit
wireless communications including GPS data over a satellite frequency band,
and a
mobile terminal that is configured to receive the wireless communications
including
the GPS data from the terrestrial wireless network over the satellite
frequency band,
and to perform pseudo-range measurements using the GPS data that is received
over
the satellite frequency band. Accordingly, a terrestrial wireless network that
uses
satellite frequency bands for terrestrial communications may be used to
provide a
GPS assist to mobile terminals.
In other embodiments of the present invention, the mobile terminal is further
configured to transmit the pseudo-range measurements to a network operations
center.
In other embodiments, the network operations center is configured to determine
a
position of the mobile terminal using the pseudo-range measurements. In yet
other
embodiments, the network operations center is further configured to transmit
the
position of the mobile terminal to the mobile terminal. In some embodiments,
the
mobile terminal is configured to transmit the pseudo-range measurements to the
network operations center via the terrestrial wireless network. In other
embodiments,
the mobile terminal is configured to transmit the pseudo-range measurements to
the
network operations center via a space-based component.
In still other embodiments of the present invention, a mobile terminal is
configured to perform pseudo-range measurements by receiving GPS
coarse/acquisition (C/A) signals from a plurality of GPS satellites,
estimating Doppler
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shifts in the GPS coarse acquisition (C/A) signals and estimating received
code phases
of the GPS C/A signals using the Doppler shifts that are estimated. The
estimated
received code phases and/or the estimated Doppler shifts of the GPS C/A
signals,
along with the time of measurement, can provide the pseudo-range measurements.
In
these embodiments, by using the estimated Doppler shifts to estimate the
received
code phases of the GPS C/A signals, reduced code searching time may be
obtained. It
also will be understood that the embodiments described in this paragraph may
be used
by mobile terminals independent of the use of a terrestrial network to
transmit GPS
data over a satellite frequency band, and/or may be used by standalone GPS
receivers
with cellular data receiving capabilities.
More specifically, mobile terminals according to some embodiments of the
present invention include a receiver that is configured to receive GPS C/A
signals
from a plurality of GPS satellites and a processor that is configured to
estimate
Doppler shifts in the GPS C/A signals and to estimate received code phases of
the
GPS C/A signals using the Doppler shifts that are estimated. In some
embodiments,
the receiver is further configured to receive from a wireless network, a
Doppler shift
that is measured at the wireless network, and a code phase that is measured at
the
wireless network. The processor is further configured to estimate residual
Doppler
shifts in the GPS C/A signals due to mobile terminal motion, using the Doppler
shift
and code phase that are measured at the wireless network, and to estimate the
received
code phases of the GPS C/A signals using the Doppler shift that is estimated.
It also
will be understood that embodiments described in this paragraph may be used by
mobile terminals independent of the use of a terrestrial network to transmit
GPS data
over a satellite frequency band, and/or may be used by GPS processors with
cellular
data receiving capabilities.
In particular, in some embodiments, the processor is configured to estimate
the
residual Doppler shifts in the GPS C/A signals due to mobile terminal motion
by
bandpass filtering the GPS C/A signals into frequency slices (i.e., slices in
the
frequency domain), despreading the slices and estimating the Doppler shifts
from the
frequency slices that are despread. In other embodiments, the processor is
configured
to bandpass filter the GPS C/A into frequency slices by frequency translating
the GPS
C/A signals, low pass filtering the GPS C/A signals that are frequency
translated and
downsampling the low pass filtered, frequency translated GPS C/A signals. In
other
embodiments, the processor is configured to despread the frequency slices by
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generating an internal reference code sequence approximately matched to the
code
sequence used by the C/A signal, frequency translating the internal reference
code
sequence for each frequency slice, low pass filtering the frequency translated
reference code sequences and multiplying by the downsampled low pass filtered,
frequency translated, GPS C/A signal slices.
In still other embodiments, the processor is configured to estimate the
Doppler
shifts from the frequency slices that are despread by frequency-translating
the
frequency slices that are despread by the Doppler shift frequency that is
measured at
the wireless network, to obtain the residual Doppler shift complex frequency
due to
mobile terminal motion, transforming the despread frequency translated slice
sample
points to the frequency domain, converting frequency domain complex values to
magnitude values and adding the magnitude values on a point-by-point basis
across
the frequency slices. Moreover, in still other embodiments, the processor is
configured to estimate the code phases of the GPS C/A signals by removing a
total
Doppler shift by frequency-translating the GPS C/A signals by the sum of the
residual
Doppler shifts that are estimated plus the Doppler shift that is measured at
the
wireless network, summing segments of the GPS C/A signals from which the total
Doppler shift has been removed, correlating the summed segments with an
internally
generated code frame and determining a time offset corresponding to the
location of
the peak magnitude squared value. Finally, in still other embodiments, the
processor
is configured to estimate the code phases of the GPS C/A signals by Fourier-
transforming the despread slice signals to the frequency domain, determining
phase
angles corresponding to the estimated Doppler shift frequency for each of the
slices,
and determining a code phase from the phase angles.
In some embodiments, as described above, the wireless network is a terrestrial
wireless network. In other embodiments, the terrestrial wireless network
comprises a
terrestrial cellular network, an ancillary terrestrial network and/or a
wireless local
and/or wide area network. In other embodiments, the wireless network is a
satellite
wireless network, and the Doppler shift and C/A code phase are measured at the
satellite wireless network. In some embodiments, the Doppler shift and C/A
code
phase that are measured at the satellite wireless network are referenced to a
geographic point on the earth determined by measuring relative differences in
received signal levels between adjacent spot beams at the mobile terminal. In
other
embodiments, the Doppler shift and C/A code phase that are measured at the
satellite
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wireless network are referenced to a geographic point on the earth determined
by
measuring path delays between the mobile terminal and a satellite gateway via
at
least two satellite transmission paths.
According to an aspect of the present invention, there is provided a
wireless communications system comprising:
a terrestrial wireless network that is configured to transmit wireless
communications including Global Positioning System (GPS) data over a satellite
frequency band that is outside a GPS frequency band; and
a mobile terminal that is configured to receive the wireless
communications including the GPS data from the terrestrial wireless network
over
the satellite frequency band that is outside the GPS frequency band and to
perform
pseudo-range measurements using the GPS data that is received over the
satellite
frequency band that is outside the GPS frequency band.
According to another aspect of the present invention, there is provided a
terrestrial wireless network for a cellular wireless communications system
comprising:
a plurality of terrestrial base stations that are configured to transmit
wireless communications including global Positioning System (GPS) data to
mobile terminals over a satellite frequency band that is outside a GPS
frequency
band.
According to another aspect of the present invention, there is provided a
mobile terminal comprising:
a receiver that is configured to receive wireless communications including
Global Positioning System (GPS) data over a satellite frequency band that is
outside a GPS frequency band; and
a processor that is configured to perform pseudo-range measurements
using the GPS data that is received over the satellite frequency band that is
outside
the GPS frequency band.
According to another aspect of the present invention, there is provided a
mobile terminal comprising:
a receiver that is configured to receive Global Positioning System (GPS)
C/A signals from a plurality of GPS satellites; and
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a processor that is configured to estimate Doppler shifts in the GPS C/A
signals and to estimate received code phases of the GPS C/A signals using the
Doppler shifts that are estimated;
wherein the receiver is further configured to receive from a wireless
network a Doppler shift that is measured at the wireless network and a code
phase
that is measured at the wireless network and wherein the processor is further
configured to estimate residual Doppler shifts in the GPS C/A signals due to
mobile terminal motion using the Doppler shift and code phase that are
measured
at the wireless network and to estimate the received code phases of the GPS
C/A
signals using the Doppler shift that is estimated.
According to another aspect of the present invention, there is provided a
wireless communications method comprising:
transmitting wireless communications including Global Positioning
System (GPS) data over a terrestrial wireless network using a satellite
frequency
band that is outside a GPS frequency band;
receiving the wireless communications including the GPS data from the
terrestrial wireless network at a mobile terminal over the satellite frequency
band
that is outside the GPS frequency band; and
performing pseudo-range measurements at the mobile terminal using the
GPS data that is received over the satellite frequency band that is outside
the GPS
frequency band.
According to another aspect of the present invention, there is provided a
terrestrial wireless communications method comprising:
terrestrially transmitting wireless communications including Global
Positioning System (GPS) data to mobile terminals over a satellite frequency
band
that is outside a GPS frequency band.
According to another aspect of the present invention, there is provided a
mobile terminal operating method comprising:
receiving Global Positioning System (GPS) C/A signals from a plurality of
GPS satellites;
estimating Doppler shifts in the GPS C/A signals; and
estimating received code phases of the GPS C/A signals using the Doppler
shifts that are estimated;
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wherein receiving further comprises receiving from a wireless network a
Doppler shift that is measured at the wireless network and a code phase that
is
measured at the wireless network;
wherein estimating Doppler shifts comprises estimating residual Doppler
shifts in the C/A signals due to mobile terminal motion using the Doppler
shift
and code phase that are measured at the wireless network; and
wherein the estimating received code phases comprises estimating the
code phases of the GPS C/A signals using the Doppler shift that is estimated.
According to another aspect of the present invention, there is provided a
wireless communications system comprising:
a satellite network that is configured to transmit wireless communications
including Global Positioning System (GPS) data over a satellite frequency band
that is outside a GPS frequency band; and
a mobile terminal that is configured to receive the wireless
communications including the GPS data from the satellite network over the
satellite frequency band that is outside the GPS frequency band and to perform
pseudo-range measurements using the GPS data that is received over the
satellite
frequency band that is outside the GPS frequency band.
According to another aspect of the present invention, there is provided a
satellite network for a wireless communications system comprising:
at least one satellite that is configured to transmit wireless communications
including global Positioning System (GPS) data to mobile terminals over a
satellite frequency band that is outside a GPS frequency band.
According to another aspect of the present invention, there is provided a
mobile terminal comprising:
a receiver that is configured to receive Global Positioning System (GPS)
C/A signals from a plurality of GPS satellites; and
a processor that is configured to estimate Doppler shifts in the GPS C/A
signals and to estimate received code phases of the GPS C/A signals using the
Doppler shifts that are estimated.
According to another aspect of the present invention, there is provided a
wireless communications method comprising:
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transmitting wireless communications including Global Positioning
System (GPS) data over a satellite network using a satellite frequency band
that is
outside a GPS frequency band;
receiving the wireless communications including the GPS data from the
satellite network at a mobile terminal over the satellite frequency band that
is
outside the GPS frequency band; and
performing pseudo-range measurements at the mobile terminal using the
GPS data that is received over the satellite frequency band that is outside
the GPS
frequency band.
According to another aspect of the present invention, there is provided a
wireless communications method comprising:
transmitting wireless communications including Global Positioning
System (GPS) data from at least one satellite to mobile terminals over a
satellite
frequency band that is outside a GPS frequency band.
According to another aspect of the present invention, there is provided a
mobile terminal operating method comprising:
receiving Global Positioning System (GPS) C/A signals from a plurality of
GPS satellites;
estimating Doppler shifts in the GPS C/A signals; and
estimating received code phases of the GPS C/A signals using the Doppler
shifts that are estimated.
Finally, it will be understood that, although embodiments of the invention
have been described above in connection with systems and mobile terminals,
other
embodiments of the present invention can provide cellular wireless
communications methods, wireless networks and methods, and mobile terminal
processing methods.
Brief Description of the Drawinas
Figure 1 is a schematic diagram of cellular radiotelephone systems and
methods according to embodiments of the invention.
Figure 2 is a block diagram of adaptive interference reducers according to
embodiments of the present invention.
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Figure 3 is a spectrum diagram that illustrates satellite L-band frequency
allocations.
Figure 4 is a schematic diagram of cellular satellite systems and methods
according to other embodiments of the present invention.
Figure 5 illustrates time division duplex frame structures according to
embodiments of the present invention.
Figure 6 is a block diagram of architectures of ancillary terrestrial
components according to embodiments of the invention.
Figure 7 is a block diagram of architectures of reconfigurable
radiotelephones according to embodiments of the invention.
Figure 8 graphically illustrates mapping of monotonically decreasing
power levels to frequencies according to embodiments of the present invention.
Figure 9 illustrates an ideal cell that is mapped to three power regions and
three associated carrier frequencies according to embodiments of the
invention.
Figure 10 depicts a realistic cell that is mapped to three power regions and
three associated carrier frequencies according to embodiments of the
invention.
Figure 11 illustrates two or more contiguous slots in a frame that are
unoccupied according to embodiments of the present invention.
Figure 12 illustrates loading of two or more contiguous slots with lower
power transmissions according to embodiments of the present invention.
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Figure 13 illustrates cellular wireless communications systems and methods,
including terrestrial wireless networks, mobile terminals and associated
methods
according to some embodiments of the present invention.
Figure 14 illustrates cellular wireless communications systems and methods,
including terrestrial wireless networks, mobile terminals and associated
methods
according to other embodiments of the present invention.
Figure 15 is a flowchart of operations that may be performed by a mobile
terminal according to some embodiments of the present invention.
Figure 16 is a flowchart of operations that may be performed by a mobile
terminal according to other embodiments of the present invention.
Figure 17 is a block diagram of operations that may performed by a mobile
terminal according to still other embodiments of the present invention.
Figure 18 is a block diagram illustrating detailed embodiments of Figure 17.
Figure 19 is a flowchart of operations that may be performed to estimate
residual Doppler shift in embodiments of Figure 17.
Figure 20 is a frequency spectrum representation of filtering of received
samples into frequency slices according to embodiments of the present
invention.
Figure 21 is a block diagram illustrating additional detailed embodiments of
Figure 19.
Figure 22 graphically illustrates low pass filter gold code autocorrelation
peaks vs. offset chip periods according to some embodiments of the present
invention.
Figure 23 is a block diagram illustrating additional detailed embodiments of
Figure 19.
Figure 24 is a block diagram illustrating still other detailed embodiments of
Figure 17.
Figure 25 graphically illustrates interpolation for estimating code phase
according to some embodiments of the present invention.
Figure 26 is a block diagram illustrating yet other detailed embodiments of
Figure 17.
Figure 27 graphically illustrates a spot beam gain pattern as a function of
distance from beam center.
Figure 28 graphically illustrates an example of determining a mobile terminal
location within spot beam coverage according to some embodiments of the
present
invention.
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Figure 29 illustrates an example of an estimate of mobile terminal location
using gain contours according to some embodiments of the present invention.
Figure 30 illustrates an example of geo-location using ranging from two
diversity satellites according to some embodiments of the present invention
Detailed Description of Preferred Embodiments
The present invention now will be described more fully hereinafter with
reference to the accompanying drawings, in which embodiments of the invention
are
shown. However, this invention should not be construed as limited to the
embodiments set forth herein. Rather, these embodiments are provided so that
this
disclosure will be thorough and complete, and will fully convey the scope of
the
invention to those skilled in the art. Like numbers refer to like elements
throughout.
Figure 1 is a schematic diagram of cellular satellite radiotelephone systems
and methods according to embodiments of the invention. As shown in Figure 1,
these
cellular satellite radiotelephone systems and methods 100 include at least one
Space-
Based Component (SBC) 110, such as a satellite. The space-based component 110
is
configured to transmit wireless communications to a plurality of
radiotelephones
120a, 120b in a satellite footprint comprising one or more satellite
radiotelephone
cells 130-130" over one or more satellite radiotelephone forward link
(downlink)
frequencies fp. The space-based component 110 is configured to receive
wireless
communications from, for example, a first radiotelephone 120a in the satellite
radiotelephone cell 130 over a satellite radiotelephone return link (uplink)
frequency
fu. An ancillary terrestrial network, comprising at least one ancillary
terrestrial
component 140, which may include an antenna 140a and an electronics system
140b
(for example, at least one antenna 140a and at least one electronics system
140b), is
configured to receive wireless communications from, for example, a second
radiotelephone 120b in the radiotelephone cell 130 over the satellite
radiotelephone
uplink frequency, denoted fu, which may be the same as fu. Thus, as
illustrated in
Figure 1, radiotelephone 120a may be communicating with the space-based
component 110 while radiotelephone 120b may be communicating with the
ancillary
terrestrial component 140. As shown in Figure 1, the space-based component 110
also undesirably receives the wireless communications from the second
radiotelephone 120b in the satellite radiotelephone cell 130 over the
satellite
radiotelephone frequency fu as interference. More specifically, a potential
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interference path is shown at 150. In this potential interference path 150,
the return
link signal of the second radiotelephone 120b at carrier frequency fu
interferes with
satellite communications. This interference would generally be strongest when
fu =
fu, because, in that case, the same return link frequency would be used for
space-
based component and ancillary terrestrial component communications over the
same
satellite radiotelephone cell, and no spatial discrimination between satellite
radiotelephone cells would appear to exist.
Still referring to Figure 1, embodiments of satellite radiotelephone
systems/methods 100 can include at least one gateway 160 that can include an
antenna 160a and an electronics system 160b that can be connected to other
networks
162 including terrestrial and/or other radiotelephone networks. The gateway
160 also
communicates with the space-based component 110 over a satellite feeder link
112.
The gateway 160 also communicates with the ancillary terrestrial component
140,
generally over a terrestrial link 142.
Still referring to Figure 1, an Interference Reducer (IR) 170a also may be
provided at least partially in the ancillary terrestrial component electronics
system
140b. Alternatively or additionally, an interference reducer 170b may be
provided at
least partially in the gateway electronics system 160b. In yet other
alternatives, the
interference reducer may be provided at least partially in other components of
the
cellular satellite system/method 100 instead of or in addition to the
interference
reducer 170a and/or 170b. The interference reducer is responsive to the space-
based
component 110 and to the ancillary terrestrial component 140, and is
configured to
reduce the interference from the wireless communications that are received by
the
space-based component 110 and is at least partially generated by the second
radiotelephone 120b in the satellite radiotelephone cell 130 over the
satellite
radiotelephone frequency fu. The interference reducer 170a and/or 170b uses
the
wireless communications fu that are intended for the ancillary terrestrial
component
140 from the second radiotelephone 120b in the satellite radiotelephone cell
130 using
the satellite radiotelephone frequency fu to communicate with the ancillary
terrestrial
component 140.
In embodiments of the invention, as shown in Figure 1, the ancillary
terrestrial
component 140 generally is closer to the first and second radiotelephones 120a
and
120b, respectively, than is the space-based component 110, such that the
wireless
communications from the second radiotelephone 120b are received by the
ancillary
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terrestrial component 140 prior to being received by the space-based component
110.
The interference reducer 170a and/or 170b is configured to generate an
interference
cancellation signal comprising, for example, at least one delayed replica of
the
wireless communications from the second radiotelephone 120b that are received
by
the ancillary terrestrial component 140, and to subtract the delayed replica
of the
wireless communications from the second radiotelephone 120b that are received
by
the ancillary terrestrial component 140 from the wireless communications that
are
received from the space-based component 110. The interference reduction signal
may
be transmitted from the ancillary terrestrial component 140 to the gateway 160
over
link 142 and/or using other conventional techniques.
Thus, adaptive interference reduction techniques may be used to at least
partially cancel the interfering signal, so that the same, or other nearby,
satellite
radiotelephone uplink frequency can be used in a given cell for communications
by
radiotelephones 120 with the satellite 110 and with the ancillary terrestrial
component
140. Accordingly, all frequencies that are assigned to a given cell 130 may be
used
for both radiotelephone 120 communications with the space-based component 110
and with the ancillary terrestrial component 140. Conventional systems may
avoid
terrestrial reuse of frequencies within a given satellite cell that are being
used within
the satellite cell for satellite communications. Stated differently,
conventionally, only
frequencies used by other satellite cells may be candidates for terrestrial
reuse within
a given satellite cell. Beam-to-beam spatial isolation that is provided by the
satellite
system was relied upon to reduce or minimize the level of interference from
the
terrestrial operations into the satellite operations. In sharp contrast,
embodiments of
the invention can use an interference reducer to allow all frequencies
assigned to a
satellite cell to be used terrestrially and for satellite radiotelephone
communications.
Embodiments of the invention according to Figure 1 may arise from a
realization that the return link signal from the second radiotelephone 120b at
fu
generally will be received and processed by the ancillary terrestrial
component 140
much earlier relative to the time when it will arrive at the satellite gateway
160 from
the space-based component 110 via the interference path 150. Accordingly, the
interference signal at the satellite gateway 160b can be at least partially
canceled.
Thus, as shown in Figure 1, an interference cancellation signal, such as the
demodulated ancillary terrestrial component signal, can be sent to the
satellite
gateway 160b by the interference reducer 170a in the ancillary terrestrial
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140, for example using link 142. In the interference reducer 170b at the
gateway
160b, a weighted (in amplitude and/or phase) replica of the signal may be
formed
using, for example, adaptive transversal filter techniques that are well known
to those
having skill in the art. Then, a transversal filter output signal is
subtracted from the
aggregate received satellite signal at frequency fu that contains desired as
well as
interference signals. Thus, the interference cancellation need not degrade the
signal-
to-noise ratio of the desired signal at the gateway 160, because a regenerated
(noise-
free) terrestrial signal, for example as regenerated by the ancillary
terrestrial
component 140, can be used to perform interference suppression.
Figure 2 is a block diagram of embodiments of adaptive interference
cancellers that may be located in the ancillary terrestrial component 140, in
the
gateway 160, and/or in another component of the cellular radiotelephone system
100.
As shown in Figure 2, one or more control algorithms 204, known to those
having
skill in the art, may be used to adaptively adjust the coefficients of a
plurality of
transversal filters 202a-202n. Adaptive algorithms, such as Least Mean Square
Error
(LMSE), Kalman, Fast Kalman, Zero Forcing and/or various combinations thereof
or
other techniques may be used. It will be understood by those having skill in
the art
that the architecture of Figure 2 may be used with an LMSE algorithm. However,
it
also will be understood by those having skill in the art that conventional
architectural
modifications may be made to facilitate other control algorithms.
Additional embodiments of the invention now will be described with reference
to Figure 3, which illustrates L-band frequency allocations including cellular
radiotelephone system forward links and return links. As shown in Figure 3,
the
space-to-ground L-band forward link (downlink) frequencies are assigned from
1525
MHz to 1559 MHz. The ground-to-space L-band return link (uplink) frequencies
occupy the band from 1626.5 MHz to 1660.5 MHz. Between the forward and return
L-band links lie the GPS/GLONASS radionavigation band (from 1559 MHz to 1605
MHz).
In the present application, GPS/GLONASS will be referred to simply as GPS
for the sake of brevity. Moreover, the acronyms ATC and SBC will be used for
the
ancillary terrestrial component and the space-based component, respectively,
for the
sake of brevity.
As is known to those skilled in the art, GPS receivers may be extremely
sensitive since they are designed to operate on very weak spread-spectrum
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radionavigation signals that arrive on the earth from a GPS satellite
constellation. As
a result, GPS receivers may to be highly susceptible to in-band interference.
ATCs
that are configured to radiate L-band frequencies in the forward satellite
band (1525
to 1559 MHz) can be designed with very sharp out-of-band emissions filters to
satisfy
the stringent out-of-band spurious emissions desires of GPS.
Referring again to Figure 1, some embodiments of the invention can provide
systems and methods that can allow an ATC 140 to configure itself in one of at
least
two modes. In accordance with a first mode, which may be a standard mode and
may
provide highest capacity, the ATC 140 transmits to the radiotelephones 120
over the
frequency range from 1525 MHz to 1559 MHz, and receives transmissions from the
radiotelephones 120 in the frequency range from 1626.5 MHz to 1660.5 MHz, as
illustrated in Figure 3. In contrast, in a second mode of operation, the ATC
140
transmits wireless communications to the radiotelephones 120 over a modified
range
of satellite band forward link (downlink) frequencies. The modified range of
satellite
band forward link frequencies may be selected to reduce, compared to the
unmodified
range of satellite band forward link frequencies, interference with wireless
receivers
such as GPS receivers that operate outside the range of satellite band forward
link
frequencies.
Many modified ranges of satellite band forward link frequencies may be
provided according to embodiments of the present invention. In some
embodiments,
the modified range of satellite band forward link frequencies can be limited
to a
subset of the original range of satellite band forward link frequencies, so as
to provide
a guard band of unused satellite band forward link frequencies. In other
embodiments, all of the satellite band forward link frequencies are used, but
the
wireless communications to the radiotelephones are modified in a manner to
reduce
interference with wireless receivers that operate outside the range of
satellite band
forward link frequencies. Combinations and subcombinations of these and/or
other
techniques also may be used, as will be described below.
It also will be understood that embodiments of the invention that will now be
described in connection with Figures 4-12 will be described in terms of
multiple mode
ATCs 140 that can operate in a first standard mode using the standard forward
and
return links of Figure 3, and in a second or alternate mode that uses a
modified range
of satellite band forward link frequencies and/or a modified range of
satellite band
return link frequencies. These multiple mode ATCs can operate in the second,
non-
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standard mode, as long as desirable, and can be switched to standard mode
otherwise.
However, other embodiments of the present invention need not provide multiple
mode
ATCs but, rather, can provide ATCs that operate using the modified range of
satellite
band forward link and/or return link frequencies.
Embodiments of the invention now will be described, wherein an ATC
operates with an SBC that is configured to receive wireless communications
from
radiotelephones over a first range of satellite band return link frequencies
and to
transmit wireless communications to the radiotelephones over a second range of
satellite band forward link frequencies that is spaced apart from the first
range.
According to these embodiments, the ATC is configured to use at least one time
division duplex frequency to transmit wireless communications to the
radiotelephones
and to receive wireless communications from the radiotelephones at different
times.
In particular, in some embodiments, the at least one time division duplex
frequency
that is used to transmit wireless communications to the radiotelephones and to
receive
wireless communications from the radiotelephones at different times, comprises
a
frame including a plurality of slots. At least a first one of the slots is
used to transmit
wireless communications to the radiotelephones and at least a second one of
the slots
is used to receive wireless communications from the radiotelephones. Thus, in
some
embodiments, the ATC transmits and receives, in Time Division Duplex (TDD)
mode, using frequencies from 1626.5 MHz to 1660.5 MHz. In some embodiments,
all ATCs across the entire network may have the stated
configuration/reconfiguration
flexibility. In other embodiments, only some ATCs may be reconfigurable.
Figure 4 illustrates satellite systems and methods 400 according to some
embodiments of the invention, including an ATC 140 communicating with a
radiotelephone 120b using a carrier frequency f'u in TDD mode. Figure 5
illustrates
an embodiment of a TDD frame structure. Assuming full-rate GSM (eight time
slots
per frame), up to four full-duplex voice circuits can be supported by one TDD
carrier.
As shown in Figure 5, the ATC 140 transmits to the radiotelephone 120b over,
for
example, time slot number 0. The radiotelephone 120b receives and replies back
to
the ATC 140 over, for example, time slot number 4. Time slots number 1 and 5
may
be used to establish communications with another radiotelephone, and so on.
A Broadcast Control CHannel (BCCH) is preferably transmitted from the
ATC 140 in standard mode, using a carrier frequency from below any guard band
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' exclusion region. In other embodiments, a BCCH also can be defined using a
TDD
carrier. In any of these embodiments, radiotelephones in idle mode can, per
established GSM methodology, monitor the BCCH and receive system-level and
paging information. When a radiotelephone is paged, the system decides what
type of
= 5 resource to allocate to the radiotelephone in order to establish the
communications
link. Whatever type of resource is allocated for the radiotelephone
communications
channel (TDD mode or standard mode), the information is communicated to the
radiotelephone, for example as part of the call initialization routine, and
the
radiotelephone configures itself appropriately.
It may be difficult for the TDD mode to co-exist with the standard mode over
the same ATC, due, for example, to the ATC receiver LNA stage. In particular,
assuming a mixture of standard and TDD mode GSM carriers over the same ATC,
during the part of the frame when the TDD carriers are used to serve the
forward link
(when the ATC is transmitting TDD) enough energy may leak into the receiver
front
end of the same ATC to desensitize its LNA stage.
Techniques can be used to suppress the transmitted ATC energy over the 1600
MHz portion of the band from desensitizing the ATC's receiver LNA, and thereby
allow mixed standard mode and TDD frames. For example, isolation between
outbound and inbound ATC front ends and/or antenna system return loss may be
increased or maximized. A switchable band-reject filter may be placed in front
of the
LNA stage. This filter would be switched in the receiver chain (prior to the
LNA)
during the part of the frame when the ATC is transmitting TDD, and switched
out
during the rest of the time. An adaptive interference canceller can be
configured at
RF (prior to the LNA stage). If such techniques are used, suppression of the
order of
70 dB can be attained, which may allow mixed standard mode and TDD frames.
However, the ATC complexity and/or cost may increase.
Thus, even though ATC LNA desensitization may be reduced or eliminated, it
may use significant special engineering and attention and may not be
economically
worth the effort. Other embodiments, therefore, may keep TDD ATCs pure TDD,
with the exception, perhaps, of the BCCH carrier which may not be used for
traffic
but only for broadcasting over the first part of the frame, consistent with
TDD
protocol. Moreover, Random Access CHannel (RACH) bursts may be timed so that
they arrive at the ATC during the second half of the TDD frame. In some
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embodiments, all TDD ATCs may be equipped to enable reconfiguration in
response
to a command.
It is well recognized that during data communications or other applications,
the forward link may use transmissions at higher rates than the return link.
For
example, in web browsing with a radiotelephone, mouse clicks and/or other user
selections typically are transmitted from the radiotelephone to the system.
The
system, however, in response to a user selection, may have to send large data
files to
the radiotelephone. Hence, other embodiments of the invention may be
configured to
enable use of an increased or maximum number of time slots per forward GSM
carrier
frame, to provide a higher downlink data rate to the radiotelephones.
Thus, when a carrier frequency is configured to provide service in TDD mode,
a decision may be made as to how many slots will be allocated to serving the
forward
link, and how many will be dedicated to the return link. Whatever the decision
is, it
may be desirable that it be adhered to by all TDD carriers used by the ATC, in
order
to reduce or avoid the LNA desensitization problem described earlier. In voice
communications, the partition between forward and return link slots may be
made in
the middle of the frame as voice activity typically is statistically
bidirectionally
symmetrical. Hence, driven by voice, the center of the frame may be where the
TDD
partition is drawn.
To increase or maximize forward link throughput in data mode, data mode
TDD carriers according to embodiments of the invention may use a more
spectrally
efficient modulation and/or protocol, such as the EDGE modulation and/or
protocol,
on the forward link slots. The return link slots may be based on a less
spectrally
efficient modulation and/or protocol such as the GPRS (GMSK) modulation and/or
protocol. The EDGE modulation/protocol and the GPRS modulation/protocol are
well known to those having skill in the art, and need not be described further
herein.
Given an EDGE forward/GPRS return TDD carrier strategy, up to (384/2) = 192
kbps
may be supported on the forward link while on the return link the
radiotelephone may
transmit at up to (115/2) 64 kbps.
In other embodiments, it also is possible to allocate six time slots of an
eight-
slot frame for the forward link and only two for the return link. In these
embodiments, for voice services, given the statistically symmetric nature of
voice, the
return link vocoder may need to be comparable with quarter-rate GSM, while the
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forward link vocoder can operate at full-rate GSM, to yield six full-duplex
voice
circuits per GSM TDD-mode carrier (a voice capacity penalty of 25%). Subject
to
this non-symmetrical partitioning strategy, data rates of up to (384)(6/8) =
288 kbps
may be achieved on the forward link, with up to (115)(2/8) 32 kbps on the
return
link.
Figure 6 depicts an ATC architecture according to embodiments of the
invention, which can lend itself to automatic configuration between the two
modes of
standard GSM and TDD GSM on command, for example, from a Network Operations
Center (NOC) via a Base Station Controller (BSC). It will be understood that
in these
embodiments, an antenna 620 can correspond to the antenna 140a of Figures 1
and 4,
and the remainder of Figure 6 can correspond to the electronics system 140b of
Figures 1 and 4. If a reconfiguration command for a particular carrier, or set
of
carriers, occurs while the carrier(s) are active and are supporting traffic,
then, via the
in-band signaling Fast Associated Control CHannel (FACCH), all affected
radiotelephones may be notified to also reconfigure themselves and/or switch
over to
new resources. If carrier(s) are reconfigured from TDD mode to standard mode,
automatic reassignment of the carrier(s) to the appropriate standard-mode
ATCs,
based, for example, on capacity demand and/or reuse pattern can be initiated
by the
NOC. If, on the other hand, carrier(s) are reconfigured from standard mode to
TDD
mode, automatic reassignment to the appropriate TDD-mode ATCs can take place
on
command from the NOC.
Still referring to Figure 6, a switch 610 may remain closed when carriers are
to
be demodulated in the standard mode. In TDD mode, this switch 610 may be open
during the first half of the frame, when the ATC is transmitting, and closed
during the
second half of the frame, when the ATC is receiving. Other embodiments also
may
=
be provided.
Figure 6 assumes N transceivers per ATC sector, where N can be as small as
one, since a minimum of one carrier per sector generally is desired. Each
transceiver
is assumed to operate over one GSM carrier pair (when in standard mode) and
can
thus support up to eight full-duplex voice circuits, neglecting BCCH channel
overhead. Moreover, a standard GSM carrier pair can support sixteen full-
duplex
voice circuits when in half-rate GSM mode, and up to thirty two full-duplex
voice
circuits when in quarter-rate GSM mode.
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When in TDD mode, the number of full duplex voice circuits may be reduced
by a factor of two, assuming the same vocoder. However, in TDD mode, voice
service can be offered via the half-rate GSM vocoder with almost imperceptible
quality degradation, in order to maintain invariant voice capacity. Figure 7
is a block
diagram of a reconfigurable radiotelephone architecture that can communicate
with a
reconfigurable ATC architecture of Figure 6. In Figure 7, an antenna 720 is
provided,
and the remainder of Figure 7 can provide embodiments of an electronics system
for
the radiotelephone.
It will be understood that the ability to reconfigure ATCs and radiotelephones
according to embodiments of the invention may be obtained at a relatively
small
increase in cost. The cost may be mostly in Non-Recurring Engineering (NRE)
cost
to develop software. Some recurring cost may also be incurred, however, in
that at
least an additional RF filter and a few electronically controlled switches may
be used
per ATC and radiotelephone. All other hardware/software can be common to
standard-mode and TDD-mode GSM.
Referring now to Figure 8, other radiotelephone systems and methods
according to embodiments of the invention now will be described. In these
embodiments, the modified second range of satellite band forward link
frequencies
includes a plurality of frequencies in the second range of satellite band
forward link
frequencies that are transmitted by the ATCs to the radiotelephones at a power
level,
such as maximum power level, that monotonically decreases as a function of
(increasing) frequency. More specifically, as will be described below, in some
embodiments, the modified second range of satellite band forward link
frequencies
includes a subset of frequencies proximate to a first or second end of the
range of
satellite band forward link frequencies that are transmitted by the ATC to the
radiotelephones at a power level, such as a maximum power level, that
monotonically
decreases toward the first or second end of the second range of satellite band
forward
link frequencies. In still other embodiments, the first range of satellite
band return
link frequencies is contained in an L-band of satellite frequencies above GPS
frequencies and the second range of satellite band forward link frequencies is
contained in the L-band of satellite frequencies below the GPS frequencies.
The
modified second range of satellite band forward link frequencies includes a
subset of
frequencies proximate to an end of the second range of satellite band forward
link
frequencies adjacent the GPS frequencies that are transmitted by the ATC to
the
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radiotelephones at a power level, such as a maximum power level, that
monotonically
decreases toward the end of the second range of satellite band forward link
frequencies adjacent the GPS frequencies.
Without being bound by any theory of operation, a theoretical discussion of
the mapping of ATC maximum power levels to carrier frequencies according to
embodiments of the present invention now will be described. Referring to
Figure 8,
let V = X(p) represent a mapping from the power (p) domain to the frequency
(V)
range. The power (p) is the power that an ATC uses or should transmit in order
to
reliably communicate with a given radiotelephone. This power may depend on
many
factors such as the radiotelephone's distance from the ATC, the blockage
between the
radiotelephone and the ATC, the level of multipath fading in the channel,
etc., and as
a result, will, in general, change as a function of time. Hence, the power
used
generally is determined adaptively (iteratively) via closed-loop power
control,
between the radiotelephone and ATC.
The frequency (v) is the satellite carrier frequency that the ATC uses to
communicate with the radiotelephone. According to embodiments of the
invention,
the mapping X is a monotonically decreasing function of the independent
variable p.
Consequently, in some embodiments, as the maximum ATC power increases, the
carrier frequency that the ATC uses to establish and/or maintain the
communications
link decreases. Figure 8 illustrates an embodiment of a piece-wise continuous
monotonically decreasing (stair-case) function. Other monotonic functions may
be
used, including linear and/or nonlinear, constant and/or variable decreases.
FACCH
or Slow Associated Control CHannel (SACCH) messaging may be used in
embodiments of the invention to facilitate the mapping adaptively and in
substantially
real time.
Figure 9 depicts an ideal cell according to embodiments of the invention,
where, for illustration purposes, three power regions and three associated
carrier
frequencies (or carrier frequency sets) are being used to partition a cell.
For
simplicity, one ATC transmitter at the center of the idealized cell is assumed
with no
sectorization. In embodiments of Figure 9, the frequency (or frequency set) fl
is taken
from substantially the upper-most portion of the L-band forward link frequency
set,
for example from substantially close to 1559 MHz (see Figure 3).
Correspondingly,
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the frequency (or frequency set) fm is taken from substantially the central
portion of
the L-band forward link frequency set (see Figure 3). In concert with the
above, the
frequency (or frequency set) fo is taken from substantially the lowest portion
of the L-
band forward link frequencies, for example close to 1525 MHz (see Figure 3).
Thus, according to embodiments of Figure 9, if a radiotelephone is being
served within the outer-most ring of the cell, that radiotelephone is being
served via
frequency fo. This radiotelephone, being within the furthest area from the
ATC, has
(presumably) requested maximum (or near maximum) power output from the ATC.
In response to the maximum (or near maximum) output power request, the ATC
uses
its a priori knowledge of power-to-frequency mapping, such as a three-step
staircase
function of Figure 9. Thus, the ATC serves the radiotelephone with a low-value
frequency taken from the lowest portion of the mobile L-band forward link
frequency
set, for example, from as close to 1525 MHz as possible. This, then, can
provide
additional safeguard to any GPS receiver unit that may be in the vicinity of
the ATC.
Embodiments of Figure 9 may be regarded as idealized because they associate
concentric ring areas with carrier frequencies (or carrier frequency sets)
used by an
ATC to serve its area. In reality, concentric ring areas generally will not be
the case.
For example, a radiotelephone can be close to the ATC that is serving it, but
with
significant blockage between the radiotelephone and the ATC due to a building.
This
radiotelephone, even though relatively close to the ATC, may also request
maximum
(or near maximum) output power from the ATC. With this in mind, Figure 10 may
depict a more realistic set of area contours that may be associated with the
frequencies
being used by the ATC to serve its territory, according to embodiments of the
invention. The frequency (or frequency set) fi may be reused in the
immediately
adjacent ATC cells owing to the limited geographical span associated with fi
relative
to the distance between cell centers. This may also hold for fm.
Referring now to Figure 11, other modified second ranges of satellite band
forward link frequencies that can be used by ATCs according to embodiments of
the
present invention now will be described. In these embodiments, at least one
frequency in the modified second range of satellite band forward link
frequencies that
is transmitted by the ATC to the radiotelephones comprises a frame including a
plurality of slots. In these embodiments, at least two contiguous slots in the
frame
that is transmitted by the ATC to the radiotelephones are left unoccupied. In
other
embodiments, three contiguous slots in the frame that is transmitted by the
ATC to the
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radiotelephones are left unoccupied. In yet other embodiments, at least two
contiguous slots in the frame that is transmitted by the ATC to the
radiotelephones are
transmitted at lower power than remaining slots in the frame. In still other
embodiments, three contiguous slots in the frame that is transmitted by the
ATC to the
radiotelephones are transmitted at lower power than remaining slots in the
frame. In
yet other embodiments, the lower power slots may be used with first selected
ones of
the radiotelephones that are relatively close to the ATC and/or are
experiencing
relatively small signal blockage, and the remaining slots are transmitted at
higher
power to second selected ones of the radiotelephones that are relatively far
from the
ATC and/or are experiencing relatively high signal blockage.
Stated differently, in accordance with some embodiments of the invention,
only a portion of the TDMA frame is utilized. For example, only the first four
(or last
four, or any contiguous four) time slots of a full-rate GSM frame are used to
support
traffic. The remaining slots are left unoccupied (empty). In these
embodiments,
capacity may be lost. However, as has been described previously, for voice
services,
half-rate and even quarter-rate GSM may be invoked to gain capacity back, with
some
potential degradation in voice quality. The slots that are not utilized
preferably are
contiguous, such as slots 0 through 3 or 4 through 7 (or 2 through 5, etc.).
The use of
non-contiguous slots such as 0, 2, 4, and 6, for example, may be less
desirable. Figure
11 illustrates four slots (4-7) being used and four contiguous slots (0-3)
being empty
in a GSM frame.
It has been found experimentally, according to these embodiments of the
invention, that GPS receivers can perform significantly better when the
interval
between interference bursts is increased or maximized. Without being bound by
any
theory of operation, this effect may be due to the relationship between the
code
repetition period of the GPS C/A code (1 msec) and the GSM burst duration
(about
0.577 msec). With a GSM frame occupancy comprising alternate slots, each GPS
signal code period can experience at least one "hit", whereas a GSM frame
occupancy
comprising four to five contiguous slots allows the GPS receiver to derive
sufficient
clean information so as to "flywheel" through the error events.
According to other embodiments of the invention, embodiments of Figures 8-
10 can be combined with embodiments of Figure 11. Furthermore, according to
other
embodiments of the invention, if an fl carrier of Figures 9 or 10 is
underutilized,
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because of the relatively small footprint of the inner-most region of the
cell, it may be
used to support additional traffic over the much larger outermost region of
the cell.
Thus, for example, assume that only the first four slots in each frame of fI
are
being used for inner region traffic. In embodiments of Figures 8-10, these
four f1 slots
are carrying relatively low power bursts, for example of the order of 100 mW
or less,
and may, therefore, appear as (almost) unoccupied from an interference point
of view.
Loading the remaining four (contiguous) time slots of fl with relatively high-
power
bursts may have negligible effect on a GPS receiver because the GPS receiver
would
continue to operate reliably based on the benign contiguous time interval
occupied by
the four low-power GSM bursts. Figure 12 illustrates embodiments of a frame at
carrier fi supporting four low-power (inner interval) users and four high-
power (outer
interval) users. In fact, embodiments illustrated in Figure 12 may be a
preferred
strategy for the set of available carrier frequencies that are closest to the
GPS band.
These embodiments may avoid undue capacity loss by more fully loading the
carrier
frequencies.
The experimental finding that interference from GSM carriers can be
relatively benign to GPS receivers provided that no more than, for example, 5
slots
per 8 slot GSM frame are used in a contiguous fashion can be very useful. It
can be
particularly useful since this experimental finding may hold even when the GSM
carrier frequency is brought very close to the GPS band (as close as 1558.5
MHz) and
the power level is set relatively high. For example, with five contiguous time
slots
per frame populated, the worst-case measured GPS receiver may attain at least
30 dB
of desensitization margin, over the entire ATC service area, even when the ATC
is
radiating at 1558.5 MHz. With four contiguous time slots per frame populated,
an
additional 10 dB desensitization margin may be gained for a total of 40 dB for
the
worst-case measured GPS receiver, even when the ATC is radiating at 1558.5
MHz.
There still may be concern about the potential loss in network capacity
(especially in data mode) that may be incurred over the frequency interval
where
embodiments of Figure 11 are used to underpopulate the frame. Moreover, even
though embodiments of Figure 12 can avoid capacity loss by fully loading the
carrier,
they may do so subject to the constraint of filling up the frame with both low-
power
and high-power users. Moreover, if forward link carriers are limited to 5
contiguous
high power slots per frame, the maximum forward link data rate per carrier
that may
be aimed at a particular user may become proportionately less.
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Therefore, in other embodiments, carriers which are subject to contiguous
empty/low power slots are not used for the forward link. Instead, they are
used for the
return link. Consequently, in some embodiments, at least part of the ATC is
configured in reverse frequency mode compared to the SBC in order to allow
maximum data rates over the forward link throughout the entire network. On the
reverse frequency return link, a radiotelephone may be limited to a maximum of
5
slots per frame, which can be adequate for the return link. Whether the five
available
time slots per frame, on a reverse frequency return link carrier, are assigned
to one
radiotelephone or to five different radiotelephones, they can be assigned
contiguously
in these embodiments. As was described in connection with Figure 12, these
five
contiguous slots can be assigned to high-power users while the remaining three
slots
may be used to serve low-power users.
Other embodiments may be based on operating the ATC entirely in reverse
frequency mode compared to the SBC. In these embodiments, an ATC transmits
over
the satellite return link frequencies while radiotelephones respond over the
satellite
forward link frequencies. If sufficient contiguous spectrum exists to support
CDMA
technologies, and in particular the emerging Wideband-CDMA 3G standard, the
ATC
forward link can be based on Wideband-CDMA to increase or maximize data
throughput capabilities. Interference with GPS may not be an issue since the
ATCs
transmit over the satellite return link in these embodiments. Instead,
interference may
become a concern for the radiotelephones. Based, however, on embodiments of
Figures 11-12, the radiotelephones can be configured to transmit GSM since ATC
return link rates are expected, in any event, to be lower than those of the
forward link.
Accordingly, the ATC return link may employ GPRS-based data modes, possibly
even EDGE. Thus, return link carriers that fall within a predetermined
frequency
interval from the GPS band-edge of 1559 MHz, can be under loaded, per
embodiments of Figures 11 or 12, to satisfy GPS interference concerns.
Finally, other embodiments may use a partial or total reverse frequency mode
and may use CDMA on both forward and return links. In these embodiments, the
ATC forward link to the radiotelephones utilizes the frequencies of the
satellite return
link (1626.5 MHz to 1660.5 MHz) whereas the ATC return link from the
radiotelephones uses the frequencies of the satellite forward link (1525 MHz
to 1559
MHz). The ATC forward link can be based on an existing or developing CDMA
technology (e.g., IS-95, Wideband-CDMA, etc.). The ATC network return link can
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also be based on an existing or developing CDMA technology provided that the
radiotelephone's output is gated to cease transmissions for approximately 3
msec once
every T msec. In some embodiments, T will be greater than or equal to 6 msec.
This gating may not be needed for ATC return link carriers at approximately
1550 MHz or below. This gating can reduce or minimize out-of-band interference
(desensitization) effects for GPS receivers in the vicinity of an ATC. To
increase the
benefit to GPS, the gating between all radiotelephones over an entire ATC
service
area can be substantially synchronized. Additional benefit to GPS may be
derived
from system-wide synchronization of gating. The ATCs can instruct all active
radiotelephones regarding the gating epoch. All ATCs can be mutually
synchronized
via GPS.
Network-Assisted Global Positioning Systems, Methods and Terminals Including
Doppler Shift Estimates
Figure 13 illustrates cellular wireless communications systems and methods,
including wireless networks, mobile terminals and associated methods according
to
some embodiments of the present invention. As shown in Figure 13, these
cellular
wireless communications systems and methods may communicate with one or more
mobile terminals 1310 via a plurality of cells 1330 served by a terrestrial
wireless
network that includes a plurality of terrestrial base stations 1320. Although
only one
cell 1330 is shown in Figure 13, a typical cellular wireless communications
system/method may comprise hundreds of cells and may serve thousands of mobile
terminals 1310. It also will be understood by those having skill in the art
that, in
some embodiments of the present invention, the cells 1330 may correspond to
cells of
an Ancillary Terrestrial Network (ATN), as were described in the preceding
section,
the terrestrial base stations 1320 may correspond to an Ancillary Terrestrial
Component (ATC) 140 in an ancillary terrestrial network, as was described in
the
preceding section, and the mobile terminals 1310 may correspond to a
radiotelephone
120, as was described in the preceding section.
Still referring to Figure 13, the terrestrial wireless network including the
plurality of cells 1330 and the plurality of terrestrial base stations 1320,
are
configured to transmit wireless communications including GPS data 1322 over a
satellite frequency band Fs, such as one or more portions of the L-band
satellite
frequency spectrum illustrated in Figure 3. In some embodiments, the satellite
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frequency band Fs is outside the GPS frequency band illustrated in Figure 3.
The
mobile terminal 1310 is configured to receive the wireless communications
including
the GPS data 1322 from the terrestrial wireless network, including the
terrestrial base
station 1320, over the satellite frequency band Fs, for example using a
receiver of a
transceiver (transmitter/receiver) 1314. The mobile terminal 1310 also is
configured
to receive GPS signals 1342 from a plurality, shown in Figure 13 as three, of
GPS
satellites 1340. The mobile terminal 1310 also is configured to perform pseudo-
range
measurements using the GPS signals 1342 that are received from the GPS
satellites
1340 and the GPS data 1322 that is received over the satellite frequency band
Fs, for
example, using a processor 1316. Accordingly, a satellite frequency band is
used
terrestrially to provide GPS assist to a mobile terminal. It also will be
understood
that, in some embodiments, the terrestrial base station 1320 is a terrestrial
base station
of a conventional terrestrial cellular communications system that uses at
least one
satellite frequency band Fs for transmission.
Figure 14 illustrates cellular wireless communications systems and methods
including wireless networks, mobile terminals and associated methods according
to
other embodiments of the present invention. Referring now to Figure 14, these
embodiments of the present invention also include a network operations center
(NOC)
1454 that is connected to the terrestrial base stations 1320, either directly
or via one or
more intermediary network elements, such as a Mobile Telephone Switching
Office
(MTSO). As shown in Figure 14, in some embodiments, the mobile terminal 1310
is
further configured to transmit the pseudo-range measurements 1312 to the
network
operations center 1454 via the terrestrial base station 1320, using a
terrestrial cellular
frequency band, a satellite cellular frequency band and/or any other frequency
band.
The network operations center 1454 is configured to receive the pseudo-range
measurements 1312, and to determine a position of the mobile terminal 1310
using
the pseudo-range measurements 1312. Accordingly, in some embodiments, as shown
in Figure 14, the mobile terminal 1310 is configured to transmit the pseudo-
range
measurements 1312 to the network operation center 1454 via the terrestrial
wireless
network, including the terrestrial base stations 1320.
In still other embodiments of the present invention, as also illustrated in
Figure
14, a space-based component 1450 also is included that is configured to
wirelessly
communicate with the mobile terminal 1310 over the satellite frequency band F.
A
feeder link 1456 is used for communications between the space-based component
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1450 and a gateway 1452. The gateway 1452 may be connected to the network
operations center 1454 either directly or via one or more intermediary network
elements. A service link also is provided for transmission from the space-
based
component 1450 to the mobile terminal 1310 via a forward service link 1458a,
and to
receive communications from the mobile terminal 1310 via a return service link
1458b. Moreover, in some embodiments, the mobile terminal is configured to
transmit the pseudo-range measurements to the network operation center 1454
via the
return service link 1458b, the space-based component 1450, the feeder link
1456, and
the gateway 1452. In still other embodiments, combinations of the terrestrial
base
stations 1320 and the space-based component 1450 may be used to receive the
pseudo-range measurements from the mobile terminal 1310 and to transmit the
pseudo-range measurements to the network operations center 1454.
Figure 15 is a flowchart of operations that may be performed by a mobile
terminal, such as a mobile terminal 1310 of Figures 13 or 14, to provide
terrestrial
wireless network-assisted GPS measurements. It will be understood by those
having
skill in the art that these operations may be performed, for example, by the
radio
frequency transceiver 1314 and the processor 1316 that is included in mobile
terminals 1310.
It also will be understood by those having skill in the art that the
operations of
Figure 15 may be performed based on GPS data from a wireless network that may
be
a terrestrial and/or satellite wireless network using terrestrial and/or
satellite
frequency bands. Accordingly, embodiments of the present invention that are
described in Figure 15 may be performed independent of the use of a satellite
frequency band by a terrestrial wireless network, to convey GPS data to the
mobile
terminal.
Referring now to Figure 15, at Block 1510, GPS C/A signals are received at
the mobile terminal such as the mobile terminal 1310, for example as part of
the GPS
signals 1342 that are received from the plurality of GPS satellites 1340. At
Block
1520, the processor of the mobile terminal, such as the processor 1316 of the
mobile
terminal 1310, estimates the Doppler shift in the C/A signals. At Block 1530,
the
received code phases of the GPS C/A signals are then estimated using the
estimated
Doppler shifts that were obtained at Block 1520. Finally, at Block 1540, the
estimated Doppler shifts and/or the estimated received code phases are
transmitted by
the mobile terminal, for example as part of the transmission of pseudo-range
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measurements 1312, to the network operations center (NOC) 1454 using the
terrestrial
base stations 1320 and/or the space-based component 1450, over a satellite,
terrestrial
and/or other frequency band, as was described above. Accordingly, the
estimated
Doppler shifts (Block 1520) are used to estimate the received code phases of
the GPS
C/A signals (Block 1530). This can reduce the amount of signal processing used
to
estimate the received code phases of the GPS C/A signals.
It will be understood that in some embodiments of the present invention,
operations of Figure 15 may be performed by a mobile terminal that includes a
GPS
processor and a cellular data transceiver (but not a cellular voice
transceiver) therein,
wherein the estimated Doppler shifts are used by a GPS processor to reduce the
amount of signal processing to estimate the received code phases of the GPS
C/A
signals. In other embodiments of Figure 15, the mobile terminal may include a
GPS
receiver, a terrestrial cellular voice and data transceiver and/or a satellite
cellular
voice and data transceiver. In some of these embodiments, the mobile terminal
may
be configured to operate in conjunction with an ancillary terrestrial network,
as was
described above.
Figure 16 is a flowchart of other operations that may be performed in the
mobile terminal 1310 according to other embodiments of the present invention.
These embodiments use GPS data, such as the GPS data 1322 of Figures 13 or 14,
that may be received from a wireless network, such as the terrestrial wireless
networks of Figures 13 or 14 and/or a satellite wireless network, to allow the
computations for computing the estimated code phases of the GPS signals to be
reduced.
In particular, referring to Figure 16, at Block 1610, the measured Doppler
shifts and measured code phases are received from the terrestrial and/or
satellite
wireless network. These parameters may be received as part of the GPS data
1322
that is conveyed from the terrestrial wireless network of Figures 13 or 14
and/or a
satellite wireless network. At Block 1510, the GPS C/A signals are received,
as was
already described. At Block 1620, Doppler shifts in the GPS C/A signals are
estimated using the measured Doppler shifts and C/A code phases conveyed from
the
wireless network. Then, at Block 1630, the code phases of the received GPS C/A
signals are estimated using the estimated Doppler shifts of Block 1620.
Finally, at
Block 1540, the estimated Doppler shift and/or the estimated received code
phases are
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transmitted to the NOC. The NOC can compute a position fix and transmit the
position of the mobile terminal to the mobile terminal.
It will be understood that in some embodiments of the present invention,
operations of Figure 16 may be performed, for example, by a mobile terminal
that
includes a GPS receiver, a terrestrial cellular transceiver and/or a satellite
cellular
transceiver for receiving the measured Doppler shifts and the measured code
phases
from a terrestrial wireless network. In still other embodiments, the
operations of
Figure 16 may be performed in connection with an ancillary terrestrial
network, as
was described above. In yet other embodiments, the operations of Figure 16 may
be
performed in connection with a satellite and/or terrestrial wireless network,
as was
described above.
Some embodiments of the present invention can support both outdoor and in-
building position reporting for mobile terminals 1310 communicating with a
terrestrial base station 1320 such as an ATC and/or communicating with a
satellite
wireless network using terrestrial and/or satellite wireless frequency bands.
Some
embodiments can exploit the relatively small size of the terrestrial network
cells 1330
to reduce or minimize the computational requirements at the mobile terminal
1310,
while reducing or minimizing the amount of pseudo-range data transferred to
the
NOC 1454. Some embodiments can preprocess the received GPS signals at the
mobile terminal 1310, which can enable further computational reductions.
Some embodiments of the present invention can be applied to any cellular
network architecture and/or local and/or wide area network architecture,
wherein the
base stations can comprise terrestrial cellular network base stations,
ancillary
terrestrial network base stations and/or access points of a wireless local
and/or wide
area network. In some embodiments, the cell sizes may be on the order of a few
miles
radius or less. Embodiments of the invention may also be used with cellular
satellite-
mode operation by providing mechanisms to determine the approximate location
(for
example within a few miles) of the mobile terminal within its assigned
satellite beam.
Two embodiments for estimating an approximate mobile terminal position within
the
satellite beam will be described in detail below in connection with Figures 27-
30.
In some embodiments of the invention, the Doppler shift and code phase of the
received coarse/acquisition (C/A) signals 1342 from each GPS Satellite 1340,
also
referred to herein as a Space Vehicle (SV), are estimated at the mobile
terminal 1310.
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The small cell size can be taken advantage of to reduce the maximum a priori
uncertainty in these parameters as follows:
Reduced Code Phase Range: The C/A code is a repeating 1023-chip
Gold Code sequence with chip rate of 1.023 Mcps. Thus, the chip "length"
(relative to the speed of light) is 3.0e+8 m/s / 1.023e+6 = 293.26 meters.
Approximating the cell coverage as a flat circle of radius R, then the
maximum range variation to the SV, denoted "A", from the center of the cell
to the cell edge is:
A = R cos(E1), where El is the elevation angle to the SV. (1)
The code phase is the offset, in chip periods, of the received C/A Gold Code
frame relative to the start of the frame, measured at a particular observation
time. In this approach, each terrestrial base station, also referred to herein
as a
BTS, is equipped with a GPS receiver with which it can accurately measure
the code phase of each C/A signal in view and download this information to
the mobile terminal. Referring to Equation (1), for the worst-case elevation
angle of 00 and a typical cell radius R of 1000m, the value of A in terms of
chip lengths is equal to 1000 / 293.26 = 3.41 chips. Since the BTS is
typically
located near the center of the cell, then the measured code phase at any
location within the cell generally will lie within about 4 chips of the code
phase measured at the BTS. Code phase range uncertainty is thereby reduced.
Reduced Doppler Shift Range: By similar analysis to the above, the
precise Doppler shift of the received C/A signal due to SV motion (which
generally is the largest contributor to Doppler shift) is also measured at the
BTS and downloaded to the mobile terminal. Because of the cell's small size
compared to the distance to the GPS satellite, this Doppler shift component
generally will be approximately the same for any location within the cell
1330.
Thus, the Doppler frequency search at the mobile terminal can be reduced to
just a small range about this known SV Doppler component. Any residual
frequency offset (after subtracting out the SV Doppler shift), which may be
referred to herein as "residual Doppler shift", will be primarily due to the
Doppler shift caused by the motion of the mobile terminal, and may also
possibly include any local oscillator frequency offset error and multi-path
fading effects. This analysis assumes that the mobile terminal's local-
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oscillator is sufficiently accurate so that down-conversion frequency errors
can
be ignored.
Assume an operating scenario where the mobile terminal is inside a
moving vehicle with no direct line-of-sight to any SV, which may approximate
the signal-to-noise ratio of the in-building case. Then, the maximum residual
Doppler shift due to the vehicle motion should be accommodated. Given a
vehicle velocity v (m/s), it can be shown that the maximum residual Doppler
shift Afv due to vehicle motion is:
Aft, (Hz) = fsource = ( 1, = 1.57542e+9 = ( v / 3.0e+8) (2)
For a vehicle traveling at 60 mi/hr (26.8 rn/sec) directly toward an SV
located
near the horizon (worst case), the Doppler shift due to vehicle motion Afv =
141 Hz. Thus a residual Doppler search range of about 160 Hz should be
accommodated for this case.
Figure 17 is a block diagram of operations that may be performed in a mobile
terminal, such as a mobile terminal 1310, according to some embodiments of the
present invention. As shown in Figure 17, received GPS signals 1342, that
generally
include noise, are received at a receiver RF section 1750 which may be part of
a
transceiver 1314. At Block 1710, a snapshot of time samples of the received
GPS
signals plus noise is collected, which may correspond to Block 1510 of Figure
15. At
Block 1720, the time samples are processed to estimate the Doppler shift of
each
received C/A signal, which may correspond to Block 1520 of Figure 15. At Block
1730, the Doppler shift estimate from Block 1720 is used to reprocess the time
samples to estimate the code phase of each received C/A signal, which may
correspond to Block 1530 of Figure 15. The code phase estimate and Doppler
shift
estimate are then transmitted to the NOC at Block 1740, which may correspond
to
Block 1540 of Figure 15. As will be described in detail below, the
determination of
the Doppler shift in Block 1720 can allow for computationally efficient
calculation of
C/A code phase in Block 1730.
Upon collecting the received signal plus noise samples in Block 1710, the
mobile terminal transmits to the NOC (for example, via the BTS 1320) the
precise
time at which the GPS signal samples in Block 1710 were measured. Using the SV
ephemeris (carried by GPS signals 1342), the NOC 1454 (for example via the BTS
1320) informs the mobile terminal 1310 of which SVs 1340 were in view at the
mobile terminal's approximate location at the time of its measurement in Block
1710.
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In some embodiments, before proceeding to Blocks 1720 and 1730, the mobile
terminal downloads the following information from the BTS 1320 for each C/A
signal
to be processed:
1. The Doppler frequency shift due to SV motion measured at the BTS,
corresponding to the mobile terminal's observation period. Since the BTS is
not moving, this can simply be the total Doppler shift measured at the BTS.
(See Block 1610.)
2. The code phase 'measured at the BTS corresponding to the mobile terminal's
measurement time. (See Block 1610.)
3. The 50 bps ephemeris data bit sequence transmitted during the mobile's
observation period. Each C/A signal is overlaid with 50 bps binary data
whose bit transitions reverse the sign of the underlying Gold Code chips.
Before the C/A signals are processed, the received samples are first
multiplied
by an exact replica of the 50 bps bit sequence to strip out this data
modulation.
To reduce processing time and/or for the in-building case, where the SV signal
strength may be too low to allow the mobile to recover the 50 bps data
directly
from the received C/A signals, this data may be downloaded to the mobile
from the BTS.
Embodiments of Blocks 1710, 1720 and 1730 in Figure 17 now will be
described in detail. Specific parameter values used herein are presently
believed to be
representative, based on computer simulation results.
Figure 18 is a block diagram illustrating detailed embodiments of Block 1710
of Figure 17. As shown in Figure 18, the C/A signals are received and
amplified by
the mobile terminal receiver's RF section, Block 1750, then band-pass filtered
at
Block 1810 to increase or maximize received signal-to-noise ratio. After
filtering, the
signals plus received noise may be down-converted to the complex baseband at
Block
1820, for example, using quadrature mixer components that separately down-
convert
the in-phase (I) and quadrature phase (Q) components. The baseband I and Q-
outputs
can be mathematically equated to a complex signal, for example, the complex
envelope of the received signal (hence the term "complex baseband"), where the
I-
component is the real part, and the Q-component is the imaginary part. It
should be
noted that, in some embodiments, from this point onward, the signal flow
actually
may be comprised of parallel I- and Q-paths. The received C/A signals plus
noise
may be further filtered at Block 1825 and time-sampled at Block 1830, for
example at
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a rate of 1.024 MHz, and then quantized at Block 1840, converting them from
analog
to discrete-time digital samples. Finally, a "snapshot" of received time
samples
"x(n)" is collected and stored in memory at Block 1850. For the in-building
environment, analysis has shown that about 1-million time samples may be
needed to
reliably recover the weak C/A signals from the received noise. For this
illustration, a
total of 1.024 million complex time samples, representing exactly 1-second of
observation time, are collected and stored in memory at Block 1850.
Figure 19 is a flowchart illustrating operations of Block 1720 of Figure 17
according to some embodiments of the present invention. As shown in Figure 19,
in
these embodiments, residual Doppler shift estimation includes two operations.
At
Block 1910, bandpass filtering of the received input samples x(n) into narrow
frequency bands called "slices" is performed, and then the filtered, frequency-
sliced
samples are despread using an internally generated reference code matched to
the
received C/A code. Then, at Block 1920, the residual Doppler shifts are
estimated
from the slices that are despread.
Figure 20 illustrates bandpass filtering GPS C/A signals into frequency
slices,
as was generally described in Block 1910 of Figure 19. More particularly, the
one
second snapshot x(n) of received signal plus noise samples from Block 1710 is
first
filtered by a bank of K bandpass filters, as illustrated in Figure 20. In some
embodiments, each filter has a bandwidth of, for example, approximately 32
kHz.
Each filter passes a frequency band or "slice" of the received signal plus
noise. The
total number of slices K to be processed may depend on the signal-to-noise
ratio of
the input samples. For example, for an outdoor case with direct line of sight
to the
SVs, simulation results indicate that a single frequency slice centered at 0
Hz (in the
complex baseband) may be sufficient. For the indoor case where significant
attenuation of the SV signals may exist, up to, for example, seven frequency
slices
(K=7) may be used to obtain an accurate estimate of the Doppler frequency. The
filtering of input sequence x(n) into frequency slices may be performed once
for all
C/A signals to be processed.
Figure 21 is a block diagram of operations that may be performed in filtering
and despreading the input samples (Block 1910 of Figure 19). These operations
may
be performed in parallel for each frequency slice. At Blocks 2010, the 1.024
million
complex received samples x(n) from Block 1850 are first frequency-translated
by
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multiplication with complex sinusoid Cimkn, (1 < k < K), where cok is the
center
frequency of the "kth" frequency slice. Fourier Transform theory states that
multiplication of a signal by Cia)kn in the time domain is equivalent to
translating its
spectral components by an amount col, in the frequency domain. The frequency
translated sequences are passed through digital low-pass filters 2020 having a
filter
bandwidth of, for example, 16 kHz. A choice of filter bandwidth is explained
below. This filtering passes that part of the frequency content of the
original x(n)
samples contained within a 16 kHz frequency slice centered at frequency cok.
The
filtered output is then down-sampled at Blocks 2030 using decimation-by-32,
resulting in a sequence 2040 containing, for example, 32000 complex elements
for
each frequency slice. Down-sampling can greatly reduce the computations for
the
downstream processes, in some embodiments of the invention.
Still continuing with the description of Figure 21, for each C/A signal, the
receiver generates an internal reference code sequence 2050 set to the same
Gold code
used by the received C/A signal. The reference code phase is synchronized to
the
code phase of the C/A signal as measured at the local BTS. A very small timing
adjustment (a few parts per million or less), due to the SV-induced Doppler
shift, may
be applied to the reference Gold Code sample period at Block 2050 to maintain
alignment with the received C/A code over the entire 1-second observation
period.
For each slice, the internal reference code sequence 2050 is frequency-
translated by multiplication at Blocks 2060, with complex sinusoid ekn. Note
that
the exponent +jokn has the opposite sign from the translation e--*n (Block
2010)
applied to the input samples. This may be used so that the translation
sinusoids e-ju)kn
and e+ja)kn cancel each other when multiplied together later at Blocks 2090.
The
frequency-translated reference code samples are next low-pass filtered at
Blocks 2070
and down-sampled at Block 2080 using, for example, decimation by 32. This
produces a frequency "slice" of the reference code corresponding to that of
the input
samples, but in the opposite frequency direction. Finally, the filtered
received
sequence 2040 for each frequency slice is despread by multiplication at Block
2090
with its corresponding filtered reference code to produce a despread output
sequence
Yk(n) (1 < k < K) containing, for example, 32000 complex elements, as shown in
Fig.
21.
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It can be shown mathematically that the despread slice output sequence yk(n)
in Figure 21 is given by:
yk(n) = C d(n) e j[copn +0 +AOk]
(+ higher frequency terms), [1 k K] (3)
where:
d(n) = 50 bps data modulation overlay ( 1),
con = received signal Doppler frequency,
= a constant,
0 = a constant phase angle, and
AOk = a variable phase angle component, proportional to the
product of the
slice center frequency cok and the time offset between the received C/A
code and internal reference code phases.
The "higher frequency terms" in Equation (3) are filtered out in subsequent
processing, and therefore may not be relevant to the solution.
Accordingly, Figure 21 illustrates embodiments of the present invention where
the processor of the mobile terminal is configured to bandpass filter the GPS
C/A
signals into frequency slices by frequency translating the GPS C/A signals,
low pass
filtering the GPS C/A signals that are frequency translated and downsampling
the low
pass filtered, frequency translated GPS C/A signals. Figure 21 also
illustrates
embodiments of the present invention wherein the processor of the mobile
terminal is
further configured to despread the frequency slices by generating an internal
reference
code sequence, frequency translating the internal reference code sequence for
each
frequency slice, low pass filtering the frequency translated reference code
sequences,
and multiplying by the downsampled low pass filtered, frequency translated GPS
C/A
signals.
A detailed description of the use of filtering prior to despreading, as was
generally described in Figure 21, now will be provided. Specifically, the
internally
generated reference code 2050 in Figure 21 is synchronized to the received C/A
code
in order to despread the received signal 1850. However, the precise code phase
of the
received C/A signal is still unknown at this point, and may be several chips
offset
from the internal reference code phase (which is synchronized to the code
phase
measured at the BTS). As explained below, both the input x(n) and reference
code
may be filtered into narrow frequency "slices" to allow sufficient correlation
between
the two to enable despreading to occur.
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The effect of the filtering is shown in Figure 22. In this figure, the
autocorrelation of a 1023-chip Gold sequence with chip rate of 1.023 Mcps is
plotted
for several values of low-pass filtering. For a filter bandwidth of 511 kHz,
the
autocorrelation peak falls off to near-zero at an offset of one chip position,
indicating
that two identical Gold sequences would be almost completely decorrelated with
just
one chip offset between them. When the filter bandwidth is reduced to 256 kHz,
decorrelation occurs at an offset of about 2 chip periods. When the filter
bandwidth is
further reduced to 32 kHz, the auto-correlation peak becomes very broad,
remaining
relatively flat over an offset range of 15 chip periods.
Recall that the reference code phase is set equal to the C/A code phase
measured at the BTS. It was previously shown that the code phase measured at
any
location within a cell of 1 km radius may lie within about 14 chip periods of
the code
phase measured at the BTS. Therefore, Figure 22 shows that the C/A code
received at
the mobile terminal, when filtered to a 132 kHz bandwidth or narrower, can
maintain
a high degree of correlation to the reference code (also filtered to the same
bandwidth)
for code phases that can be received within the coverage area of the cell.
A potential benefit of this filtering technique is that it can provide
sufficient
correlation between the received and reference codes to perform despreading,
without
having to do an exhaustive search to precisely align the two sequences. This
can
significantly reduce the computational load for operations of Figure 21.
Figure 23 is a block diagram of operations that may be performed to estimate
residual Doppler shifts from the slices that are despread, which may
correspond to
Block 1920 of Figure 19. These operations may be repeated for each received
C/A
signal. Referring now to Figure 23, the despread yk(n) 2090 are first
multiplied by the
50 bps ephemeris data at Blocks 2310 to strip off the 50 bps modulation. The
50 bps
data bit sequence may be supplied by the BTS to the mobile terminal. The
samples
are then multiplied at Blocks 2320 by the complex sinusoid where cos, is
the
received Doppler frequency component due to SV motion measured at the BTS. The
multiplication at Block 2320 removes the SV-induced Doppler shift by frequency-
translating yk(n) by the negative of the SV Doppler frequency component. After
this
frequency translation, only the residual Doppler shift due to user motion may
remain.
Still referring to Figure 23, the samples are next low-pass filtered at Blocks
2330 to a bandwidth of, for example, 1 kHz to remove the higher frequency
terms in
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Equation (3), and down-sampled at Blocks 2335 by a factor of, for example, 64.
The
resulting sequences zk(n) Blocks 2340 contain, for example, 500 samples for
each
frequency slice. The zk(n) are "zero-padded", for example, to 2048 samples,
and then
Fast Fourier Transformed (FFT) at Blocks 2345 to the frequency domain. The
resulting complex frequency elements Zk(m) are converted to magnitude (or, in
some
embodiments, magnitude-squared) values at Blocks 2350, and then added together
across all K-frequency slices on a point-by-point basis at Block 2355.
The frequency peaks corresponding to the residual Doppler frequency add
coherently at Block 2355 since they occur at the same place for all slices.
However,
random peaks caused by channel noise add non-coherently because the noise
contained within the slices are uncorrelated (since the slices are assumed to
not
substantially overlap in frequency). Thus, the summation process across
multiple
frequency slices at Block 2355 can enhance the ratio of true Doppler peak to
the false
peaks caused by the channel noise. The frequency peak and several adjacent
points
on either side are selected at Block 2360, and then interpolated at Block 2365
using,
for example, a parabolic best-fit to estimate the exact value of the residual
Doppler
shift ow due to user motion 2370. This value 2370, when added to the SV-
Doppler
component cosy downloaded to the mobile terminal from the BTS, provides an
estimate of total Doppler shift an for the received C/A signal, which can be
accurate
to within a fraction of a Hertz.
Accordingly, Figure 23 illustrates embodiments of the present invention
wherein the processor is configured to estimate the Doppler shifts from the
frequency
slices that are despread by frequency-translating the frequency slices that
are despread
by the Doppler shift frequency that is measured at the wireless network to
obtain the
residual Doppler shift due to mobile terminal motion, transforming the
despread
frequency-translated slices to the frequency domain (for example by fast-
Fourier
transform), converting complex frequency domain values to magnitude (and/or
magnitude-squared) values, and adding the magnitude values on a point-by-point
basis across the frequency slices.
Figure 24 is a block diagram of operations that may be performed to estimate
the received code phase, which may correspond to Block 1730 of Figure 17.
Referring to Figure 24, the received samples x(n) from Figure 18 are first
multiplied
at Block 2410 by the 50 bps ephemeris data sequence to strip off the data
modulation.
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The total Doppler shift is then removed by multiplying the received samples by
the
complex sinusoid ei'Dn at Block 2420, where the frequency cop is the total
Doppler
shift comprised of the residual Doppler estimated in Figure 23, plus the SV-
Doppler
component downloaded from the BTS. A very small effect of the Doppler shift on
the
received chip duration may also be compensated by first up-sampling the
sequence by
a factor of, for example, 8 at 2430, and then adjusting the time scale of the
samples
periodically in 1/8-chip steps by incrementing or decrementing the tap delay
output of
the shift register 2440. The resulting true baseband samples 2450 (having all
Doppler
shift removed in some embodiments) contain two components: a signal term
comprised of identical repeating Gold Code frames at a frame period of 1-
msec., and
a noise component. Successive 1-msec. segments are then summed at Block 2460
point-by-point to produce a single frame of, for example, 8192 sample points,
thereby
compressing, for example, the entire 1-second snapshot into a single
equivalent 1-
msec. frame. The signal components from each of the 1000 original segments
2450
combine coherently, while the uncorrelated noise components combine non-
coherently. This can produce a factor of 1000 (30 dB) improvement in the
signal-to-
noise ratio of the equivalent frame 2460.
Still referring to Figure 24, the 1-msec. frame 2460 is then correlated with
an
internally generated reference code frame identical the received C/A Gold Code
at
Block 2470. The correlation at Block 2470 can be based on a Fast Fourier
Transform
(FFT) in some embodiments. This FFT can use the correspondence between
multiplication in the frequency domain and convolution in the time domain to
produce
a very computationally efficient correlation in some embodiments. The complex
output values from the correlation Block 2470 are converted to magnitude-
squared
values at 2480, and the location of the peak value corresponds to the code
phase of the
received C/A signal. The peak value 2490 together with several adjacent points
on
either side are interpolated at 2495, for example by generating a least-
squares-fit to a
parabolic curve in the same manner as was done to interpolate the Doppler
frequency
at Block 2365. The code phase value corresponding to the vertex of the
parabola is
chosen as the final interpolated code phase estimate for the received C/A
signal.
Figure 25 graphically illustrates an embodiment of this interpolation.
Accordingly, Figure 24 illustrates embodiments of the present invention
wherein the processor of the mobile terminal is configured to estimate the
code phases
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of the GPS C/A signals by removing a total Doppler shift by frequency-
translating the
GPS C/A signals by the sum of the residual Doppler shifts that are estimated
plus the
Doppler shift that is measured at the wireless network, summing segments of
the GPS
C/A signals from which the total Doppler shift has been removed, correlating
the
summed segments with an internally generated code frame, and determining the
time
offset corresponding to the peak magnitude squared value.
Operations of Figures 21, 23, and 24 may be repeated for each C/A signal as
was shown in Figure 17, and the resulting estimates of code phase and residual
Doppler shift for each SV determined from Figures 23 and 24, respectively, are
transmitted from the mobile terminal to the NOC (Block 1740). At the NOC, they
are
combined with SV ephemeris and cell location information to generate a final
position
fix. The processing performed at the NOC may be similar to that used in
conventional assisted-GPS methods, such as described in U.S. Patents 5,663,734
and/or 6,169,514, and need not be described further herein. The final position
fix may
then be downloaded to the mobile terminal from the NOC and displayed to the
mobile
user.
In some embodiments of the invention, the complex frequency translation
sinusoids eic'xn shown in Figures 21, 23, and 24, where cox generically
represents the
specific translation frequencies given in the Figures, may in some cases be
replaced
by complex sinusoidal approximations such as a square wave or other suitable
periodic complex waveforms having frequency cox.
In some embodiments of the invention, the combination of removing the total
Doppler shift (determined in Figure 23) and stripping off the 50 bps data
modulation
from the received input samples can enable the entire 1-second snapshot
collected in
Figure 18 to be compressed down to an equivalent single 1-msec. code frame of
8192
points (Block 2460). This can greatly reduce the number of computations to
implement the FFT-based correlation (Block 2470) in Figure 24 to determine the
code-phase of the received C/A signals, according to some embodiments of the
invention.
Other embodiments for determining the code phase of the received C/A
signals now will be described. These embodiments can replace and/or be used
with
the embodiments described in Figure 24.
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In particular, referring to the Doppler estimation in Figure 21, it was shown
in
Equation (3) that the despread outputs yk(n) for each of the K frequency
slices contain
a variable phase angle AOk associated with the Doppler frequency term. It can
be
shown analytically that AOk is related, to a very close approximation, to the
difference
between the received C/A signal code phase and the internal reference code
phase as
follows:
AOk = ¨tokAt (4)
where:
ok = center frequency of "kth" slice (radians/sec.), and
AT = received C/A signal code phase minus internal reference code phase
(sec.).
Denote AT as the "residual code phase", because it represents the time
difference
between the code phase measured at the mobile terminal and the code phase
measured
at the local BTS (to which the mobile terminal's internal reference code is
synchronized).
Equation (3) shows that the Doppler frequency term in yk(n) also contains a
constant phase angle 0 that is the same for all frequency slices. Let Ototk be
the total
phase angle of the Doppler frequency term in yk(n). Then from Equations (3)
and (4):
Ototk = 0 + AOk = 0 ¨ okAt. (5)
The difference in total phase angle between the kth and jth frequency slices
is given by:
Ototk ¨ Ototj = wiAt cokAt = (coj cok)At. (6)
Equation (6) can be solved explicitly for the residual code phase AT as
follows:
AT = ¨ (Ototk ¨ OtOti) /(COk CO) (7)
From Equation (7), it can be seen that if Ototk is plotted versus slice center
frequency
cok for all K frequency slices, the resulting points will align in a straight
line whose
slope is equal to the negative of the residual code phase At. In this way, AT
can be
determined as an extension of the Doppler shift estimation in Figure 23 with
very
little additional computational effort, in some embodiments of the invention.
These
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embodiments can represent a significant reduction in computations compared to
the
code phase estimation process described in Figure 24.
Figure 26 is a block diagram of these alternate embodiments to estimate the
received code phase, which may also correspond to Block 1730 of Figure 17.
Referring to Figure 26, the phase 4(m) of the complex FFT output elements
Zk(m)
2610 from Figure 23 are calculated at Blocks 2615 using the relation:
4k(111) = Arctan[Im{Zk(m)} / Re {Zk(m)} I, (8)
where Im{Zk(m)} and Re {4(m)} are imaginary and real parts of Zk(m),
respectively.
The resulting phase values 41k(m) 2620 are interpolated at Blocks 2630 to
determine
the phase angle 4)k(o)D) corresponding to the precise Doppler shift frequency
cop
determined in Figures 23 and 24. The Ok(coD) values as a function of slice
center
frequency cok are least-squares fit to a straight line at Block 2640. Since
the range of
(I)k(0D) values may vary by more than 2rc radians, this operation may involve
"unwrapping" the (1)k(coD) terms by adding or subtracting multiples of 27t so
that the
results fall in an unbroken straight line over all slice frequencies wk. The
residual code
phase At is then determined as the negative slope of the linear best-fit to
the (1)1(0)0
vs. wk data points. Finally, the total code phase estimate for the given C/A
signal is
equal to the residual code phase At plus the internal reference code phase
(i.e., the
code phase measured at the local BTS).
Accordingly, Figure 26 illustrates other embodiments of the present invention,
wherein the processor of the mobile terminal is configured to estimate the
code phases
of the GPS C/A signals by calculating phases of the frequency slice values
that have
been converted to the frequency domain, determining phase angles corresponding
to
the estimated Doppler shift frequency for each of the slices, and determining
a
residual code phase from the phase angles.
Embodiments of the invention that were described above assume that the
received C/A signal code phase at the mobile terminal is within a relatively
small
number of chips offset from the code phase measured at the local BTS. This may
constrain the maximum distance between the mobile and BTS to the order of a
few
miles. This potential constraint may not pose a problem in a terrestrial
cellular
network (conventional and/or ATC-based) where the cell radius may typically be
on
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the order of or less than one mile. -However, in a satellite network, a
satellite beam
coverage may extend over hundreds of miles, which may far exceed the assumed
maximum distance. Consequently, it may be desirable for the satellite network
to be
able to estimate the location of the mobile terminal to within, for example, a
few
miles uncertainty inside its beam, and then transmit to the mobile the
pertinent C/A
code phase and Doppler information referenced to the mobile's estimated
location.
Using this information, the mobile terminal can then proceed to measure the
precise
C/A code phase and Doppler shift as described above. Two embodiments for
determining the mobile terminal's approximate location within a satellite beam
will
now be described.
In general, in first embodiments, the mobile terminal's approximate location
within a satellite beam is determined by measuring relative differences in
received
signal levels between adjacent satellite beams at the mobile terminal. In
second
embodiments, the mobile terminal's approximate location within a satellite
beam is
determined by measuring path delays between the mobile terminal and a
satellite
gateway via at least two satellites.
More specifically, the first embodiments compare signal levels from adjacent
beams and may be suitable when satellite coverage includes multiple spot beams
with
coverage overlap between adjacent beams. In these embodiments, the signal
levels
received in the mobile terminal's assigned spot beam as well as the
immediately
adjacent beams are compared, and an estimate of the mobile terminal's location
is
determined by analyzing the relative differences in received signal levels
between the
beams.
Figure 27 graphically illustrates an example of a typical spot beam antenna
pattern. The pattern generally is short-term invariant and is assumed to be
known to
the satellite network.
Consider a mobile terminal located within the spot beam coverage map of
Figure 28. The mobile terminal transmits a signal to the satellite, which is
simultaneously received at the satellite gateway over the mobile terminal's
assigned
beam and, for example, six adjacent beams. The received signal levels vary
between
the beams, due to the different spot beam gains that occur at distances dl
through d7
from the mobile terminal to each beam center. The NOC measures these signal
levels
and makes an initial guess as to the radial distance dl from the mobile
terminal to the
center of its assigned beam. Then, the spot beam gain corresponding to radial
CA 02512271 2005-06-30
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distance dl is calculated using the known spot beam pattern data, such as that
shown
in Figure 27. The differences in measured signal levels between the mobile
terminal's
assigned beam and adjacent beams are then used to produce spot beam gain
estimates
for beams 2 through 7, which in turn produce estimates of radial distances d2
through
d7 for the adjacent beams.
As shown in Figure 29, circles of radius dl through d7 centered on their
respective spot beams are projected onto the coverage map. These circles
represent
contour lines of the spot beam gains that were estimated from the signal
measurements. If the contour circles do not intersect at at least one common
point, a
new radial distance dl is chosen and the above operations may be repeated
iteratively.
When all the contour circles intersect, the area of intersection represents
the mobile
terminal's estimated location. This case is illustrated in Figure 29.
As an alternate embodiment to the above, the mobile terminal may measure
multiple signals received from the satellite gateway over each of the beams in
Figure
28. For example, a satellite network typically broadcasts a common control
channel
in each beam. The mobile terminal can make near-simultaneous measurements of
the
received control channel level in each beam, and transmit this information
over the
satellite link to the NOC. The NOC then can use the same operations described
above
to estimate the mobile terminal's location by finding an intersection region
of the
estimated spot beam gain contour circles as illustrated in Figure 29.
As noted above, second embodiments can use multi-satellite diversity. These
embodiments may be more suitable when two or more satellites are used to
provide
transmission path diversity for the network. The path distances between the
satellites
and the gateway station may be assumed to be known precisely by the network.
The
satellite gateway broadcasts a periodic time mark to all mobile terminals in
the
network over one of the satellites. The mobile terminal receives the time
mark, and
after a fixed wait time known to the network, transmits a response signal back
to the
gateway. The response signal is simultaneously received at the gateway over
the
plurality of diversity satellites. The total elapsed time from transmission of
the time
mark to reception of the mobile terminal response signal over each of the
satellite
paths is measured. The fixed wait time at the mobile terminal, as well as the
satellite's
internal transmission delay and any other fixed delays are subtracted out. The
remaining times represent the round-trip path delays from gateway to mobile
terminal
and back to gateway for each satellite.
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By knowing the speed of light and the precise distances from the gateway
station to the satellites, the path distance "Pn" between satellite "n" and
the mobile
terminal can be estimated for each satellite. The set of equidistant points at
a distance
"Pn" from satellite "n" forms a spherical surface of radius Pn centered at
satellite "n".
The intersection of this spherical surface with the surface of the earth forms
an arc on
the earth's surface. The mobile terminal is constrained to lay somewhere on or
very
close (accounting for altitude uncertainties) to this arc. The arcs produced
by the
plurality of satellites are offset from each other due to their different
satellite orbital
locations, and therefore intersect each other at a finite number of points.
Figure 30 illustrates the ranging position arcs produced by two diversity
satellites, superimposed on the coverage area of the mobile terminal's
assigned beam.
The intersection of the two arcs is chosen as the estimate of the mobile
terminal's
location.
After the mobile terminal's location within its satellite beam has been
estimated using one of the two embodiments described above and/or other
embodiments, the SV code phase, Doppler shift, and 50 bps ephemeris data
sequence
that would be observed from that location are transmitted to the mobile
terminal.
However, unlike the terrestrial cellular network, there may be no local BTS to
measure these parameters directly. Instead, the satellite network may have
complete
knowledge of the positions, velocities, and directions of all SVs in view at
the time of
the mobile's measurement, as well as their code phases and 50 bps data
sequences.
With this information, the NOC can calculate the code phase, Doppler shift,
and 50
bps data bit transition times projected to any given observation point on the
earth,
replacing the direct measurement performed by the terrestrial BTS. The mobile
terminal uses this information supplied by the satellite network to perform
its SV code
phase and Doppler shift measurements, as was previously described for the
terrestrial
case.
As will be appreciated by one of skill in the art, the present invention may
be
embodied as a method and/or an apparatus. The present invention may take the
form
of an entirely hardware embodiment, an entirely software embodiment or an
embodiment combining software and hardware aspects. Furthermore, the present
invention may be embodied as a computer program product on a digital storage
medium having computer-readable instructions embodied in the medium. Any
suitable digital storage medium may be utilized, including a memory device,
hard
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disk, CD-ROM, optical storage device, transmission medium, such as a wireless
transmission medium and/or those supporting the Internet or an intranet,
and/or a
magnetic storage device.
The present invention was described with reference to block diagrams and/or
flowchart illustrations of methods, apparatus (systems), mobile terminals,
and/or
computer program products according to embodiments of the invention. It is
understood that a block of the block diagrams and/or flowchart illustrations,
and
combinations of blocks in the block diagrams and/or flowchart illustrations,
can be
implemented by computer program instructions. These computer program
instructions may be provided to a processor of a general purpose computer,
special
purpose computer, a Digital Signal Processor (DSP) and/or other programmable
data
processing apparatus to produce a machine, such that the instructions, which
execute
via the processor of the computer and/or other programmable data processing
apparatus, create means for implementing the functions/acts specified in the
block
diagrams and/or flowchart block or blocks.
These computer program instructions may also be stored in a computer-
readable memory that can direct a computer, DSP, or other programmable data
processing apparatus to function in a particular manner, such that the
instructions
stored in the computer-readable memory produce an article of manufacture
including
instructions which implement the function/act specified in the block diagrams
and/or
flowchart block or blocks.
The computer program instructions may also be loaded onto a computer, DSP,
or other programmable data processing apparatus to cause a series of
operational steps
to be performed on the computer or other programmable apparatus to produce a
computer-implemented process such that the instructions which execute on the
computer or other programmable apparatus provide steps for implementing the
functions/acts specified in the block diagrams and/or flowchart block or
blocks.
It should also be noted that in some alternate implementations, the
functions/acts noted in the blocks may occur out of the order noted in the
flowcharts.
For example, two blocks shown in succession may in fact be executed
substantially
concurrently or the blocks may sometimes be executed in the reverse order,
depending upon the functionality/acts involved.
In the drawings and specification, there have been disclosed typical preferred
embodiments of the invention and, although specific terms are employed, they
are
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used in a generic and descriptive sense only and not for purposes of
limitation, the
scope of the invention being set forth in the following claims.
44
