Note: Descriptions are shown in the official language in which they were submitted.
CA 02527157 1993-O1-07
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MULTICHANNEL OUADRATURE MODULATION COMPATIBLE
WITH STANDARD TELEVISION CHANNEL PLANS
This application is a divisional of Canadian patent application Serial
No. 2,127,686 filed internationally on January 7, 1993, and entered nationally
on
July 8, 1994.
FIELD OF THE I1WENTION
?he present invention relates generally to television transmission and
receiving systems
and more particularly to CATV or over-the-air multichannel systems wherein a
plurality
of standard television signals are scrambled by suppression of normal
synchronizing
pulses or otherwise modified during the active video time and subseguently
selectively
descrambled at authorized subscriber locations in a manner which
simultaneously renders
a plurality of authorized channels to be provided unscrambled to the
subscriber's
television set.
BACKGROUND OF TIRE ll~lVEN1'ION
In subscription television, or pay television, programming signals of premium
channels are typically transmitted in a scrambled form so that unauthorized
viewers, chat
is, viewers who do not pay to receive the specific programming, are not able
to view the
transmitted signal on their television receivers. Authorized viewers are
provided with
means to descramble or decode the scrambled television signal and usually such
descrambling means are capable of being selectively enabled to descramble
specific
channels. This enablement can be done remotely by the service provider or the
CAT'V
. operator, by selective addressing of individual dexrambling nurans
associated with
subscribers who paid to receive the scrambled programming.
A technique commonly used for television scrambling is that of sync-
suppression
wherein the RF level of the horizontal and vertical synchronizing pulses is
suppressed
to a level below that of the video so that the standard television receiver is
unable to
establish regular synchrony and instead erratically locks on RF peaks in the
active video
CA 02527157 1993-O1-07
thereby creating unviewable picture on the television set. In addition, the<
abilzty of the
television receives to use the color reference burst associated with the
horizontal
synchronizing pulse is severely degraded thereby causing distorted color
reproduction
or disabling the color circuitry of the television set. Exemplary prior art
sync
suppression systems. are disclosed in U:S. Pat. Nos. 3, I84,53T to Court et
al; 3,48,166
to Reiter et al; 3,530,232 to Reiter et al and 4,222,068 to Thompson. In these
systems,
the horizontal synchronizing levels are suppressed or reduced to gray level
and an
additional keying control signal is normally transmitted together with the
scrambled
television signal for controlling reconstruction or regeneration of the proper
lyric levels
at the authorized descrambler, thereby providing unscrambled normal television
signal
at the receiver. For example, in U.S. Pat. No. 3,184,537 the audio subcarrier
is
amplitude modulated with a suitable sync insertion control signal. In U.S.
Pat. No.
4,222,068 horizontal sync reconstruction at the authorized descrambler is
effected by
transmitting normal sync signals during the Vertical Blanking Interval ("VBI")
of the
television signal for enabling a timing circuit at the receiver to lock on the
horizontal
components thereof. The timing circuit is then used to gate the Horizontal
BlanIang
Interval ("HBI") thereby facilitating the restoration of the horizontal
synchronizing pulses
in the composite baseband video signal.
Sync suppression systems normally operate in one of two ways: The first
method, as in U.S: Pat. No: 3,184,53T,-is based on attenuating the modulated
RF
television signal during the HBI (and in some cases during the VBI as well) by
'a known
fixed amount such as 6 Db: At the authorized deserambler, the signal is
deserambled by
attenuating the RF received signal during the active: video portion by the
same k~iown
amount. This method is known as "RF sync suppression" since the scrambling and
descrambling processes occur at RF stages. after the modulation process at the
transmitter
and before the demodulation. p~cess at the receiver respectively: The second
sync
suppression method is known as "Baseband Sync Suppression" wherein a known
gated
offset level is added to the baseband video signal during the HBI; thereby
suppressing
the horizontal synchronizing signal. At the authorized descrambler, our equal
offset level
is subtracted from the demodulated video signal during the HBI, thereby
re$toring the
CA 02527157 1993-O1-07
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horizontal synchronizing signal levels to their normal values. An example of
Baseband
sync suppression system is disclosed in U.S. Pat. No. 4,222;068,
Both prior art RF and baseband sync suppression descrambling systems described
above are typically accomplished by incorporating the timing and descrambling
circuitry
with a television tuner and demodulator, often termed a converter, that can
cover the
entire CATV band or over-the-air channels used by the subscription television
service
provider. In RF sync suppression systems, descrambling is effected by gated
attenuation
at the IF stage of the converter or at its fixed RF output channel frequency
such as TV
channel 3 or 4. In baseband sync suppression systems, descrambling is effected
at the
demodulated video stage of the converter and the resultant normal video signal
is
remodulated by means of a fixed frequency television modulator operating on
channel
3 or 4. Thus, in both examples, the subscriber is equipped with a converter-
descrambler
device capable of tuning and descrambling a single channel at a lime and
converting it
to channel 3 or 4 (output channel), whichever is unused by a local broadcast
station. The
subscriber's television set may permanently remain tuned to the converter-
descrambler
output channel while channel selection is done by means of the converter-
descrambler
tuner.
In a subscription television service, a plurality of scrambled channels may be
transmitted simultaneously on several channels: However, as explained above,
prior art
converter-descramblers can only process one channel at a time. This means that
subscriber households who are authozized to receive several premium scrambled
channels, cannot receive these services descrambled simultaneously. So for
example, if
dad wishes to watch a "pay-per-view" event such as a championship fight, while
the kids
watch the Disney Channel on the second television set, the household would
require two
converter-descramblers. Furthermore, if mom wanted to record a movie on HBO at
the
same time, a third converter-descrambler must be employed in conjunction with
the
Video Cassette Recorder ("VCR"j. This is clearly one of the disadvantages of
prior art
single channel descrambling techniques.
Most television households in America are equipped with "Cable ready"
television sets or VCR's. The term "Cable ready equipment" ~ should be
understood. as ,
consumer television equipment with expanded frequency tuning capability that
allows a
CA 02527157 1993-O1-07
subscriber to tune not only the off air broadcast channel frequencies but also
all.-CATV ~ '
channels. A significant portion of the cable ready equipment sold in America
contains
additional features such as Multichannel Television Sound ("MTS ") allowing
reception
of stereo TV programming or second audio programming ("SAP"); remote control
to
control sound level or sound muting as well as channel tuning. Still another
feature of
some cable ready television sets is the Picture In-Picture ("PIP") display
capability
allowing the subscriber to simultaneously watch one pmgram while observing
additional
video progzam(s) in a picture inset displayed within the main picture. When
this cable
ready equipment is ' connected directly to the cable drop entering the
household, the
subscriber is usually able to receive-and record alI non-scrambled chancels
without the
aid of a converter and thus use his equipment's special features described
above to
receive such non-scrambled signals. The emblem occurs when the CATV system
employs signal scrambling as means of controlling access to the premium
channels.
Normally in these circumstances, being fed by the converter-descrambler's
output
channel, the television set is permanently tuned to channel 3 (or 4) while the
converter-descrambler is tuned to the desired channel. Thus, the remote
control
purchased with the television set is unused, and an additional remote control
unit must
be provided for the: converter-descrambIer. Moreover, many prior art baseband
converter-descramblers cannot pass the composite MTS audio program material
and thus
the stereo signal and the SAP are lost despite the fact the television set may
be capable
of receiving MTS. Recent attempts. to address this problem typically involve a
stereo
bypass operating mode as disclosed in U.S.. Pat. No. 4,630,113 to, Long, which
unfortunately disables the remote volume control of the converter; or
alternatively by
dematrixing and reprocessing the, stereo signal at the converter in order to
effect volume
control with,resultant degradation of stereo separation and compander
performance. Even
prior art RF sync suppression descramblers tend to introduce MTS. audzo
performance
degradations since they pass the audio subcatzier with additional amplitude
modulation
at the horizontal line rate thereby causing AM to FM effects at the television
receiver.
These audio degrading effects are discussed in an article by J. O. Farmer
entitled
"Operational Characteristics of Modern Sat Top Tercninais" published in the
Technical
Papers of the 33rd Annual Convention of the NCTA in Las Vegas, June 3-6, 1384
and
CA 02527157 1993-O1-07
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in the IEEE Transactions on Consumer Electronics, Vol. CE-30; pages 489-502,
August
1984. Finally, the PIP feature of new television sets requires that the TV set
be able to
receive multiple channels simultaneously, a requirement that obviously cannot
be met by
current prior art converter-descramblers. Clearly, all these problems
constitute yet
another set of significant disadvantages of the prior art single channel
descrambling
techniques.
Collectively, the foregoing deficiencies discussed above are known in the
industry
as the "CATV consumer interface" problem. There is a growing body of evidence
that
these consumer interface problems present severe hardships on consumers and
CATV
operators alike. Specific evidence and indication of the public interest in
this regard is
expressed by legislation cited as the "Cable Ready Equipment Act of 1991"
recently
proposed by Senator Patrick Leahy in U.S: Senate Bill S. 2063, published in
the
Congressional Record - Senate, pages S18377 - 518380, November 26, 1991.
Leahy's
bill seeks to encourage solutions to the CATV consumer interface problems
described
above.
It may be easily understood that the most desirable solution to the problems
discussed above is to provide the subscriber with a cable drop which carries
all the
channels to which he/she is entitled in unscrambled form ("in the clear"). The
subscriber can then connect by proper broadband RF signal splitting means
multiple
cable ready television sets and VCR's and freely enjoy all of the consumer
features that
were purchased with that equipment. Since the CATV system may be caxrying
other
channels not purchased by the subscriber, an effective rnean~ is , required to
unit a
subscriber's access only to those channels and special events for which
subscription fees
have been paid.
If one attempts to employ prior art single channel descrambling techniques in
order to provide each subscriber with simultaneous multichannel descrambling,
one
would have to provide each subscriber with a multiplicity of prior art
descrambling
means wherein each descrambler is tuned and dedicated to one scrambled
channel. The
outputs of such descramblers must then be assigned distinct channel
frequencies and must
be combined by means of a diplexer with all non-scrambled channels to form a
broadband multichannel clear signal service entering the subscriber home.
Clearly; the
CA 02527157 1993-O1-07
_~_.
cost per subscriber for such an arrangement could reach the prohibitive value
of the cost
of one descrambler times the number of processed channels.
There thus exists the need for an effective low cost system for simultaneously
descrambling an arbitrary subset of a plurality of scrambled channels, said
arbitrary
channel subset being the channels to which the subscriber has subscribed, and
supply
them together with the basic non-scrambled channels into the home in such a
manner that
all authorized channels are supplied 'in the clear' and all scrambled channels
not
purchased by the subscriber are passed through into the home either in their
original
scrambled form, or preferably with additional security added to further deny
any
unauthorized access by "piratical" devices.
CA 02527157 1993-O1-07
SUMMARY OF THE I1WENTION
The present invention is directed to subscription television system employing
video scrambling on a plurality of channels. A broadband multichannel
simultaneous
processing system is disclosed which performs descrambling and other access
control
functions on a plurality of addressable selected channels in accordance with
the
subscriber authorization to receive each channel.
It is an object of the present invention to provide a simultaneous
multichannel
television descrambling system of the type described, enabling the cable ready
and other
consumer features contained in the subscriber's television sets and VCR's to
be fully
utilized by the subscriber.
It is a further object of the present invention to provide a scrambling and
descrambling system of the type described which introduces virtually na
artifacts oz
distortion in either the audio or video signals on descrambled and non-
scrambled
channels, wherein video quality is not compromised and functions such as MTS
stereo
are not degraded or affected.
It is still another object of the invention to provide a subscription
television access
control system that offers CATV system operators the ability to nugr~te
currently used
prior art systems to a new encryption based multichannel video scrambling
method with
an irriproved head end originated security scheme providing an enhanced
security. Such
system shall allow migration to the new encryption based security system on
any channel
for which all authorized subscribers are served by an installed access control
device of
the present invention.
To: that end and other objectives, a method for processing a broadband signal
comprised. of a plurality of amplitude modulated television signals comprising
the steps
of generating a plurality of radio frequency (RF) signals in phase synchrony
with at least
a portion of said television signals; each of said plurality of of generated
RF signals
having phasor values; combining said broadband signal with said generated RF
signals,
and adjusting said phasor values of each of said plurality of generated RF
signals in
accordance with corresponding phasor values of said television signals so as
to maintain
a substantially constant relationship therebetween.
CA 02527157 1993-O1-07
-~a-
Accordingly, in one aspect, the present invention provides a method of
generating a btoadband signal comprising up to 2N+1 RF signals where N is an
integer value, each of the RF signals having a frequency that is related to a
center
frequency by an integral multiple of a positive or negative increment of a
fundamental frequency, each of the RF signals having an amplitude and phase
characterized by a phasor value, said method comprising the steps of
generating a
first and second baseband signal each comprising a linear combination of up to
N
sinusoids, each of the sinusoids having a frequency that is an integral
multiple of the
fundamental frequency; and modulating in quadratme the first and second
baseband
signals on a third signal having a frequency equal to the center frequency.
In a further aspect, the present invention provides an apparatus for
generating
a broadband signal comprising up to 2N+1 RF signals each having a frequency
separated from a center frequency by an integral multiple of a positive or
negative
inerexnent of a fundamental frequency, said apparatus comprising: memory means
, for storing a sequ~ce of digital samples representing first and second
baseband
signals, each of the baseband signals comprising a linear combination of up to
N
sinusoids, each of the sinusoids having a frequency that is as integral
multiple of the
fundamental frequency; address generator means, coupled to an address port of
said
memory means, for generating addresses accessing the sequence of digital
samples
from said memory means; clock signal generator means, coupled to said address
generator means, for varying the address values generated by said address
generator
means, said clock signal generator means having a frequency equal to an
integral
multiple of the fundamental frequency; digital to analog conversion means,
coupled to said memory means, for converting the accessed digital sample to
the
first and second baseband signals; signal generation means for gea~erating a
local
signal having a frequency equal to the center frequency; and modulator means,
coupled to the first and second_baseband signals and the local signal, for
modulating
the first and second baseband signals on the local signal to produce the
broadband
signal.
CA 02527157 1993-O1-07
In a further aspect, the present invention a method of generating a brnadband
signal comprising up to 2N+1 RF signals where N is an imeger, each of the RF
signals
having a frequency that is related to a center frequency by a frequency
difference that is a
positive or negative integral multiple of a fundamental frequency, each of the
IZF signals
having an amplitude and phase characterized by a phasor value; said method
comprising the
steps of generating a first and second baseband signal each comprising a
linear
combination of up to N sinusoids, each of the sinusoids having a frequency
that is an
integral multiple of the fundamental frequency; and modulating in quadrature
the first and
second baseband signals on a local signal [146] having a frequency equal to
the center
frequency.
In another aspect, the present invention provides an apparatus for generating
a
broadband signal comprising up to 2N+1 RF signals where N is an integer, each
of the RF
signals having a frequency separated from a center frequency by a frequency
difference that
is a positive or negative integral multiple of a fundamental frequency, said
apparatus
comprising: means for sting digital samples representing first and second
baseband
signals, each of the baseband signals comprising a linear combination of up to
N sinusoids,
each of the sinusoids having a frequency that is an integral multiple of the
fundamental .
frequency; means, coupled to said means for generating digital samples, for
converting the
digital samples to the first and second baseband signals; means for generating
a clock signal
coupled to said means fior generating digital samples and bo said mans for
corning;
means for generating a local signal having a frequency equal to the center
frequency; and
means, coupled to the first and second baseband eigaals and the local signal,
for modulating
in quadrature the fnst and second baseband signals on the local signal to
produce the
bTOadband signal.
CA 02527157 1993-O1-07
_g_
BRIEF DESCRIPTION OF DRAVYINGS
To the accomplishment of the above and such further objects as may hereinafter
appear, the present invention relates to a television access control system
substantially
as defined in the appended claims, and as described in the following detailed
S specification as considered in conjunction with the accompanying drawings;
in which:
Figure la is a baseband sync suppression waveform;
Figure 1b is the coherent RF injected signal for descrambling the signal of
Figure
la in accordance with the present invention;
Figure 2a is the spectrum of the television channels entering the multichannel
descrambler of the present invention;
Figure 2b is the spectrum of the infected signals generated by the broadband
generator of the present invention;
Figure 2c is the descrambled broadband spectrum supplied to the subscriber
from
the multichannel descrambler of the present invention;
IS Figure 3 is the first preferred embodiment of the present invention;
Figure 4 is a plot of two baseband quadrature components required for VSB
spectral shaping of the injected signals used for multichannel descrambling in
accordance
with the present invention;
~.
Figure 5 is the computational digital filter used to generate .the quadrature
components of Figure 4;
Figure 6a describes the broadband coverage by channel group time sharing of
the
~j~t~ ~gnals of the present invention;
Figure 6b is the frequency spectrum partitioning by channel block grouping;
Figure fx is the frequency spectrum partitioning by frequency interlaced
grouping;
Figure 7 is the timing diagram of coherent injection groups and their
respective
RAM address schedule;
Figure 8 is a phasor diagram describing the received signals and the injected
signals;.
Figure 9 is an active calibration and analysis timing diagrarrr in the
preferred
embodiment of the present invention;
CA 02527157 1993-O1-07
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Figure 10 is another active calibration and analysis timing diagram including
RAM address space trajectory;
Figure 11 is an analysis and measurement timing diagram describing the phasor
tracking mechanism;
Figure 12 is an overall video timing diagram describing pertinent events and
processes utilized in the present invention;
Figure 13 is an example of a CATV head end configuration desired for the
practice of the present invention;
Figure 14 is a further detail of the comb generator system utilized at the
headend.;
Figure 15a is a .baseband representation of an RF sync suppression waveform;
Figure 15b is the coherent RF inj ected signal for descrambling the signal of
Figure 15a in accordance with the present invention;
Figure 16 is a digital broadband generator for the second preferred embodiment
of the present invention;
Figure 17 is a second preferred embodiment of the present invention directed
towards use in multiple dwelling units;
Figure l8 is a timing and memory diagram for an enhanced security video
scrambling using the injection methods of the present invention.
CA 02527157 1993-O1-07
DETAB.ED DESCRIPTION OF PREFERRED' EMBODI~~NTS
It should be appreciated that since the object of the present invention is to
provide
all channels to the subscribers in a broadband form and that only certain
channels must
be deserambled, and hence modified, the approach taken according to the
present
5. invention is to avoid any single channel filtering or tuning techniques but
rather focus
on techniques that involve broadband signal addition in such a manner as to
modify
(descramble) only selected channels. Froze the outset, it is instructive first
to observe
that both baseband sync suppression and RF sync suppression scrambling
constitute a .
Iinear modifying process in the radio frequency domain. This linear process is
active
IO only during the blanking intervals in which suppression occurs. This means
that for each
of these two scrambling process, there- exists an additive RF signal pulse of
the
appropriate onset time, duration, amplitude envelope, frequency and phase,
such that
when added to the scrambled RF signal, results in an RF television signal with
normal
synchronizing signals; and hence unscrambled.
15 Figure la depicts the baseband video signal during the 13BI in a baseband
sync
suppression system. As can be seen, the normal sync signal 10 is suppressed by
adding
at the scrambler a baseband offset signal of magnitude 20 during the HBI which
results
in suppressed sync signal 12, while the active video signal 14 is unmodified.
Furthermore, as can be seen in Figure Ia, the color reference burst signal I6
is
20 unmodified bit merely superimposed on an offset level of magnitude 20 in
the figure.
'1'o those skilled in the art of television modulation, it is well known that
the sync level
IO (-401RE units} corresponds to peak RF amplitude while video white Ievel 22
(I00
IRE level) corresponds to an RF amplitude that is 12.5% of peak RF amplitude.
Thus,
in this inverted modulation scale of the RF domain, the HBI gated offset level
20 at
25 baseband is equivalent to the subtracfion of an RF pulse shown in Figure lb
frozri an
otherwise non-scrambled RF television signal. The frequency and phase of this
RF pulse
is equal to those of the picture carrier of the television signal Hence, in
order to restore
a baseband sync suppression signal corresponding to 12 to it's unscrambled
mode, one
can add a coherent RF . pulse train with the horizontal synchronizing
repetition rate,
30 coinciding in time with the HBI as shown in Figure lb; witlx the
appropriate amplitude
30 and in phase with the picture carrier so as to obtain an unscrambled
signal, the
CA 02527157 1993-O1-07
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baseband representation of which corresponds to 10. During the active video
time 34,
essentially no RF signal must be present.
Of course such addition at RF, hereinafter termed "Coherent RF Injection",
mush
be effected with appreciable amplitude, phase and timing precision so as to
obtain
essentially an unscrambled signal. For example; a common baseband sync
suppression
system used in the American CATV market is essentially in accordance with U.S.
Pat.
No, 4,222,068 wherein the baseband sync level shift 20 is set to 70 IRE units.
Thus, in
this case; using the standard television modulation scale, it can be shown
that the
inphase coherent RF injection level 30 required to effect descrambling must
have an
amplitude corresponding to 44.75 % of the peak RF level of the non-scrambled
portion
of the signal. Moderate errors in the level of such injected RF pulse may not
affect the
proper synchronizing of the television receiver for satisfactory viewing of
unscrambled
programs, but may cause errors in the black reference level of the. television
set. This
is because most sets establish their black video reference level by sampling
the 0 IRE
level 24 during the color burst 1G period. However, generally accepted video
specifications embodied in the well known RS-170A standard allow for fixed
errors up
to ~ 2 IRE in black level. These error levels are permissible and are
undetected for all
practical proposes.
Similarly, the injection phase of the RF pulse must be within approximately
~6°
of the picture carrier phase. Because the injected RF amplitude is nearly
equal to that
of the suppressed sync amplitude, these injection phase errors would result in
total false
phase modulation of the picture carrier by no more than ~3° during the
HBI. This
requirement limiting the phase error to f 3° stems from two different
considerations: The
first is related to the fact that good broadcast television engineering
practice requires that
Incidental Phase Modulation ("ICPM") shall not exceed 3° so as not to
cause perceptible
intercarrier audio buzz or MCS stereo distortion. The second reason for this
accuracy
requirement stems from the fact that this phase error only exists duzing the
HBI, within
which the color reference burst signal 16 is received, and not during the
active video
time I4. The RF phase shift during the HBI causes an equal baseband phase
shift in the
detected color reference signal, thereby causing color errors in the detected
chrominance
signal during the active video. Subjective tests evaluating the permissible
color reference
CA 02527157 1993-O1-07
-12-
phase errors in NTSC television signals were conducted by Bernard D. Loughlin
of the
Hazeltine Corporation and reported in a chapter entitled "Nonlinear Amplitude
Relations
and Gamma Correction" on pages 63-91 in a book edited by Ted Rzeszewsla
entitled
Color Tetevision published by the IEEE, 1984. Figure 11-27 in this chapter
shows that
phase errors of ~3° are not perceptible by at least 8090 of observers:
Based on the permissible amplitude and phase error magnitudes discussed above,
it can be shown that the required accuracy of coherently injected RF signals
for
descrambling can be characterized by an ideal errorless injection signal
accompanied by
an injection RF phasor error of magnitude not exceeding -20 Db as compared to
the
1Q desired injected RF signal. As will be subsequently appreciated, this
permissible error
level facilitates a low cost embodiment of the present invention for
simultaneous
injection of a plurality of channels.
Reference is now made to Figure Ia in which the onset and termination of sync
suppression for every horizontal. video line at baseband is shown as level
transitions 26
and 27 respectively. These baseband transitions at the scrambler take place
over a time
duration not exceeding two hundred nanoseconds, consistent with a standard
television
video- bandwidth of 4.2 Mhz. Upon descrambling of such signals, it is
preferable to
match these transition times when the sync offset level is removed at the
descrambler in
accordance with the present invention by coherent injection of an RF pulse
shown in
Figure ib. Here, the onset and termination periods of the injected RF pulse
are
designated by 32 and 33 respectively: 'Thus, their durations would preferably
each be no
longer than two hundred nanoseconds. '
If the pulse shape of the injected RF envelope of Figure lb is attempted by
means
of an amplitude pulse modulator with video bandwidth, the fast. rise-times 32
and 33
would generate double sideband spectral. broadening of up to 4.2 Mhz above and
below
the picture carrier frequency. The upper sideband content of this injected
signal will be
contained within the desired, normal television bandwidth as transmitted in
Vestigial
Sideband ("VSB") Modulation: Hovcrever, since the injected signal is assumed
to be
combined with the received signal in a broadband combiner, the lower sideband
of the
injected signal may interfere with a lower adjacent television channel and in
particular
with it's audio subcarrier located only 1.5 Mhz below the picture carrier of
the
CA 02527157 1993-O1-07
_13_
descrambled channel. If, instead, longer transition times are assigned for 32
and 33 so
as to limit the spectral broadening of the amplitude modulated pulse to less
than 1.5
MHz, the descrambled video signal will contain front and back porch transients
with
durations in the microsecond range, which may invade the leading edge of the
horizontal
synchronizing signal 12 or the active video time 14 of the descrambled signal,
thereby
delivering .a degzaded video signal to the subscriber which may cause false
horizontal
sync or' unstable video clamping action by the television set.
These two conflicting requirements in the frequency domain and in the time
domain as discussed above, can -be resolved by fuming to the very method which
allow
fast video transitions in television transmission without lower sideband
spectral
expansion, namely; the use of VSB modulation techniques for the transitions 32
and 33
prior to the RF combining with the scrambled signal. This will allow the
television
receiver to process the upper sideband containing up to 4.2 MHz wide energy
associated
with fast baseband transitions while limiting the lower sideband expansion
below 1.5
MHz and thus preventing the associated interference to the lower adjacent
channel. An
elaboration on the digital embodiments which facilitate such VSB spectral
shaping of the
,injected signal will, be found in subsequent sections below:
The coherent gated RF injection described above can be useful for purposes
other
than descrambling. Provisions are made in the present invention for
implementing signal
2Q denial .techniques for non authorized subscribers on a channel by channel
basis. In
contrast to the descrambling case, in .these signal denial cases, the coherent
gated RF
injection of Figure lb can be effected in opposite phase to that of the
picture carrier of
the television signal, thereby further suppressing or nulling the
synchronizing signals and
optionally, with sufficient injection level, even reversing the resultant RF
phase of the
received television signal during the HBI. This method results in enhanced
security for
unauthorized subscribers since it denies "pirate" decoders the ability to
reconstruct the
sync signals and further causes phase discontinuity in the intercarrier audio
detector of
the television set; thereby introducing disturbing audio buzz and further
audio noise
masking in some television sets. It should be understood that coherent RP
injection for
denial .of the type described above must also be gated during the HBI using
VSB
CA 02527157 1993-O1-07
.I4_
transition modulation so as to prevent interference to the lower adj scent
channel that may
otherwise be clear or authorized for descrambling.
Thus far, processing of the television signals at RF by coherent injection was
discussed in a context of a single channel. Clearly, the main reason to use RF
injection
techniques is to offer simultaneous processing of this novel type for a
plurality of
channels. This means that coherent injection should take place at each
frequency for
which either descrambling or further sync denial is required-
Figure 2a depicts a portion of the bzoadband spectrum of the incoming signal
carried on the CATV system entering the multichannel descrambler of the
present
invention. Each television channel contains a.picture carrier 40 and an audio
subcarrier
42 separated by 4.5 MHz from the picture carrier 40. The channel spacing 48 is
6 MHz.
As can be seen, the spectra of the television signals appears asymmetric about
its gicture.
carrier frequency as it is upper sideband VSB modulated. In Figure 2a one can
observe
two types of channels being transmitted down the CATV distribution. The first
constitutes alh the non scrambled channels 46 for which all subscribers are
authorized
and thus. we assume no processing is required. The second type of channels
carried is
the sync suppressed scrambled channels 44, designated with the letter S. In
this example
the assumption is made that the subscriber has purchased subscriptions for a
portion of
the premium scrambled channels 44, said channel portion designated here by
44':
Consequently, the required RF injection spectrum corresponding to this
subscription
configuration is depicted schematically in Figure 2b: Each signal in Figure 2b
is assume&
to be generated within the multichannel deserambler of the present invention
and
combined in phase lock with its respective transmitted counterpart picture
carrier of the
v sage frequency in Figure 2a. The broadband composite signal of Figure 2b is
coherently injected and thus linearly combined with the broadband incoming
signal of
Figure 2a to form the composite broadband signal depicted. in Figare 2c that
is
subsequently provided to the subscriber for his/her viewing pleasure. The
injected signals
50,52 consists of VSB injection signals with time domain profiles of the type
depicted
in Figure lb. The injected RF signals 50 are injected in phase with respect to
their
corresponding incoW ia~ scrambled Signals 44', thereby effecting simultaneous
descrambling and resulting in clear channels 60 of Figure 2c being provided to
the
CA 02527157 1993-O1-07
-15-
subscriber. In contrast, the injected RF signals S2 are injected out of phase,
thereby
causing sync null or other signal denial effects such as sync phase reversal:
The resultant
television signals 62 are not viewable or otherwise useful to the unauthorized
subscriber.
However, 'because of the broadband characteristics of this combining system,
they are
fed to the subscriber along with all other clear signals 60.
Because the injected signals 50 and 52 all have identical temporal profile as
that
of Figure lb, their simultaneous generation as a group can be made much
simpler if they
are all required at the same time, that is if the HBI of all television
signals in the channel
gmup of Figure 2a coincides in time repeatedly. This relative tinning
coincidence
condition among this gmup of video channels is well known to those skilled in
the art
of video engineering as °frame synchronous" video sources. This
condition can be
accomplished at the CATV headend before each channel is modulated by means of
video
frame synchronizers providing video outputs genlocked to a master video
synchronizing
source.
Tt will become apparent in following discussions regarding preferred
embodiments
of the present invention that another appreciable subscriber injection
generator
simplification can be realized in CATV ~ systems that employ picture carrier
frequency
assignments and control in accordance with an Incrementally Related Carrier
("IRC")
or Harmonically Related Carrier ("HRC") channel plan, at least within that
channel
- 20 portion of the band that is being processed in.accordance. with the
present invention. This
simplification is related to the fact that under such conditions, the
composite RF signal
of Figure 2b can be derived from a periodic signal with fundamental
periodicity equal
to the Incremental frequency 48 separating any two nearest channels within the
group.
This RF synchrony condition can be effected at the CATV headend by phase
locking all
modulators in the channel group to an appropriate IRC or HRC comb signal as
disclosed
for example by the present inventor in. U.S. Patent No. 5,125,100 and in the
cited
references therein, or in U.S. Patent No. 4;099,123 to Finlay et al.
A first preferred embodiment of the present invention is depicted in Figure 3.
The assumption is made that the headend video frame synchronization and TItC
or .3RC
3U carrier phase locking is effected as discussed above. The broadband
incoming signal
containing all transmitted channels arrives on line 100. A portion of the
power of the
CA 02527157 1993-O1-07
-16-
broadband signal is coupled to a broadband amplifier 104 via a power sputter
102. When
the subsarber is authorized, RF switch 106 is closed and the amplified
broadband signal
corresponding to Figure 2a feeds the directional coupler 10&. The injected
signals
corresponding to Figure 2b, each having a temporal profile in accordance with
Figure
lb, are combined through Iine I14 with the amplified broadband signal by
directional
coupler 108, resulting in the composite signal of Figure 2c on line 109. The
broadband
signal on line 109 is fed to the subscriber port lI2 through directional tap
110. The
injected signals 50 and 52 of Figure 2b forming together the. composite signal
injected
at line I14 are generated by a digital broadband generator I36 feeding RF
switch 116.
If During the time in which injection is required, RF switch ll6 is at
position 117 as
shown in the figure.
The Digital broadband generator 136 is capable of generating signals 50 and 52
situated on arbitrary subset of frequency points on a 6 MHz grid centered
about a center
local oscillator frequency fed on line 146. These signals are generated by
means of
quadrature modulator 138 being fed with appropriate baseband signals on Iines
I61 and
162 and having a one sided bandwidth not exceeding the cutoff frequency of low
pass
filters 164 and 166. The signals thus generated are amplified by broadband
amplifier 124
and fed through the appropriate SAW band pass filter in filter bank 17f., to
broadband
amplifier 120 and to RF switch 1I6. The baseband signals feeding the two
quadratuze
2~ inputs 161 and 162 are functions of time designated hereinafter aS BI(~)
and BZ(t) and are
digitally generated by Digital to Analog Converters ("DAC") I68 and I70
respectively,
based on the contents of their respective RAM's 171 and 172. Low pass filters
164,166
provide antialiasing filtering as well as out of band harmonic rejection. The
clock
frequency driving DACs I68,170 and RAMS 17I,17~ is fed at line I48: This clock
signal also advances the RAM address generator 150, causing the appropriate
RAM data
contents to be loaded sequentially into DACs 168 and I70 respectively. The
clock
frequency is set so that it is a sufficiently large integral multiple: of the
fundamental
frequency increment 48. between two adjacent channels. For a 6 MHz channel
spacing
one might select the clock frequency to be 72 MHz, thereby allowing digital
generation
of baseband signals approaching the Iqyquist rate of 36 MHa. With practical
low pass
CA 02527157 1993-O1-07
_1~_
filter designs, one can easily generate injection signals for a 9 channel
group as shown
in Figure 2.
Referring again to Figure lb, it should be noted that during the time interval
30,
the amplitude and phase of the injected signals are constant, thereby
requiring a CW
sinusoidal signal to be injected at each carrier frequency for which additive
processing
is required. The assumption is made that the local oscillator feeding the
quadrature
modulator 138 on line 146 is phase locked to a channel group center carrier
arriving on
line 140 by an Injection Locked Oscillator ("ILO") or Phase Locked Loop
("PLL")
oscillator contained in subsystem 142. A further assumption is made that the
clock signal
on line I48 (72 MHz in this example) is similarly phase locked to a pilot
signal arriving
on line 1411 which is an integral multiple of the fundamental incremental
frequency 48
(6 Iv~z in this example). Due to the linearity of the quadrature modulation
process in
138, the required signals BI(t) and Bi(t) must be each a linear combination of
sinusoids
with frequencies that are integral multiples of 6 MHz. Hence, during that
period of time,
BI (tJ and BZ(t) are periodic waveforms with a 6 MHz periodicity, thereby
allowing a
discrete time representation of each signal with M samples (wherein in the
present
example M = 72 MHz/6 MHz = 12 ). Therefore, the RAM address generator 150 need
only repetitively scan through 12 RAM addresses during period 30 in order to
generate
an .arbitrary CW frequency grid at 134. The specific amplitude and phase of
the
generated injection carriers will depend on the values ~of the 12 scanned
samples in the
B1 RAM and the 12 scanned samples in the B2 RAM.
At the RF domain,' each generated carrier output on line I34 can be resolved
into
two quadrature components 1 and Q. The index k shall be used here to designate
the
baseband order of the 6 MHz harmonic corresponding to the separation of the
specific
carrier in question from the center frequency carrier on which the group ' l~
oscillator
is locked. Thus, for any given k ,(up to a value of I~, there are two sideband
components
that can be generated, corresponding to two distinct: channels. Given their
desired
quadrature components lk+,Qk+,Ik and Qk represented by a 4 entry column
vector, the
bas~band signals BI(tJ and Ba(t) (represented by a 2 entry column vector for
each sample
time t) can. be shown to be givemby the following vector equation:
CA 02527157 1993-O1-07
-18-
BOt)
B2(t)
0 1 0 I Ix
cos(2~k~) sin(2uk~) 0 0 I O _I 0 Q,~
k~N
r~° _ 0 p ~s(2~rk~ sin(2~rk~) I 0 I 0 1~
0 -I 0 I
(I)
~~ ~ ~itraxy RF phase reference coordinate system in Equation (1),1,x+ and
Qk~'
correspond respectively to the in phase and quadrature components of the upper
sideband
RF injection signals of frequency index k, while Ik and Qk correspond to those
of the
lower sideband of frequency index k. The parameter t is. the discrete time
variable
assuming integer values of I thzough M. In the case k=Oy both sidebands
degenerate into
two halves of a single center carrier, for which Equation (1) reduces to an
equation for
2 entry vectors and a 2x2 matrix.
By pmper calculations essentially in accordance with Equation (1?, based on
the
required generated values of Ik+,Q,E+,Ik and Qk, the Digital Signal Processor
and Micro
Controller ("DSPIuC") I52 evaluates the required samples B~(tJ and BZ(tJ, and
stores
them in RAMS 17I,1~2 via data-bus I54. As indicated above, the injected
signals
corresponding to the column vector in the extreme right hand side of Equation
(1) must
be fed in proper phase and amplitude so that when observed on line 109 they
match the
phasor values of the incoming signals also observed on the same line I09. To
this end,
IS the present ~~nvention provides for phasor measurement and analysis means
that
preferably may be comprised of essentially the same apparatus means that
effect the
coherent injection, namely, digital broadband generator I36, whereby these two
functions are time shared is accordance with a predetermined video timing
schedule: Tt
is possible to time share and perform measurements dW ing time periods in
which no
descrambling or RF injection is rerluired. These measurement or phasor
analysis periods
would preferably be within the VBI, during time intervals in which no active
video
CA 02527157 1993-O1-07
-19-
information is present, therefore providing a fixed received reference for the
picture
carrier level and phase.
Turning now to these phasor analysis time periods for which no injection is
required, an RF switch 116 is set in position lI8 and RF switch 126 is turned
to
position 128. In this way, a sample of the signal to be analyzed on line 109
is fed
through a directional tap 110 then amplified by the broadband amplifiers 132,
124
feeding the appropriate filter , in filter bank 1'74. This filtered signal is
then further
amplified by the broadband amplifiers 120,130 then fed to the quadrature
modulator 138
at line 134. It is parenthetically noted that the assumption made here is that
the
quadrature modulator 138 is reciprocal, that is, that it is constructed from
passive
devices allowing its internal power combiner to act as a power splitter and
that both it's
balanced mixers are bidirectional. Under these assumptions, device 138 may
also be
referred to as a .quadrature demodulator, for which the quadrature outputs are
on lines
161,162. An example of a reciprocal quadrature modulator 138 that may be
employed
in accordance with the teaching of the .gresent invention is Model QMC-170
manufactured by the Mini-Circuits corporation of Brooklyn, New York. It can be
shown
that by providing a particular sinusoidal signal pair of a given frequency k
at the output
of the low, piss filters .164 and 166 and feeding them to the mixers of the
quadrature
demodulator 138 at lines 161, and.162 respectively, double mixing will occur,
so that
DC components related to the secondary mixing at baseband will result at lines
161, and
162. This operation is essentially that of a double conversion quadrature
receiver where
the first local oscillator is supplied at line 146 while the second "local
oscillatorsu are
supplied in quadrature by pure sinusoidal signals digitally generated by the
DACs 168
and 170 through Iines I6I and 162. The DC components on these same lines
constitute
the detected signals coupled through high impedances 176 and 178 to a gated
Integrate
and Dump (''I&D") circuits 180 and 182, the output .of which can then be
sampled
seqnentiatly using switch 184 that is connected via line 185 to the analog to
digital
("AID") converter 188. The values acquired by the A/D 188 are communicated to
the
DSPluC 152 via data bus 190. It should be noted that the output impedance of
the Iow
pass filters 164 and 166 should be sufficiently large in order that the DC
components
CA 02527157 1993-O1-07
-20-
generated by the balanced mixers in 138 be measurable. For every frequency k,
two
independent measurements are performed by setting the RAM contents so that the
DACs
168 and l7U generate in the first measurement the values
Bl(t)=a cos(2~rktllil) ; B2(t}=a sin(2xktlN~
and in the second measurement the values
BI(r)=a sin(2~rktl3~ ; B2(t)=a cos(2~rkt/~ .
where a is an appropriate amplitude scale factor adapted to the DACs dynamic
range.
The measured scalars obtained in I&Drs 180 and I82 during the first
measurement (acquired sequentially using both states of switch 184} are
designated here
by xk, vk respectively, while those similarly obtained in the second
measurement, are
designated as yk, uk respectively. It can be shown that the first measurement
corresponds
to analysis of the upper sideband, a step hereinafter denoted by AN+, while
the second
measurement corresponds to an analysis of the lower sideband channel, denoted
here by
AN . Hence, four parameters for each frequency k are obtained, thereby
enabling the
determination of the inconung signal phasors for Goth the upper (-t-) aad
lower
sideband channels. These are given by a relation related to the inverse of
Equation (I),
namely:
~k 0 1 1 0, x~
AQk 1 1 0 0 -1 Yk ~)
~~ 2 0 -1 1 0 u~
AQk I 0 0 1 v
k
The phasor values obtained in Equation (2) durixtg the measurement periods
A,N+
~d ~ , hereinafter termed analysis periods, are used bx the D~P/uC 1;52 for
computing and adjusting the content of RAMS In and 1'72 for the ejection
period in
a manner to be further discussed below, so as to provide for the precise phase
and
amplitude relationship required of the injected signals during the HBI baswi
on the
CA 02527157 1993-O1-07
received phasor analysis. For the center carrier analysis (k=~; the system
degenerates
into a single conversion receiver, in which one applies 0 values for Bl(tJ and
BZ(tJ and
with a single measurement, one directly obtains the two quadrature component
estimates
at the I&D's 180 and 182 outputs.
In order to perform the injection and analysis in coordination with the group
video timing, a vertical frame reference data sequence is inserted in the VBI
of all, or
at least one, of the channels within the group. The apparatus of Figure 3
provides for
vertical frame synchronization by the use of data matched filter 194 and sync
and data
detector 196, resetting the timing chain circuitry 210 via sync detect line
202. The
timing chain circuitry 210 provides all timing signals (tl to t7) controlling
the RF
switches, the I&D's and the A/D strobe. The address generator 150 is provided
with the
proper reset and preload signals from the timing chain circuit 210 so as to
scan the
proper RA,M memory locations containing the appropriate data records for the
synthesis
of the injection signals and the analysissignals as required. As explained
above,
based on the reciprocity property of the broadband generator 136, it also
serves as a
broadband quadrature synchronous receiver which is used for phasor analysis
and data
and sync acquisition on selected channels. A particular channel can therefore
be received
by the additive combination of the quadrature signals at 161 and 162 via the
summing
resistors 193 and 192 that are connected to the matched filter 194. The
channel that is
being received can be determined by the proper choice of the sinusoidal
components
being generated by the DAC pair 168,170 so that the appropriate sinusoids are
applied
in quadrature at 161 and 162 respectively, essentially in the same manner as
that
described for the analysis measurements AN+ or AN-, depending whether an upper
side
or lower side channel is being received. For example, if data in the VBI of an
upper
sided channel corresponding to frequency index k is being received, the RAM
values of
B,(tJ and BZ(tJ are set to
Bl (t) _a cos( 2~ +B) ~ BZ(t) =a sin( 2M +e) . ;
CA 02527157 1993-O1-07
-zz-
where B is the phase angle required to receive this channel in-phase based on
the
measured components AIk+ and AQx+. It can be shown that the above
trigonometric
expressions for the RAM values required in the receive mode, correspond to'
the values
which would be obtained according to Equation (1) with only one term in the
sum
(corresponding to frequency index k), wherein Ix and Qx are set to zero.
Stated another
way; the RAM contents required in order to receive a selected channel is the
same as
than which is required to generate a single carrier, in-phase coherence with
that channel's
received carrier.
With the proper RAM signals set as described above, data inserted in the ~Bf
of
IO any channel within the group can be selectively detected by the matched
filter I94 and
the data detector 196. This detected data stream is used to address the
subscriber units
in a manner known in the art of addressable subscription control The identity
of the
specific channels authorized for descrambling and those denied from each
subscriber will
then be communicated to the microcontroller section of 152 of each subscriber.
Based
1~ on this subscription configuration information, the DSP/uC portion of 152
calculates the
spec ~~ ~mbination required for B1 (t) and BZ(t) in accordance with Equation
(1)
for each frequency k.
To , see this in the specific example of Figure 2, one first designates the
Iower
sided channel corresponding to frequency index k as ° K" and the upper
sided channel
2U of that freqneacy index as "+K".' Accordingly, note that Figure 2 contains
nine
channels- with the . designations -4,-3, 2,-1,0, -!-I,-f-z, +3; +4. Given this
channel
designation method, Figure 2b indicates a specific subscription configuration
which calls
for DESCRAMBLING (in phase injection) of Channels -3,-1 and +3, shown as
injected
signals S0, and further DENIM: (out-of phase injection) of Channels 2 and +z;
shown
25 as injected signals 52 and finally NO PROCESSING (no injection) on Channels
-4,0,+I
and +4, shown at 54. It is assumed that the values of AIk~,AQ,~~',AI,~ and AQk
for the
channels in need of injection have been obtained by the Analysis measurement
sessions
AN+ and AN as described above and in accordance with Equation (2): Based on
this
subscription configuration, the following vector entries should be used for
each k in the
30 right hand side of Equation (1) in order to generate the required
subscription
configuration of Figure 2:
CA 02527157 1993-O1-07
0 d A1,: -8'A12 d ~A13 0
p d.AQi -S'A~z d'ASZs 0
k=0: ; k=1: ; k=2: ; k=3: ; k=4:
0 0 -g ~AIZ d ~AI3 0
0
0 .B.AQi d.AQ3 0
where d and g are appropriate positive scale factors relating the relative
magnitude of
the measured carrier during the analysis period and the injection phasor
magnitudes
required for descrambling and out-of-phase injection.denial respectively.
~s recalled from a previous discussion, the injected signals must be VSB
modulated. This can be accomplished by providing digital samples to DACs 168
and 170
in accordance with digitally precalculated sequences which specifically yield
injection
pulses onset and terminations with VSB spectra. For a single carrier systems,
there
exists known methods of providing two quadrature components with certain
relationships
that yield a VSB modulated signal. A discussion of such methods can be found
in the
book entitled Principles of Data Communications by R. Lucky, J. Salt and E.J.
Weldon
published, by McGraw-Hill, New York 1968. It can be mathematically shown that
two
such signals used for quadrature modulation of a carrier yielding a desired
VSB pulse
shaping are given by C(t) and S(t) shown in Figure 4 whereiwC(t) is the
modulating
function of the in-phase component and S(t) is that of the quadrature
component. Recall
that the case of CW injection 30 of Figure lb is characterized by a time
invariant
column phasor vector on the right hand side of Equation (1). Thus, for the
inclusion of
VSB transitions 32,33, the time invariant phasor column in Equation (1) must
be
replaced by the time varying vector Rk(t) given by:
C(t~ S(t) 0 0 Ik .... ..
_S(t) C(t) 0 0 Q~ (3)
R~(t) _ 0 0 C(t) S(t) Ik
~ p -'~(t~ c(t) Qk
CA 02527157 1993-O1-07
24.-
Since this expression is used in Equation (1) for each frequency k, and since
C(t) and
S(t) are not periodic functions, generally, the values of Br(t) and B2(tJ will
not be
periodic in 12 samples, but rather a longer data record will be required for
the synthesis
of these spectrally shaped signals. Hence, with an appropriately increased
value for the
$ sample number M, and the numerical values for the functions C(tJ and S(tJ
stored in a
ROM inside the DSP/uC 152, the RAM record values BI(tJ and B2(tJ are
calculated by
the DSPIuC 152 in accordance with Equation (1-A):
(1-A)
Bi(t)
BZ(t) _
~.N ~s(2~k t ) sin(2~rk t ) 0 0 ~ 1 0 l '
g~~t)
_~ M M 1 0 -1 0
~° 0 ~ cos(2xkM) sin(2~rk~) 1 ~ I 0
0 -1 0 I
It should be further noted that a multiplicity of incrementally spaced VSB
signals
similar to those in Figure 2b would thus be generated on line 134. If, for
example,
lV--4, Equation (I-A) w~71 yield 9 independent VSB modulated phasors generated
with
nominal phasor values determined by the selected phasor infection values fOr
I,~~',Qk+,I~
and Q,E in Equation (3~, with k = O,I,2,3,4. These phasor values will be set
in
accordance with the desired processing action on each channel, based on the
subscriber's
subscription status.
There exists a tradeoff between the RAM record length required for
representing
transients 32 and 33 of Figure lb and the equivalent channel passband
frequency
response and transition bandwidths of the VSB spectra.: In the example of the
first
preferred embodiment of the present invention, the transition intervals. 402
and 403 in
Figure 4, each correspond..to a record lengtlu of 109 samples at 72 MSps. for
each of
records 32 and 33 respectively: Figure 5 depicts the frequency response 'S00 _
of an
equivalent digital filter that was computationaly applied to a squaFe
modulated pulse from
CA 02527157 1993-O1-07
_25_
which the baseband quadrature functions C(t) and S(t) shown in Figure 4 are
derived.
As can be seen in Figure 5, one can achieve an acceptable VSB spectral shaping
about
the picture carrier with this relatively short sample sequence. Another
feature provided
by the digital signal processing capability of the present invention is the
ability to design
the functions C(t) and S(t) based on a precise computational digital filter of
Figure 5,
wherein specific spectral nulls are provided around the audio subcarrier
frequencies 501,
thereby ensuring that the broadband injected VSB signals do not interfere with
the audio
portion of the signals. These strong attenuation features will in general not
be required
on frequencies which only affect the video portion of adjacent channels; since
the
resultant crosstalk effects are video frame synchronous for a contiguous group
of
channels and thus introduce only small transient artifacts at the edges of the
HBI which
are unviewable. Referring again to Figure 5, a specific provision far a
spectral null at
the chroma subcamer frequency 502 of the lower adjacent channel is made so as
to
minimize chroma interference to the lower adjacent channel bordering the frame
synchronous group of channels, should it's video timing be horizontally
shifted in
accordance with a video staggering method to be explained below.
Summarizing the discussions above, it, is this specific ability of generating
.
multichannel signals with essentially arbitrary amplifude and phase in video
synchrony
and RF coherence with incoming signals that allows one to provide simultaneous
processing of a group of channels.
Because of practical limitations such as DAC clock rates and DAC dynamic range
and linearity, the number of channels that can simultaneously and accurately
be
generated in the broadband generator 136 of Figure 3 is limited. It is
therefore desirable
to be able to use the broadband generator 136 in a timesharing mode, whereupon
several
channel.groups may be processed sequentially, thereby increasing the total
number of
channels processed by switching the .local oscillators feeding the broadband
generator
I36 to the center frequency channels of each group. This operation mode is
generally
possible in this video application since the required processing (and
injection) time per
lane is limited to the HBI which is less than one forth of the total
horizontal line time.
Thus, at the headend, groups of channels are video synchronized in a staggered
manner
so that their HBI do not overlap. This is shown schematically in Figure 6a.
Time
CA 02527157 1993-O1-07
-2ti-
intervals b00 provide a guard interval; during which no injection is required,
for
switching the center carrier local oscillators among the four channel groups A
through
D (switch 144 in Figure 3) and the appropriate group band pass filter in
filter bank 1?4
using switch 122. Tyro possible frequency arrangements for channel assignments
for
each group are shown in Figures 6b and tx. The first preferred embodiment of
Figure
3 utilizes the frequency block grouping of Pigure 6b, s~~ it requires lower
DAC speeds
foz the same number of channels per group. However, at the expense of faster
DACs
and RAMS, the frequency interlaced grouping of Figure 6c offers potential
savings in
the filter bank,-since in this case only a single (and wider) harmonic filter
is required.
In the preferred embodiment of channel frequency grouping of Figure 6b, the
local
oscillator of frequency Fa 602 is selected by switch 144 during the interval
labeled as
GROUP A in Figure 6a. Subsequently, switch 144 is similarly moved to select
the Iocal
oscillator corresponding to the center channel of channel GROUP B,. at
frequency Fb
604 and so on for groups C and D, to fill out the rest of the horizontal video
time,
sec~nentially dwelling on frequencies Fc 606 and Fd b08 for equal periods of
time,
corresponding to the HBI duration: In this manner; the same broadband
generator
performs the required processing for each channel in the four groups by
coherent
injection during the HBI of every channel.
Figure 7 shows the method of digital generation of the injected waveforms
during
one horizontal video line far all channel groups. As can be seen,. in each
channel .group
period; there are three RAM segments each corresponding to sequence of samples
required for the respective time segments 32,30,33 of Figure Ib. The address
value
schedule generated by the address generator 150 is shown by 700. The RAM
address
value at each time is shown on the vertical axis range ?02. it is generated by
the address
generator 150 and supplied to B1 and B2 RAMS 171 and I?2 simultaneously via
RAM
address bus 156. The address generator ISO is advanced by the clock signal
online 148,
said clock .signal frequency is preferably an integral multiple of the channel
incremental
frequency which is 6 MHz in North America. Hence, in the example of the
present
invention the clock frequency is 72 MHz.
1fie numerical values stored in Bl and B2 RAMs; that are fed to DACs 168 and
170, are prepared by the DSPIuC 152 based on calculations specified in
Equation (I-A)
CA 02527157 1993-O1-07
-27-
for each channel group, thus filling in the values in the RAM VSB "rise"
section 732,
CW section 730 and VSB "fall" section 733 for channel GROUP A and similarly
for the
other three groups.
'I~rning to the first embodiment of the present invention, the system of
Figure
3 maintains its injection and analysis waveforms in a fixed phase and
amplitude
relationship to the incoming signals by closing a phasor control loop for each
processed
channel by making successive analysis and corrections, thereby tracking any
slow
-alive phase or amplitude drifts ~n the CATV distribution system or any of the
:riponents in the subscriber unit sr r as power splitt - 142; directional
coupler 10~:
the broadband amplifier 104 or any v: the componen.~ within the broadband
gener4
136 which may affect the relative injected phasors as compared to the incoming
channel
phasors.
In order to better understand the phasor tracking and calibration method of
the
present invention, reference is made to Figure 8. It is assumed that the
phasor diagrams
in Figure 8 correspond to phasors observed on line 109 using the analysis
method
discussed above for an arbitrary channel within the controllable channel
group. S,,
ghasor 300, represents the picture carrier phasor of an incoming carrier with
black level
transmission of 0 IRE during a line portion of the VBI for which no injection
is
required. At this point, an analysis period can be commenced to yield an
estimate of the
phasor Sl. If one assumes that the sync signal is suppressed by a baseband
equivalent
shift of 70 IRE, one may wish to measure a portion of the signal thaf
picovides identical
suppression but at a line portion which does not require reconstruction by
injection,
hence making available the broadband generator as a phasor analyzer. This may
be done
by modifying the video signal at the headend to have a 70 IRE gray pedestal in
a line
portion of the VBI which otherwise would be at 0 IRE blanking level. This
ituation is
depicted by 5~, (phasor 302) suppressed by the 70 IRE pedestal corresponding
to an RF
amplitude amount 303. In theory; an analysis of S~ and S2 should suffice to
determine
their phasor difference so that it may be generated and .injected by the
broadband
generator 136. The difficulty with this approach stems from the fact that the
signal paths
foz the measured signals and the injected signals are unequal . and thus no
direct
measurement of the injected phasor as it is imparted on line 109 is available.
This
CA 02527157 1993-O1-07
-28-
primarily stems from the fact that during the injection period; the generator
must
generate a plurality of signals which cannot be useful for phasor analysis and
vice versa.
That is, only one function can be achieved at a time. It would be preferable
to measure
S, the combination of the incoming phasvr S~ and the injected phasor S;g;
which would
then be compared to Sl. Using these two measurements., one then derives the
error
phasor dS based on the vector equation
aS = Sl -S (4)
From the above equation, a phasor increment is calculated by DSPIu~ I5Z and
accumulated so as to modify the RAM records for injection. This process is
constructed
iteratively so convezgence to phasor matching is effected:
The assumption made above regarding the measurability of S is that both an
injection and analysis sources exist simultaneously. It is possible to obtain
a measure for
S in a sequential manner by taking advantage of sufficient SAW filter delay in
filter bank
I74. The method is based on an injection of the required composite RF signal
during a
short pulse and the subsequent reception of the delayed version of the pulse
while the
broadband generator becomes an analyzer. This is done by manipulating RF
switches
I16. and 126 during a full line of the. YBI wherein a 70 IRE pedestal is
inserted at the
headend. In the first preferred embodiment, such calibration and analysis is
performed
dur;.ag video lines 17 and 280. Figure 9 shows the pulsed injection and
analysis timing
as well as the state schedule of RF switches lI6 and 126. Figure 10 shows the
related
RAM addressing schedule in which injection sectors designated by Rec. A,B,C
and
analysis sectors AN+ and AN are periodically alternating through the control
of the
gddzess generator I50. The samples obtained by the AID I88 in this process may
be
averaged in the DSP/uC 152 in order to provide additional precision for the
phasor
estimates thus derived. The procedure described above can be employed
sequentially on
all channels for which injection is required.
The above paragraph described the measurement of the combined phasor S of
Figure 8. In order to perform the adjustrrient in accordance with Et~nation
(4), Sl, may
be obtained freely during much bf the U 1RE blanking level of the VBI. Tn this
case,
multiple channels and groups can be processed within one .BBL frame. This is
simply
CA 02527157 1993-O1-07
-29-
done by subjecting the system to sequential CW analysis sessions AN+ and AN-
as
shown for multiple groups in Figure 11.
The overall video timing schedule of the first embodiment of the present
invention is shown in Figure 12. As can be seen, the periods for injection,
analysis, sync
and data detection are non-overlapping and thus provide an efficient use of
the
broadband generator. Upon completion of each video field or frame, correction
to all
injecfed phasors can be made in accordance with increments OS yielding for all
frequencies the following increments forBl(t) and BZ(t):
~l~t)
~2(t) -
0 1 0 1 ~Ik
,~=N cos(2~kM) sin(2~kM) 0 0 1 0 -1 0 OQk
x=o 0 0 cos(2~rk~) sin(Z~rkM) 1 0 1 ~ Olk
0 -1 0 1 ~
4lCk
Figure 13 provides a schematic diagram of a CATV headend configured in
accordance with the present invention in which only two groups of channels
(groups B
and D) contain scrambled channels to be processed by the subscriber unit of
Figure 3.
A comb generator supplies independent comb signals to each group, thereby
allowing
partial IRC operation. A more detailed view of the comb generator is provided
in Figure
14. The two broadband comb generators feeding group B comb and group D comb
are
of similar construction to the subscriber broadband generator :X36. However,
here, all
carriers are constructed in a CW periodic mode. Thus, for a 72 MHz clock
frequency,
they address generator is essentially a modulo 12 counter providing the 6 MHz
periodicity. The 72 MHz pilot signal is also feti into the CATV system in
order to
provide all subscriber units with synchronous clock. Note also that the master
video
timing supplied by the genlock reference signal is locked to the 72 MHz clock,
thereby
guaranteeing, the rigid relationship between line rate and RF clock rate that
eliminates
fhe need to change the RF RAM records for every video line at the subscriber
unit. The
CA 02527157 1993-O1-07
-30-
relative phases of the comb signals may be changed by downloading other RAM
records
through data bases W$ and WD. lfiese phase values are practically irrelevant
to the
operation of the system of the present invention but it may be desirable tv
adjust them
in order to reduce total transmission distortion as disclosed in the above
cited U.S.
Patent to the present inventor.
Figure ISa shows a baseband representation of an RF sync suppression HBI.
Figure 15b depicts tile required coherent injection signal for descrambIing.
Here, due
to the attenuation at RF,' all signal components are, in need of injection
including a
portion of the color burst. As can be seen, six RAM records are required for
generation
IO of the picture carrier portion of the injection signal. These are 1032,
1040, I03$, 1030,
1036, and 1033. Atthough it may be possible to generate the missing burst
signal by
VSB generation about the picture carrier, this approach is prohibitively
expensive in
memory because of the poor congruence between the color subcarrier frequency
and 6
NlHz.
IS An alternative solution is based on the fact that all upper sideband color
subcarriers in a frame synchronized video sources are incrementally related
with 6 MHz
spacing. Hence they can be generated separately with a 3:5$ MHz offset local
oscillator
locked on the color burst of the center carrier channel as shown in the
description of the
second embodiment of the present invention provided by Figures lb and 17. In a
manner
20 similar to the first embodiment, all injected phasors are measured and
tracked preferably
during the VBI. Figure I7 shows the arrangement for a Multiple Dwelling Unit
- (n~~n~ yubscziber group. Here,. the digital broadband .generator of
subscriber A
prlo~lvliJdes the analysis function during calibrated injection of subscriber
B and vice versa.
Finally, it is worth noting that if a broadband generator is dedicated to one
group
25 during the entire video line, it can generate additional fixed injection
signals of varying
levels during the active video time so as to provide an additional denial
security based
qn random "video folding". This method cart be implemented by effecting gated
coherent
injection at the headend using some fixed set of injection values governed by
cryptographic keystream control and selectively null them out (therefore
descrantble) at
~. 30 the subscriber location based on precise phasor adjustment. Because the
headend injected
signal can be in opposite phase, video inversion on onl~r a sector of the
screen may take
CA 02527157 1993-O1-07
_31 _
place, further frustrating any existing "pirate" descramblers. Figure 18 shows
the R:AM
timing diagram for a descrambler utilizing the video folding desczambling and
signal
denial technique:
