Note: Descriptions are shown in the official language in which they were submitted.
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TITLE OF THE INVENTION
ECHO CANCELLER EMPLOYING DUAL-H ARCHITECTURE
HAVING VARIABLE ADAPTIVE GAIN SETTINGS
CROSS-REFERENCE TO RELATED APPLICATIONS
The following are related US Patents:
Patent No. 6,181,793 "Echo Canceller Employing Dual-H Architecture Having
Improved Coefficient Transfer"; Patent No. 6,266,409, "Echo Canceller
Employing
Dual-H Architecture Having Improved Double-Talk Detection"; Patent No.
6,028,929,
"Echo Canceller Employing Dual-H Architecture Having Improved Non-Linear Echo
Path Detection"; Patent No. 6,198,819, "Echo Canceller Employing Dual-H
Architecture
Having Improved Non-Linear Processor"; Patent No. 6,240,180, "Echo Canceller
Employing Duel-H Architecture Having Split Adaptive Gain Settings".
STATEMENT REGARDING SPONSORED RESEARCH
OR DEVELOPMENT
Not Applicable
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BACKGROUND OF THE INVENTION
Long distance telephone facilities usually comprise four-wire transmission
circuits between switching offices in different local exchange areas, and two-
wire
circuits within each area connecting individual subscribers with the switching
office. A
call between subscribers in different exchange areas is carried over a two-
wire circuit in
each of the areas and a four-wire circuit between the areas, with conversion
of speech
energy between the two and four-wire circuits being effected by hybrid
circuits.
Ideally, the hybrid circuit input ports perfectly match the impedances of the
two and
four-wire circuits, and its balanced network impedance perfectly matches the
impedance
of the two-wire circuit. In this manner, the signals transmitted from one
exchange area
to the other will not be reflected or returned to the one area as echo.
Unfortunately,
due to impedance differences which inherently exist between different two and
four-
wire circuits, and because impedances must be matched at each frequency in the
voice
band, it is virtually impossible for a given hybrid circuit to perfectly match
the
impedances of any particular two and four-wire transmission circuit. Echo is,
therefore, characteristically part of a long distance telephone system.
Although undesirable, echo is tolerable in a telephone system so long as the
time
delay in the echo path is relatively short, for example, shorter than about 40
milliseconds. However, longer echo delays can be distracting or utterly
confusing to a
far end speaker, and to reduce the same to a tolerable level an echo canceller
may be
used toward each end of the path to cancel echo which otherwise would return
to the far
end speaker. As is known, echo cancellers monitor the signals on the receive
channel
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of a four-wire circuit and generate estimates of the actual echoes expected to
return
over the transmit channel. The echo estimates are then applied to a subtractor
circuit in
the transmit channel to remove or at least reduce the actual echo.
In simplest form, generation of an echo estimate compiises obtaining
individual
samples of the signal on the receive channel, convolving the samples with the
impulse
response of the system and then subtracting, at the appropriate time, the
resulting
products or echo estimates from the actual echo on the transmit channel. In
actual
practice generation of an echo estimate is not nearly so straightforward.
Transmission circuits, except those which are purely resistive, exhibit an
impulse response that has amplitude and phase dispersive characteristics that
are
frequency dependent, since phase shift and amplitude attenuation vary with
frequency.
To this end, a suitable known technique for generating an echo estimate
contemplates
manipulating representations of a plurality of samples of signals which cause
the echo
and samples of impulse responses of the system through a convolution process
to obtain
an echo estimate which reasonably represents the actual echo expected on the
echo
path. One such system is illustrated in FIG. 1.
In the system illustrated in FIG. 1, a far end signal x from a remote
telephone
system is received locally at line 10. As a result of the previously noted
imperfections
in the local system, a portion of the signal x is echoed back to the remote
site at line 15
along with the signal v from the local telephone system. The echo response is
illustrated here as a signal s corresponding to the following equation:
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where h is the impulse response of the echo characteristics. As such, the
signal sent
from the near end to the far end, absent echo cancellation, is the signal y,
which is the
sum of the telephone signal v and the echo signal s. This signal is
illustrated as y at line
15 of FIG. 1.
To reduce and/or eliminate the echo signal component s from the signal y, the
system of FIG. 1 uses an echo canceller having an impulse response filter h
that is the
estimate of the impulse echo response h. As such, a further signal s
representing an
estimate of echo signal s is generated by the echo canceller in accordance
with the
following equation:
s=h*x
The echo canceller subtracts the echo estimate signal y from the signal y to
generate a signal e at line 20 that is returned to the far end telephone
system. The
signal e thus corresponds to the following equation:
e=s+v-s'z~v
As such, the signal returned to the far end station is dominated by the signal
v of the
near end telephone system. As the echo impulse response lh more closely
correlates to
the actual echo response h, then s-bar more closely approximates s and thus
the
magnitude of the echo signal component s on the signal e is more substantially
reduced.
The echo impulse response model h may be replaced by an adaptive digital
filter having an impulse response h. Generally, the tap coefficients for such
an
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adaptive response filter are found using a technique known as Normalized Least
Mean
Squares adaptation.
Although such an adaptive echo canceller architecture provides the echo
canceller with the ability to readily adapt to changes in the echo path
response h, it is
highly susceptible to generating sub-optimal echo cancellation responses in
the presence
of "double talk" (a condition that occurs when both the speaker at the far end
and the
speaker at the near end are speaking concurrently as determined from the
viewpoint of
the echo canceller).
To reduce this sensitivity to double-talk conditions, it has been suggested to
use
both a non-adaptive response and an adaptive response filter in a single echo
canceller.
One such echo canceller is described in USPN 3,787,645, issued to Ochiai et al
on
January 22, 1974. Such an echo canceller is now commonly referred to as a dual-
H
echo canceller.
Although the dual-H echo canceller architecture of the '645 patent provides
substantial improvements over the use of a single filter response
architecture, the '645
patent is deficient in many respects and lacks certain teachings for
optimizing the use of
a such a dual-H architecture in a practical echo canceller system. For
example, the
present inventors have recognize that the adaptation gain used to adapt the
tap
coefficients of the adaptive filter may need to be altered based on certain
detected
conditions. These conditions include conditions such as double-talk, non-
linear echo
response paths, high background noise conditions, etc.. The present inventors
have
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recognized the problems associated with the foregoing dual-H architecture and
have
provided solutions to such conditions.
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BRIEF SUMMARY OF THE INVENTION
An echo canceller circuit for the use in an echo canceller system is set forth
that
provides sensitive double-talk detection. The echo canceller circuit comprises
a second
digital filter having adaptive tap coefficients to simulate an echo response
occurring
during the call. The adaptive tap coefficients of the second digital filter
are updated over
the duration of the call using a Least Mean Squares process having an adaptive
gain a. A
channel condition detector is used to detect channel conditions during the
call. The
channel condition detector is responsive to detected channel conditions for
changing the
adaptive gain a during the call. For example, the channel condition detector
may detect
the presence of a double-talk condition and set the adaptive gain a to zero.
Similarly, the
channel condition detector may detect the occurrence of a high background
noise
condition and set the adaptive gain a to a level less than 1 that is dependent
on the
detected level of the background noise. Other similar channel conditions and
corresponding adaptive gain settings may likewise be utilized.
According to one aspect of the invention, there is provided an echo canceller
circuit comprising a digital filter having adaptive tap coefficients that
simulate an echo
response, the adaptive tap coefficients being updated at times using during
each such time
an adaptive process having an adaptive gain and a channel condition detector
that lowers
the adaptive gain a variable amount depending on the amount of background
noise, the
echo canceller circuit being responsive to the adaptive tap coefficients to
generate an
echo-compensated signal.
According to another aspect of the invention, there is provided an echo
canceller
circuit comprising a digital filter having adaptive tap coefficients that
simulate an echo
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response, the adaptive tap coefficients being updated at times using during
each such time
an adaptive process having an adaptive gain and a channel condition detector
that lowers
the adaptive gain in response to a narrow band signal condition, the echo
canceller circuit
being responsive to the adaptive tap coefficients to generate an echo-
compensated signal.
According to a further aspect of the invention, there is provided an echo
canceller
circuit comprising a digital filter having adaptive tap coefficients that
simulate an echo
response, the adaptive tap coefficients being updated at times using during
each such time
an adaptive process having an adaptive gain and a channel condition detector
that lowers
the adaptive gain in response to a non-linear echo path condition, the echo
canceller
circuit being responsive to the adaptive tap coefficients to generate an echo-
compensated
signal.
According to another aspect of the invention, there is provided an echo
canceller
circuit comprising a digital filter having adaptive tap coefficients that
simulate an echo
response, the adaptive tap coefficients being updated at times using during
each such time
an adaptive process having an adaptive gain and a channel condition detector
that lowers
the adaptive gain in response to substandard performance of the echo
canceller, the echo
canceller circuit being responsive to the adaptive tap coefficients to
generate an echo-
compensated signal.
According to a further aspect of the invention, there is provided a method for
processing an echo response, comprising simulating the echo response with
adaptive tap
coefficients updating the adaptive tap coefficients at times with an adaptive
process
having an adaptive gain and lowering the adaptive gain depending on an amount
of
background noise.
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According to another aspect of the invention, there is provided a method for
processing an echo response of a far-end signal, comprising simulating the
echo response
with adaptive tap coefficients; updating the adaptive tap coefficients at
times with an
adaptive process having an adaptive gain and lowering the adaptive gain in
response to a
narrow band signal condition.
According to a further aspect of the invention, there is provided a method for
processing an echo response, comprising simulating the echo response with
adaptive tap
coefficients; updating the adaptive tap coefficients at times with an adaptive
process
having an adaptive gain and lowering the adaptive gain in response to a non-
linear echo
path condition.
According to another aspect of the invention, there is provided a method for
processing an echo response, comprising simulating the echo response with
adaptive tap
coefficients; updating the adaptive tap coefficients at times with an adaptive
process
having an adaptive gain and adjusting the adaptive gain during substandard
echo
cancellation performance.
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BRIEF DESCRIPTION OF THE SEVERAL
VIEWS OF THE DRAWINGS
Figure 1 is a block diagram of a conventional canceller.
Figure 2 is a schematic block diagram of an echo canceller that operates in
accordance with one embodiment of the present invention.
Figure 3 is a flow chart illustrating one manner of carrying out coefficient
transfers wherein the transfer conditions may be used to implement double-talk
detection in accordance with one embodiment of the present invention.
Figure 4 is a flow chart illustrating a further manner of carrying out
coefficient
wherein the transfer conditions may be used to implement the double-talk
detection an
accordance with one embodiment of the present invention.
Figure 5 illustrates an exemplary solution surface for the adaptive filter
whereby
the desired result is achieved at the solution matching the echo response of
the channele
Figure 6 illustrates one manner of checking for various echo canceller
conditions and responding to these conditions using a change in the adaptive
gain
setting of the adaptive filter of the echo canceller.
Figure 7 illustrates one manner of implementing an echo canceller system
employing the present invention.
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DETAILED DESCRIPTION OF THE INVENTION
Figure 2 illustrates one embodiment of a dual-h echo canceller suitable for
use
in implementing the present invention. As illustrated, the echo canceller,
shown
generally at 25, includes both a non-adaptive filter h and an adaptive filter
h to model
the echo response h. Each of the filters h and h are preferably implemented as
digital
finite impulse response (FIR) filters comprising a plurality of taps each
having a
corresponding tap coefficient. The duration of each of the FIR filters should
be
sufficient to cover the duration of the echo response of the channel in which
the echo
canceller 25 is disposed.
The output of the non-adaptive filter h is available at the line 30 while the
output of the adaptive filter h is available at line 35. Each of the signals
at lines 30
and 35 are subtracted from the signal-plus-echo signal of line 40 to generate
echo
compensated signals at lines 50 and 55, respectively. A switch 45, preferably
a
software switch, may be used to selectively provide either the output signal
at the line
50 or the output signal at line 55 to the echo canceller output at line 60.
A transfer controller 65 is used to transfer the tap coefficients of filter h
to
replace the tap coefficients of filter h. As illustrated, the transfer
controller 65 is
connected to receive a number of system input signals. Of particular import
with
respect to the present invention, the transfer controller 65 receives the
signal-plus-echo
response y and each of the echo canceller signals e and e at lines 50 and 55,
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respectively. The transfer controller 65 is preferably implemented in the
software of
one or more digital signal processors used to implement the echo canceller 25.
As noted above, the art is substantially deficient of teachings with respect
to the
manner in which and conditions under which a transfer of tap coefficients from
h to h
is to occur. The present inventors have implemented a new process and, as
such, a
new echo canceller in which tap coefficient transfers are only made by the
transfer
controller 65 when selected criterion are met. The resulting echo canceller 25
has
substantial advantages with respect to reduced double-talk sensitivity and
increased
double-talk detection capability. Further, it ensures a monotonic improvement
in the
estimates h .
The foregoing system uses a parameter known as echo-return-loss-enhancement
(ERLE) to measure and keep track of system performance. Two ERLE parameter
values are used in the determination as to whether the transfer controller 65
transfers
the tap coefficients from h to h. The first parameter, E, is defined in the
following
manner:
E=y
e
Similarly, the parameter E is defmed as follows:
y
e
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Each of the values E and E may also be averaged over a predetermined number of
samples to arrive at averaged E and E values used in the system for the
transfer
determinations.
Figure 3 illustrates one manner of implementing the echo canceller 25 using
the
parameters E and E to control tap coefficients transfers between filter h to
h. As
illustrated, the echo canceller 25 provides a default h set of coefficients at
step 80
during the initial portions of the call. After the tap coefficients values for
h have been
set, a measure of E is made at step 85 to measure the performance of the tap
coefficient values of filter h.
After the initialization sequence of steps 80 and 85, or concurrent therewith,
the
echo canceller 25 begins and continues to adapt the coefficients of h to more
adequately match the echo response h of the overall system. As noted in Figure
3, this
operation occurs at step 90. Preferably, the adaptation is made using a
Normalized
Least Mean Squares method, although other adaptive methods may also be used
(e.g.,
LMS and RLS).
After a period of time has elapsed, preferably, a predetermined minimum period
of time, the echo canceller 25 makes a measure of E at step 95. Preferably,
this
measurement is an averaged measurement. At step 100, the echo canceller 25
compares the value of E with the value of E. If the value of E is greater than
the
value of E, the tap coefficients of filter h are transferred to replace the
tap coefficients
of filter h at step 105. If this criterion is not met, however, the echo
canceller 25 will
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continue to adapt the coefficients of the adaptive filter /h at step 90,
periodically
measure the value of E at step 95, and make the comparison of step 100 until
the
condition is met.
If the echo canceller 25 finds that E is greater than E, the above-noted
transfer
takes place. Additionally, the echo canceller 25 stores the value of E as a
value E..
This operation is depicted at step 110 of the Figure 3. The value of E,n,,, is
thus the
maximum value of ERLE that occurs over the duration of the call and at which a
transfer has taken place. This further value is used thereafter, in addition
to the E and
E comparison, to control whether the tap coefficients of h are transferred by
the
transfer controller 65 to replace the tap coefficients of h, This further
process is
illustrated that steps 115, 120, and 125 of Figure 3. In each instance, the
tap
coefficient transfer only occurs when both of the following two conditions are
met: 1)
E is greater than the current , and 2) E is greater than any previous value of
E used
~
during the course of the call. (E is greater than Eõ,.). Each time that both
criteria are
met, the transfer controller 65 of echo canceller 25 executes the tap
coefficient transfer
and replaces the previous Em~ value with the current E. value for future
comparison.
Requiring that E be greater than any E value used over the course of the call
before the coefficient transfer takes place has two beneficial and desirable
effects.
First, each transfer is likely to replace the prior tap coefficients of filter
h with a better
estimate of the echo path response. Second, this transfer requirement
increases the
double-talk protection of the echo canceller system. Although it is possible
to have
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positive ERLE E during double-talk, the probability that E is greater than
E,.,,, during
double-talk decreases as the value of Eincreases. Thus an undesirable
coefficient
transfer during double-talk becomes increasingly unlikely as the value of
E,n,,, increases
throughout the duration of the call.
The echo canceller 25 may impose both an upper boundary and a lower
boundary on the value of E,,.. For example, En,,,, may have a lower bounded
value of 6
dB and an upper bounded value of 24 dB. The purpose of the lower bound is to
prevent normal transfers during double-talk conditions. It has been shown in
simulations using speech inputs that during double-talk, a value of greater
than 6 dB
ERLE was a very low probability event. The upper bound on E. is used to
prevent a
spuriously high measurement from setting Em~ to a value at which further
transfers
become impossible.
The value of E,,,ar should be set to, for example, the lower bound value at
the
beginning of each call. Failure to do so will prevent tap coefficient
transfers on a new
call until the echo cancellation response of the echo canceller 25 on the new
call
surpasses the quality of the response existing at the end of the prior call.
However, this
criterion may never be met during the subsequent call thereby causing the echo
canceller 25 to operate using sub-optimal tap coefficients values. Resetting
the E,n.
value to a lower value increases the likelihood that a tap coefficient
transfer will take
place and, thereby, assists in ensuring that the h filter uses tap
coefficients for echo
cancellation that more closely correspond to the echo path response of the new
call.
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One manner of implementing the Evalue change is illustrated in the echo
canceller operations flow-chart of Figure 4. When all transfer conditions are
met
except E greater than Eand this condition persists for a predetermined
duration of
time, the echo canceller 25 will reset the E,,. value to, for example, the
lower bound
value. In the exemplary operations shown in Figure 4, the echo canceller 25
determines whether E is greater than the lower bound of E,,. at step 140 and
less than
the value of E,õ,,, at step 145. If both of these conditions remain true for a
predetermined period of time as determined at step 150, and all other transfer
criterion
have been met, the echo canceller 25 resets the E,,,,,, value, at step 155.
This lowering
of the E. value increases the likelihood of a subsequent tap coefficient
transfer.
Choosing values for the lower and upper bound of E,,. other than 6 dB and 24
dB, respectively, is also possible in the present system. Choosing a lower
bound of
E,,. smaller than 6 dB provides for a relatively prompt tap coefficient
transfer after a
reset operation or a new call, but sacrifices some double-talk protection. A
value
greater than 6 dB, however, inhibits tap coefficient transfer for a longer
period of time,
but increases the double-talk immunity of the echo canceller. Similarly,
varying the
value of the predetermined wait time T before which E. is reset may also be
used to
tweak echo canceller performance. A shorter predetermined wait time T produces
faster reconvergence transfers, but may sacrifice some double-talk immunity.
The
opposite is true for larger predetermined wait time values.
A further modification of the foregoing echo canceller system relates to the
value stored as E,õ,, at the instant of tap coefficient transfer. Instead of
setting Em,,,
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equal to the E value at the transfer instant, Emax may be set to a value equal
to the value of
E minus a constant value (e.g., one, three, or 6 dB). At no time, however,
should the Emax
value be set to a value that is below the lower bound value for Emax.
Additionally, a
further condition may be imposed in that a new softened Emax is not less than
the prior
value of Emax. The foregoing "softening" of the Emax value increases the
number of
transfers that occur and, further, provides more decision-making weight to the
condition
of E being larger than E. Further details with respect to the operation of the
echo
canceller coefficient transfer process are set forth and the co-pending patent
application
titled "Echo Canceller Employing Dual-H Architecture Having Improved
Coefficient
Transfer", (Now US Patent No. 6,181,793).
Preferably, the adaptive filter h uses a Normalized Least Mean Square (NLMS)
adaptation process to update its tap coefficients. In accordance with the
process,
coefficients are adapted at each time n for each tap m= 0, 1, . . . , N - 1 in
accordance with
the following equation:
a
hõ+i (m) = h,, (m) + N-1
Tx2
i
i=a
where hõ(m) is the mth tap of the echo canceller, xõ is the far-end signal at
time n, e,z is the
adaptation error of time n, and aõ is the adaptation gain at time n.
The foregoing adaptation process will converge in the mean-square sense to the
correct solution the echo path response h if 0< a" < 2. Fastest convergence
occurs
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when a = 1. However, for 0< a_< 1, the speed of convergence to h is traded-off
against steady-state performance.
Figure 5 is provided to conceptualize the effect of the adaptation gain on the
filter response. The graph or Figure 5 includes an error performance surface
185
defined to be the mean square error between h and h, to be a V dimensional
bowl.
Each point in the bowl corresponds to the mean-square error for each
corresponding h
(of length M. The bottom of the bowl is the h which produces the least mean-
square
error, i.e., h. The NLMS process alternatively moves the h towards h at the
bottom of
the performance surface as shown by arrow 190. When a 1, h moves to the
bottom of the bowl most quickly, but one the bottom is reached, the adaptation
process
continues to bounce h around the true h bottom of the bowl, i.e., EIhI= h but
h* h.
If a small a is used, then the steady-state error is smaller (h will remain
closer to h),
but h requires a longer time to descend to the bottom of the bowl, as each
step is
smaller.
In some cases, as the present inventors have recognized, the performance
surface will temporarily change. In such situations, it becomes desirable to
suppress
the h from following these changes. This presents a challenge to choose the
best a for
each scenario.
Figure 6 illustrates operation of the echo canceller 25 in response to various
detected scenarios. It will be recognized that the sequence of detecting the
various
conditions that is set forth in Figure 6 is merely illustrative and may be
significantly
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varied. Further, it will be recognized that the detection and response to each
scenario
may be performed concurrently with other echo canceller processes. Still
further, it
will be recognized that certain detected scenarios and their corresponding
responses
may be omitted.
In the embodiment of Figure 6, the echo canceller 25 entertains whether or not
a
double-talk condition exists at step 200. Double talk, as noted above, is
defined as the
situation when both far-end and near-end talkers speak at the same time during
a call.
In such a scenario, the adaptive error signal is so severely corrupted by the
near-end
speaker that it is rendered useless. As such, if a double-talk condition is
detected, the
echo canceller 25 responds by freezing the adaptation process at step 205,
i.e., set a
0, until the double talk ceases.
There are several methods that the echo canceller 25 can use for detecting a
double-talk condition. One is to compare the power of the near-end signai to
the far-
end signal. If the near-end power comes close enough to the far-end power
("close
enough" can be determined by the system designer, e.g., within 0 or 6 or
10dB), then
double talk can be declared. Another method is to compare the point-by-point
magnitudes of the near-end and far-end signals. This search can compare the
current
ixl with the current ryl, the current Ixl with the last several lyl, the
current jyl with the
last several lxl, etc. In each case, the max lxl and jyl over the searched
regions are
compared. If
maxl y, > Double Talk Threshold
maxlxl
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where max lxl indicates the maximum Jxl over the search region (tyl is
similarly defined),
then a double-talk condition is declared.
A still further manner of detecting a double-talk condition is set forth in US
Patent
No. 6,266,409, titled "Echo Canceller Employing Dual-H Architecture Having
Improved
Double-Talk Detection". As set forth in that patent, a double-talk condition
is declared
based on certain monitored filter performance parameters.
It may be possible to further condition the double-talk declaration with other
measurements. For example, the current Echo Return Loss (ERL) may be used to
set the
Double Talk Threshold noted above herein. The short-term power of either the
far-end,
the near-end, or both, may also be monitored to ensure that they are larger
than some
absolute threshold (e.g. -50dBm or -40dBm). In this manner, a double-talk
condition is
not needlessly declared when neither end is speaking.
Once a double-talk condition is declared, it may be desirable to maintain the
double-talk declaration for a set period of time after the double-talk
condition is met.
Examples might be 32, 64, or 96 msec. After the double-talk condition ceases
to exist,
the adaptive gain value may be returned to the value that existed prior to the
detection of
the double-talk condition, or to a predetermined return value.
At step 210, the echo canceller 25 determines whether a high background noise
condition is present. A low level of constant background noise can enter from
the near-
end, for example, if the near-end caller is in an automobile or an airport.
Its effects are
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noise is frequently constant, thus setting a= 0 until the noise ends is not
particularly
advantageous. Also background noise is usually of lower power than double-
talk. As
such, it corrupts the adaptation process but does not render the resulting
adaptation
coefficients unusable.
As illustrated at step 215, it is desirable to choose a gain 0< a < 1, i.e.
lower
the gain from its fastest value of 1 when a high background noise condition is
present.
While this will slow the adaptation time, the steady state performance
increases since
the effects of noise-induced perturbations will be reduced. In other words,
the tap
variance noise is reduced by lowering the adaptation gain a.
Preferably, the background noise is measured as a long-term measurement of
the power of when the far-end is silent. As this measurement increases, a
decreases.
One schedule for setting the adaptive gain a as a function of background noise
level is
set forth below.
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Background Noise (dBm) a
> -48 e125
> -54 _ -48 .25
> -60 z -54 .5
< -60
It will be readily recognized that there are other schedules that would work
as well, the
foregoing schedule being illustrative.
A further condition in which the adaptive gain may be altered from an
otherwise
usual gain value occurs when the adaptive filter h is confronted with a far-
end signal
that is narrow band, i.e. comprised of a few sinusoids. In such a scenario,
there are an
infinite number of equally optimal solutions that the LMS adaptation scheme
can find.
Thus it is quite unlikely that the resulting cancellation solution h will
properly identify
(i.e. mirror) the channel echo response h. Such a situation is referred to as
under-
exciting the channel, in that the signal only provides information about the
channel
response at a few frequencies. The echo canceller 25 attempts to determine the
existence of this condition that step 220.
Consider a situation where the far-end signal varies between periods in which
a
narrow band signal is transmitted and wide band signal is transmitted. During
the wide
band signal periods, the h filter should adapt to reflect the impulse response
of the
channel. However, when the narrow band signal transmission period begins, the
h
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filter will readapt to focus on canceling the echo path distortion only at the
frequencies
present in the narrow band signal. Optimizing a solution at just a few
frequencies is
likely to give a different solution than was found during transmission of the
wide band
signal. As a result, any worthwhile adaptation channel inforrnation gained
during wide
band transmission periods is lost and the lh filter requires another period of
adaptation
once the wide band signal returns.
When the far-end signal is narrow band, the adaptation can and should be
slowed considerably, which should discourage the tendency of the coefficients
to
diverge. Specifically, when a narrow band signal is detected, a may be upper-
bounded
by either 0.25 or .125. This operation is illustrated at step 225.
Narrow band signal detection may be implemented using a fourth order
predictive filter. Preferably, this filter is implemented in software executed
by one or
more digital signal processors used in the echo canceller system 25. If it is
able to
achieve a prediction gain of at least 3 to 6 dB (user defined) over the h
filter, then it is
assumed that the received signal is a narrow band signal.
An amplitude threshold for the far-end signal is also preferably employed in
determining the existence of a narrow band signal. If the far-end power is
greater than
-40 dBm, the current far-end sample is sent to the fourth order predictive
filter, which
determines whether or not the far-end signal is narrow band. If the far-end
power is
less than -40 dBm, the predictive filter is re-initialized to zero.
A further scenario in which it is desirable to alter the gain of the adaptive
filter
h is when the echo path response is non-linear. The presence of non-
linearities in the
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echo path encourages constant minor changes in the coefficients h in order to
find short-
term optimal cancellation solutions. The detection of non-linearity of the
echo path
response preferably proceeds in the manner set forth in US Patent No.
6,028,929, titled
"Echo Canceller Employing Dual-H Architecture Having Improved Non-Linear Echo
Path Detection". The presence of a non-linear echo path is determined at step
230.
In a non-linear echo path scenario, it is desirable to choose the adaptive
gain
constant a large enough that h can track these short-term best solutions.
However,
choosing a = 1 may be suboptimal in most non-linear scenarios. This is due to
the fact
that the gain is too large and, thus, short-term solutions are "overshot" by
the aggressive
adaptation effort. Accordingly, as shown at step 235, choosing a gain lower
than I is
preferable. Choosing a = 0.25 was found to be the best trade off between
tracking and
overshooting short term optimal solutions. The gain constant a may be further
reduced if
large background noise is measured, as discussed above.
A still further scenario in which the adaptive gain may be varied relates to
the
convergence period of the adaptive filter h. As noted above, a large gain
constant a is
desired during convergence periods while a smaller a is desired in steady
state conditions
after the filter has converged. In other words, there seems little lost and
perhaps some
potential gain to reduce a after an initial period of convergence is
completed. This
appears to be especially valuable if the long-term performance is found to be
substandard.
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(e.g., at either .25 or .125). This mode is detected at step 240 and is
entered at step
245 if the ERLE remains below a predetermined threshold value (e.g., either
6dB or
3dB) after a predetermined period of adaptation. The adaptation time is
preferably
selected as a value between 100 to 300 msec. This amount of time will
generally
prevent the echo canceller 25 from entering the reduced gain mode during
convergence
periods.
As will be readily recognized, the echo canceller of the present invention may
be implemented in a wide range of manners. Preferably, the echo canceller
system is
implemented using one or more digital signal processors to carry out the
filter and
transfer operations. Digital-to-analog conversions of various signals are
carried out in
accordance with known techniques for use by the digital signal processors.
Figure 7, illustrates one embodiment of an echo canceller system, shown
generally at 700, that maybe used to cancel echoes in multi-channel
communication
transmissions. As illustrated, the system 700 includes an input 705 that is
connected to
receive a multi-channel communications data, such as a T1 transmission. A
central
controller 710 deinterleaves the various channels of the transmission and
provides them
to respective convolution processors 715 over a data bus 720. It is within the
convolution processors 715 that a majority of the foregoing operations take
place. Each
convolution processor 715 is designed to process at least one channel of the
transmission at line 730. After each convolution processor 715 has processed
its
respective channel(s), the resulting data is placed on the data bus 720. The
central
controller 710 multiplexes the data into the proper multichannel format (e.g.,
T1) for
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retransmission at line 735. User interface 740 is provided to set various user
programmable parameters of the system.
Numerous modifications may be made to the foregoing system without departing
from the basic teachings thereof. Although the present invention has been
described in
substantial detail with reference to one or more specific embodiments, those
of skill in
the art will recognize that changes may be made thereto without departing from
the
scope and spirit of the invention as set forth in the appended claims.
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