Note: Descriptions are shown in the official language in which they were submitted.
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A WAVEGUIDE FILTER
TECHNICAL FIELD
This invention relates to waveguide filters. More particularly, this invention
relates to
substrate integrated waveguide bandpass filters.
BACKGROUND OF THE INVENTION
An electrical bandpass filter is a fundamental element used for selecting an
electrical
signal in a frequency passband while suppressing electrical signals in a
frequency stopband of
the filter. Microwave and millimeter-wave bandpass filters are often used in
modem radio-
frequency transceivers. Filters having low in-band insertion loss, high
spectral selectivity, and a
wide stopband are commonly required. As an example, in a typical ground
terminal for
communication with satellites in the K, frequency band, a filter is required
to suppress signals at
transmission frequencies in a 29.5GHz - 30GHz frequency range while conveying
the signals at
reception frequencies in a 19.2GHz - 21.2GHz frequency range. An insertion
loss of less than
1dB and a stopband suppression level of at least 45dB are desired to select
the signal while
avoiding self-jamming effects during simultaneous reception and transmission
of
electromagnetic signals by the ground terminal.
Microwave bandpass filters can be implemented as bulk waveguide structures.
These are
relatively heavy, bulky, and expensive: due to their size and weight,
integration of bulk
waveguide filters with planar components and electronic circuits can be a
challenging task.
Substrate integrated waveguides (SIWs) are waveguide structures formed in a
substrate
of an electronic circuit. SIWs allow easy integration of planar circuits on a
single substrate using
a standard printed circuit board (PCB) or low-temperature co-fired ceramic
(LTCC) process, or
any other process of planar circuit fabrication. By using SIWs in an
electronic circuit, the
interconnection loss between components can be reduced. The size and the
weight of the entire
circuit can also be reduced.
SIW filters are known in the art. They offer a low-cost, low mass and compact
size
alternative to conventional waveguide filters, while maintaining high
performance. Although
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various techniques have been implemented to improve the stopband performance
of conventional
i-ectangular waveguide filters, these techniques often utilize E-plane
discontinuities that are
difficult to realize for SIW filters implemented on a single-layer substrate.
The SIW filters of the
prior art have often been limited to resonant structures based on physical
coupling elements to
achieve a pre-selected spectral shape of the filter response function and/or
high levels of
stopband suppression. For example, a SIW filter designed to block an
electromagnetic signal at
a frequency fo has a slit in the top or bottom conducting layer to provide an
attenuation pole at
the frequency fa.
Transmission zeros (TZs) in the insertion loss response of a microwave filter
can be used
to improve the spectral selectivity and the stopband attenuation of the
filter. To generate the
TZs, an "extracted pole" technique can be implemented to construct so called
"bandstop"
resonators. Alten7atively, electrical couplings can be introduced between non-
adjacent
resonators, wherein the TZs are generated due to a phenomenon of multipath
interference of
electromagnetic waves propagating inside the resonators. However, such filters
are usually
constructed using conventional waveguide technology, which tends to use bulky
and complex
filter structures. Furthermore, the TZs implemented using these prior-art
methods cannot be far
away from the desired passband due to the limitation of the physical structure
of a prior-art
waveguide filter.
The present invention overcomes the above stated problems of the prior art. It
provides a
low-cost, high-performance SIW filter that is easy to integrate with planar
circuits.
Advantageously, the spectral shape of the SIW filter of the present invention
can be adapted to
provide a high level of attenuation away from a desired passband. Furthermore,
SIW filters can
offer a significant improvement in passive intermodulation performance over
conventional
filters.
SUMMARY OF THE INVENTION
According to the present invention, a substrate integrated waveguide (SIW)
filter includes
a chain of sequentially coupled conterminous multimode SIW cavities, of which
the first and the
last multimode SIW cavities can be directly excited by a transmission line.
The entire filter is
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implemented using arrays of metalized via holes on a dielectric substrate. The
via holes are
produced by using a standard printed circuit board (PCB) or other planar
circuit manufacturing
process. The diameter of the via holes and the pitch between neighboring via
holes are selected
so as to suppress radiation losses in the SIW cavities. A desired passband is
generated by the
fundamental mode of propagation in the SIW cavities. The finite transmission
zeros (TZs) are
generated by destructive interference between the fundamental and a higher-
order
electromagnetic mode of the SIW cavities. The size and the shape of the SIW
cavities are
selected so that the TZs are far away from the passband, for high out-of-band
rejection. The
position of every finite TZ is independently controllable. The freedom of
positioning the TZs is
achieved by changing the inter-cavity coupling ratios and the size of
corresponding multimode
SIW cavities. According to the present invention, no other mode discriminating
physical
structures within the SIW cavities, such as openings in a conductive layer of
the PCB, are
required to control the position of the TZs.
In accordance with the invention there is provided a filter having a passband
and a
stopband, for conveying passband frequency components of an electromagnetic
signal, while
suppressing stopband frequency components of the electromagnetic signal, the
filter comprising:
an SIW formed in a planar dielectric layer sandwiched between first and second
opposing
planar conductive layers, the SIW having a chain of sequentially coupled
conterminous
multimode SIW cavities defined on their perimeters by an array of conductive
vias
connecting the first and the second conductive layers through the dielectric
layer, the
chain having first and second ends;
an input transmission line coupled to the first end of the chain, for coupling
the
electromagnetic signal to the first end of the chain; and
an output transmission line coupled to the second end of the chain, for
outputting the
passband frequency components of the electromagnetic signal from the second
end of the
chain;
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wherein a distance between neighboring vias of the array of conductive vias is
small
enough to suppress radiation losses of the SIW, for example less than half of
a shortest
wavelength of the electromagnetic signal in the SIW cavities.
BRIEF DESCRIPTION OF THE DRAWINGS
Exemplary embodiments will now be described in conjunction with the drawings
in
wliich:
FIG. 1 is a three-dimensional view of a single-cavity substrate integrated
waveguide
(SIW) filter having opposing input and output microstrip transmission lines;
FIG. 2 is a three-dimensional view of a single-cavity SIW filter having input
and output
microstrip transmission lines disposed at 90 with respect to each other;
FIG. 3 is an equivalent circuit model for the mode coupling in the SIW
cavities of FIGs.
1 and 2;
FIGs. 4A and 4B are magnetic field distributions of the fundamental mode and a
higher-
order mode, respectively, of the SIW filter of FIG. 1;
FIG. 5 is an insertion loss spectral plot for the SIW filter of FIG. 1,
superimposed with
electric field distribution patterns in the SIW cavity corresponding to a
first transmission
maximum, a first transmission zero (TZ), and a second transmission maximum;
FIGs. 6, 7, and 8 are three-dimensional views of SIW filters of the pi-esent
invention,
having four sequentially coupled conterminous multimode SIW cavities;
FIGs. 9A and 9B are electric field distribution patterns in a four-cavity SIW
filter at a
fundamental passband and a spurious passband frequency of a signal,
respectively;
FIGs. 10 to12 are spectral plots of transmission and reflection of the SIW
filters of FIGs.
6 to 8, respectively;
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FIGs. 13 to 15 are plan views of SIW filters of FIGs. 6 to 8, respectively,
showing
dimension notations of the filters;
FIG. 16 is a comparative spectral plot of simulated and measured insertion
loss of a SIW
filter of FIG. 7; and
FIG. 17 is a comparative spectral plot of simulated and measured insertion
loss of a SIW
filter of FIG. 8.
DETAILED DESCRIPTION OF THE INVENTION
While the present teachings are described in conjunction with various
embodiments and
examples, it is not intended that the present teachings be limited to such
embodiments. On the
contrary, the present teachings encompass various alternatives, modifications
and equivalents, as
will be appreciated by those of skill in the art. In FIGs. 6, 7, 8, 9A, and
9B, like numerals refer
to like elements.
A waveguide filter of the present invention uses at least two electromagnetic
modes,
propagating or evanescent. A passband of the filter is defined by a frequency
range at which
only the fundamental mode appears at an output port of the filter. A stopband
of the filter is
defined by all frequencies outside of the passband. Within the stopband,
higher-order modes
may create spurious passbands. By careftilly selecting the dimensions of the
substrate integrated
waveguide (SIW) cavity, one transmission zero (TZ) or multiple TZs can be
generated at specific
locations in the stopband to suppress these spurious passbands.
In general, the insertion loss of a filter is proportional to the number of
resonators n,
inversely proportional to the unloaded quality factor Qu of the resonator, and
also the relative
bandwidtll FBJV of the filter. For a small-ripple, less than 0.1dB, Chebyshev
filter, the increase
in insertion loss dS21 at a center frequency 4 is given by
" (1)
AS 20B) w=wo - 4.FBW343 ~ g;
Qu;
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wherein gi is a generalized low-pass prototype element (inductor or capacitor)
value for an i`h
resonator.
The Qu of an SIW cavity is determined by three Q-factors, namely, the Q-factor
related
to lossy conducting walls Qc, the Q-factor related to dielectric loss D: Qd =
1/tan(D), and the Q-
factor related to energy leakage via gaps in the SIW cavity Qr. The unloaded
quality factor is
then expressed as
Qu - Q, + Qd+QY (2)
As is known in the art, by properly selecting the SIW substrate materials and
the shape of
the filter, the radiation loss represented by 1/Qr can be made much smaller
than the dielectric
and conductive losses represented respectively by 1/Qd or 1/Qc. At Ka-band,
the SIW cavity
based on a conventional microwave dielectric substrate with a height of 20mil
and a dielectric
loss tangent tan(D) of 0.0012 has a Qu of about 350, which is a typical
quality factor of finline
waveguide resonators. Therefore, a small number of SIW cavities, preferably
four cavities, are
used in a filter of the present invention to minimize insertion loss. The
spectral selectivity of a
filter of the present invention is improved by selecting SIW cavities of
certain size and shape as
will now be described.
Referring to FIG. 1, a single-cavity SIW filter 10 is presented having a
dielectric layer 11
sandwiched between a top planar conductive layer 12 and a bottom planar
conductive layer 13.
A SIW cavity 19 of the filter 10 is defined on the perimeter of the cavity 19
by an array of
conductive vias 14 connecting the top and the bottom conductive layers 12 and
13 through the
dielectric layer 11. The SIW cavity 19 is directly excited by one of
symmetrical 5052, microstrip
lines 15 or 16. Due to the symmetry of the SIW cavity 19, it supports only
TEõo,,, modes of
propagation, wherein m is a positive number and n is an odd positive number.
Preferably, the
SIW cavity 19 is shaped and sized so as to support only two modes of
propagation of the
intended signal, the TEiol mode and the TE301 mode. The SIW filter 10 can be
manufactured at
a low cost using a standard printed circuit board (PCB) manufacturing process,
or a low-
temperature co-fired ceramic (LTCC) manufacturing process.
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Throughout the specification, multimode SIW cavities are called,
interchangeably,
"oversized" cavities. This means that the size of the cavities can support
more than one mode of
propagation of an incoming signal. The SIW cavity 19 is termed herein as
"oversized TE101 /
TE30, SIW cavity".
The distance b between neighboring vias 14 is small enough to suppress
radiation losses
of the SIW cavity 19. As a rule, the distance b should be less than one half
of the shortest
wavelength of the electromagnetic signal in the SIW cavity 19. The distance b
for the cavity 19
of FIG. 1 is lmm, and the diameter d of the vias 14 is 0.5mm. The overall size
of the SIW cavity
19 is approximately 4.5mm x 10.5mm for the given passband frequency range and
the selected
dielectric layer material Rogers RT/DuroidTM 6002. A central frequency f of
the passband is
related to effective width aeff and length lef~ of the SIW cavity 19 as
follows:
co (
1 3)
2 ~
o = 7+~I_fj
- ~~a:nwhere co is the speed of light in air, a~ff = a - dy 95b' h~ d~ 0.95b ~
and where a and 1 are
tlie geometrical width and length of the SIW cavity 19, respectively.
Referring to FIG. 2, a single-cavity SIW filter 20 has the same elements as
the filter 10 of
FIG. 1, but the microstrip line 16 is at 90 w.r.t. the microstrip line 15. An
oversized cavity 29
of the filter 20 supports two modes of propagation of an electromagnetic
signal, the TEIOI mode
and the TE201 mode. The SIW cavity 29 is termed herein as "oversized TEIOI /
TEZOI SIW
cavity". The coupling between the input and the output microstrip lines 15 or
16 and the higher-
order TE201 mode can reverse when the relative position of the lines 15 and 16
changes from the
same half of the SIW cavity 29 to the opposite half of the cavity 29. This
coupling, which
reaches a maximum when the input and the output are at an angle of 90 , can be
adjusted by
changing the relative position of the input and the output microstrip lines 15
and 16 and the size
of the SIW cavity 29. Therefore, a finite TZ can be on the lower-frequency
side or the higher-
frequency side of the resonance of the higher-order TE201 mode, and can be
positioned slightly
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closer to the resonance of the fundamental TEzOI mode, to further improve the
stopband
performance of the filter 20.
Turning now to FIG. 3, an equivalent circuit model 30 for the mode coupling in
the SIW
cavities 19 and 29 of FIGs. 1 and 2 is illustrated. The model 30 shows, in a
symbolic form,
signal paths between a source port S and a load port L. The fundamental
resonant mode TE]ol
generates a transmission pole in the desired passband. A second-order resonant
mode TE301
provides a different path for the signal flow between the two ports S and L
corresponding to
inicrostrip lines 15 and 16 of the SIW filter 10 from a path corresponding to
the fundamental
resonant mode TE101. Similarly, a second-order resonant mode TE201 provides a
different path
for the signal flow between the two ports S and L corresponding to microstrip
lines 15 and 16 of
the SIW filter 20 as compared to a path provided by the fundamental resonant
mode TEIO,.
Because all the couplings J, ', Jz', J3', and J4' in an oversized SIW cavity
of the present invention
have the same sign, and Jl' and Jz' are much larger than J3' and J4' close to
the resonant
frequency of the second-order mode TE201 or TE301, a TZ between the resonant
frequency of the
TE I mode and the resonant frequency of the TE201 or TE301 mode is generated.
The location of
the TZ can be approximately determined by using the following relationship:
3 4 Bi (
~~~- J~ J~ rF~,,, ITezo1 l4)
, wherein (o', is the generalized angular frequency of the TZ, JI' and J2' are
the generalized
coupling admittances between the source port S and the load port L and TEiol
mode, and J3' and
J4' are the generalized coupling admittances between the source port S and the
load port L and
one of TE201 or TE301 modes, as is denoted in FIG. 3. B'TE2O11TE301 is the
generalized constant
susceptance of one of the TE)oj or TE301 modes. In general, the TZ is shifted
in frequency
relative to the transmission pole of the fundamental mode TEIor because the
product of Ji' and
J2' is much larger than the product of J3' and J4' close to the resonance
frequency of the TE201 or
TE301 mode. For the oversized SIW cavity 19, the location of the TZ can be
slightly tuned by
changing the width of the SIW cavity 19 with little effect on the desired
passband response
generated by the TElol mode. The location of the TZ in the oversized SIW
cavity 29 can be
tuned by changing the relative position of the microstrip lines 15 and 16, as
noted above.
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Turning now to FIGs. 4A and 4B, magnetic field distributions 40A and 40B of
the
fundamental mode TEIOI and the higher-order mode TE301 are illustrated. The
modes TEIOI and
TE301 are symmetrically excited in the SIW cavity 19 by the 5052 microstrip
line 15. The mode
couplings between the microstrip line 15 and the modes TEiol and TE3o, are
both positive, the
coupling between the microstrip line 15 and the TEIoi mode being significantly
stronger than the
coupling between the microstrip line 15 and the TEjol mode. Thus, a TZ above
the resonance of
the TErol mode is generated; this TZ is shifted far away from the resonance of
the TEJo, mode
because the coupling between the microstrip line 15 and the TElol mode is much
stronger than
the coupling between the microstrip line 15 and the TE301 mode.
Referring to FIG. 5, a simulated spectral plot 50 of the insertion loss of the
single-cavity
SIW filter 10 is shown, having superimposed thereupon electric field
distributions in the SIW
cavity 19 of the filter 10 corresponding to a first transmission maximum 54, a
first TZ 55, and a
second transmission maximum 56. A pattern 51 denotes the electric field
distribution at the
resonance point 54 in the SIW cavity 19 of the filter 10 excited by the input
microstrip line 15.
The pattern 51 corresponds to an electric field distribution of a transmission
pole, when the TEIoI
rnode is in resonance. Similarly, patterns 52 and 53 denote the electric field
distribution at the
TZ 55 and at the transmission pole 56, respectively. At the point 55, the
TE301 mode is close to
being in resonance, at which point it is of a sufficient strength to cancel
the off-resonance mode
TElo, at the output microstrip line 16. One can see that the TZ 55 is
generated at about 30GHz,
while the point of maximum transmission 54 is at 20GHz. Advantageously, such a
large
distance between the TZ 55 and the transmission pole 54 is generated without
resorting to
placing any discriminating physical structures inside the cavity 10, such as
openings in the top
conductive layer 12 or the bottom conductive layer 13 of the SIW cavity 10.
Referring now to FIG. 6, a three-dimensional view of an SIW filter 60 of the
present
invention is sllown. Similar to the single-cavity SIW filter 10 of FIG. 1, the
SIW filter 60 of
FIG. 6 has a dielectric layer 61 sandwiched between top and bottom opposing
planar conductive
layers 62 and 63, respectively. An array of the conductive vias 14 connects
the conductive layers
62 and 63 through the dielectric layer 61 thereby forming a chain of four
sequentially coupled
conterminous multimode SIW cavities 69, to 694 defined on their perimeters by
an array of the
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vias 14 as shown. The neighboring cavities 69, and 692; 692 and 693; and 693
and 694 are
coupled to each other by a via-free opening 101 in a common wall therebetween.
The SIW
cavity 69, is directly excited by an input signal coupled to a transmission
line 65, and a
transmission line 66 is used to output the signal. The lines 65 and 66 are
preferably microstrips,
however striplines or coplanar waveguides can also be used. Inside the outer
SIW cavities 69,
and 694, the lines 65 and 66 are defined by non-conductive slots 67 and 68,
respectively. The
slots 67 and 68 have ends perpendicular to the lines 65 and 66, which
facilitates improvement of
the stopband performance without deteriorating the passband performance of the
filter 60.
Preferably, the slots 67 and 68 and the microstrips 65 and 66 are formed by
patterning the top
conductive layer 62. The electromagnetic signal is coupled into the first SIW
cavity 69i by the
line 65 having slots 67, and then is coupled into the next cavities 692; 693;
and 694 by the via-free
openings, or "post-wall irises" 101 as shown in FIG. 6. The via-free openings
are defined by
eight conductive vias 14 common to perimeters of neighboring SIW cavities. At
least two vias
can be used for this purpose. The line 66 is used to output the
electromagnetic signal from the
last cavity 694 of the filter 60.
According to the present invention, the size and the shape of the SIW cavities
69, to 694
of the filter 60 are selected to support at least two modes of propagation for
passband frequency
components and for stopband frequency components of the electromagnetic
signal. At least two
modes of each stopband frequency component cancel each other at TZs upon
propagating
through the chain of the SIW cavities 69i to 694, thereby suppressing the
stopband frequency
components. Preferably, the output transmission line 66 is positioned at one
of these TZs, so that
the two modes of each stopband frequency component cancel each other upon
propagating
through the filter 60. The output transmission line 66 may be disposed co-
planar with the top
conductive layer 62, as is shown in FIG. 6, or, alternatively, it may be co-
planar with the bottom
conductive layer 63.
The position of the TZs is dependent on the position of the input transmission
line 65 and
the shape of the SIW cavities 69i to 694. A specific example of dimensions of
the filter 60
suitable for Ka-band performance will be given below. Spatial distributions of
the electric field
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in a filter having similar geometry as the filter 60 are shown in FIGs. 9A and
9B, to be discussed
later.
The stopband frequency components are suppressed at the prescribed finite TZs
produced
by corresponding oversized SIW ca~ities. Preferably, each SIW cavity 69i to
694 is of such
shape and size that the two modes of at least a fraction of the stopband
frequency components
cancel each other upon propagating through a corresponding SIW cavity.
Shifting the
frequencies of TZs of the SIW cavities 69, to 694 relative to each other
results in broadening of
the stopband of the filter 60, while still attaining high levels of
attenuation in the stopband.
Turning to FIGs. 7 and 8, three-dimensional views of SIW filter 70 and 80 of
the present
invention are shown, respectively. The SIW filter 70 has SIW cavities 79, to
794, and the SIW
filter 80 has SIW cavities 89i to 894. What is different between the SIW
filters 60, 70, and 80 of
FIGs. 6, 7, and 8, is the position of the input microstrip lines 65 and the
output microstrip lines
66 relative to a longitudinal axis 102. Specifically, in the SIW filter 60,
the microstrip lines 65
and 66 are parallel to the axis 102; in the SIW filter 70, the microstrip line
65 is parallel to the
axis 102 while the microstrip line 66 is perpendicular to the axis 102; and in
the SIW filter 80,
both microstrip lines 65 and 66 are perpendicular to the axis 102.
Accordingly, the SIW cavities
69i to 694; 79, to 793; and 892 and 893 are oversized TElol / TE301 SIW
cavities; and the SIW
cavities 794, 891, and 894 are oversized TElol / TE201 SIW cavities. Varying
orientations of the
microstrip lines 65 and 66 allow fine tuning of the TZ frequencies of a first
and a last SIW cavity
in a chain of consecutively coupled SIW cavities, in a similar manner to
tuning the TZ
frequencies of the SIW cavity 29 of FIG. 2.
Referring now to FIGs. 9A and 9B, simulated electric field distribution
patterns 91A and
91B in the SIW cavities 991 to 694 of the filter 90 are shown. The filter 90
has the same general
geometry as the filter 60 of FIG. 6, having input and output microstrip lines
95 and 96,
respectively, and TEIOI / TE301 STW cavities 99i to 994. The patterns 91A and
91B correspond to
electromagnetic signals at a fundamental passband frequency and a spurious
passband frequency,
respectively. The resonant mode of the fundamental passband is the TEJo, mode,
while the
resonant mode of the spurious passband is the TE301 mode.
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Turning now to FIGs. 10 to 12, simulated transmission and reflection response
characteristics of the SIW filters 60, 70, and 80 of FIGs. 6, 7, and 8 are
shown, respectively. The
filters 60, 70, and 80 are exemplary embodiments of a Ka -band filter. In a Ka-
band satellite
communications ground terminal, the transmission occurs at 29.5 to 30GHz,
while the reception
occurs within 19.2 - 21.2GHz. A receiving filter is normally used for
suppressing a 29.5-30GHz
transmission signal to prevent self-jamming, while conveying a 19.2 - 21.2GHz
signal to be
received by a receiver. One can see that the stopband rejection over the
satellite transmit
frequency band of 29.5-30GHz, seen in FIG. 10, is close to 45dB. Furthermore,
in FIGs. 11 and
12, the stopband rejection of the filters 70 and 80 over the satellite
transmit frequency band of
29.5-30GHz is better than 50dB, although only four multimode SIW cavities are
used to arrive at
a low insertion loss of 0.5 - 0.7dB. An alternative way of defining the
performance of the filters
60, 70, and 80 as seen from FIGs. 10 to 12, is to define a 3dB passband and a
35dB stopband.
The 3dB bandwidth of the passband in FIGs. 10 to 12 is at least 10% of a
center frequencyfp =
20.2 GHz of the passband, that is, a middle frequency of the 3-dB points
defining the passband.
The 35dB bandwidth of the stopband is at least 2% of a center frequency fs =
29.75 GHz of the
stopband, that is, a middle frequency of the 35-dB points defining the
stopband. This
performance is achieved at the stopband located away from the passband, so
thatfs - fp > 0.3 *
fp.
Referring to FIGs. 13 to 15, plan views of SIW filters of the present
invention are
presented. The views of FIGs. 13, 14, and 15 show notations of the main
dimensions of the
filters 60, 70, and 80, respectively. Tables 1 to 3 below show example
dimensions of the
corresponding Ka -band filters, in accordance with the notations of FIGs. 13
to 15.
TABLE 1 for FILTER 60
w,,, 3.22mm h 4.46mm
W12 3.19mm 12 4.54mm
w23 2.99mm aslW 10.5mm
wms 1.28mm wsLo 2.56mm
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TABLE 2 for FILTER 70
w,,,s. 1.28mm W12 3.19mm
wio 3.22mm W23 2.99mm
Wi 2.56mm W34 3.24mrn
1.48rnrn a, 10.66mm
4.4611un a2 6.60rnnt
4.54inm wo 3.14nam
13 4.53mm l0 1.6mm
14 5.35mm
TABLE 3 for FILTER 80
fv,,s 1.28mm W23 2.99mm
w,o 3.08mm W34 3.24mm
tiy; 2.88mm al 6.6mm
/; 1.50innr az 10.751111n
t~ Ii 5.43n1m u4 6.6rnnn
12 4.47mm wo 3.14mm
13 4.52mm 10 1.6mm
14 5.35mm ol 3.14mm
w72 3.46mm o4 2. l l mm
A skilled artisan will realize that the filter shapes and sizes, defined by
the sets of
dimensions tabulated in Tables 1 to 3, are not the only possible shapes and
sizes of a Ka -band
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filter of the present invention. Furthermore, for another passband and
stopband frequency and
attenuation level specification, as well as for another dielectric layer
material, the dimensions can
be different. It is to be understood, however, that the invention encompasses
various sizes and
shapes of SIW cavities that support two modes, so that the two modes cancel
each other upon
propagating through the sequential chain of the SIW cavities, thereby
suppressing the stopband
frequency components at defined TZ locations. As is appreciated by one skilled
in the art, the
above described "mode cancelling" ftinction will determine the shape and size
of SIW cavities.
In particular, one can observe from the Tables I to 3 that individual SIW
TErol / TE301 cavities
arc nlore than twice as wide as they are long. One can also observe that the
individual SIW
cavities are more than three times as wide as the width of the corresponding
via-free openings.
As for the size of the SIW cavities, for a Kd band application, the TElol /
TE301 cavities are
preferably 8mm to 14mm wide, the TEioI / TE201 cavities are between 5mm to 8mm
wide, with
the total length of the entire chain of four cavities being in the range of
16mm to 22mm. The
size of the cavities may vary and depends on the dielectric constant of the
substrate material
used.
The filters 60, 70, and 80 are preferably manufactured in a PCB having linear
arrays of
metalized via holes with a diameter of 0.5mm and a center-to-center pitch of
1mm, although
other pitch dimensions that are fine enough to prevent radiation losses may be
used. For the
PCB, a 20mi1 thick RT/DuroidTM 6002 or 20mil thick RT/Duroid 5880 PCB material
may be
used. Botll materials are supplied by Rogers Corp.. having headquarters in
Rogers, CT, USA. In
thcory, the unloaded quality factor Qu of an SIW resonator based on 20mil
thick Rogers
RT/Duroid 5880 is about 500, while the Qu of an SIW resonator based on 20mi1
thick Rogers
RT/Duroid 6002 is only about 350. Hence, the RT/Duroid 5880 substrate is
expected to be
beneficial from the insertion loss standpoint. In reference to Eq. (2) above,
both Qd and Qc of an
SIW cavity made of RT/Duroid 5880 are higher than Qd and Qc of an SIW cavity
made of
RT/Duroid 6002. The Qd is higher because of a lower loss tangent tan(D). The
Qc is higher for
the RT/Duroid 5880 because of larger cavity dimensions, due to a lower
dielectric constant as
compared to Rogers RT/Duroid 6002.
14
CA 02660553 2009-03-27
Doc No: 102-41 CA Patent
Both abovementioned Rogers substrates use a similar fabrication process and
have a
similar fabrication cost. However, RT/Duroid 6002 has better mechanical
properties than
RT/Duroid 5880. The RT/Duroid 6002 material is suitable for laser drilling,
and via holes of a
wide i-ange of diameters can be drilled by this method. The RT/Duroid 5880
material must be
mechanically drilled, and mechanical drilling generally has a lower degree of
precision than laser
drilling. The better suitability for machining of the RT/Duroid 6002 material
makes it preferable
over the RT/Duroid 5880 material, even though the 5880 material has a better
electrical
performance as explained above. The filters 60, 70, and 80 were designed and
fabricated using
20mi1 thick Rogers RT/Duroid 6002 material.
Turning now to FIG. 16, spectral plots of simulated and measured insertion
loss of the
SIW filter 70 of FIG. 7 are presented. A variation of the dielectric constant
of the substrate and a
fabrication error led to a slight frequency shift of about 1.5% between the
simulated and the
measured responses. The measured minimum in-band insertion loss is
approximately 0.9dB,
which is slightly higher than the simulated loss of 0.75dB due to the
additional loss of a 90
microsti-ip bend, not sliown, and an additional section of microstrip line,
not shown. There is a
maximum variation of about 0.6dB in the insertion loss across the passband.
The attenuation in
the frequency band of 25.3GHz - 31.7GHz is better than 40dB, while in the
transmission (Tx)
band of 29.5GHz - 30 GHz it is better than 58dB. There is a spike around
31.7GHz due to
higher-order resonances of the TE201 mode and TE301 mode.
Referring now to FIG. 17, spectral plots of simulated and measured insertion
loss of the
SIW filter 80 of FIG. 8 are presented. Similar to the spectral plot of Fig.
16, a slight frequency
shift of about 1.3% between the simulated and measured responses occurs due to
the variation of
the dielectric constant of the substrate, as well as due to fabrication
tolerances. The measured
inininium in-band insertion loss is around 0.8dB, which is very close to the
simulated loss of
0.77dB. The attenuation in the frequency band of 23.94GHz - 31.48GHz is better
than 40dB,
while in the Tx band of 29.5GHz - 30GHz it is better than 52dB. There is a
spike around
31.6GHz due to the higher-order resonances of the TEZoI mode and TE301 mode.
