DC/DC POWER CONVERTER
FIELD OF THE INVENTION
The present invention relates to DC/DC converters, more particularly, to a
DC/DC
power converter.
BACKGROUND OF THE INVENTION
DC/DC converters typically receive an input voltage from a voltage source and
produce a DC output voltage at their output. Input voltages provided by those
voltage sources may vary widely. Furthermore, the desired DC output voltage
may
be variable over a certain range rather than fixed to a certain value, in
accordance
with the particular application of the converter. For example, the DC output
voltage
may range from 250V to 410V (e.g. for Lithium-ion battery charger). Therefore,
it is
generally desired for DC/DC converters to provide output regulation over wide
load
and line variations, while preserving high efficiency.
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SUMMARY OF THE INVENTION
In a first aspect of the present invention, there is provided a DC/DC power
converter
comprising: input terminals being connected to receive an input voltage;
a pulse wave generator comprising of a plurality of switches being arranged in
a
bridge configuration, to generate a pulse wave voltage from the input voltage;
a transformer comprising a primary winding, a secondary winding and a
magnetizing
inductance in parallel with the primary winding;
a DC blocking capacitor connected in series with said primary winding, and
their
series connection is coupled to receive the pulse wave voltage;
at least one resonant inductor connected in series with the secondary winding;
a filter capacitor connected in series with the series connection of the
secondary
winding and the at least one resonant inductor;
a rectifier being connected in series between the filter capacitor and the
series
connection of the secondary winding and the at least one resonant inductor;
a resonant capacitor being either connected in parallel with the rectifier, or
connected in parallel with the series connection of the rectifier and the
filter;
output terminals being connected across the filter capacitor; and
a control unit configured to: (i) operate the plurality of switches at a
constant
frequency with complementary duty cycles D and (1-D), under ZVS condition,
such
that the duty cycle of the pulse wave voltage is D; and (ii) vary the duty
cycle D
within a duty cycle range in which the fall transitions of the pulse wave
voltage occur
while the rectifier is conducting and the rise transitions of the pulse wave
voltage
occur while a reverse voltage across the rectifier is resonantly falling from
a ringing
peak, thus causing an output power across the output terminals to responsively
vary
in positive correlation.
In some embodiments of the first aspect of the present invention, the at least
one
resonant inductor is integrated with the transformer.
In some embodiments of the first aspect of the present invention, the
rectifier is a
switch being operated as a synchronous rectifier.
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In some embodiments of the first aspect of the present invention, the
rectifier is a
diode.
In a second aspect of the present invention, there is provided a DC/DC power
converter comprising: input terminals being connected to receive an input
voltage;
a pulse wave generator comprising of a plurality of switches being arranged in
a
bridge configuration, to generate a pulse wave voltage from the input voltage;
a transformer comprising a primary winding, a secondary winding and a
magnetizing
inductance in parallel with the primary winding;
at least one resonant inductor connected in series with the primary winding;
a DC blocking capacitor connected in series the series connection of the
primary
winding and the at least one resonant inductor, and the series connection of
the
DC blocking capacitor, the primary winding and the at least one resonant
inductor is
coupled to receive the pulse wave voltage;
a filter capacitor connected in series with the secondary winding;
a rectifier being connected in parallel with the series connection of the
filter
capacitor and the secondary winding;
a resonant capacitor being either connected in parallel with the rectifier, or
connected in parallel with the transformer;
output terminals being connected across the filter capacitor; and
a control unit configured to: (i) operate the plurality of switches at a
constant
frequency with complementary duty cycles D and (1-D), under ZVS condition,
such
that the duty cycle of the pulse wave voltage is D; and (ii) vary the duty
cycle D
within a duty cycle range in which the fall transitions of the pulse wave
voltage occur
while the rectifier is conducting and the rise transitions of the pulse wave
voltage
occur while a reverse voltage across the rectifier is resonantly falling from
a ringing
peak, thus causing an output power across the output terminals to responsively
vary
in positive correlation.
In some embodiments of the second aspect of the present invention, the at
least
one resonant inductor is integrated with the transformer.
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In some embodiments of the second aspect of the present invention, the
rectifier is
a switch being operated as a synchronous rectifier.
In some embodiments of the second aspect of the present invention, the
rectifier is
a diode.
In a third aspect of the present invention, there is provided a DC/DC power
converter comprising: input terminals being connected to receive an input
voltage;
a pulse wave generator comprising of a plurality of switches being arranged in
a
bridge configuration, to generate a pulse wave voltage from the input voltage;
a transformer comprising a primary winding, a secondary winding and a
magnetizing
inductance in parallel with the primary winding;
at least one resonant inductor connected in series with the primary winding;
a DC blocking capacitor connected in series the series connection of the
primary
winding and the at least one resonant inductor, and the series connection of
the
DC blocking capacitor, the primary winding and the at least one resonant
inductor is
coupled to receive the pulse wave voltage;
at least one further inductor connected in series with the secondary winding;
a filter capacitor connected in series with the series connection of the
secondary
winding and the at least one further inductor;
a rectifier being connected in series between the filter capacitor and the
series
connection of the secondary winding and the at least one further inductor;
a resonant capacitor being either connected in parallel with the rectifier, or
connected in parallel with the series connection of the rectifier and the
filter;
output terminals being connected across the filter capacitor; and
a control unit configured to: (i) operate the plurality of switches at a
constant
frequency with complementary duty cycles D and (1-D), under ZVS condition,
such
that the duty cycle of the pulse wave voltage is D; and (ii) vary the duty
cycle D
within a duty cycle range in which the fall transitions of the pulse wave
voltage occur
while the rectifier is conducting and the rise transitions of the pulse wave
voltage
occur while a reverse voltage across the rectifier is resonantly falling from
a ringing
peak, thus causing an output power across the output terminals to responsively
vary
in positive correlation.
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In some embodiments of the third aspect of the present invention, the
rectifier is a
switch being operated as a synchronous rectifier.
In some embodiments of the third aspect of the present invention, the
rectifier is a
diode.
In some embodiments of the third aspect of the present invention, the DC/DC
power
converter further includes a first doubling diode, a second doubling diode and
a
doubling capacitor; the first doubling diode is connected in series between
the
rectifier and the second doubling diode; the series connection of the first
doubling
diode and the second doubling diode being connected in parallel with the
output
terminals; the doubling capacitor is connected in parallel with the series
connection
of the first doubling diode and the rectifier.
In some embodiments of the third aspect of the present invention, the pulse
wave
generator is a half bridge inverter.
In some embodiments of the third aspect of the present invention, the pulse
wave
generator is a full bridge inverter.
BRIEF DESCRIPTION OF THE DRAWINGS
The invention may be more clearly understood upon reading of the following
detailed description of non-limiting exemplary embodiments thereof, with
reference
to the following drawings, in which:
Fig. 1 is a circuit diagram of a DC/DC power converter in accordance with
embodiments of the present invention;
Fig. 2 shows an electrical-equivalent circuit of a pair of coupled inductors;
Figs. 3-5 are circuit diagrams of DC/DC power converters in accordance with
embodiments of the present invention;
Fig. 6 is a circuit diagram of a DC/DC power converter with an exemplary
feedback
loop circuit for providing regulation, according to embodiments of the present
invention;
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Fig. 7 shows timing diagrams illustrating drive signals transmitted by a
control unit
and a pulse wave voltage generated by a pulse wave generator;
Fig. 8 is a circuit diagram of a DC/DC power converter having a full bridge
inverter
as a pulse wave generator, according to embodiments of the present invention;
Fig. 9 is a circuit diagram of a DC/DC power converter having two DC blocking
capacitors, according to embodiments of the present invention;
Fig. 10 shows timing diagrams illustrating an operation mode of the DC/DC
power
converter according to embodiments of the present invention;
Fig. 11-13 illustrate the limits of the operation mode;
Fig. 14 illustrates curves representing thresholds for satisfying ZVS of
switches in
the pulse wave generator;
Fig. 15 is a circuit diagram of a DC/DC power converter with a voltage
doubling
circuit, according to embodiments of the present invention; and
Fig. 16 shows timing diagrams illustrating the operation of a DC/DC power
converter
with a voltage doubling circuit, according to embodiments of the present
invention.
The following detailed description of embodiments of the invention refers to
the
accompanying drawings referred to above. Dimensions of elements and waveforms
shown in the figures are chosen for convenience or clarity of presentation and
are
not necessarily shown to scale or indicate component specification values.
Wherever possible, the same reference numbers will be used throughout the
drawings and the following description to refer to same and like elements.
DESCRIPTION OF THE PREFERRED EMBODIMENTS
The terminology used herein is for the purpose of describing particular
embodiments
only and is not intended to be limiting of the disclosure. Although particular
features
may be described with reference to one or more particular embodiments and/or
drawings, it should be understood that such features are not limited to usage
in the
one or more particular embodiments or drawings with reference to which they
are
described, unless expressly specified otherwise. It will be also understood
that the
use of any and all examples, or exemplary language (e.g., "such as") provided
herein, does not limit the scope of the disclosure.
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It will be further understood that, although the terms first, second, etc. may
be used
herein to describe various elements, these elements should not be limited by
these
terms. These terms are only used to distinguish one element from another. Any
description, of certain embodiments as "preferred" embodiments, and other
recitation of embodiments, features, or ranges as being preferred, or
suggestion that
such are preferred, is not deemed to be limiting the scope of the disclosure.
The
abstract is submitted with the understanding that it will not be used to
interpret or
limit the scope of the disclosure.
As used herein, the singular forms "a", "an" and "the" are intended to include
the
plural forms as well, unless the context clearly indicates otherwise. As used
herein,
the terms "comprising", "having", "including", and "containing" are to be
construed as
open-ended terms. As used herein the term "and/or" includes any and all
combinations of one or more of the associated listed items.
When two components are referred to as being "connected", it means that there
are
no components electrically between the components, the insertion of which
materially affect the function or functions provided by the DC/DC power
converter.
For example, two components can be referred to as being connected, even though
they may have a negligible DC impedance (e.g. sense resistor, current
transformer
etc.) between them which does not materially affect the function or functions
provided by the device; likewise, two components can be referred to as being
connected, even though they may have an additional electrical component
between
them which allows the DC/DC power converter to perform an additional function,
while not materially affecting the function or functions provided by a DC/DC
power
converter which is identical except for not including the additional
component;
similarly, two components which are directly connected to each other, or which
are
directly connected to opposite ends of a wire or a trace on a circuit board or
another
medium, are connected. In contrast, when two components are referred to as
being
"coupled", it means that the two components are either connected, or that
there is at
least one passive component intervening between the two components.
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The term "capacitor" as used herein should be understood broadly as any
component, including one or more elements, with a capacitive property. As
such,
the term "capacitor" may be used to refer to a lumped capacitive element
and/or to a
distributed capacitive element. Similarly, the term "inductor" as used herein
should
be understood broadly as any component, including one or more elements, with
an
inductive property. As such, the term "inductor" may be used to refer to a
lumped
inductive element and/or to a distributed inductive element.
Fig. 1 shows a circuit of a DC/DC power converter, having input terminals 18
for
receiving an input voltage Vin, and output terminals 20 for outputting a DC
output
voltage Vo. A DC output current lo may be drawn, if an output load (not shown)
is
connected across output terminals 20. Thus, the output power of the DC/DC
power
converter is the product of the DC output voltage Vo and the DC output current
lo.
Typically being variable, the input voltage Vin may be received, for example,
from a
power factor correction stage. Other sources for the input voltage Vin
include, but
not limited to, a battery voltage, a solar cell voltage etc.
A pulse wave generator 10 is connected to input terminals 18 and includes a
plurality of switches that are arranged in a bridge configuration, for
generating a
pulse wave voltage. Typically, pulse wave generator 10 can be a half bridge
inverter, which includes a first switch 24 and a second switch 26. Switches 24
and
26 may have diodes 25 and 27 placed across them, respectively. If switches 24
and
26 are MOSFETs (as shown in Fig. 1) then respective diodes 25 and 27 represent
integral body diodes. Capacitors 29 and 30 represent inherent stray
capacitances of
switches 24 and 26, respectively. External snubber capacitors (not shown) may
be
added across switch 24 and switch 26 to reduce their turn-off switching
losses.
A control unit 12 transmits drive signals Va and Vb to switches 24 and 26,
respectively. Control unit 12 may include a PWM (Pulse Width Modulation)
controller, which can be an analog and/or digital controller (for example
ASIC,
FPGA, etc.). As can be seen from Fig. 7, the drive signals Va and Vb are non-
overlapping and complementary of each other, such that switch 24 is operated
with
a duty cycle D while switch 26 is operated with a duty cycle (1-D).
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The drive signals Va and Vb do not necessarily have exact duty cycles D and (1-
D),
respectively. Between the fall transition of Vb and the rise transition of Va
there can
be a slight dead time for allowing diode 25 to conduct. Similarly, between the
fall
transition of Va and the rise transition of Vb there can be a slight dead time
for
allowing diode 27 to conduct. Thus, switches 24 and 26 switch on and off in a
complementary manner, generating a pulse wave voltage Vg across switch 26.
Assuming that switches 24 and 26 are ideal and that the forward voltages of
diodes
25 and 27 are negligible, the pulse wave voltage Vg equals to the input
voltage Vin
when switch 24 is on and switch 26 is off, and equals to zero volts when vice
versa.
Thus, the pulse wave voltage Vg is a pulse wave voltage waveform with a period
T,
a duty cycle D and a peak-to-peak amplitude that is approximately equal to the
input
voltage Vin. As will be shown later, pulse wave generator 10 can also be a
full
bridge inverter, which can be configured to generate a pulse wave voltage with
identical period T and identical duty cycle D, but with a peak-to-peak
amplitude that
is approximately equal to twice the input voltage Vin. As a consequence, a
generated current Ig is being drawn from pulse wave generator 10.
In a typical operation of switches 24 and 26, the period T is fixed by control
unit 12,
causing the instantaneous frequency of pulse wave generator 10 to be constant.
Nonetheless, control unit 12 may apply a frequency spreading operation, under
which the instantaneous frequency of pulse wave generator 10 varies within a
constant frequency range. Thus, the expression "constant frequency" as used
herein with regards to the plurality of switches, means either fixing the
period T, or
applying frequency spreading operation. Due to the internal resistances of
switches
24 and 26, forward voltages of diode 25 and diode 27 and/or external
impedances
that may be present, the peak-to-peak amplitude of the pulse wave voltage Vg
may
deviate (within 1%) from being exactly equal to the input voltage Vin, and
thus the
expression "approximately equal" as used herein should be understood to
encompass such deviation. Furthermore, the total combined rise and fall time
of the
pulse wave voltage Vg may be prolonged up to 50% of the period T, due to the
stray
capacitances of switches 24 and 26 and/or external snubber capacitors that may
be
present, and thus it should be understood that the term "pulse wave voltage"
as
used herein is intended to encompass such prolonging.
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Referring to Fig. 1, the DC/DC power converter further includes a DC blocking
capacitor 28, a transformer 32, a magnetizing inductor 38, a resonant inductor
48, a
resonant capacitor 46, a rectifier 44 and a filter capacitor 56. Transformer
32
includes a primary winding 34 and a secondary winding 36, which are
magnetically
coupled to each other. The turn ratio of transformer 32, denoted by N, is the
ratio of
the number of turns of primary winding 34 to the number of turns of secondary
winding 36. Although Fig. 1 shows that resonant inductor 48 is connected in
series
with secondary winding 36, resonant inductor 48 can be connected in series
with
primary winding 34. Thus, resonant inductor 48 can be an external component or
implemented as the leakage inductance of transformer 32. Furthermore,
additional
resonant inductors may be connected in series with secondary winding 36.
Although Fig. 1 shows that resonant capacitor 46 is connected in parallel with
the
series connection of filter capacitor 56 and rectifier 44, resonant capacitor
46 can be
connected in parallel with rectifier 44. Thus, resonant capacitor 46 can be an
external component or implemented as the inherent stray capacitance of
rectifier 44.
DC blocking capacitor 28 is coupled in series with primary winding 34 to block
the
DC component of the pulse wave voltage Vg from transformer 32. Magnetizing
inductor 38 is connected in parallel with primary winding 34 and can be an
external
inductor or implemented as the magnetizing inductance of transformer 32.
The operation of rectifier 44 is similar to that in a class E low dv/dt
resonant rectifier
circuit. During off-state of rectifier 44, resonant capacitor 46 resonates
with resonant
inductor 48. At the turn-on transition of rectifier 44, the initial current
through it rises
in a step change. During conduction of rectifier 44, the voltage across
resonant
capacitor 46 is substantially constant. Minor ringing may occur due to the
stray
inductance of rectifier 44 during its conduction. The turn-off transition of
rectifier 44
is at low dv/dt and under zero current switching (ZCS) condition. Although
Fig. 1
shows a diode serving as a rectifier 44, a switch such as a MOSFET can be
operated in a synchronous rectification for serving as rectifier 44. Filter
capacitor 56
is connected in series with rectifier 44 and connected in parallel with output
terminals 20, to smooth the voltage ripple of the DC output voltage Vo.
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Fig. 10 shows waveforms illustrating the operation of the DC/DC power
converter
according to embodiments of Fig. 1. The first (top) waveform of Fig. 10 shows
the
pulse wave voltage Vg across switch 26. The second and third waveforms of Fig.
10
show the currents la and lb passing through switches 24 and 26, respectively.
The
fourth waveform of Fig. 10 shows the generated current Ig that is drawn from
pulse
wave generator 10. The fifth waveform of Fig. 10 represents the current Id
passing
through rectifier 44. The sixth (bottom) waveform of Fig. 10 represents the
reverse
voltage Vdr across rectifier 44.
The operation mode of the DC/DC power converter according to embodiments of
Fig. 1 will now be described with reference to Fig. 10. The turn-off
transition of
rectifier 44 initiates a half resonant interval, denoted by T. During this
half resonant
interval, resonant capacitor 46 discharges and the reverse voltage Vdr across
rectifier 44 rises resonantly. At the end of the half resonant interval, the
reverse
voltage Vdr reaches a ringing peak and starts to fall resonantly, as resonant
capacitor 46 starts to recharge. While the reverse voltage Vdr is strictly and
resonantly falling from its ringing peak, the pulse wave voltage Vg rises to
the level
of Vin, triggering faster charging of resonant capacitor 46 until the turn-on
transition
of rectifier 44. Assuming that the forward voltage of rectifier 44 is
negligible, the
reverse voltage Vdr is equal to zero volts during conduction interval of
rectifier 44.
While rectifier 44 conducts, the pulse wave voltage Vg falls back to zero
volts,
triggering the fall of the current Id until the turn-off of rectifier 44.
As can be seen from Fig. 10, the reverse voltage Vdr across rectifier 44 is
characterized by a ringing peak, caused by the resonance of resonant inductor
48
with resonant capacitor 46. In contrast, a full bridge rectifier typically
clamps the
peak level of the reverse voltage across the rectifying elements to the level
of the
output voltage. Nonetheless, the full bridge and/or other rectifier circuits
can be
additionally incorporated with tertiary windings, for providing an auxiliary
power. This
auxiliary power can be used, for example, to activate control unit 12 and/or
switches
24 and 26. If provided, such auxiliary is comparably negligible to the output
power
that rectifier 44 provides, and thus have no effect on the ringing peak of the
reverse
voltage Vdr.
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As can also be observed from Fig. 10, the near-linear waveform of the
generated
current Ig during conduction of rectifier 44 is attributed to DC blocking
capacitor 28,
which decouples the DC component of the pulse voltage Vg without participating
in
resonance. Thus, for embodiments of Fig. 1, the half resonant interval t and a
characteristic impedance Z can be given by:
T c== TC X IX/71,1-
Z ,'"-- VL,./C,
Lr is the inductance of resonant inductor 48 and Cr is the capacitance of
resonant
capacitor 46. Further defined herein are three variables a, (3 and G:
OC = Z10/V0
13 = 2T/T
G = Vo/Vir,
where G is in fact, the voltage gain of the DC/DC power converter. Fig. 11
illustrates
the boundaries of the operation mode by plotting curves of the voltage gain G
as a
function of the duty cycle D. For =0.5 and N=1, each curve in Fig. 11
represents a
different value of a. At the points that are marked with circle, the fall
transitions of
the pulse wave voltage Vg occur just when rectifier 44 turns off (as shown in
box
200). At the points that are marked with X, the rise transitions of the pulse
wave
voltage Vg occur just when the reverse voltage Vdr starts to fall (as shown in
box
202). At the points that are marked with square, the rise transitions of the
pulse
wave voltage Vg occur just when the reverse voltage Vdr ends to fall (as shown
in
box 204). The dashed parts of the curves in Fig. 11 represent operating
outside of
the operation mode. Fig. 12 and Fig. 13 illustrate the limits of the operation
mode for
p=0.4 and 13=0.6, respectively. As evident from Fig. 11, Fig. 12 and Fig. 13,
either
the DC output voltage Vo or the DC output current lo can be regulated against
fluctuations of the output load and/or the input voltage Vin, by configuring
control
unit 12 to vary the duty cycle D within the duty cycle range in which the
operation
mode applies.
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As shown in Fig. 11-13, the output power can reach a maximum when:
1 ¨ (VT) > D > 0.5
By decreasing the duty cycle D toward zero, the output power can be reduced.
Therefore, the output power can effectively vary in positive correlation with
the duty
cycle D. That is, the output power can increase when the duty cycle D
increases
and vice versa. Upon regulating the DC output voltage Vo, two noticeable
ramifications can be derived from this effect: (i) as the output power varies,
the peak
level of the reverse voltage Vdr also varies in positive correlation with the
output
power; and (ii) as the input voltage Vin varies and the output power remains
constant, the peak level of the reverse voltage Vdr also remains substantially
constant (within 10%), regardless of the duty cycle D. The peak level of the
reverse
voltage Vdr can be given by the following equation (only for embodiments of
Fig. 1):
Vdr(max) (2DVin/N) 2V0
As shown in Fig. 10, at the fall transitions of the pulse wave voltage Vg,
rectifier 44
is conducting, the generated current Ig is positive and the sum of currents
(Ig+Im) is
positive; At the rise transitions of the pulse wave voltage Vg, the generated
current
Ig is negative, the current through resonant capacitor 46 is positive and the
sum of
currents (Ig+Im) is positive. Both switches 24 and 26 have conditions for
their
operation under ZVS, however, the condition for ZVS of switch 24 is harder to
satisfy. For illustrating the condition for ZVS operation, further two
variables y and 5
are defined herein:
y = GN
TVoN2
= _____________
27cLmIo
Lm is the inductance of magnetizing inductor 38. Fig. 14 illustrates curves of
threshold values of 5 as a function of a, for satisfying ZVS condition of
switches 24
and 26. The curves of Fig. 14, each representing different value of y, refer
only to
embodiments of Fig. 1, and are only of approximation (within 20%).
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The following empirical formula (only for embodiments of Fig. 1) is provided
herein
for satisfying ZVS condition of switch 24:
T> (PmaxLm) v ( 45
NVinVo 8c ¨ 0.4)
E = _______________________________________________
(3V0 /2 Pmax) + [Lin I I (LrN2)1/CbN4
Cb is the capacitance of DC blocking capacitor 28 and Pmax is the maximum
output
power. In consideration that most magnetic cores have a finite magnetic energy
storage capability, the inductance Lm of magnetizing inductor 38 is preferably
small
enough, such that its magnetic core doesn't fully saturates. The capacitance
of DC
blocking capacitor 28 may affect ZVS operation. If the ripple voltage (RMS)
across
DC blocking capacitor 28 is higher than the RMS value of the pulse wave
voltage
Vg, then ZVS operation of switches 24 and 26 may not be possible under all
load
and line conditions. Incorporation of snubber capacitors across switch 24 or
switch
26 may have an undesired influence on ZVS operation.
The efficiency of the DC/DC power converter is influenced by the value of p.
The
closer 0 is to 1, the higher the voltage gain and the circulating currents
from the
resonance of resonant capacitor 46 and resonant inductor 48. The closer 0 is
to 0,
the higher the DC bias of magnetizing inductor 38, and the higher the voltage
stress
on rectifier 44. Therefore, a trade-off can be made between components sizes
of the
DC/DC power converter. Best mode of operation can be obtained when 0.2<p<0.9.
Referring now to Fig. 6, shown is a feedback loop circuit for providing
regulation.
The feedback loop circuit may include an error amplifier 35 and a signal
isolator 37.
Error amplifier 35 can produce an error signal that is based on the difference
between a reference voltage Vref and the DC output voltage Vo. The reference
voltage Vref can be generated, for example, by an LDO regulator. The error
signal
can be either isolated by signal isolator 37 before being fed to the control
unit 12, or
fed directly to control unit 12. Signal isolator 37 can then be placed between
control
unit 12 and pulse wave generator 10, to isolate the drive signals Va and Vb.
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Signal isolator 37 can be, for example, a signal transformer or an opto-
coupler.
Control unit 12 may include a triangle wave oscillator, a comparator and a
digital
inverter. The triangle wave oscillator can feed a triangle wave voltage into
one input
of the comparator, while the error signal can be fed into the other input of
the
comparator. As a result, the comparator can output a PWM signal. The PWM
signal
can be inverted by the digital inverter. Thus, the drive signals Va and Vb can
be
received from the PWM signal and the inverted PWM signal, respectively. The
upper and lower limits on the excursion of the duty cycle D can be set, for
example,
by biasing and scaling the triangle wave voltage accordingly.
If control unit 12 is a digital controller, the feedback loop circuit may
include an
analog-to-digital converter (not shown) for converting the error signal into a
digital
signal, which then can be fed to control unit 12. For regulation of the DC
output
current lo, a sense resistor (not shown) may be connected in series with the
output
load. The voltage that is produced on the sense resistor can be fed to error
amplifier
35. Frequency spreading can be implemented, for example, by incorporating a
variable frequency oscillator, which can be used in conjunction with the
comparator
to provide such pulse frequency modulation. Dead time for ZVS operation can be
set, for example, by feeding the drive signals Va and Vb through RCD networks.
An example according to embodiments of Fig. 1 will now be provided herein:
DC output voltage: 12 V
Maximum output power: 300 W
Minimum input voltage: 400 V
Maximum input voltage: 480 V
Frequency of pulse wave generator: 200 kHz
DC blocking capacitor: 1 p,F
Magnetizing inductor: 200 pH
Resonant inductor: 380 nH
Resonant capacitor: 420 nF
Turn ratio of the transformer: 16:1
Filter capacitor: 1 mF
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In the above example, the DC output voltage Vo can be regulated against
fluctuations of the output power at the minimum input voltage, by varying the
duty
cycle D from 0.46 at 75 W, up to 0.6 at 300 W; While operating at the maximum
output power, the DC output voltage Vo can be regulated against fluctuations
of the
input voltage by varying the duty cycle D from 0.6 at 400 V, down to 0.42 at
480 V.
Fig. 2 shows an equivalent model 100 which is the electrical-equivalent
circuit of a
pair of coupled inductors. Equivalent model 100 comprises a parallel
inductance L ,
a primary series inductance Li, a secondary series inductance L2 and an ideal
transformer having a turn ratio M. The two equations (in frequency domain)
relating
terminal voltages (V1 and V2) and terminal currents (11 and 12) are given by:
L Li
= s X X
1V21 LLig M 112
+ L2
(m2)
Since these two equations contain four variables (LI, L2, L ,, and M), it is
possible to
derive an infinite number of specific equivalent models from equivalent model
100.
Thus, the embodiments of Fig. 1 realize a specific equivalent model, in which
the
primary series inductance L1 is set to zero.
Fig. 3 shows a DC/DC power converter according to embodiments, in which
resonant inductor 48 is connected in series with primary winding 34.
Additional
resonant inductors may be connected in series with primary winding 34.The
embodiments of Fig. 3 realize another specific equivalent model, in which the
secondary series inductance L2 is set to zero and thus can have the same
terminal
voltages and terminal currents as those of the embodiments of Fig. 1. For the
embodiments of Fig. 3, the half resonant interval T and the characteristic
impedance
Z can be given by:
T IC X VCr(Lmill-ir)/N2
Z V(LmilLr)/C N2
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Fig. 4 shows a DC/DC power converter according to embodiments, in which a
further resonant inductor 49 is connected in series with primary winding 34.
Additional resonant inductors may be connected in series with primary winding
34
and/or secondary winding 34.The embodiments of Fig. 4 realize another specific
equivalent model in which both the primary series inductance L1 and the
secondary
series inductance L2 are non-zero. Thus, the embodiments of Fig. 4 can have
the
same terminal voltages and terminal currents as those of the embodiment of
Fig. 1.
For the embodiments of Fig. 4, the half resonant interval T and the
characteristic
impedance Z can be given by:
t =--- TE X V[Cr(Lmi I La)/N21 + CrLr
Z ''''' V [(Lm 1 1 La)/CrN2i + Lr/C,
La is the inductance of further resonant inductor 49, which can be implemented
as
an external component or as the leakage inductance of transformer 32. Fig. 5
shows
a DC/DC power converter according to embodiments, in which resonant inductor
48
is connected in series with primary winding 34 and resonant capacitor 46 is
connected in parallel with primary winding 34. Thus, for the embodiments of
Fig. 5,
the half resonant interval 'I and the characteristic impedance Z can be given
by:
t r-z TC X A/Cr (Lm I ILO
Z -,-,- -ALA I Lr)/Cr
Fig. 8 shows a DC/DC power converter according to embodiments, with a full
bridge
inverter serving as pulse wave generator 10. The full bridge inverter
comprises
further two switches 64 and 66, in addition to switches 24 and 26. Switches 64
and
66 may have diodes 65 and 67 placed across them, respectively. Capacitors 69
and
70 are the respective inherent stray capacitances of switches 64 and 66.
Switch 66
turns on and off synchronously with switch 24, while switch 64 turns on and
off
synchronously with switch 26. Thus, pulse wave generator 10 can generate a
pulse
wave voltage with a peak-to-peak amplitude that is approximately twice as that
of
the pulse wave voltage Vg.
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Fig. 9 shows a DC/DC power converter according to embodiments, having an
additional DC blocking capacitor 31 being connected in series with DC blocking
capacitor 28, while their series combination is connected in parallel with
input
terminals 18.
Fig. 15 shows a DC/DC power converter according to embodiments, which further
includes a voltage doubling circuit, formed by a first doubling diode 51, a
second
doubling diode 52 and a doubling capacitor 53. Doubling capacitor 53 resonates
with resonant inductor 48 when rectifier 44 is at off-state. Thus, the voltage
doubling
circuit provides further rectification, lowers the DC magnetization of
magnetizing
inductor 38 and reduces the peak level of the reverse voltage Vdr. In
alternative
configurations, additional snubber capacitors (not shown) can be also
connected in
parallel with first doubling diode 51 or in parallel with second doubling
diode 52, thus
either partially or wholly forming the capacitance of resonant capacitor 46.
Alternatively, a voltage multiplier circuit can be included, by further
cascading
voltage doubling circuits.
Fig. 16 shows waveforms that illustrate the operation of the DC/DC power
converter
of Fig. 15 at maximum output power. The first (top) waveform of Fig. 16 shows
the
pulse wave voltage Vg across switch 26. The second waveform of Fig. 16 shows
the
generated current lg. The third waveform of Fig. 16 shows the current Id
passing
through rectifier 44. The fourth waveform of Fig. 16 shows the reverse voltage
Vdr
across rectifier 44. The fifth waveform of Fig. 16 shows the current lw
passing
through doubling capacitor 53. The sixth waveform of Fig. 16 shows the current
Icr
passing through resonant capacitor 46. The seventh (bottom) waveform of Fig.
16
shows the reverse voltage Vw across second doubling diode 52.
Although the present invention may have been described in terms of the present
embodiments, it will be understood that various modifications thereof are
possible
within the principles outlined above and will be evident to those skilled in
the art, and
thus the invention is not limited to the preferred embodiments but is intended
to
encompass such modifications.
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