Note: Descriptions are shown in the official language in which they were submitted.
78929
This invention relates to a digital regenerator and more
particularly to one which provides improved noise immunity (such as to
near-end crosstalk) in the transmission of partial-response signals.
Background of the Invention
In a typical digital transmission system, a digital signal
is transmitted along a transmission path such as a pair of wires in a
telephone cable having digital repeaters periodically interposed which
regenerate the signal. Due to the proximity of the wires in the cable,
crosstalk interference is generated of which the three most significant
types are: near-end crosstalk (NEXT), far-end crosstalk (FEXT) and
near-end-near-end interaction crosstalk (NENEIXT). Of these three the
most disturbing is generally NEXT which results when high-level
regenerated signals are coupled directly into the paths carrying low-level
signals in the opposite direction. An interesting discussion on crosstalk
interference can be found in a paper entitled: "Engineering TlC Carrier
Systems" by J.P. Fitzimmons and W.J. Mayback, Conference Record, ICC 75,
June 16-18 San Francisco, Vol.III, pp.39-5 to 39-9.
In many applications, binary data is transmitted over the
cable pairs using ternary partial-response signals such as bipolar,
duobinary and modified duobinary signals. References to such systems
are "Correlative Level Coding For Binary-Data Transmission" by Adam Lender,
IEEE Spectrum, February 1966, pp.104-115, "Transmission Systems For
Communications" 4th Edition by Bell Telephone Laboratories, Inc.,
pp.666-673, and "Principles of Data Communication" by R.W. Lucky,
J. Salz and E.J. Weldon Jr., McGraw-Hill 1968, pp.83-92.
The basic form of a digital repeater used to regenerate
such a transmitted signal is discussed in an article entitled: "A New
3.152Mb/s Digital Repeater" by A. Anuff et al, Conference Record, ICC 75,
June 16-18 San Francisco, Vol.3, pp.39-10 to 39-13. In such a repeater,
the signal is first passed through an equalizer which compensates for the
transfer characteristics of the telephone cable. A regenerator then
1~78g~9
reconstructs the signal in its original form for transmission along the
following section of the transmission line.
In many systems, the interfering power density of the noise
varies with frequency. For instance in cable circuits, it is well known
that NEXT is frequency dependent and usually increases with frequency
at 4.5dB/octave rate; ibid BTL text, pp.286-288. To minimize the NEXT
effect, the bandwidth of the equalizer should be as narrow as possible.
However, the response of the equalizer has in the past been such as to
minimize intersymbol interference thereby restricting its minimum
bandwidth. This has frequently led to the use of a raised-cosine
transmission characteristic, ibid BTL text, pp.715-718. In other systems,
the power density of the interfering noise may decrease with frequency
such as when it is coupled from an adjacent power line. In still other
systems, the noise appears only at the band edges. An example of this
is the noise which results from inter-channel interference in a digital
radio system.
Statement of the Invention
It has been discovered that particular types of noise
interference having a power density which varies with frequency, can be
decreased by introducing a controlled amount of intersymbol interference
in a band limiting filter prior to regeneration of the partial-response
signal without increasing the number of signal levels. After regeneration,
an inverse function filter can restore the signal to its original form so
that the digital repeater may appear transparent to the transmission line.
Thus, the present invention provides an improved regenerating
circuit, such as a digital repeater for partial-response signals subject
to noise interference having a power density which varies with frequency.
The regenerating circuit generally comprises an equalizer for equalizing
the transfer characteristics of the incoming transmission line and a
regenerator for regenerating the partial-response signals. The improvement
comprises a filter located prior to the regenerator to introduce
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intersymbol interference so as to transform the partial-response signal
to a modified partial-response signal having not more than the same
number of levels as the received partial-response signal. The system
parameters are such that the filter also reduces the total power of the
noise interference thereby reducing the error rate of the regenerated
signal. If the circuit is functioning as a digital repeater, it will
generally include a second filter located after the regenerator and
having an inverse characteristic to the first filter so as to restore
the signal to its original form. However, the second filter may not be
required in all instances, e.g. such as where the signal is demodulated
directly from its modified form in an appending receiver. Since the
number of signal levels of the modified partial-response signal is the
same or fewer than that of the received signal, the filters may be
readily added to an existing system with little or no modifications
thereto.
In order to achieve better noise immunity, the first
filter response must be such as to decrease the overall noise power.
Thus, where the power density of the noise increases with frequency
(e.g. NEXT or FEXT), the initial filter will in general have a high
frequency roll-off characteristic such as obtained from a cosine filter.
Conversely, where the power density of the noise decreases with frequency
(e.g. power line interference), the initial f~lter will usually have a
low frequency roll-off characteristic such as obtained from a sine filter.
Since the initial filter response is dictated by the type of transmitted
signal, use of the invention to obtain an improvement is limited to
particular types of interference in each case. In all cases, flat or
white noise will not benefit from use of the filter, since it will not
reduce the overall noise power relative to that of the signal.
To obtain a modified ternary partial-response signal at
the output of the first filter, the product R(j~), of the partial-response
signal's frequency response S(j~), and the transfer function of the first
-- 3 --
1~)7~g;~9
filter H(j~), ;s equal to:
R(j~) = S(j~) Htj~)
1 -jN~iT (1)
where:S(j~) = N ~ 1 ~-jn(~iT ~ a~)
n = o
H~ iT + a~)]
T = pulse repetition period of the partial-response
signal,
iT = unit delay of each filter tap,
N = total number of filter taps,
a = O or 1. ~ -
As stated earlier the invention must be selectively applied
in order to obtain improved noise immunity. One important application is
to the reduction of NEXT in a cable transmission system using bipolar
(or alternate mark inversion (AMI)) signals. However in such a system
the improved noise immunity to NEXT will not accrue if the partial-response
signal is converted to a binary signal prior to regeneration in the
repeater as the required filter would only further emphasize the higher
frequency signal energy, as discussed in a paper entitled: "On The
Application Of Some Digital Sequences To Communication" by Jack K. Wolf,
IEEE Trans. Communications Systems, December 1963, pp.422-427.
During transmission, a code violation introduced into the
system can cause a series of errors to be generated at the output of the
second filter. In a particular embodiment of the invention, additional
circuitry is included in the second filter to restore the signal to its
correct form in response to the detection of preselected pulse sequences
in the repeater.
In many applications,ternary partial-response, e.g. bipolar,
duobinary and modified duobinary signals, are transmitted. In these
lV78929
embodiments of the invention, a ternary signal is maintained throughout
the repeater even though content of the signal is transformed from one
type to another due to the intersymbol interference which is first
introduced and later removed by the two filters.
Brief Description of the Drawings
Example embodiments of the invention will now be described
with reference to the accompanying drawings in which:
Figure 1 is a block schematic diagram of a digital repeater;
Figures 2, 3 and 4 illustrate schematically the basic form
of various filters used in the digital repeater of Figure 1 for bipolar,
duobinary and modified duobinary signals, respectively;
Figures 5 and 6 illustrate the effect of the filter shown
in Figure 2, on the transmission response in a bipolar transmission
system having a raised-cosine transmission characteristic;
Figure 7 illustrates the reduction in noise achieved
utilizing the filter shown in Figure 2 for the bipolar transmission
system having a raised-cosine transmission characteristic; --
Figure 8 is a schematic diagram of a digital version of
a filter for transforming a modified ternary signal back to its original
bipolar form for use in ~he digital repeater of Figure l;
Figure 9 is the truth table for the filters illustrated
in Figures 2 and 8,
Figure 10 is an example of how the introduction of a
single error into the pulse stream can cause a series of errors to be
produced at the output of the digital repeater;
Figure 11 illustrates the effect of preselected input
sequences on the output of the digital filter illustrated in Figure 12; and
Figure 12 is an expanded version of the digital filter
found in Figure 8, having additional circuitry for correcting for digital
errors introduced in the transmission system.
~078~29
Description of the Preferred _bodiments
Figure 1 illustrates a digital repeater, which is used to
regenerate a partial-response signal from a transmission line. In a
typical application, the transmission line comprises a pair of w;res 10
in a telephone cable which is driven by a signal such as one having a
Tl or TlC format, as discussed in a paper entitled: "The TlC System"
by J.A. Lombardy et al, Conference Record, ICC 75, June 16-18,
San Francisco, Vol.III, pp.39-1 to 39-4. While various multi-level
partial-response signals can be transmitted, one common type is a ternary
level bipolar or AMI signal.
Due to the proximity of the wires 10 to others in the
telephone cable, various forms of crosstalk are introduced. In addition,
the transfer characteristics of the cable pair 10 cause a deterioration
in the amplitude and shape of the transmitted pulse signals. Consequently
in the repeater, the signal is first coupled from an input transformer 11
to an equalizer 12 prior to regeneration. As is well known, the function
of the equalizer is to equalize the transfer characteristics of the
cable pair 10. In the prior art, the signal is then coupled to a
regenerator 13 which is synchronized by a clock signal derived from a
clock recovery circuit 14. Again in the prior art, the output from the
regenerator 13 is fed through an output transformer 15 to an on-going
section 16 of the transmission line. These circuits are described in
considerable detail in the above-mentioned article by A. Anuff et al.
The crux of the present invention is the inclusion of
two filters, having inverse characteristics from each other, before and
after the regenerator 13. As shown in Figure 1, a first filter 20 is
interposed between the equalizer 12 and the regenerator 13, to introduce
a controlled amount of intersymbol interference so as to derive at its
output a modified partial-response signal. After regeneration, the
modified partial-response signal is then transformed back to its original
form by a second filter 21 having an inverse characteristic to the first.
-- 6 --
1()78929
In applications such as digital terminals, where it is not necessary to
replicate the input signal, the second filter 21 may be eliminated and
the signal demodulated directly.
In the following description, elements performing the same
function as previously described elements are given the same base number
with an added reference character to distinguish between them. In
addition corresponding reference characters are used to locate the pulse
sequences, shown in Figures 9, 10 and 11, in the schematic drawings.
Figure 2 illustrates two basic filter circuits 20A and 21A, for use with
10 a bipolar signali Figure 3 filters 20B and 21B for a duobinary signal;
and Figure 4 filters 20C and 21C for a modified duobinary signal. In
the first two cases, the incoming signals to the filters 20A and 20B
are delayed by an interval T (which is equal to the pulse repetition
period of the partial-response signal) before being summed as shown;
while in the third case the signal to the filter 20C is delayed by an
interval of 2T. Similar delays are found in the inverse filters 21A, 21B
and 21C. In all three cases, the signal remains at a ternary level
throughout even though transformed by the filters. In the case of the
bipolar and duobinary signals, they are both transformed to a modified
20 duobinary form. Functionally each of the filters consists of a delay
network 23 and an adder 22 which either sums the signals directly or
differentially as indicated, in a well known manner. The actual
realization of the filters 20 may however be linear such as achieved
using passive components, or digital such as achieved using logic elements.
On the other hand the filters 21 will in general be realized using digital
techniques otherwise the circuit will tend to oscillate due to the positive
feedback required to achieve the transfer functions.
As stated earlier, the total noise power in any one
system will be a function of the type of interference and required
30 equalization, thereby limiting any noise improvement achieved by inclusion
of the filters 20 and 21 to differing types of interference in different
A~`~
lV7~929
systems. In one such system, the noise improvement achieYed by the
inclusion of the filters 20A and 21A in the digital repeater can better
be illustrated by reference to Figures 5, 6 and 7 which illustrate typical
results achieved for a bipolar signal. Figures 5 and 6 illustrate the
response of two channels having different raised-cosine amplitude
characteristics. In both figures, the dashed line shows the noise
introduced by NEXT increasing at the well known 4.5dB/octave rate. In
Figure 5, the equalized channel response has a 100% raised-cosine
amplitude characteristic, ~ = 1, while in Figure 6 the raised-cosine
amplitude characteristic is 50%, ~ = 0.5 (ibid BTL text p.717).
In both cases, the general transfer function of
equation (3) for the filter 20A reduces to:
H(j~) = (1 _ f-i~T~ = 2 cos ~ f-(i~T/2)) (4)
which results in a null at a frequency f = 21T as shown by the broken lines.
In both figures, R is the combined response of the equalized channel and
the filter 20A, thus:
R f( ) 2 ~ T/2) ~ ( ~
It can be seen that in both instances, the resultant R falls below that of
the equalized channels (i.e. curves ~ = 1 and ~ = 0.5) between the
frequencies 31T and 32T which results in less noise contribution o~er this
portion of the frequency spectrum. Outside this band however, there is an
increased noise contribution. If the noise contributed by crosstalk were
flat with frequency, the inclusion of the filter 20A would have a
detrimental affect on the overall system performance. However, because of
the noise contributed by NEXT which rises at a 4.5dB/octave rate (as shown
in Figures 5 and 6) and of the high frequency gain in the equalizer 12,
the overall result is that an improvement in noise immunity to NEXT can be
obtained by introduction of the filter, for most raised-cosine characteristics
of the channel response. While not shown in Figures 5 and 6, the response
of filter 21A is an inverse characteristic to that of filter 20A thus:
1()7~929
H l(j~) = 1/(1 ~-j~T) = -sec ~ T/2)) (6)
Figure 7 illustrates the relative NEXT noise for differing
raised-cosine channel response characteristics before and after the
filter 20A. The graph shows that the prefilter NEXT noise is a minimum
at ~ ~ 0.22. For values of < 0.92, some noise reduction can be obtained
as a result of the introduction of the filter, ranging up to 9dB as ~ ~ 0.
The small post-filter increase in noise when a > 0.92 results from the very
high contribution of NEXT noise introduced at frequencies greater than 32T
as shown in Figure 5. However, with lower values of ~, the channel ~ -
response characteristic is such that the higher frequency noise is cut off
as shown in Figure 6. Note in Figure 5, R is significantly greater than
between ~2T ' f ' T~ while in Figure 6, R is only marginally greater
than ~ between 32T ' f ' 43T. In practice, an < 0.5 is not readily
achieved and consequently a realistic reduction in crosstalk noise is
in the order of 5.5dB. However, due to the nature of a digital signal,
this causes a marked decrease in the error rate of the regenerated signal.
The filters illustrated in Figures 2, 3 and 4 will provide
a perfect replica of the signal at the output of the digital repeater
providing no code violations develop in the regenerated signal. However,
such violations occur in a practical system and are due to such factors
as noise, distortion and start-up conditions. Due to the intersymbol
interference generated by the filters a single code violation can result
in a series of errors being transmitted further along the system. In
addition to this, the positive and negative pulses of a ternary level
signal are usually generated in separate channels which are then combined
at the output as illustrated in the above-mentioned paper by A.Anuff et al.
A digital realization of the equivalent filter 21A shown
in Figure 2, which uses a split output from the regenerator 13, is
illustrated in Figure 8 in conjunction with the truth table of Figure 9.
This example is for the case where the filters have a delay T
(i.e. N = 2, equation (2)). The bipolar signal which is transformed
lV7B~29
by the first filter 20A to a modified duobinary signal B, is split into
+ and - digits at the output of the regenerator 13D, and fed to the input
of the filter 21D. The basic filter 21D comprises two OR gates 30A and
30B, two AND gates 31A and 31B, two flip-flops 32A and 32B, and an
exclusive OR gate 33 connected as shown. The flip-flops 32A and 32B
provide the delay period T. The outputs are combined in the primary
of the transformer 15D to provide a reconstructed bipolar signal for
transmission along the line 16D.
Figure 9 com~oares the truth table for both filters 21A
10 (Fig. 2) and 21D (Fig. 8), and except for the invalid words, the outputs
D and Z are identical. With filter 21A the introduction of an error ~;
during start-up or by the regenerator 13 results in an invalid output
(i.e. 2+ or 2-). However, with filter 21D, an error at one of the
inputs Q or R results in a continuous stream of errors at the output Z
until a + - or - + sequence occurs at the output of the regenerator.
An example of this is shown in Figure 10 where it is
evident from the bipolar signal patterns that a series of errors at
output Z will be transmitted as a result of a single error occurring
in either pulse streams S' or T' ( in this example the error is in T').
20 While the initial error cannot be corrected, the number of such errors
which occur before the correct bipolar signal is again reestablished,
can be limited by the inclusion of additional circuitry which forces
the output to a particular value whenever certain input sequences occur.
This is achieved for several different input sequences of a bipolar
signal in the circuit shown in Figure 12, which is an alternate embodiment
of the circuit shown in Figure 8. Here in addition to the basic elements
described in the previous embodiment bearing identical reference numerals,
the circuit of Figure 12 includes two logic networks 40A and 40B each
consisting of two flip-flops 41, 42 and a NAND gate 43. It can be
30 readily shown that the unique sequence X + + or X - - occurs at the
output B of the first filter 20A whenever the bipolar input signal A
- 10 -
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is O + O or O - O respectively, where X is a different level than
the latter two levels of each sequence. This results in the sequence
of pulses shown in Figure 11 at the indicated points in the filters
20A and 21E. Such a sequence can be used to advantage to correct any
errors which may have been introduced by forcing a O at output Z from
the repeater, simultaneously with the last digit of either of these
sequences. This is achieved by the networks 40A or 40B, which in
response to detection of these sequences ;nhibit the AND gates 44A
and 44B thereby forcing a O at the output Z.
A second property of the bipolar signal A is that when
there are several consecutive logical O's or l's the input energy must be
low or high respectively. This information can be used to add an
additional correction factor to the digital signal. AND gates 50A and 50B
in conjunction with OR gates 51A and 51B and flip-flop 52 provide an output
whenever there are three consecutive logical O's or l's respectively just
prior to the output Z. To determine the energy level of the incoming
bipolar signal A, it is rectified and filtered in full-wave rectifier 60
and R-C filter 61. When the signal energy level is very low there is an
output from NOR gate 62. If however, there is a simultaneous output from
AND gate 50B, it indicates there is an error in the output signal of the
digital repeater. The simultaneous outputs from gates 50B and 62 produces
a logical O at the output of NAND gate 63 which in turn, by way of AND
gates 44A and 44B, forces a O at the output Z of the digital repeater.
Similarly, when the incoming signal energy level is high an output is
obtained from OR gate 64 driven from the resistive divider 70. Again
however, if there is a simultaneous output from AND gate 50A, an error is
indicated which then produces an output from AND gate 65 thereby forcing
a - pulse at the output Z of the digital repeater through OR gate 66. It
is to be noted that the bipolar sequence will be reestablished by the
insertion of a + or - pulse at the output. Hence the OR gate 65 can be
placed in series with either of the outputs of AND gates 44A or 44B.
While the concept of forcing a particular output pulse,
1 1
7~g29
in response to the detection of particular sequences of pulses in the
repeater, is shown only for the bipolar case, it is evident that similar
techniques cou1d be applied to other partial-response signals and in
particular ternary level signals such as duobinary and modified
duobinary signals.
