Note: Descriptions are shown in the official language in which they were submitted.
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The present invention relates to radio frequency
antenna structures and more specifically to resonant micro-
strip radiator elements.
In general, microstrip radiators are specially shaped
S and dimensioned conductive surfaces formed on one surface of
a planar dielectric substrate, the other surface of such sub-
strate having formed hereon a further conductive surface com-
monly termed the "ground plane~. Microstrip radiators are
typically formed, either singly or in an array, by conventional
photoetching processes from a dielectric sheet laminated be-
tween two conductive sheets. The planar dimensions of the
radiating element are chosen such that one dimension is on
the order of a predetermined portion of the wavelength of a
predetermined frequency signal within the dielectric sub-
strate, and the thickness of the dielectric substrate chosento be a small fraction of the wavelength. A resonant cavity
is thus formed between the radiating element and ground plane,
with the edges of the radiating element in the non-resonant
dimension defining radiating slot apertures between the
;20 radiating element edge and the underlying ground plane sur-
face.
A dilemma arises in the prior art with respect to
constraints on the minimum size of antenna elements. By
definition, the effective resonant dimension of the resonant
cavity, defined by the radiating element (commonly called the
"E-plane dimension") must be approximately a predetermined
portion of a wavelength of the operating frequency signal in
the dielectric. The prior art has generally attempted to
reduce the size of the antenna elements by utilizing substrates
with hi~h dielectric constants to, in effect, reduce the
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wavelength of the resonant frequency within the dielectric
substrate and thereby allow for a smaller resonant dimension.
Such an approach, however, is disadvantageous in that the use
of a high dielectric substrate increases the loss conductance
of the cavity and results in a larger non-resonant dimension,
as will be explained, or significantly lower efficiency of
the antenna or both.
The non-resonant dimension, commonly termed the "H-
plane dimension", is determined in major part by the beam
width and efficiency of he antenna. The efficiency of the
antenna is typically expressed as a ratio of the power actu-
ally radiated to the power input, where the power input is
(neglecting any reflected components) substantially equal to
the sum o~ the power radiated and the power loss through heat
dissipation in the dielectric. The equivalent circuit of the
antenna element, with respect to power dissipation, may be
expressed as a parallel combination of a radiation resistance
and a dielectric loss resistance where the radiation and
dielectric loss resistances are respectively defined as the
resistances which, when placed in series with the antenna
element, would dissipate the same amount of power as actually
radiated by the element and as dissipated by the dielectric,
respectively. The radiation power and dielectric loss are
thus inversely proportional to the respective values of the
radiation and loss resistances. The radiation resistance,
however, is inversely proportional to the non-resonant dimen-
sîon of the element. For a given dielectric, a required
efficiency therefore prescribes the minimum non-resonant
`; ~ dimension of the element. Thus, conflicting criteria for
reducing the respectlve dimensions of an antenna element
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existed in the prior art, in that the required effective
resonant dimension of the element is determined by the wave-
length of the resonant fxequency signal in the dielectric
- and substrates having high dielectric constant to reduce such
wavelength typically present a low loss resistance, requiring,
therefore, a wider non-resonant dimension.
It should be appreciated that minimum size constraints
can cause significant problems in applications where a large
multiplicity of radiating elements are required, but limited
space is available for antenna area, for example, a communica-
tion system antenna for use on an astronaut's backpack.
The present invention provides an antenna assembly
for operation at a predetermined frequency comprising: a
first conductive element having a plurality of projecting
sections; a second conductive element having a plurality of
projecting sections; the projecting sections of said first
and second conducting elements being disposed in an inter-
digitated manner, and being interspaced with a dielectric
material, said first and second conductive elements defining
a resonant cavity between the projecting sectians thereof
such that said resonant cavity is of a length no longer than
approximately one wavelength of a signal at said predetermined
frequency in said dielectric material and equal to approxi-
mately a predetermined multiple of one-quarter of a wavelength
2~ of said signal of said predetermined frequency în said
dielectric material.
The present invention further provides an antenna
assembly for operation at a predetermined frequency, of the
type including a resonant cavity defined between at least
two conductive sheets also defining a radiating aperture
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between the edge of at least one of the sheets and the re-
maining sheet, said resonant cavity having a longitudinal
resonant dimension approximately equal to a predetermined
. multiple of one-quarter of a wavelength at said predetermined
frequency, said cavity resonant dimension being no larger
than one wavelength of said predetermined frequency and where-
in said resonant cavity is folded along at least one axis
. transverse to said resonant dimension such that the longi-
tudinal size of said antenna is reduced.
The present invention-further provides an antenna
assembly for operation at a predetermined frequency of the
type including first and second conductive sheets; said con-
ductive sheets being disposed substantially in parallel and
being separated by a dielectric material, ~aid conductive
; 15 sheets defining a resonant cavity therebetween having radiat-
ing slot apertures longitudinally spaced apart at a predeter-
mined distance approximately e~ual to one-half wavelength of
said predetermined frequency, wherein said conductive sheets
and said dielectric material are folded along at least one
~axis parallel to at least one of said radiating apertures,
folding thereby said resonant cavity such that the longitudinal
dimension of said antenna assembly is reduced.
: The present invention further provides an antenna assem-
bly of the type including a first conductive element separated
;~ 25 from a second conductive element by a dielectric material, said
first and second conductive elements deflning a half-wave reson-
ant cavity therebetween and at least one radiating aperture, and
wherein said first and second conductive elements each comprise
a pIurality of conductive sheets interconnected by at least one
further conductive sheet, the plurality of conductive sheets of
first and second
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conductive elements being disposed alternately in an over-
laying manner, separated by said dielectric material.
A description of the preferred embodiment follows
with reference to the accompanying drawing, wherein like
numerals denote like elements, and:
Figure 1 is a perspective view of a microstrip
radiating element with narrowed non-resonant dimension in
accordance with one aspect of the present invention.
Figures 2 and 3, respectively, are sectional and
perspective vie~s of a folded microstrip radiating element
in accordance with another aspect of the present invention;
Figure 4 is a sectional view of an interdigitated
antenna structure utilizing standoffs; and
Figure 5 shows a microstrip radiator adapted to
radiate circularly polarlzed signals.
With reference to Figure 1, a planar conductive
radiating element 10 is insulated from a conductive ground
plane 12, disposed parallel thereto, by a dielectric sub-
strate 14. Signals of a predetermined operating frequency
are applied to radiating element 10 and ground plane 12, for
example, by a coaxial cable 16. Coaxial cable 16 is prefer-
ably coupled to radiating element 10 at a point 18 where the
; impedance of element 10 matches the impedance (typically 50
ohms) of the cable. Radiating element 10 is generally rec-
tangular, having planar dimensions such that one set of edges
20 and 22 defined a resonant dimension approximately equal
to one-half of the wavelength of the predetermined frequency
signal in dielectric substrate 14, for example, 0,45 of the
free space wavelength of the signal. Dielectric substrate 14
is à fraction of a wavelength, for example, 0.002 times the
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free space wavelength of the resonant frequency. A resonant
cavity is formed between radiating element 10 and gro~md
plane 12 with radiation emanating from radiating aperture
slots 28 and 30 formed between edges 24 and 26 and ground
plane 12.
Dielectric substrate 14 is preferably a low density,
low loss expanded dielectric substance such as a honecombed
or foamed structure. Briefly, such expanded dielectric
comprises, in substantial portion, voids to provide a rigid,
low weight, low density, low loss structure. Expanded dielec-
trics, however, typically present a lower dielectric constant
than non-expanded dielectric substrates, such as teflon-
fiberglass typically used in the prior art. Thus, use of an
expanded dielectric generally requires an elongation of the
effective resonant dimension. Ilowever, the present inventors
have discovered that the loss resistance of such expanded
dielectric substrate is far greater than the loss resistance
of non-expanded dielectric substrate, providing for a reduc-
tion in the minimum non-resonant dimension, substantially
exceeding the increase in the resonant dimension required due
to decreased dielectric constant. For example, the non-
resonant dimension can be chosen to be 0.1 times the free
space wavelength of the applied signal, as compared with 0.3-
0.9 times free space wavelength typical for the prior art.
Thus, in accordance with one aspect of the present invention,
a radiating element of reduced planar area can be constructed
by utilizing an expanded dielectric substrate, and narrowing
the non-resonant dimension. For example, a radiator of
given efficiency utilizing a *Teflon-*Fiberglass substrate is
0.15 times the square of the free space wavelength, while a
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typical radiating element of such efficiency utilizing an ex-
panded dielectric substrate and narrowed non-resonant dimen-
sion in accordance with the present invention is 0.05 times
the square of the free space wavelength,-a reduction in area
by a factor on the order o~ 3.
The planar area of a radiating element can be further
reduced in accordance with the present invention by, in effect
folding the resonant cavity. For example, the cavity can be
folded along one or more axes perpendicular to the resonant
~dimension to create a tiered or layered structure. Alter-
nately, a reduction in the planar size of the resonant cavity
can be effected by folding or bending the microstrip into,
for example, a "V" or "U" shape. Figures 2 and 3 depict an
antenna wherein an interdigitated structure is utilized to
effect a folded resonant cavity. Referring to Figures 2 and
3, generally ground plane 12 includes a pluralitv of longi-
tudinally disposed planar conductive sheet sections 31-35
electrically connected by vertical side members 36 and 38.
Radiating element 10 comprises a plurality of generally planar,
longitudinally disposed conductive sheets 40-42 disposed in
an interdigitated manner with respect to ground plane sections
31-35 separated therefrom by dielectric 14, and electrically
connected by a vertical member 44, disposed parallel to side
members 36 and 38. Apertures 28 and 30 are defined by the
vertical most edges of radiating element 10. The cumulative
distance from aperture 28 to aperture 30, through dielectric
14, is approximately equal to one-half wavelength of the
operative frequency within the dielectric. Thus, radiating
element 10 and ground plane 12 define a resonant cavity hav-
ing radiating slot apertures 28 and 30 defined by edges 24
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and 26 of radiating element 10 on opposite longitudinal sides
of the antenna structure.
Such an interdigitated structure is, in effect, a
planar microstrip element, for example such as shown in
Figure 1, folded from each end toward the middle, then folded
again back toward the end along axes perpendicular to the
¦ resonant dimension and parallel to radiating apertures 28 and
30, such folding sequence repeated four times to provide a
! five tiered structure. It should be appreciated that inter-
digitated structures may be utilized to provide resonant
cavities folded along a greater or lesser num~er of axes,
with axes not necessarily parallel to the radiating aperture
nor perpendicular to the resonant dimension. While it is not
necessary, it is preferred that an odd number of tiers be
effected such that the apertures are on opposite longitudinal
sides of-the antenna structure.
An input signal is applied to the radiating element
via coaxial cable 16, with the center conductor connected to
radiating element 10 at a point 18 of appropria-te impedance.
While cable 16 is shown coupled through the side of the
antenna element in Figures 2 and 3, it should be appreciated
that connection can be made in any appropriate manner such as,
for example, through the bottom of ground plane 12 or from
the resonant dimension side.
The planar length (L) of a five-tiered interdigitated
structure, such as shown in Figures 2 add 3 having a non
resonant dimension W on the order of 0.1 times the free space
wavelength of the operating frequency, is also on the order
of 0.1 times he free space wavelength, as opposed to 0.45
times the free space wavelength typical in a non-folded
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structure such as shown in Figure 1. The height or thickness
(H) of the interdigitated structure is on the order of 0.01
times the free space wavelength, as opposed to 0.002 times
the free space wavelength in the unfolded element.
It should be appreciated that, while Figures 2-and 3
show an interdigitated structure wherein both of the side
members are formed by ground plane 12, folded resonant cavi- -
ties can be effected by interdigitated structures wherein one
or both of the side members are formed by radiating element
- 10, and by interdigitated structures wherein a plurality of
vertically disposed conductive elements are connected by
longitudinally disposed members. Further, the conductive
sheets need not be planar, but can be curved, nor need all
the conductive sheets be of the same planar size. Moreover,
the spacing between sheets need not be uniform or constant.
It should be appreciated that dielectric 14 can com-
prise a void with radi~ting element 10 being isolated from
ground plane 12 by standoffs. Such a structure is shown in
Figure 4. Non-conductive standoffs 46 and 48 are disposed
between ground plane element 35 and radiator element 40, to
effect spatial separation between ground plane 12 and radiat-
ing element 10. The conductive sheets of radiator 10 and
ground plane 12, in an embodiment utilizing standoffs, must
be rigid enough to maintain the interdigitated separation.
Where a solid or honeycombed or otherwise expanded dielectric
is used, the conductive sheets can be extremely thin, with
the dielectric providing structural support.
The interdigitated structure depicted in Figures 2
and 3 is particularly advantageous in the generation of
3Q circular or elliptically polarized signals. Circular or
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elliptical polarization is generated utilizing a flat radiat-
ing element by applying e~ual amplitude signals, 90 out of
¦ phase, to adjacent (intersecting), perpendicular edges of the
I element. Such a technique is not feasible for use with
~ 5 folded or interdigitated elements. To provide circular or
I elliptical polarization, two interdigitated or folded elements
, are, in effect, stacked and rotated with respect to each other
j by 90 as shown in--Figure 5. Quadrature signals, as generated
¦ by, for example, a quadrature hybrid 50, are applied to re-
- spective stacked elements 52 and 54 via coaxial cables 56 and
58. Due to a masking effect by the upper element, it was found
desirable to utilize cavities of approximately a half wave-
length, and that the cavities maintain two radiating apertures
on opposite sides of the element. It should be appreciated
that, where the coaxialcables are coupled through vertical
sides in the non~resonant dimension of the respective elements,
the coaxialcables can be routed straight downward without
interfering with the operation of radiating apertures 60-63.
The thickness (T) of such stacked elements are typically on
the order of 0.02 times the freespace wavelength of the operat-
ing frequency.
Radiating elements utilizing folded resonant cavities
- in accordance with the present invention have been built for
operational frequency of between 259.7 MHz to 296.8 MHz. The
elements constructed were interdigitated structures similar
to that shown in ~igures 2 and 3, and were stacked as shown
in Figure 5 to provide circular polarization. A radiation
pattern of -10 db gain was achieved over approximately 80%
spherical coverage. The physical package was 6" x 18" x 3"
and weighed less than 0.45 Kg. The conductive sheets were
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formed of aluminum 0.005-.020 inch thick. ~he sheets were
set in an interdigitated arrangement, furnace brazed and
then sealed with tin. The structure was then set in a mold
and the space between the conductive sheets filled with
s li~uid e-xpanding insulating resin. The resin hardened to
provide rigidity.
An interdigitated antenna structure has also been
constructed on alayer-by-layer approach, sandwiching a layer
of honeycomb material between conductive sheets.
A seven tiered interdigitated antenna structure
utilizing a dielectric comprising standoffs and a void has
also been constructed. The conductive sheets were formed of
brass on the order of 0.020 inch thick, and spacing between
the interdigitated elements was maintained at 0.1 inch by
transverse nylon screws running through the interdigitated
elements.
It should be appreciated that folded cavities in ac-
cordance with the present invention can also be of lengths
other than one half wavelength. For example, quarter-wave
cavities have been constructed with an appropriate impedance
termination (e.g. a short circuit~ in the cavity opposite
the radiating aperture. Similarly, a full wavelength
resonant cavity can be utilized. Other modifications of the
exemplary embodiment may also be apparent and are to be in-
cluded within the scope of the appended claims.
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