Note: Descriptions are shown in the official language in which they were submitted.
DIRECT CURRENT BALLASTIN~ AND STARTING
CIRCUITRY FOR GASEOUS DISCHAR~E LAM~S
Background Of The Invention
This invention relates in general to ballasting and
starting circuitry for operating gaseous discharge lamps from D.C,
and more particularly to such circuitry with a current-regulated
output characteristic wherein series-pass switching means is
alternatively switched between on and off conductive states and
to A.C. and D.C. power conversion circuits for supplying direct
current to a plurality of ballasting and starting circuits.
At the present time, use of A.C. power sources to power
gaseous discharge ~ub~s,especially those used in 1uorescent
lighting, by far exceed the use of D.C. power sources. This is
not particularly surprising because AoC~ power sources are usually
more readlly available than D.C. power sources. However, operation
of fluorescent lamps from A.C. power sources has a number of dis-
advantages. One of these problems is that fluorescent lamps generate
and radiate radio frequency interference (RFI). RFI is a form
of electro-magnetic radiation, which among other things is known
for interferring with the performance of communications systems,
e.g. radio and television. The RFI is generated because, as
the A.C. changes or reverses polarity during a portion of each
cycle, the arc between the electrodes of the gaseous discharge
tube extinguishes~ The tube must then be restarted for current
flow in the opposite direction and much of the RFI is generated
when the arc between the electrodes of the gaseous discharge tuhe
begins to restrike.
The constant polarity reversal of voltage and current
in an A.C.`power source also requires that heating be provided at
both electrodes of the gaseoas discharge tube. To condition a
gaseous discharge tube for the striking of an arc between the
electrodes, it is neces'sary to he'at the cathodic electrode to
facilitate electron emission. However, in an A~C. system, the
electrode of the tube'which is the' cathodic electrode is con-
stantly changing as the polarity of the voltage and current
change. This necessitates the heating of both terminals.
Because the arc between the terminals of the gaseous
discharge tube operating from an A.C. power source is constantly
being extinguished and then reignited, lighting from a fluores-
cent lamp is not continuous. Instead, the lamp actually flickers.This flickering phenomenom is not noticeable to the unaided eye
because the frequency of most A.C. power sources is somewhat above
a frequency level which is perceptible. Nonetheless, recent
behavorial and physiological studies have indicated that the inher-
ent flickering has undesireable side-effects. Behavior and
activity of children tending to be hyperactive are believed to
be aggravated by the flickering. The flickering is also believed
to hasten fatigue, which is a serious problem to medical per-
sonnel when attempting to differentiate between the various shadings '
of x-ray films.
The flickering phenomenom further causes stroboscopic
effects when any movement is related to a harmonic of the A.C.
power source frequency. This can present a safety hazard because
- the stroboscopic effects cause rotating machinery to appear to be
either stationary or slowly rotating. '`
Even when operatiny from an A.C. power source, ~luores-
cent lamps require circuitry to power and control the lamp because
of the unusual load characteristics of gaseous discharge tubes.
To achieve arcing between the electrodes of a gaseous discharge
tube, the str1king voltage of the tube must be exceeded. The
striking voltage is often twice the voltage at which the tube will
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operate once striking of the arc between the terminals of the tube
occurs~ Circuitry must be provided to generate a voltage pulse
of sufficient magnitude and duration to achieve striking. However,
when striking of the arc occurs, current to the tube must then be
limited. The current limiting function is often provided in A.C.
circuits by a high leakage reactance transformer, which may con-
stitute the bulk of the weight and expense in a fluorescent system.
Gaseous discharge tubes are not susceptible to voltage regulation
once arcing between the electrodes thereof is initiated because
of the negative impedance characteristic of the tube. The tube
will conduct an excessive amount of current to the point o~ sel~-
destruction Therefore the current to a
fluorescent lamp must be limited and the particular load charac-
teristic of the lamp will determine the voltage at which it operates
at a regulated current level. For a given current, the operating
voltage of the tube is a function of the lenyth of the tube, its
diameter, the types of gases within the tube and a number of other
factors.
Some prior art efforts have been concerned with operating
gaseous discharge tu~es in conjunction with D.C. ballasting and
starting circuits. These efforts have been generally centered
aro~md biasing a series pass semiconductor, usually a transistor,
such that the current therethrough is li~ited ~o the d~ d ~urre~t
through the gaseous discharge tube. TO compensate ~or a number
of variables in such circuit deslgn and the expected variations in
the D.C. power source, a relatively lar~e volta~e is usually drop-
ped across the series pass transistor. Hence,the series-pass
transistor must dissipate a significant amount of power. This power
is wasted energy and Ieads to a low eficiency of operation for
the circuit. The power diss1pation also requires the use of
larger and more expensive semiconductors for the series pass element.
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This further requires a heat sink to ~issipate the hea~ from ~he
transistor, and some inskances, forced air ventilation thereof.
Thus, such prior art ballasting and starting circuits have not
met with much acceptance or commercial success, except in quite
limited or specialized applications.
Summary Of The Invention
The direct current ballasting and starting circuitry
of the present invention employs a series-pass switching means
in one o a pair of input lines. The circuit provides a regulated
output current. The switching means or semiconductor is alter-
nately switched between on and off conductive states. During the
off state, no current is conducted through the series-pass semi- `~
conductor an~ during the on stage the voltage drop thereacross
is very small. The power lost in the series--pass semiconductor
is very minimal. Thus there is no need for massive heat sinks or `
~or large semiconductors capable of withstanding and dissipating
higher power losses, as in the prior art circuits. Means of
limiting the current conducted by the switching means is accom-
plished by current sensing means in series connection with switch-
ing means. Filtering mean~ are in series with the current sensing
means for smoothing pulses of energy delivered by the switching
means ~or the D.C. voltage source. The filter means has a direct
current output with a comparatively small alternating current
component thereon.
~ The starting circuitry is in se~ies connection ~etween the
filter means and an output terminal of the circuitry for sensing
the nonionized condition of a lamp connectible to the output
terminal. The starting circuitry further provides a voltage
pulse of sufficient magnitude and duration to initiate ionization
within the Iamp. Once ionization is achieved, the starting
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circuit becomes inactive and does not impede -the supply of
direct current from the filter means to the lamp.
Various means for controlling the on and off conduc-
tive states of the series-pass switching semiconductor may be
utilized. In one embodiment, an oscillator is momentarily res-
ponsive to excitation provided by the starting voltage pulse
at one of the output terminals and therea~ter interacts with
the switching means. The oscillator has a resonant tank circuit
for controlling the series pass semiconductor and the tank cir-
cuit is also phase-corrected to provide the series pass semi-
conductor with a 70 per cent duty cycle to insure that sufficient
power is available ~rom the circuit.
Another embodiment of the control means utilizes a
~ree-running multivibrator; the output of which is pulse-width
modulated to control the on-off states of the series-pass semi-
conductor.
Where a plurality of b~llasting and startiny circuits
are used, a single starting circuit may be used to generate the
striking voltage pulses for all of the ballasting circuits.
Secondary windings of pulse transformers are in series connection
with an output line o~ each ballasting circuit. Primary windings
are connected in series between an energy storage means and a
voltage responsive means. A diode from the output line of each
ballasting circuit is poled in a logic "or" configuration such
a nonionized condition in any lamp associate with its ballasting
circuit will charge the energy storage means to a voltage level
which wlll render the voltage responsive means conductive, thereby
discharging the energy storage means through the primary windings
of the pulse transformers and starting the desired lamp.
A power conversion circuit for converting A.C. voltage
to a suitable D.C. voltage for operation of the ballasting and
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starting circuitry is also disclosed. A pOrtion of a primary
win~ing of a transformer is tapped for applying the A.C. voltage
source thereacross. The entire primary winding is applied to
rectification means for rectifying the A.C. ~oltage and supply-
ing the same to a filter means. The filter means supplies an
elevated D.C. voltage level to a plurality of D.C. ballasting
and starting circuits. A secondary winding of the transformer
supplies a considerably lower A.C. voltage level -to the cathodic
electrode of the lamp associated with each ballasting and start-
ing circuit for heating the same. The secondary winding of thetransformer is center-tapped, with the center-tap referenced to
a second of the input lines of the ballasting and starting cir-
cuitry. Thus the cathode of each lamp is heated with a small
A.C. voltage which is balanced with respect to the second input
line thereby eliminating A.C. modulation of the D~Co current
supplied to each lamp.
Various other objects, features and advantages of the
invention will become apparent from the following detailed dis-
closure when taken in conjunction with the drawings.
Brief Description Of The Drawings
In the drawings:
Figure 1 is a schematic diagram of the A.C. to D~C~
power conversion circuit for supplying D.C. power to a plurality
of ballasting and starting circuits, each of which supplies
regulated direct current to a gaseous discharge tube;
Figure 2 is a schematic circuit diagram, mostly in
block form, of a D.C. ballasting and starting circuit as illus-
trated in Figure l;
Figure 3 is a schematic circuit diagram of the preferred
embodiment of a D.C. ballasting and starting circuit for supply-
ing power to a gaseous discharge tube;
Figure 4 is a schematic circuit diagram of a D.C.
ballasting circuit with an alternative embodiment oE con-
trolling the conductive state of a series-pass semiconductor;
Figure 5 is a schematic cixcuit diagram of an alter-
nate pulse-width modulation technique for controlling the con-
ductive state of the series-pass semiconductor;
Figure 6 is a schematic circuit diagram illustrating
an alternate embodiment of a starting circuit;
Figure 7 is a schematic circuit diagram illustrating
use of a single starting circuit in con unction with a plurality
of ballasting circuits.
Description Of The Preferred Emhodiments
Referring to Figure 1, there is shown an A.C. to D.C.
power conversion circuit, generally designated as 10. A single
circuit 10 is capable of supplying D.C. power of a suitable vol-
tage level to a plurality of D.C. ballasting and starting circuits
ll. As will be hereinafter presented, the ballasting and start-
ing circuit ll supplies regulated current to at least one gaseous
discharge tube or fluorescent lamp 12.
The power conversion circuit lO has a power transformer,
generally designated as 13, with a core 14 of iron or other suit-
able magnetic material. A primary winding 15 of the transformer
13 has a tap, at a point 16. A pair of leads ~7, 18, one of
which is connected to the tap at point 16, supply voltage from
an A.C. power source to a portion of the primary winding 15.
A pair of leads 19, 20 interconnect opposite ends of
the primary winding 15 to opposite ~erminals 21, 24 of a rectifi-
cation means, i.e. a diode rectification bridge 22. A fuse 23,
in series with the lead 19, protects the circuit 10 and the cir-
cuits ll from overload or malfunction which could result in ~
excessive current demand~ -
To minimize the amount of filtering required to filter
the rectified A.C. voltage, the diode bridge 22 is preferably of
the full-wave type. The diodes in the bridge 22 are poled to
provide a positive D. C. potential at a terminal 25 with respect
to a common terminal 26. The potential between the te~minals
25, 26 will typically be in the range of 145 to 190 volts D.C.
Connected in parallel across a pair of leads 27, 28, which are
respectively connected to the terminals 25, 26, is at least one
capacitor 29 for filtering the rectified A.C. vol-tage from the
bridge 22. Because of the magni~ude of D.C. voltage between
the leads 27, 28, it may be more economical to provide a plurality
of capacitors 29.
A bleeder resistor 30 is usually provided in parallel
with the capacitors 29 to reduce the voltage across the leads
27, 28 within a specified period of time a~ter A.C. voltage has
been removed from the leads 17, 18. The phosphor coating in
some fluorescent lamps 12 will continue to fluoresce at a reduced
level of illumination until the voltage across the lamp 12 is
insufficient to maintain arcing between an anodic electrode
(not shown) and cathodic electrode (not shown) of the lamp 12.
The bleeder resistor 30 insures that within a specified period
of time the vol-tage levels within the circuit 10 and hence the
circuit 11 will be reduced to a level at which the fluorescent
phenomenom will terminate.
The positive D.C. voltage lead 27 is connected to an
input line 32 o~ each o~ the ballasting and starting circuits 11.
l'he common lead 28 is similarly connected to a second input line
33 of the circuits 11~ An ou~put terminal 34 of each of the
circuits 11 provides suitable power, sensing and controlling
~unctions to power and control at least one fluorescent lamp 12
of the circuits 11, as is hereinafter described.
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The power transformer 13 also has a secondary ~7inding
35 with an A~C. voltage thereacross which is considerably lo~es
in magnitude than that across the primary winding 15. The
secondary winding 35 is center-tapped and connected by a lead
36 to the common line 28 of the circuit 10 and to the terminal
26 of the diode bridge 22. A pair of leads 38, 39 are connected
to opposite ends o~ the secondary winding 35 and to terminals
40, 41 to heat the cathodic electrode within the lamp 12 to pro-
vi.de electron emission therefrom. Because the secondary winding
35 is center~tapped and referenced by a lead 36 to the common
line 2g, the average voltage on the cathodic electrode of any
lamp 12 will be zero. Thus, ~.C. voltage modulation of the
potential across any lamp 12 is minimized or eliminated. Use of
a center-tapped secondary winding 35 avoids the need for supply-
ing a small D.C. voltage ~or heating ~he cathodic electrode in
the lamps 12, which would re~uire additional com~onents such as
rectifying diodes and filtering capacitors to eliminate ripple.
The secondary winding 35, while providiny an A.C.
voltage for heating the tube 12 also ~orms part of the retun
~a path for the D.C. current for the lamps 12. D.C. current
supplied by the ballasting and starting circuitry 11 at the
output terminal 34 flows through the tube-12 and returns to the
power conversion circuit 10 through both of the leads 38j 39,
the secondary winding 35, and the lead 36 to the-terminal 26 of
the diode bridge 22.
As previously noted, in operating fluorescent lamps
12 from an A.C. power source, both ends of the lamp lZ must be
heated because the cathodic electrode in khe lamp 12 changes as
the polarity of the voltaye and current in the lamp 12 reverse.
3Q However, in operating a lamp 12 from a D.C. power SOurGe, the
polarity o the voltage and current applied to the lamp 12
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emains constant. Thus, only the end of -the lamp 12 which is
to be the cathodic end needs to be heated. As illustrated in
Figure 1, the output terminal 34 of the ballastiny and starting
circuit 11 need only be applied to one terminal on the anodic
end of the lamp 12 ~o provide electrical connection there-to.
Turning now to Figure 2, there is shown a circuit dia-
gram, mostL~ in block form, of one of the D.C. ballasting and
starting circuits 11 of Figure 1. A pair of input lines 32, 33
supplies a source of D.C. voltage to the circui~ ll. An output
terminal 34 supplies power to the gaseous discharge tube 12, and
controls and senses the condition of the gaseous discharge tube.
In series with one of the input lines 32 is switching means in
the form of a switching series-pass transistor 44. The switching
transistor 44 is alternately switchable between on and off con-
ductive states for periodically supplying pulses of energy from
the sources of D.C. voltage. An oscillator circuit 45 in comhina-
tion with base drive 46 provides a means of controlling the con--
ductive state of switching transistor 44. Current sensing means
47 limits the maximum current through the transistor 44. One
means of limiting current through the transistor 44 is by divert-
ing the base drive 46 therefrom, to immediately switch the trans-
istor 44 to an off conductive state. Filter means 55 includes an
inductor 48 and a capacitor 49. The inductor 48 receives pulses
of energy from the switching transistor 44 and in combination with ~ -
the capacitor 49 filters the pulses of energy into direct current
with a small alternating current component, in the form of ripple,
superimpos-ed on the direct current at a junction 50. The cap-
acitor 49 is connected between the junction 50 and the second
input line 33. A commutating diode 51 is connected between the
second input 33 and a junction 52 with the diode 51 poled to main-
tain current continuity in the inductor 43 when the switching
transistor 44 is in an off conductive state.
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A starting pulse circuit 53 is biassed between the
output terminal 34 and the second input 33. Because the ballast-
ing and starting circuit 11 does not regulate voltage at the
output terminal 34, but only regulates current deliverable thereto,
the potential at the output terminal 34 will rise to a level
similar to that at the input line 32 when the circuit 11 is first
energized. The starting pulse circuit 53 senses this higher
voltage level as indicative of a nonionized or nonconductive
condition of a lamp connectible to the ou~put terminal 34 and
thereupon generates a voltage pulse of sufficient magnitude and
duration to initiate ionization in a gaseous discharge lamp.
Once striking of the arc within the lamp has occurred, the vol-
tage at the output terminal 34 drops and the output voltage is
controlled and determined by the load characteristics of the
particular gaseous discharge tube connected thereto. The circuit
11 will then begin operating in a current-regulated output mode.
As the potential at the terminal 34 drops, the startiny pulse
circuit 53 will become inactive. However, should some occurrence
cause a loss of arcing within the lamp, the above starting pro
cess will automatically repeat.
The oscillator 45 is connected by a lead 54 to the
output terminal 34 such that the starting voltage pulse initially
excites the oscillator 45, and thereafter the oscillator 45
interacts with the transistor 44 to remain in an oscillatory
condition. The oscillator circuit 45 alternately aids and opposes
the base drive 46 to the switching s3ries-pass transistor 44,
thereby alternately switching the transistor 44 between on and
off conductive states.
Figure 3 illustrates the preferred embodiment of the
D.C. ballastlng and starting circuit 11 of Figures 1 and 2. A
series-pass switching transiStGr 44~ of the NPN type, is connected
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in series between the input line 32 and a junction 56. The
collector terminal of the transistor 44 is connected to the
junction 56. A resistor 57 is connected from the input line 32
to a junction 58. The base terminal of the transistor 44 is
also connected to the junction 58 such that the transistor
receives base drive from the resistor 57 to normally bias the
transistor 44 in an on conductive state.
Connected in series between the junctions 52, 56 is a
resistor 59, of low ohmic value, for sensing the current delivered
by the transistor 44. A series combination of a zener diode
60 and a rectifying diode 61 are connected between the junction
52 and the junction 58. The zener diode 60 has its cathode
terminal connected to the junction 58 while the rectifying diode
61 has its cathode terminal at the junction 52. When the current
through the current sensing resistor 59 establishes a potential
thereacross which exceeds the zener voltage of the zener diode ~ .
60, the zener diode 60 begins conduction and diverts base cur- :~
rent drive delivered by the resistor 57 and by an oscilla-tor ~.
winding 86 away from the base of the transis~or 44. The diode ~
61 compensates for both the potential drop by the forward-biassed
~ase-emitter junction of the transistor 44 and also for tempera-
ture variation thereof. Thus, the combination of the resistor
59, the zener diode 60 and the rectifying diode 61, limits the
maximum current which the switching transistor 44 may conduct.
In fact, in steady-state operation of the circuit 11, the switch-
ing transistor 44 is repeatedly switc~ed ~o the off conductive
state upon reaching a predetermined current level. The ~ransistor
44 however receives sufficient hase drive that it operates near
: the saturation region before delivering the limited current.
Thus, although the transistor 44 delivers significant power to
the junction 56 during its on conductive states, because of the
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low collector to emitter drop across the transistor 44, power
dissipation in the transistor 44 is minimal. Of course, during
off conductive states of the transistor 44, no current is con~
ducted therethrough and no power di~sipation therein occurs.
A series combination of another zener diode 62 and
another rectifying diode 63 are connected between the junctions
56, 58 across the base-emi~ter junc~ion of the transistor 44.
The zener diode 62 has its cathode terminal connected to the
emitter of the transistor 44 while the rectifying diode has its
cathode terminal connected to the base of the transistor 44.
l'he diodes 62, 63 pre~ent reverse voltage breakdown of the base-
emitter junction of the transistor 44 due to signals applied to
the base-of the trans.istor 44 by the oscillator circuit. The
zener diode 62 also reduces oscillator loading during off con-
ductive states of the transistor 44.
An inductor 48 is connected in series with the current .:
sensing resistor S9 between the junctions 5~ and 50. A capa-
citor 49 is connected between the junction 50 and the second .
input line 33. The combination of the series inductor 48 and the
parallel capacitor 49 comprise a filter to smooth the pulses of
energy delivered by the switching transistor 44 at the junction
52. The voltage at the junction 50 is primarily D.C. with a
small amount of ripple superimposed th~reon~ The small voltage
ripple at the point 50 is due to the fact that the filter com-
prised of the inductor 48 and the capacitor 49 is not an ideal
filter.
Connected between the junction 52 on the opposite side
of the inductor 48 and the second input lead 33 is a commutating
diode 51. The commutating diode 51 has its cathode terminal
connected to the junction 52. Continuity of current through the
inductor 48 during the off conductive state of the transistor 44
is provided by the commutating diode 51. However, duriny the
on conductive state of the transistor 44 the commutating diode
51 is reverse-biassed and non-conductive.
Due to consideration5 of power ef~iciency and rapid
switching, as are more fully discussed hereinafter, the commutat-
ing diode 51 must have fast recovery times when switching between
conductive states. A suitable diode is commercially available
from Varo, Inc., Garland, Texas 74040, as part number V334X
and has 3 ampere, 400 volt ratings.
ld Connected in series between the junction 50 and the
output terminal 34 is a secondary winding 64 of a pulse trans-
former 65. The secondary winding 64 does not interfere with
passage of direct current therethrough to th~ lamp 12. However,
the secondary winding 64 is capable of delivering a starting
pulse of sufficient magnitude and duration which, when added to
the potential already at the junction 50, provides a sufficient
potential at the output terminal 34 to initiate ionization and
the stxiking of an arc between the electrodes o the lamp 12.
The starting circuit further has a pair of voltage dividing
resistors 66 and 67 connected between the junction S0 and the
second input line 33. ~nergy storage means in the form o~ a
capacitor 68 is connected in parallel with the resistor 66. A
primary winding 69 o~ the pulse transformer 65 has one end con-
nected to the junation 50. Another end of the primary winding
69 is connected through a spark gap 70 to another junction 71
between ~he voltage dividing resistors 66 and 67. Connected in
parallel across the spark gap 10 is a second capacitor 72 which
is of greater capacitance than the energy storage capacitor 6~.
The spark gap 70 is a v~ltage threshold sensitive device, which
is nonconductive for voltages across the resistor 66 which are
below its threshold voltage. Upon exceeding its threshold voltage,
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the spark gap 70 assumes a very low impedance characteristic if
provided with su~ficient current during initial conduction. The
capacitor 72 initia].ly provides suf~icient current to insure that
the spark gap 70 assumes a low impedance condition to completely
discharge the energy storage capacitor 68 through the primary
winding 6V of the pulse transformer 65, thereby generating a
starting pulse across the secondary winding 64. After discharging
the capacitor 68 to a low voltage level, the arcing in the spark
gap 70 will extinguish whereupon the spark gap 70 will resume its
high impedance, nonconductive sta~e. A suitable spark gap 70
with a threshold voltage level of approximately 90 volts is
commercially available from the Siemens Corp, Iselin, New Jersey
08830, as part number BI-F90.
If the starting pulse generated across the secondary
winding 64 is unsuccessful in striking an arc in the lamp 12, the
voltage at a junction 50 will nearly equal that at the input lead
32. The energy storage capacitox 68 will again recharge in
approximately one second to the point at which the voltage there-
across exceeds the threshold voltage of the spark gap 70. Thus
the starting circuit will continue to generate starting pulses
until ionization is established in the lamp 12. Unless the lamp
12 is defective or some other circuit malfunction is present, the
starting circuit will usually energize the lamp 12 when the
first starting pulse is generated. When arcing in lamp 12 com-
mences, the D.C. potential at the junction 50 and at the output
terminal 34 will drop to a potential which is determined by the
load characteristics of the lamp 12. That is, the ballasting and
starting circuit 11 beyins to operate in a current-regulated out-
put mode.
: 30 The means for controlling the on and off states of ~.
the series-pass transistor 44 is provided by an oscillator circuit.
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As shown in Figure 3, the oscillator circuit has a resonant
tank circuit consisting of a trans~ormer 75 and a capacitor
76. A primary winding 77 of the transformer 75 is connected
in series with the capacitor 76 between the junction 56 and the
second .input line 33. A transistor 78 momentarily excites the
resonant tank circuit when the starting pulse is generated at the
output terminal 34. Thereafter, the tank circuit interacts with
the transistor 44 to remain in a self-oscillating condition.
The transistor 78 has a collector terminal connected to ajunction
79 between the primary winding 77 and the capacitor 76~ An
emitter terminal of the transistor 78 is referenced to the second
input line 33 through a resistor 80. A diode 81 is connected
between the emitter and base terminals of the transistor 78~ with
the cathode of the diode 81 connected to the base of the tran-
sis~or 78, to prevent reverse voltage breakdown of the base-emitter
junction of the transi.stor 78 and to recharge a capacitor 83
through the resistor 80 to prepare the circuit for a subsequent
starting pulse, if necessary. A series comb.ination of a resistor
82 ana a capacitor 83 are connec~ed hetween the base of the tran- ;
sistor 78 and the output terminal 34. The oscillator circuit is
thus insen~itive to tke D- C. level at the output terminal 3~, but
is instanteously excited when ~he starting voltage pulse appears
at the output terminal 34. The transistor 78 momentarily conaucts
current through winding 77 thereby generatin~ base drive for the
transistor 44 across the winding 86. Resistor 80 limits peak
current conducted through the winding i7. Transistor 7g only
conducts during the starting pulse which, however, has the after~
effect of causing the~resonant tank circuit to ring for a suffi-
cient number of cycles to cause transistor 44 to begin self-os-
:cillatin~ at the natural resonant frequency of the tank circuitincluding winding 77 and capacitor 76. A parallel combination of
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a capacitor 84 and a resistor 85 are connec~ed in series with
a secondary winding 86 of the resonant transformer 75, whieh is
in turn eonneeted across the base and emitter terminals of the
serles-pass switching transistor 44. The secondary winding 86
applies the oseillator output signal aeross the base-emitter
~unetion of a series-pass switching transistor 44, thereby
alternately driving the transistor 44 into an on conductive
state. When the level of the transistor output signal across
the secondary winding 86 rises, the oseillator will reverse bias
10 the base-emitter junetion of the transistor 44 whieh will eause
the transistor 44 to assume an off conductive state. At the
same time, the secondary winding 86 will conduct any base drive
through the resistor 57 away from the transistor 44.
A parallel combination of the capacitor 84 and the
resistor 85 provide an R~C phase-shifting network in series with
the secondary winding 86. The R-C network phase-eompensates
the oseillator output signal for phase-shifts eaused by other
eireuit eomponents, and under normal operatin~ conditions pro-
duces 70 per cent on period and 30 per cent off period for the
switehing transistor 44.
The resonant frequency of th~ oseillator cireuit will
determine the frequency at whieh the switching resistor 44
operates. Higher operating frequencies are preferred because
the induetor 48 may be of less induetanee and the capaeitor 49
of less eapaeitance and still maintain the peak~o-peak rippIe
voltage appearing at the output terminal 34 below permissible
levels. Lower inductive and capacitive values mean that the
induetor 48 and the eapaeitor 49 of the filter means will be of
smaller physieal ~size and usually of a lower cost. However,
3Q Iimitation on the maximum frequeney which the oseillator should
operate is imposed by power loss eonsiderations in the switching
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transistor 44 and in the commutating diode 51. As previously
noted, very little power dissipation occurs in the series-pass
transistor 44 when the transistor 44 is in an on conductive
state because it is operating in the saturation region, i,e.
a very low collector to emitter voltage drol~. No power dis-
sipation occurs in the series-pass transistor 44 when it is in
an off conductive state, because there is no curren~ passing
therethrough. Similarly, the commutating diode 41 experiences
no power dissipation when in an off conductive mode, and very
little power dissipation when in an on conductive mode because
the voltage drop thereacross is only that of a forward-biassed
diode junction.
However, both the transistor 44 and the diode 51 have
finite turn-on and turn-off time periods. When the operating
frequency of the circuit becomes high enough that the turn-on
and turn-off times of the transistor 44 and the diode 51 become
an appreciable portion of the time period associated with the
operating frequency, the power losses in both the translstor 44
and the diode 51 also become appreciable. ~ suitable operating
frequency at which the size and cost of the inductor 48 and the
capacitor ~9 are minimized but which is also low enough to avoid
appreciable switching losses in the transistor 44 and in the
diode~Sl is in the vicnityof 20 kiloHertz.
Various'other forms of control and drive means for the
series-pass switching transistor 44 will become apparent to those
skilled ln the art besides the oscillator circuit illustrated in
Fiyure 3. Shown in Figure 4 is a resonant circuit of the tuned~T -
configuratlon which is interposed between the current sensing
resistor 59 and the junction 52, at which the commutating diode
,
51 is connected to one side of the inductor 48. A resonant
transformer 90 has a pair of primary windings 91, 92 connected in
'~ ~
-18~
.
,
series between the current sensing resistor 59 and the junction
52. A capacitor 93 is connected to a junction 94 between the
two windings 91, 92 and to the second input line 33. A second-
ary winaing 95 of the transformer 90 is connected in series with
a resistor 96 across the base and emitter terminals of the tran-
sistor 44. The resistor 96 limits the amount of current deliver-
able by the secondary winding 95 to the transistor 44. As the
polarity markings associated with the windings 91, 95 would in-
dicate, the secondary winding 95 provides more drive for the
transistor 44 when the transistor 44 begins to assume an on con-
ductive state.
At a later time during the on conductive state, when
current through the primary windings 91, 92 begins to decrease ,
the secondary winding in 95 will experience a voltage reversal
which will reverse bias the base-emitter junction of the transistor
44, thereby switching the transistor 44 to an of conductive state.
At the same time, base drive for the transistor 44 through the
resistor 57 will be divertea through the secondary winding 95.
Thus, the time periods of the on and off conductive states of
the series-pass transistor 44 will be determined by the frequency
of the tuned resonant circuit comprising the transformer 90 and
the capacitor 93. Otherwise, operation o~ the circuik in Figure
4 is similar to that in Figure 3. It is, of course, realized
tha~ the circuit in Figure 4 will require a starting circuit for
initiating ionization in a gaseous discharge lamp. The start-
ing circuit will be connected across a junction 50 and the second
input line 33 in a manner similar to the circuitry in Figure 3.
A pulse-width modulation technique for control]ing
and driving the series-pass transistor 44 is illustrated in
Figure 5. A zener diode 100 has an anode terminal connected to
the second input line 33 and a cathode terminal connected through
a resistor 101 to the first input line 32. The zener diode 100
':
--19--
~ .
provides both reference voltages and biasing voltages a-t a
terminal 102 which is connected to the cathode terminal of the
zener diode 100. The terminal 102 is electrically the same as
other terminals 103 and 104.
Terminal 103 is referenced to the second input line
33 through the series combination of a resistor 105, a lead 106,
a resistor 107, a lead 108, and a resistor 109. A base terminal
of a transistor 110, which is of the NPN type, is connected to
the lead 108. An emitter terminal of the transistor 110 is
referenced to the second input line 33. A collector terminal
of the transistor 110 is connected to a base terminal of a PNP
transistor 111 through a resistor 112. The transistor 111 has
an emitter terminal connected to the first input line 32 and a
collector terminal connected to the base of the series transistor
44. A resistor 113, connected across the base~emitter junction
of the transistor 111 provides a path for collector leakage cur-
rents of the transistor 110. Similarly, a resistor 114 connected
across the base-emitter junction of the series-pass transistor i
44 provides a path for collector leakage currents from the tran-
sistor 111.
The resistors 105, 107, 109 normally bias the tran-
sistor 110 into a nonconductive state which further causes the
transistors 111 and 44 to also assume on conductive states.
However, shortly after the ballasting circuit in Figure 5
is turned on, a free-running multivlbrator will begin periodically
switching the transistors 110, 113, 44 to an off conductive state.
An ampllfier 116 has a non-inverting input 117 connected to a
reference voltage supplied by a pair of voltage divlding resistors
118j 119. The resistors 118, 119 are connected in series between
the reference voltage supplied by the zener diode 100 at the
terminal 102 and the second input lîne 33. A feedback resistor
~.......
-20-
120, connected between the non-inverting input 117 and an output
121 of the amplifier 116 is selected to fix the gain of the
amplifier circuit. The output 121 of the amplifier 116 is con-
nected to the lead 106 and is therefore capable of controlling
bias voltage supplied by the resistors 105, 107, 109 to the base
of the transistor 110 to control the conductive state thereof.
An inverting input 122 of the amplifier 116 is referenced to the
second input line 33 through a capacitor 123. The inverting
input 122 is also connected to the output 121 through the series
combination diode 124 and a resistor 125, and throu~h another
series combination o~ another diode 126 and another resistor 127.
The diode 124 and 126 are poled in opposite directions with the
anode terminal of the diode 124 connected to the input 122 and
the cathode of the diode 126 connected to the input 122.
Operation of the free-running multivibrator is as
follows. When the ballasting circuit in Figure 5 is first ener-
gi2ed across the input leads 32, 33, the reference voltage applied
to the non-inverting input 117 by the voltage dividing resistors
118, 119 causes the output of the amplifier to assume a high
output condition which enables the biasing resis~ors 105, 107,
109 to bias the transistors 110, 111, 44 into on conductive
states. At the same time, the capacitor 123 connected to the
inverting input 122 of the amplifier 116 begins to charge through
the resistor 127 and the diode 126. The diode 124, bei.ng re-
verse biased, is nonconductive. At some later time, a voltage
across the capacitor 123 will begin to exceed the reference
voltage at: the non-inverting input 117, thereby causing the
amplifier 116 to assume a low voltage output condition at the
output~121. When the output 121 of the amplifier 116 assumes
a low output condition,~the transistor 110 loses its bias and
assumes an off conductive state, thereby also causing transistors
3,.f~
111, 44 to assume off conductive states. The low output of the
ampliier 116 also causes the voltage across-the capacitor 123
to exceed voltage at the output 121. Thus, the capacitor 123
begins discharging into the output 121 of the amplifier 116
through the diode 124 and the resistor l25. Diode 126 is now
reverse-biased and therefore nonconductive. The capacitor 123
will continue to discharge until the voltage at the inverting
input 122 is less than that of the non-inverting input 117, at
which time, the output 121 of the amplifier 116 will again assume
a high voltage condition.
It will be readily appreciated that the timing of the
high and low voltage conditions at the outpu~ 121 can be con-
trolled by selection of the resistors 125 and 127. For instance,
if the resistive value of the resistor 127 exceeds that of the
resistor 125, the capacitor 123 will take a longer time period
in which to charge than the time period in which the capacitor
123 will discharge through the resistor 125. ~hus, the duty
cycle, defined as the ratio of the time in which the output 121
will remain at a high voltage condition to the sum of the time
periods of the high and low voltage conditions at the output
121, may be controlled depending upon the selection of the
resistors 125, 127. Since the output 121 directly controls the
conductive state of the series-pass switching transistor 44, the
duty cycle of the multivibrator circuit directly correlates to
the duty cycle of the series-pass transistor 44. To ensure
r~hat sufficient energy is delivered ~o the load by the series~
pass transistor 44, a duty cycle for the multivibrator circuit
of approximately 70 per cent lS preferred.
However, with a duty cycle in the vicinity of 70 per
cent, the series-pass transistor 44 may deliver more energy than
is desirable and therefore some means of controlling or overriding
-22-
.
the output of the multivibrator circuit is desired. To this
end, the resistor 129, a low ohmic value, is placed in series
with the commutating diode 51 to sense the current level there-
through. As previously presented, the commutating diode 51 main-
tains continuity of current through the inductor 48 when the
series transistor 44 assumes a nonconductive state. During
steady state of operation o~ the ballasting circuit, the current
of level through the inductor 48 will remain constant from cycle
to cycle during any instant thereof. Thus, current through the
commutating diode, while the transistor 44 is off~ is directly
related to current through the transistor 44 when the transistor
44 is in a conductive sta~e. There~ore, current sensing means
may be placed in series with the commutating diode 51 instead of
in series with the series-pass swltching transistor 44 as is done
in Figures 3 and 4.
Current threshold aetecting means is used to sense cur-
rent levels in the commutating diode 51 above a predetermined
threshold. A light emitting diode 130 is connected in series
with an adjustable resistor 131 across the current resistor 129.
The light emitting diode 130 is placed in close proximity to a
photosensitive transistor 132 such that light from the diode 130
is detected by the photosensitive transistor 132.
The electrical isolation o~ the diode 130 from the
transistor 132 provides a means of voltage shifting. The collector
o~ the transistor 132 is connected directly to the terminal 104
and the emltter terminal is referenced through a resistor 133
through the second input line 33. The emitter of the transistor
- 132 is also connected to an inverting input 134 of a voltaye
comparator 135. A non-inverting input 136 of the comparator 135
3Q lS reerenced to the second input 11ne 33 to a resistor 137 and
is also referenced to the terminal 104 through another resistor
~ '
138. The resistors 137, 138 provide a voltage reference for
the non-inverting input 136. An output 139 of the comparator
135 is connected through diode 140 to the lead 108. The diode
140 is poled such that the cathode terminal thereof is connected
to the output 139.
The operation o the current threshold detecting means
will now be considered. Current through the current sensing
means, resistor 129, and through the commutating diode 51 cause
the light emitting diode 130 to emit illumination. The adjust
able resistor 131 provides adjustment of the intensity of the
illumination from the diode 130. Illumination on the photo-
sensitive transistor 132 causes conduction of current through
the transistor 132, with higher le~els of illumination causing
larger currents to be conducted in the transistor 132. Because
the inputs 134, 136 of the comparator 135 are both of quite high
impedance, current through the transistor 132 flows through the
resistor 133 thereby establishing an analog voltage at the in-
verting input 134. The analog voltage is related in magnitude
to the level of current through the commutating diode 51. When
the current through the commutating diode 51 becomes sufficiently
high, the analog voltage at the inverting input 134 will exceed
the reference voltage at the non-inverting input 136 thereby
causing the output 139 of voltage comparator 135 to assume a
low voltage condition. ~he diode 140 which was previousl~ re-
verse biassed now becomes forward biassed and diverts bias
current away from the transistor 110. The current threshold
detecting means will thus be able to override the output of the
free-running multivibrator circuit/ thereby providing a pulse-
width modulated control means for controlling the on and of
conductive states of the series-pass transistor 44.
-24~
5~
It will be appreciated that the ballasting circuit in
Figure 5 will use a starting circuit between the junction 50 and
the second input line 33 for striking an arc in a gaseous dis-
charge lamp such that the ballasting circuit of Figure 5 will
then operate in its current-regulated output mode.
Turning to Figure 7, there is disclosed a schematic - --
diagram wherein a single starting pulse circuit is capable of
sensing a nonfunctioning lamp on any one of a plurality of bal-
lasting circuits and independently starting the nonfunctioning
lamp. The secondary winding 144 of a pulse transformer 145 is
provided in series with the output of each of a plurality of
ballasting circuits. In each of the circuits, a lead 146 con-
nects one terminal of the secondary winding 144 to a junction 150
as in Figures 3, 4 and 5, while another lead 147 connects the
other terminal of the secondary winding 144 to the output termi-
nal 34 as in Figures 3, 4 and 5. A diode 148 is provided for
each of the plurality of the circuits lI with an anode of each
of the diodes 148 connected to the leads 146. The cathode termi-
nals of all diodes 148 are connected together at a junction 149,
such that the diodes are wired in a logic l'or" confi~uration.
An energy storage capacitor 150 is referenced through a resistor
151 to the second input line 33. From the junction 149, each
primary winding~152 of the plurality of pulse transformers 145
is wired in series to a junction 153. Polarity markings must be
observed as indicated in Figure 7 when wiring the primary windings
152 in series. ~Connected in parallel ~etween the junction 153
and a lead 154 are a capacitor 155, a spark gap 156, and a
resistor 157. The lead I54 is also connected to a junction 158
between the capacitor 150 and the resistor 151.
When a nonenergized lamp is connected to any one of
the ballasting and starting circuits, the potential on one of
the leads 146 will rise toward the potential present across the
25-
,
inputs 32, 33 of the ballasting circuit. The energy storage
capacitor 150 will begin charging toward the potentials supplied
by one of the diodes 148, as determined by the voltage dividing
resistors 151, 157. The potential at the leads 146 of other
ballasting circuits with energized lamps will remain unaffected
because the diodes 138 associated with those circuits will be
reverse biassed. When the voltage across the energy storage
capacitor reaches the voltage threshold point of the spark gap
156, an arc is established in the spark gap 156 and capacitor
155 provides sufficient current when the arc is first established
in the spark gap 156 to ensure that the spark ~ap 156 assumes a
low impedance mode. The energy storage capacitor 150 is there-
upon discharged through each of the primary windings 152 o~ the
pulse transformers 1~5 through the spark gap 156. Because the
ballasting circuit with the nonionized lamp will exhibit a much
greater impedance than those circuits with ionized lamps, most
of the energy in the capacitor 150 will be presented to the pulse
transformer 145 which is associated with the nonionized lamp.
Thus, the starting circuit is capable of sensing which one of a
plurality of ballasting circuits has a nonionized lamp and is
further capable o~ delivering a starting pulse of sufficient
magnitude an~ duration to ionize said lamp.
Another embodiment of the starting circuit 53 of
Figures 2 and 3 is illustrated in Figure 6. Similar to Figures
2 and 3, a secondary winding 64 of a pulse transformer 65 is
connected between the junction 50 and the outpùt terminal 34 by
leads 160 and 161, respectively. A voltage dividing resistor
163 is connected between the lead 160 and a junction 164 which
; is in turn referenced to the second input lead 33 by another
voItage dividing resistor 165. A capacitor 166 connected between
the junction 164 and the second input lead 33 filters electrical
~26-
noise at the junction 16~. A voltage level detecting means,
such as a diac 167, monitors the voltage level at the junction
164, The diac 167 is connected in series with a resistor ].68
between the junction 164 and the gate terminal of an SCR 169.
The resistor 168 limits the amount of current which may be
received by the gate terminal of the SCR 169 when the voltage
threshold level of the diac 167 is exceeded. Another resistor
170 is connected between the gate terminal of the SCR 169 and
the second input line 33 for passing any leakage currents from
the diac 167.
The energy s~orage capacitor 162 is connected in series
with a resistor 171 to the lead 160. One end o~ the primary
winding 69 ~f the pulse transformer 65 is connected to a junction
172 between the capacitor 162 and resistor 171. The other end
of the primary winding 69 is connec~ed to the anode terminal of
the SCR 169.. The cathode terminal of -the SCR 169 is referenced
to the second input line 33.
The line 160 will rise to a potential nearly equal to
the D C. supply voltage to the inputs 32, 33 o~ the ballasti.TIg
cixcuit, as previously discussed wit.h the starting circuits i.n
Figures 2 and 3. When this occurs, the voltage at the junction 164
will rise slowly causing the threshcld voltage of the diac 167 to
be exceeded. The diac 167 will the~eupon become conductive and
discharge`the potential across the capacitor 166 into the gate
terminal of the SCR 169, thereby firing the SCR 169. The SCI~ 169
. will in turn discharge the energy storage capacitor 162 -through
the primary winding 69 of the pulse transformer 65, thereby
generating a pulse on the secondary winding 64 of suficient magni-
tude and duration to s~rike an arc in .~he lamp~ The resistor
171 is of a high resistive value such that when the SCR 169 has
discharged the capacitor 162 to a low level, the resistor 171
~27
. : ;. .. .
. . .
will not supply sufficient holding current to keep the SCR
169 in a conductive state. The SCR 169 will then cease conduc-
tion and the capacitor 162 will again begin to charge toward
the potential on the line 160 through the resistor 171. The
starting circuit 53 thus resets itself and continues to monitor
the potential at the lead 1600 Should some malfunction or other
problem arise, the starting circuit 53 is prepared to restart
the lamp.
For proper operation of the circuity disclosed above
and in the Various drawings, selection and design of the Various
magnetic components is important. For example, i:f the switching
transistor 44 is to operate near the 2û kilo~ertz range, powdered
ferrite cores are preferred. The magnetic components may utilize
pairs of "E" cores with the legs of the pairs of "E" cores
butted together. Air gaps of various widths are also employed.
For examplel the inductor 48 typically has 400 turns of #28
copper wire wound on a pair of "E" cores with a 0.020 inch air
gap between the legs of the "E" cores. Suitable cores are com-
mercially available from Ferroxcube Corp., Saugerties, New York
12477 as part number 782E272-3E2A.
Similarly, the transformer 75 has a primary winding
77 o:E 134 turns of #39 wire and a secondary winding of 8 turns
of #29 wire with a 0. 005 inch air gap between a pair of "E"
cores of part number 206F440-3E2A (Ferroxcube). The pulse trans-
formers 65, 145 have a primary winding 69~ 152 of 9 turns of
#23 wire and a secondary winding 64, 144 of 200 turns o~ #26
wire with a 0.010 inch air gap between pairs of "E" cores oi~
`~ part number 782E272~3E2A (Ferroxcube).
It will be understood that various changes and modifi~
cations can be made without departing from the spirit of the
invention as defined in the folIowing claims, and equivalents
thereof.
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