Note: Descriptions are shown in the official language in which they were submitted.
~ J
RC~ 72,717
OVERLOAD PROTECTION CI~CUIT
FOR VIDEO A~IPLIFIERS
This invention relates to an overload ~rotection
circui-t for transis-tor video amplifiers which may be
employed as output driver amplifiers in a video signal
processing system such as a television receiver or an
equivalent system. In particular, the pro-tection circuit
provides amplifier overload protection in the presence of
excessive high frequency components in the siynal supplied
to the amplifier.
Video signal processing systems such as a
television receiver typically employ one or more transistor
video signal amplifier stages for supplying video output
signals to intensity control electrodes of an image
reproducing kinescope. Although relatively high power
transistor devices have been employed in these video signal
amplifiers (e.g., Class A amplifiers), more recently such
high power stages have been replaced by relatively lower
power video output stages (e.g., including transistors
arranged for Class B or C operation) in order to reduce the
power consumption of television receivers. The lower power
; stages typically are arranged to exhibit lower quiescent
25 power dissipation compared to Class A stages, for example.
Low power transistors can be employed in the low
power output stages since stages of this type exhibit power
dissipation which is substantially proportional to the
magnitude of the signal to be amplified. ~lowever, low
30 power transistors are susceptible to overload when the
signals processed by these stages include significant
amounts of high frequency components with a high density
of duration. An overload may occur, for example, when weak
signals containing significant amounts of noise are
35 amplified, or when the receiver is switched to a vacant
channel containing no vldeo information. In these
instances, the noise is amplified by the intermediate
frequency stages and following amplifier stages, which
~` typically operate at maximum gain under these conditions
; :
.
`: . , ' :
.
,
.
.:
~$.~
1 - 2 - ~CA 72,717
due to the au-tomatic gain control (AGC) action of the
receiver. The noise typically encompasses the entire video
signal frequency spectrum, and can occur without
interruption during the entire image cycle (i.e., during
both image trace and retrace blanking intervals). This
continuous stream of noise resul-ts in the amplifier stage
conducting virtually continuously, -thereby causing the
power dissipation and operating -temperature of the
amplifier stage to increase over a period oE time. This
in turn can lead to destruction of the ampliEier due -to
the phenomenon of thermal runaway (i.e., overheating of the
transistors forming the amplifier stage). Under certain
signal conditions (e.g., vacant channel reception), the
power dissipation can be many times above that experienced
under normal signal reception conditions. Excessive power
dissipation can also occur when the signal to be amplified
is representative of complex image patterns such as may be
20 reproduced by a television receiver employed in conjunction
with a "video games" system, or when nonstandard test
patterns are used.
Current limiting circuits associated with each
amplifier subject to overload under the conditions mentioned
~5 above are considered disadvantayeous for several reasons.
Circuits of this type typically cannot distinguish between
video information and noisy signals or noise alone, and
therefore can be expected to undesirably limit peak signal
currents representative o~ video information. These
30 circuits also commonly require at least one relatively
large and costly high voltage power transistor. ~oreover,
three such circuits would be required in the case of a
; color television receiver having three driver amplifiers
for supplying amplifiedt color representative video signals
35 to respective intensity control electrodes of the kinescope.
The use of heatsinks for low and medium power
video output stages to compensate for the excessive power
dissipation under the described conditions is also
disadvantageous. ~featsinks are relatively large and costly,
,~
, ,,
. . . . .~ . : .
- . ' . .. . ' ~
.
1 - 3 - RCA 72,717
and can compromise -the high frequency response of the
output stages due -to capaci-tance loading oE the video output
signal.
An automatic galn control (A5C) voltage derlved
from AGC circuits commonly employed in television receivers
is not suitable as a means for providing an inc~ication of
the abnormal slgnal conditions likely to cause excessive
power dissipation, since this voltage typically does not
discriminate between normal ancl abnormal signal conditions
(e.g., between normal signal recep-tion and vacant channel
reception). Accordingly, -the AGC voltage is unsuitable as
a means for controlling the operation of the video ou-tput
stages to limit excessive power dissipation due to
abnormal overload conditions.
A circuit for providing amplifier overload
protection in the presence of abnormal complex signals,
noisy signals, and noise alone (e.g., due to loss of signal
or when the receiver is set to a vacant channel) desirably
should avoid the disadvantages mentioned above, while at
the same time being relatively economical and uncomplicated.
The circuit should be capable of discriminating between
normal and abnormal signals, and should react to
25 potentially destructive long term signal conditions capable
of overloading the video amplifier stage, rather than to
relatively short term signal overload conditions. A
circuit which achieves these results is provided in
accordance with the present invention.
A protection circuit in accordance with the
present invention is included in a video signal processing
system comprising a video signal transmission path including
a video signal amplifier. The video amplifier is
undesirably susceptible to excessive conduction and
35 dissipation when the input signal to the amplifier contains
high frequency components of a significant magnitude and
with a high density of occurrence. The protection circuit
includes an input network coupled to the video signal path
and selectively responsive to high frequency signals for
.
- : :
7~
RCA 72,717
derlving a signal indicative of -the presence of high
frequency signals. A controllable conduction device is
responsive to the derived signal for provicling an output
control signal when the derived signal exceeds a given
level indicative of the presence of high frequency
components of a significant magni-tude and with a high
density of occurrence. The con-trol signal is applied to
the video signal pa-th to vary the gain thereof, and thereby
the level of video signal, in a direction to reduce the
magnitude of the video signal. The conduction and
dissipation of the video amplifier in response to the high
density high frequency components is correspondingly
redueed.
In the drawing, FIGURE 1 illustrates a diagram,
partially in block form and partially in schematic clrcuit
diagram form, of a portion of a color television receiver
including a proteetion eireuit aecording to the present
invention;
FIGUR~ 2 depiets a portion of FIGURE 1 in more
detail, ineluding a circult eonstrueted in accordanee wi-th
the present invention;
FIGURE 3 shows a eircuit embodiment of an
alternative use of a protection eircuit according to the
invention.
Referring to FIGURE 1, television signal
proeessing eireuits 10 ineluding, for example, intermediate
frequeney amplifier and video deteetor stages, and
frequeney seleetion networks, provide outpu-t luminanee and
ehrominanee signal eomponents (and other appropriate
signals) to inputs of an interrnediate signal proeessing
unit 17. In this example, unit 17 eorresponds to the
TDA 2560 integrated eireuit whieh is shown and diseussed
35 in more detail in eonneetion with FI~URE 2. Chrominanee
and luminanee components from the outputs of proeessor 17
reeeive further amplifieation and processing from a
ehrominanee signal proeessor 18 and a luminance signal
` proeessor 19, respeetively. Chrominanee proeessor 18
,~
- . - , ' : ~ :
: . .: . - . . : :
- ~ ~ . .. . : , ..
'' ' ' ' '
.
:.
1 - 5 - RC~ 72,717
develops R~Y, G-Y and B-Y color diEference signals which
are combined with an amplified luminance signal Y from
unit 19 in a demodulator-matrix 20 to provide R, ~ and B
color video signals (i.e., red, green and blue color image
representative signals). These signals are then amplified
by similar, low power video driver s-tages 22, 25 and 30,
the latter stage being shown in circuit form.
~mplifier 30 comprises a pair of complementary
conductivity type transistors 32 and 34 arranged as a
push-pull video amplifier. Transistors suitable for use
as transistors 32 and 34 include the types BF 470 and
BF 469, respectively. Video signal B from the outpu-t of
15 matrix 20 is coupled to a base input of ~IPN transistor 34,
and to a base input of PNP transistor 32 via a capacitor 36.
An amplified output signal B' of stage 30 appears at the
junction of collector resistors 35 and 38 of transistors 32
and 34. A network 40 provides degenerative feedback for
amplifier 30. Feedback network 40 can comprise a resistive
voltage divider for example, and can be coupled via
appropriate circuits to the base input of transistor 34 or
to an input of a preamplifier stage (not shown) prior to
stage 30, such as may be included in unit 20. Feedback
25 network 40 may include video signal black and white level
adjusting circuits, as well as frequency selective feedback
to provide signal peaking at one or more selected video
signal frequencies. The amount of ~.C. and D.C. feedback
can be varied to adjust the circuit gain and opera-ting
30 point.
Video amplifiers 22 and 25 are similar to
amplifier circuit 30 including network 40. Amplified video
signals R', G' and B' from amplifiers 22, 25 and 30 are
respectively appIied to intensity control electrodes
(e.g., cathodes) of a color kinescope 45.
The arrangement of FIGUR~ 1 also includes a
protection circuit 50 coupled to intermediate signal
processor 17. Protection circuit 50 serves to control the
amplitude of the signals processed by the luminance signal
-
.
~ ~, ~
1 - 6 - RCA 72,717
processing portion of unit 17 in the presence of abnormal
signal conditions such as excessive noise likely to damage
the transistors comprising push pull amplifiers 22, 25 and
30 (e.g., transistors 32 and 34 of amplifier 30). Although
the receiver arrangement of FIGURE 1 includes three
push-pull amplifiers 22, 25 and 30 susceptible to damage
under these condi-tions, a single protection circuit 50
(as will be described) provides the desired protection for
all three amplifiers, since the protection circuit is
associated with -the luminance signal path which is common
to each of the driver amplifier stages.
Intermediate signal processing stage 17 and
associated protection circuit 50 are shown in greater detail
in FIGURE 2. In FIGURE 2, signal processing unit 17 is
represented by a TDA 2560 video signal processing integrated
circuit which is available from Mullard Limited of London,
~ngland. Integra-ted circuit signal processor 17 includes
a plurality of external terminals numbered 1 through 16 for
coupling various signals and operating voltages between
processor 17 and other circui-ts of the receiver, as labeled.
These numbered terminals correspond to the actual numbered
terminals of the TDA 2560. In the interest of brevity only
those portions of processor 17 which are associated with
protection circuit 50 and thereby relevant to an under-
standing of the present invention will be described.
Protection circuit 50 comprises a normally
non-conduc-tive common emitter control transistor 52 having
30 a base input for receiving a sample of the signal appearing
at terminal 15 of unit 17, and a collector output D.C.
coupled to a contrast control circuit 60. Contrast control
60 includes a manually adjustable contrast potentiometer 62
coupled between a source of direct voltage ~+12 volts) and
35 ground by means of voltage divider resistors 65 and 67.
A wiper of potentiometer 62 is coupled to a contrast control
input terminal 16 of integrated circuit processor 17, which
in turn is coupled to signal gain control circuits within
processor 17 for varying the amplitude and thereby the
.. . , ' ' '
' ' ' ' ' . '.' ~ " ' , , . -
:
- . : -
: ' ' ' ' ' .. . ~ '
. , . , ~ . . ' : ~
1 - 7 - RC~ 72,717
contrast of the luminance signals processed by unit 17 in
accordance with the setting of potentiometer 62. The
collector of control transistor 52 is coupled to the
junction of resistor 65 and potentiometer 62.
The signal appearing at terminal 15 oE unit 17
is relatively unprocessed by unit 17 (i.e., unaEfec-ted by
the contrast and brigh-tness controls associated wi-th
unit 17), and is coupled to a sync separator 42 for
separating the sync component of the luminance signal in
known fashion. The signal from -terminal 15 is coupled to
transistor 52 via an Ao C. coupling capacitor 53 and a
reetifier diode 55. Capaeitor 53 together with the
impedanees of the assoeiated eircuit elements comprises a
signal differentiating network for differentiating the
signals eoupled from terminal 15. A diode 56 coupled
between the anode of diode 55 and ground serves to clamp
negative-going amplitude portions of the differen-tiated
signal eoupled via eapaeitor 53. A eharge storage
integrator eapaeitor 57 and a bleeder resistor 58 are
eoupled between the base of transistor 52 and ground.
The signal at terminal 15 of unit 17, from whieh
the ehrominanee and sound earrier signals have been removed,
is inverted relative to the input luminance signal applied
to terminal 14, and ineludes positive-going sync pulses Vs
disposed on a pedestal level (approximating the image black
level) and oeeurring during eaeh horizontal line blanking
interval, and relatively negative-going image portions
30 between blanking intervals. In this example, the luminance
signal appearing at terminal 15 exhibits a nominal peak-to-
peak amplitude of approximately three to four volts,
ineluding a syne pulse peak-to-peak amplitude of
~ approximately one volt. The differentiated signal whieh
'~ 35 is eoupled via eapaeitor 53 is reetified by diode 55 to
provide a referenee voltage for the proteetion eireuit.
Only high frequeney signal amplitude transitions
are passed by differentiator eapaeitor 53. Thus eapaeitor
53 will pass high frequeney noise when present, as well as
'
, .
,
.
~.. h ~ L~
1 - 8 - RCA 72,717
high frequency amplitude t:ransitions of bo-th normal vi~eo
signals and video signals represen~ative oE complex
patterns or images to be reproduced in a "video games"
system. In this connection, it is noted that -the high
frequency si~nal density of a normally expected video signal
and that of noise are not the same. The na-ture of normal
high frequency components of a video signal can be
considered as being sporadic in the time domain, whereas
in contrast high frequency noise components can be
considered as relatively continuous in the time domain.
Similarly, image patterns in a video games system are often
continuous in the time domain compared to normally received
television signals.
Diode 56 (e.g., type lN914) clamps negative-going
amplitude peaks oE the differentiated signal from capacitor
53 to approximately 0.7 volts. Rectifier diode 55
preferably is a germanium type (e.g., type OA 91) having a
low threshold conduction level to minimize the voltage drop
of the rectified, clamped signal. Filter capacitor 57
integrates the rectified signal from diode 55 to develop
a D.C. reference voltage at the base of transistor 52. The
arrangement of clamp diode 56 and rectifier diode 55 with
capacitors 53 and 57 provides an appropriate reference
voltage at the base of transistor 52 indicative of the
presence of normal video signals or abnormal signals such
as high frequency components occurring with a high density
of duration (hereinafter referred to as high density
signals). Since the reference voltage developed at the
base of control transistor 52 is primarily derived from the
rectified positive going portion of the video signal (which
primarily includes the sync pulse), this arrangement
: provides a significant differential between a reference
35 voltage at the base of transistor 52 due to normal signals,
and a reference voltage attributable to the abnormal high
density signa:Ls, particularly under low level signal
conditions.
The base-emitter threshold conduction level of
.. . ., :
. -: . - . ~ . - . . : ..
. .
, . . ,, . ,' - ,
. . - . - , ~ .
' ,,',;' '';" ' , ,',,: ' ,:~,, ,",::'', "',',' ;" '-',. '.' ' ' ': ' -
: .:. . . . : :: . . .. .. :.. . . :: . . : : .. .. .
1 - 9 - RCA 72,717
normally non-conductive -transistor 52 is exceeded only in
-the presence of the high density sianals from terminal 15
of sufficiently large amplitude, since only such signals
can cause capacitor ~7 to charge sufficiently to render
transistor 52 conductive when i,t is desired to compensate
for these sisnals, as will now be discussed.
Under normal signal c:onditions, transistor 52
is non-conductive due to insufficient base bia~3.
Specifically, -the high ~requency components of the positive
portion of the video signal as rectified by diode 55 charge
filter capacitor 57 to a voltage proportional to the
average of the rectified high frequency components. Since
this average voltage is derived from relatively sporadic
high frequency video information, and due to the discharging
action of bleeder resistor 5~, capacitor 57 i5 normally
unable to charge to a voltage level sufficient to forward
bias the base-emitter junction of transistor 52 to render
transistor 52 conductive.
The voltage developed at the base oE transistor 52
increases significantly in the presence of the high density
signals when a relatively continuous stream of high
' frequency components of sufficient amplitude is present in
the output from terminal 15, since the high density high
` frequency signals charge capacitor 57 faster than it can
'~ be discharged by bleeder resistor 58. When the base voltage
exceeds the base-emitter threshold conduction level of
transistor 52 (approximately 0.7 volts), transistor 52 is
forward biased into conduction. Transistor 52 then conducts
collector current through resistor 65 of contrast control
circuit 60, and the collector voltage of transistor 52 and
thereby the contrast control voltage appearing at the wiper
~ of potentiometer 62 decrease in accordance with the level
'~ 35 of conduction of transistor 52. The reduced contrast
control voltage developed at the wiper of potentiometer 62
is of a magnitude and direction to cause the gain control
circuits coupled to terminal 16 within unit 17 -to reduce
the amplitude of the video signal by a corresponding amount.
.
, . .
, . . . ..
~ 31~ ~
1 - 10 - ~C~ 72,717
Accordingly, the amplitucle of -the output signal from
terminal 10 of unit 17 is attenuated a corresponding amount.
The amount by which the signal gain is reduced under these
conditions can be tailored by inserting a resistor of
appropriate value in series with the collector of
transistor 52.
The attenuated signal from terminal 10 of unit 17
corresponds to luminance information plus noise in -the case
of a weak video signal, or noise alone when the receiver
is tuned to a vacant channel, for example. In either case,
the a-ttenuated signal from terminal 10 serves to prevent
output amplifiers 22, 25 and 30 (FIGURE 1) Erom being
damaged due to overload under -the abnormal high frequency
signal conditions, since the signal drive to these
amplifiers is reduced. All -three video output stages are
protected in -this manner since the output signal from
unit 17 is coupled via luminance processor 19 to matrix 20
(FIGURE 1), where it is combined with the color difference
signals to generate the R, G and B signals which drive
output amplifiers 22, 25 and 30. That is, the controlled
signal from terminal 10 of unit 17 is common to all three
output amplifiers.
The arrangemen-t of protection circuit 50 also
provides a degree of temperature compensation.
Illustratively, a temperature rise common to the output
video driver stages and to protection circuit S0 increases
the gain and therefore the dissipation of the video driver
stages, and also increases the gain of control transistor
52. The latter effect in turn serves to reduce the video
signal amplitude and thereby the drive level to the video
output stages.
Referring now to FIGURE 3, there is shown an
36 alternative embodiment of the present invention in a
circuit which provides image enhancement by means of
kinescope beam current velocity modulation in a television
receiver. This method of image enhancement is not essential
to an understanding of the principles of the invention as
.
., ,., , , . . . . : . .-.
- . ~ . ~ : : .
: ~ . . . ~ - . .
.- : . . . .
, ., ~ , . . . : .
., - . . . .
. ..
.
RC~ 72,717
applied to the arranyement of FIGURE 3, and therefore will
be discussed only briefly.
In FIGURE 3, luminance signals (Y) from a source
110 are coupled to luminance s:Lgnal processing circuits of
the recei.ver via a conventiona:L luminance delay line 113
which provides a signal delay within a range of 400-700
nanoseconds. A luminance signal Y' derived from a tap on
delay line 113 is buffered by an emitter follower
transistor 116, differentiated by a capacitor 118 and
coupled via a resistor 119 to a common emitter pre-amplifier
transis-tor 122. An amplified version of the ~ifferen-tiated
luminance signal appears at a collec-tor output of tran-
sistor 122, from which it is A.C. coupled -to a low power
push-pull video amplifier 125 comprising Class C
complementary conductivity type inpu-t transistors 123, 124
and complementary conductivity type output transis-tors 126,
128 all arranged as shown.
The output signal from transistor 122 is applied
to base inputs of transistors 123 and 124, and an amplified
output signal appears a-t a point A in the interconnected
collector electrodes of output transistors 126 and 128.
This signal is then used to drive a small auxiliary yoke
coil 130 located beneath the main yoke on the neck of the
television receiver kinescope (not shown). In this
: example, the system is arranged such that the signal
: appearing at point A comprises positive going line retrace
pulses generated by the deflec-tion circuits, and amplified,
30 differentiated pulses of positive and negative polarity
(developed by the differentiator action of capacitor 118)
disposed between adjacent positive retrace pulses.
Since amplifier 125 amplifies a differentiated
luminance signal, only the black to white and the white
35 to black amplitude transitions of the luminance signal are
ampl.ified by amplifier 125. These transitions and the
associated high frequency components occur sporadically
under normal program conditions. Amplifier transistors 123,
: 124 and 126, 128 therefore conduct only for very short
: 40
-' . ~ :
., , ~ . ~' ', - :
, , . , ' ':: .: . : -
~ 7~i'(3
1 - :L2 - ~C~ 72,717
periods during each horizont~l image line, which permi-ts
low power -transistors -to be used for these transistors.
Transistors sui-table for use as transistors 123 and 124
include types 2N4126 and 2N~124, and transistors suitable
for use as transistors 126 and 128 include types MPS6531
and MPS6534, respectively.
As in the case of the video output st:ages of
10 FIGURE 2, when amplifier 125 is supplied with an abnormal
input signal such as may contain a significant amount of
non-sporadic, high frequency components occurring wi-th a
high density of duration, the amplifier transistors can be
cause~ to conduct almost continuously for the duration of
the abnormal signal. This causes the power dissipation
of these transistors to increase, thereby increasing the
likellhood that these transistors will be destroyed due
to overheating and the attendant phenomenon of thermal
runaway. A protection circuit 150 is included to prevent
this from occurring. Circuit 150 in large part is
structurally and operationally similar to protection
circuit 50 shown in FIGURE 2.
In circui~ 150, a sample of the output signal
appearing at point A is coupled via a resistor 154 to a
25 rectifier diode 155. In this example, diode 155 rectifies
the negative-going portion of the output signal ra-ther
than the more positive portion of the signal. This approach
is preferable in this example, since the negative portion
of the signal does not contain the positive retrace pulse
30 which is generated during the horizontal flyback intervals
and induced into auxiliary yoke 130. Due to the presence
of the positive retrace pulse, the more positive portion
of the signal is less useful for noise sensing purposes
compared to the system of FIGURE 2.
The rectified signal from diode 155 charges an
integrator capacitor 157 to a voltage proportional to the
average of the rectified signal. Capacitor 157 and a
bleeder resistor 158 are coupled in parallel across the
base-emitter junction of a normally non-conductive PNP
: . .. . : . . . . .
;, . . , . ~ ' ~
~ ' ':
~ .
. .
'` 'L'J~ ra~
1 ~ 13 - ~CA 72,717
control transistor 152 (e.g., type 2N4:L26). Und~r normally
expected signal conditions, the charge developed on
capaci-tor 157 at the base electrode of transistor 152 is
insufficient to forward bias transistor 152 to conduct,
due to the sporadic nature of the high frequency content
of a normal video signal and -the bleeder action of resistor
158, as mentioned in connection with FIGURE 2.
In the presence of significant amounts of the
high density signals, however, capacitor 157 charges faster
than it can be discharged via resistor 158, and develops a
voltage sufficient to cause transistor 152 -to conduct. The
collector current of conductive transistor 152 flows
through a resistor 156 and resistor 119 to supply
additional base current drive to pre-amplifier transistor
122. This additional base current causes transistor 122 to
saturate, thereby attenuating or limiting the output signal
at the collector of transistor 122. The pulsed output
signal at point A and thereby the input to protection
circuit 150 decreases until an equilibrium condition is
reached, at which transistor 152 conducts sufficiently to
hold the signal at point A at a maximum predetermined level.
This result is obtained while the abnormal condition
2S persists, after which transistor 152 reverts to the normal
cut-off state.
The level of the signal at point A is determined
by the values of resistors 154 and 158 and capacitor 157.
An increasing signal at point A causes the voltage developed
across resistors 154 and 158 to increase, which in turn
increases -the conduction of transistor 152 and transistor
122 which saturates, thereby reducing the collector output
signal of transistor 122 by the limiting action of
transistor 122. Resistors 154 and 158 comprise a voltage
divider such that when the value of resistor 154 is
increased for a given value of resistor 158, the signal at
point A is permitted to increase to a higher level before
the threshold conduction level of transistor 152 is reached.
Capacitor 157 serves to integrate the signal pulses
- ; ~ . . : ' ~ '
: : . . ~.: -
~ . - - ~' . ' '- .
.
: ~ ~ ' ' .- ......... .. . ':
.
RCA 72,717
appearing a-t the base oE transis-tor 152 -to thereby preven-t
sporadic signals from forward biasing transistor 152.
Resistor 156 serves as a current limiting
resistor to protect transistor 122 if control transistor 152
fails. For example, lf a kinescope are destroys transistor
152 by causing a short-eircuit in the base-emitter or
base-collector junctions of transistor 152, resistor 156
will limit the base current of transistor 122, which could
otherwise increase -to a potentially destructive level.
The limited output signal from transistor 122
limits the conduction of the transistors comprising
amplifier stage 125 to a level sufficient to ensure that
these transistors will not be damaged or destroyed due to
exeessive dissipation. As a specific example, under normal
signal eonditions, it has been observed that the average
power dissipation of output stage 125 is approximately
0.35 watts. With the antenna disconnected from the receiver
to simulate an interrupted signal transmission or a vacant
channel, the dissipation rises to seven watts and increases
with time in the absence of protection circuit 150. With
. . proteetion eireuit 150 installed, the power dissipation is
. limited to an aeceptable, eonstant 1.3 watts.
Temperature eompensation is also provided in this
example. Speeifically, a temperature induced increase in
the gain and attendant dissipation of video amplifier .:
. stage 125 increases the level of the signal monitored by
~ proteetion circuit 150. The current conduction of control
30 transistor 152 increases in response to this signal,
whereby transistor 152 supplies additional temperature
indueed base eurrent drive to pre-amplifier transistor 122.
This current is in a direction to oppose the effect of the
signal current supplied to transistor 122, and serves to
35 reduee the signal gain of transistor 122 and thereby the
dissipation of output stage 125.
In sum, an uncomplieated, eeonomieal and energy
effieient overload proteetion cireuit has been described.
. No speeial eomponents sueh as high power transistors are
. ' '-;. ' '
,: - ~ . '
:
~- ' ~ . ,
: , .
1 - 15 - RC~ 72,717
required, and no power is consumed under normal operating
conditions, since the protection circuit control transistor
(e.g., transistor 152 in FIGURE 3) is normally
nonconductive. Moreover, a system employing the described
protection circuit can utilize video output transistors
without heatsinks which would otherwise be required to
compensate for excessive power dissipation cause~ by the
adverse signal conditions mentioned previously~
Although the invention has been described with
reference to particular embodiments, various additional
modifications can be made within the scope of the
inven-tion.
For example, with regard to the FIGURE 2
embodiment, the protection circuit could be arranged to
monitor the output signal appearing at terminal 10 of
unit 17, instead of the signal appeaxing at terminal 15
of processor 17, in accordance with the requirements of a
20 particular system. In the latter instance, the effect of
the contrast and brightness controls upon the output video
; signal at terminal 10 does not alter the operation of the
described protection circuit.
It may also be desirable to arrange the
25 protection circuit in a manner so as to monitor noise for a
"noise squelch" application, where the signal includes
high density, high frequency components independent of
image signals during the line interval. This can be
accomplished by monitoring the output of sync separator 42,
30 which contains no image information.
;
~ 3
. .
~ ~ '
.
,
.- . . :
: ~ -- . ,
:
