Note: Descriptions are shown in the official language in which they were submitted.
3~6~i
RCA 71,111
l This invention relates to video amplifier circuits
and, in particular, to wide bandwidth, low power consump~
tion, video output stages with suppressed radio frequency
harmonic radiation suitable for driving a color kinescope.
It is desirable for a video output stage to have
a wide signal bandwidth, a linear signal response and low
quiescent currents with attendant low power consumption.
Conventional Class A video output amplifier stages for
driving a capacitive load represented by a kinescope
of a television receiver require relatively large output
currents in order to achieve required signal bandwidth
for large amplitude video signals. Reducing power con-
sumption of Class A stages by reducing output currents
tends to impair the large signal bandwidth response
of the stage by reducing the slew rate (the change in
output voltage per unit time) of the stage.
Low power, wide bandwidth transistor video out~
put stages have recently been proposed. The low power
stages typically exhibit lower power consumption than
Class A stages, for example, and do not require transistors
of large physical size, transistor heatsinks, or large
power load resistors. Lower operating temperatures also
result due to the low power consumption of these stages.
Circuit cost is reduced, and reliability is enhanced.
A low power, wide bandwidth video output stage
of this type commonly includes a pair of similar conductivity
type transistors coupled across an operating voltage supply.
Input signals are supplied to a first, common emitter
amplifier transistor having an active load circuit. The
active load circuit includes a second, high voltage
- 2 -
~ 3~6 RCA 7~
transistor, and permits an increase in the load impedance
of the first transis-tor to reduce bias currents while
preserving the bandwidth response of the stage. A dis-
connect diode, interposed be-tween an emitter of the load
transistor and a collector of the first transistor, is
biased to a non-conductive state under quiescent con-
ditions. Degenerative direct current feedback is pro-
vided from an output to an input of the stage, and opera-
ting current for the stage is provided from the operating
voltage source to a base of the load transistor and to
the collector of the input transistor. Output signals
appear at an emitter of the load transistor.
A video output stage of this type is described
in an article entitled "Complementary Push-Pull Video
Amplifiers for Television Receivers," by D. ~. Beakhust
and M. C. Gander, contained in Mullard Technical Com-
munications Bulletin, Volume 13, No. 128, October 1975
and published by Mullard Limited, London; and also
in a Consumer Design Note entitled "A Low Dissipation
Class AB Video Output Stage Using TO-92 Transistors,"
published by Motorola Semiconductors Europe (1975). A
low power video output stage employing complementary con-
ductivity type transistors is also described in the afore-
mentioned Technical Communications Bulletin published by
Mullard Limited.
Various cascode video output amplifiers are
also known (see, for example, U.S. Patents No. 3,499,104 -
Austin; 3,598,312 - Nillesen; and 3,823,264 - Haferl~.
The cascode arrangement typically includes a low voltage,
high current gain, common emitter device coupled to a
- 3 -
~.Z3~6
RCA 71,111
high vol.tage, unity current gain, common base devieefor isolating the collec-tor of the low voltage deviee
Erom load voltage variations, thereby minimizing Miller
multiplication of collector-base capacitance of the
low voltage device. The effeet on amplifier bandwidth
of the eollector-base capacitances ofthe active devices
ls therefore less in the cascode arrangement than in,
for example, a common emitter amplifier. A particularly
advantageous low power eonsumption eascode video output
stage is deseribed in U.S. Patent No. 4,096,517, entitled
"Video Amplifier," issued June 20, 1978.
In the eascode amplifier deseribed
therein, in addition to the direct benefits o eonserving
power and avoiding thermal drift of operating eharaeteris-
ties, the reduced power consumption of the output devlees
enhances the wide bandwidth eapabillty of the cascode
amplifier.
Video output stages are required to proeess
wide bandwidth video signals (e.g., zero Hertz to 4-5 MHzj
of large amplitude (e.g., 110 volts peak-to-peak) in
eolor television systems. The video output stages ean
produce non-linear signal distortions which, in the case
of some relatively high power conventional Class A stages,
result in undesired signal harmonic frequeneies. Higher
order signal harmonics developed and radiated by such
video output stages can be detected by radio frequency
~RF) signal processing circuits of the receiver and
can interfere with received image representative video
signals. The undesired radiated RF signals can appear
-- 4
., ,
. ~,
2~6 RCA 71,111
as visible interference in an image displayed by the kine-
scope.
Attempts -to suppress -the RF interference have
included placing ~' chokes in the output circuits of the
video output stages, and employing shielded cables for coup-
ling video output signals to the kinescope. These solu- -
tions, however, undesirably increase circuit cost and
complexity.
The recently proposed low power video output ;
stages tend to produce greater RF radiation than conven-
tional Class A stages. A cause of such RF radiation is
the pronounced, non-linear (exponential) conduction
characteristic of these stages due to low quiescent current
operation. The nonlinear characteristic tends to produce
signal distortions and, consequently, harmonics of the
input signal frequency. Also, the input amplifier transistor
can be operated at or near cut-off in response to rapid
input signal amplitude transitions or transients, thereby
producing significant high frequency signal harmonics -
due to the more pronounced nonlinear conduction characteris-
tic in the region of transistor cut-off.
The undesired high frequency harmonics can extend
into the very high frequency (VHF) band of television signals
(e.g., 55 MHz to 211 MHz according to United States
television system standards, and a comparable frequency
range according to European television system standards).
Suppression of the undesired RF harmonics by employing
an output RF signal choke is difficult due to the low
output impedance typically presented by the low-power
video output stages.
-- 5 --
~ 6~ RCA 71,111
an embodiment of
In accordance with/the present invention, a
wide bandwidth video amplifier for driving an image re-
producing device having an intensity control electrode
exhibi-ts suppressed RF harmonic radiation. The amplifier
comprises a semiconductor amplifier having a nonlinear
conduction characteristic in a region of low current
conduction. Input and output circuits are associated
with the amplifier, and means are included for coupling
the output circuit to the intensity control electrode
of the image reproducing device. Degenerative feedback
is provided from the ou-tput circuit to the input circuit.
In accordance wi-th a feature of the invention, an imped-
ance common to the input and output circuits comprises a
semiconductor device having a nonlinear conduction character-
istic in a region of low current conduction, or providing
degenerative feedback for the amplifier when operating
in the region of low-current conduction.
In accordance with a further feature of the
invention, the input circuit includes a low pass signal
filter network to assist suppression of the RF radiation
independent of the operation of the common impedance.
In the drawings:
FIGURE 1 shows, partially in block diagram form
and partially in schematic circuit diagram form, a portion
of a color television receiver employing a circuit con-
structed in accordance with the present invention;
FIGURE la - ld show signal waveforms useful in
understanding the operation of the circuit shown in
FIGURE l;
FIGURE 2 shows another circuit embodiment of the
-- 6
~3.Z3~66 RCA 71,111
1 present invention;
FIGURE 3 shows a further circui-t embodiment of
the present invention;
FIGURE 4 shows still another circuit embodiment
of the present invention; and
FIGURES 5 - 7 show circuit modifications which can
be incorporated in the circuits of FIGURES 1 - 4.
In the following description, circuit elements
having the same reference designation are similar.
Referring to the embodiment of the invention
shown in FIGURE 1, television signal processing circuits
10 including, for example, a video detector, provide
luminance and chrominance signal components to a demodulator-
matrix circuit 12 which, in turn, provides color video
signals (e.g., red, green and blue image-representative
signals) to respective video output amplifier circuits
14, 16, 18 (the latter two being shown in block form)
Amplified video output signals are supplied from each
of the amplifier circuits 14, 16, 18 to respective control
(e.g., cathode) electrodes 20 of a color image reproducing
cathode ray tube 22 of, for example, the in-line gun type.
Since amplifiers 14, 16 and 18 are substantially identical,
only amplifier 14 is shown and will be described in detail.
Amplifier 14 comprises a cascode arrangement
24 of a common emitter transistor 26 and a common base
transistor 28. The emitter of transistor 26 is coupled
to a point of reference voltage (e.g., +6.2 volts provided
by a zener diode 50) via a diode Dl and a resistor R2
coupled in parallel.
The base of transistor 26 is supplied with video
-- 7 -- ~
~ 3~ RCA 71,111
1 signals via a network including a shunt resistor 42,
a white level adjustment variable resistor 44, a resistive
voltage divider 46, 54, 56 (-the latter serving as a black
level adjustment) shunting capacitors 48, 58 and a
resistor Rl arranged as shown. A capacitor Cl coupled
to the base of transistor 26 forms a low pass, bandwidth
limiting filter network with resistor Rl.
A relatively low direct bias voltage (e.g., +12
volts) is coupled to the base of common base transistor 28.
An active load circuit 30 is coupled to the collector of
transistor 28 and comprises a third transistor 32
arranged as an emitter follower, a "disconnect" diode 34
coupled between the emitter of follower 32 and the
collector of transistor 28, a cross-over distortion compen-
sation diode 36 coupled between the base of follower tran-
sistor 32 and the collector of transistor 28, a bias
resistor 40 coupled between a relatively high voltage
supply (e.g., +250 volts) and the base of follower tran-
sistor 32, and a current limiting resistor 38 coupled
between the collector of transistor 32 and the voltage
supply.
Degenerative voltage dependent direct current
feedback is provided from the output of amplifier 14
(i.e., the emitter of follower transistor 32) to the
base of transistor 26 via a resistor 52. Signals are
coupled to the cathode 20 of the red electron gun (R)
of cathode ray tube 22 via a series resistor 60. A
capacitive load for stage 14 comprises stray circuit
and transistor capacitances and the capacitive load
presented by cathode ray tube 22.
-- 8
- '~
3~6 RCA 71,111
In the following discussion of the operation
of the illustrated video amplifier, it will be assumed that
resistor 56 is adjusted so that a desired "black level"
(quiescent) voltage of the order of 150 volts is provided
at the emitter oE load transistor 32. A quiescent current
(e.g., 2.5 milliamperes) is then established in resistor
40, diode 36 and transistors 28 and 26. A quiescent
emitter current (e.g., approximately 3 milliamperes) is also
established in load transistor 32. The latter current
flows in feedback resistor 52 and establishes base bias
for transistor 26. In the quiescent state, diode 36 and
the base-emitter junction of load transistor 32 are each
forward-biased so that the cathode and anode of diode 34
are each at the same voltage. Diode 34 is therefore
non-conducting in the absence of siynals.
In operation, when a negative-going signal voltage
transition is provided across resistor 42, resistors 44
and 46 convert such signal to a decrease in base current
of transistor 26. The current through cascode transistors
26, 28, diode 36 and resistor 40 then decreases relatively
rapidly from its quiescent value, and the collector ~oltage
of transistor 28 and the base voltage of transistor 32
increase. When the collector voltage of transistor 28,
and therefore the base voltage of transistor 32, rises
by approximately 0.7 volts, load transistor 32 conducts
heavily and the load capacitance of the cathode ray tube
electrode 20 is charged via the low impedance voltage source
represented by the emitter of transistor 32.
The feedback resistor 52 aids in lineari-
zing the transfer characteristic of the stage by reducing
_ g _
... .
..,
. ;Z 3 ~ ~ 6
RCA 71,111
I crossover distortion (i.e., the delay in the start of
the rise of the ou-tput voltage), as well as establishing
a stabilized gain for the video output stage.
When a positive-going signal voltage transition
lS provided across resistor 42, cascode transistors 26,
28 are driven into conduction levels greater than the
quiescent level, thereby causing the collector voltage
of transistor 28 to drop. Diode 34 turns fully on when
this voltage drop reaches about 0.7 volts and provides
a low impedance discharge path from the load capacitance
tcathode 20) through diode 34 and cascode amplifier 24
to reference potential.
The small signal behavior of the circuit provides
relatively wide bandwidth. Since the common emitter
transistor 26 need only sustain a relatively low reverse
breakdown voltage and dissipates only a small amount of
power as a result of its low collector voltage, a small
signal device readily may be selected to provide the
desired bandwidth.
Caseode stage 14~ exclusive of filter network
Rl, Cl and the network including diode Dl and resistor
R2, is deseribed in greater detail in U.S. Paten-t No.
4,096,517 mentioned previously.
Under certain conditions, video amplifier
stage 14 can produce unwanted high frequency harmonics of
the video signal applied to stage 14. The harmonics
can extend to within the frequency band of VHF signals
received and processed by television signal processing
circuits 10, causing interference with such received
signals.
-- 10 --
,
RCA 71,111
3~i6~
Signal distortion which can be produced by
the nonlinear conduction characteristic of -transistor
26 at a near cut-off cul~rent conductlon contributes
to the generation of higher order signal frequencies, or
harmonics of the fundamental frequency of the video signal.
The conduction nonlinearity o transistor 26 and attendant
signal distortion become increasingly pronounced as the
conduction of transistor 26 approaches cut-off. The
operation of transistor 26 at a relatively low quiescent
current level increases the likelihood of transistor
26 being caused to operate in a highly nonline~r region.
Transistor 26 can be caused to operate at
or near cut-off in response to a rapid, neyative-going
input signal amplitude transition or transient. A large,
corresponding positive~going signal amplitude transition
then appears at the emitter output of transistor 32. If
transistor 26 is then operating in the very nonlinear
region at or near cut-of~, excessive high frequency harmonics
extending into the VHF band are produced. Amplified high
frequency harmonics appear at the emitter output of
transistor 32, and are radiated by a conductor connecting
the output of video stage 14 to the electron gun input
of kinescope 22. Thus, the generation of RF harmonics
is facilitated by the operation of transistor 26 at low
quiescent current levels, since transistor 26 can then
be caused to operate at or near cut-off more readily than
in the case of higher quiescent current levels.
FIGURE la depicts a portion of an input video
signal waveform (e.g., .l volt peak-to-peak amplitude
or less) coupled to the end of resistor Rl remote from
-~,
RCA 71,111
1 the base of transistor 26. The input waveform has a
rapid, short time duration, ne~ative-going ampli-
tude transition tf and a shor-t duration, I~ositive- going
amplitude transition tr. Transition tf is in a
direction to reduce the conduction of transistor 26.
In this instance, it ls assumed that the fall time of
transition tf is less than the charging time constant
of the capacitive load, so -that a positive-going signal
as shown in FIGURE lb is produced at the emitter output
of transistor 32 in response to the reduced conduction
of transistor 26. A portion of the positive-going output
signal (FIGURE lc~ is coupled via resistor 52 and
combined with the input waveform (FIGURE la) at the
junction of resistors 52 and Rl to produce a combined
input signal waveform (FIGURE ld).
The combined signal waveform (FIGURE ld) contains
positive-going and negative-going overshoots developed
in response to the lagging response times of the rising
and falling amplitude transitions of the positive feed-
back signal (FIGURE lc). A negative-going amplitude
transition tlf associated with the negative-going overshoot
is in a direction to momentarily reduce the conduction
of transistor 26 toward cut-off. When this occurs, the
feedback loop is momentarily opened, and a significant
amount of nonlinear distortion and attendant high fre-
quency signal harmonics are produced.
The likelihood of transistor 26 being caused
to operate in the highly nonlinear region of at or near
cut-off is related to the amount by which the rise and fall
times of the output signal exceed the associated fall and
- 12
;~ ' .
RCA 71,111
~.23~'6Çi
rise times of the input signal. Increasing the output
load capacitance serves to increase the rise time of
the output signal and therefore the magnitude of the
negative-going overshoot of the combined input signal
5 (FIGURE ld).
The feedback signal can inhibit operation of
transistor 26 at or near cut-off in the presence of
negative-going input video signals of relatively lower
frequency, or signals having amplitude transitions of
relatively longer duration. If the duration of such
amplitude transitions is substantially equal to or
greater than the charging time constant of the capacitive
load, a positive-going output signal transition then
appear across the capacitive load in response to the
decreasing conduction of transistor 26. This output signal
transition exhibits a rising amplitude substantially in time
coincidence with the negative-going (falling~ amplitude
transition of the input signal waveform. In this event,
the combined input waveform will not contain amplitude
overshoots, and the peak amplitude levels of the combined
waveform will then correspond to positive levels Vl and
V2 (see FIGURE ld). Transistor 26 will then be maintained
in a conductive state due to the compensating effect of
the closely tracking feedback signal, since the feedback
signal is in a direction to increase the base current
drive of transistor 26 in this instance.
The undesired RF harmonics can be suppressed
by the bandwidth limiting low pass filter network including
resistor Rl and capacitor Cl. The networ]c Rl, Cl is
included in the base input circuit of transistor 26 to
- 13 -
~ z3~ RCA 71,111
I limit the bandwidth response of signals coupled to the
base of transistor 26 to the system signal frequency
bandwidth of from zero hertz to between 4-5 MHz, for
example. Resistor Rl additionally serves to increase
the driving source impedance of signals applied to the
base of transistor 26. Nonlinear signal distortion
is reduced since, as is known, the linearity of a
transistor amplifier is improved when d~iven by a high
impedance source. Resistor Rl also serves to reduce
the amplitude of the signal vol-tage appearing across
the base-emitter junction of transistor 26.
Radio frequency harmonic radiation can be
suppressed a ~reater amount by the operation of diode
Dl and resistor R2-
Diode Dl is a forward biased semiconductor PN
junction device having a nonlinear current conduction
characteristic in a region of low current conduction.
Diode Dl is coupled in series with and similarly poled
for forward current conduction with the PN base-emitter
junction of transistor 26, and is common to the input
and output circuits of transistor 26. The nonlinear
conduction characteristic of diode Dl is similar to that
of the base-emitter junction of transistor 26, so that
diode Dl provides nonlinear degenerative emitter (current~
feedback for transistor 26 to counteract the nonlinearity
which is otherwise exhibited by transistor 26, particularly
at low transistor current in the vicinity of cut-off
conduction.
The current of transistor 26 and consequently
that of diode Dl decreasesrapidly in response to the
.
.. .
~ 3~6~ RCA 7l,lll
1 negative-going amplitude transition of the combined
input signal. During this time, the impedance of diode
Dl increases in proportion to the decreasing emit-ter current
of transistor 26, thereby providing emitter degeneration
S or emitter current feedback (emitter feedback provided by
diode Dl is negligible at normal, higher currents due
to the small impedance of diode Dl in such case). sy
this mechanism diode Dl linearizes the conduction character-
istic of transistor 26 in the vicinity of very low current
conduction near cut-off. The more linear conduction
characteristic results in reduced signal distortion and
consequently RF harmonics of significantly reduced magnitude.
Thus diode Dl permits transistor 26 to tolerate
higher input signal overshoots (tlf~ in a direction to
reduce the conduction of transistor 26 toward cut-off.
The feedback loop including transistor 26 and resistor 52
remains closed and feedback remains effective. Nonlinear
signal distortions and attendant RF signal harmonics
are significantly reduced.
The adverse effects of large magnitude, negative-
going overshoots of the combined input signal are further
suppressed by filter network Rl, Cl. Resistor ~l forms
a voltage divider with the impedance presented by the base- ~:
emitter junction of transistor 26. Therefore, a combined
input signal of reduced magnitude appears across the .
base-emitter junction of transistor 26. Also, harmonic
frequencies above the system bandwidth of 4-5 MHz in this
example are associated with the rapid, negative-going
amplitude overshoot of the combined input signal (FIGURE ld~.
Such harmonic frequencies are attenuated by low pass filter
- 15 -
,
~.23~ RCA 71,111
1 network Rl, C1 as well as by input capacitances of tran-
sistor 26. Therefore, -the amplitude of the negative
overshoot and the associated tendency to reduce the
conduction of transistor 26 toward cut-off are further
diminished. Network Dl, R2 can be used either alone
or in combination with network Rl, Cl.
Resistor R2 limits the emitter impedance and
thereby the amount of emitter feedback during the interval
of decreasing transistors 26 conduction in response to
negative-going input signal transitions, since excessive
amitter feedback can impair the gain of transistor 26
and therefore upset the opera-tion of the feedback network
including transistor 26 and resistor 52. Resistor R2
can be omitted if the quiescen-t current of the video
output stage is sufficiently high such that the impedance
of diode Dl is not expected to exceed the value of resistor
R2 (220 ohms in this example).
Referring now to FIGURE 2, there is shown an
alternate form of a low current, low power consumption
common emitter video amplifier output stage 214 employing
similar conductivity type transistors. Input signals are
applied to a base electrode of an NPN common emitter
transistor 226. An NPN emitter follower transistor 232
and a "disconnect" diode 234 comprise an active collector
load for transistor 226. Output signals are developed
across a load resistor 240 and are supplied from the
emitter of transistor 232 via a resistor 260 to a capaci-
tive load as noted in connection with FIGURE 1.
~he operation of stage 214 is similar to that of
stage 14 of E'IGURE 1. Negative-going signal amplitude
- 16 -
~ ~3u6~ RCA 71,111
transitions are produced by conduction of transistor 226
through diode 234, and positive-going amplltude -transitions
are produced by conduction of emitter follower transistor
232. When the output signal appearing at the emitter
of transistor 232 is negative-going, transistor 226 is
conducting and discharges -the load capacitance of stage
214 through diode 234. When the conduction of transistor
226 decreases in response to a negative-going input signal,
its collector voltage increases, causing diode 234 to be
cut-off. The load capacitance of stage 214 retains the
voltage appearing at the emitter of transistor 232 until
the base voltage of transistor 232 is sufficiently high
to render transistor 232 conductive, at which time the
load capacitance discharges. As with the arrangement
of FIGURE 1, the value of load resistor 240 can be
greater than the load resistance of a Class A video
output stage, thereby reducing the bias current and power
consumption of the stage.
Bandpass limiting network Rl, Cl of stage 214
serves the same purpose as the corresponding network of
stage 14 of FIGURE 1, except that in stage 214 capacitor
Cl is directly connected across the base-emitter junction
of common emitter amplifier transistor 226. A diode D
and a resistor R2 of stage 214 serve the same purpose
as the corresponding elements of stage 14.
A degenerative feedback network 252 of stage
214 serves essen-tially the same purpose as the feedback
netw~rk including resistor 52 of stage 14 of FIGURE 1.
Feedback network 252 can comprise a resistive voltage
divider, for example, and can be coupled to the base input
- 17 -
~ 66 RCA 71,111
1 f transistor 226 as shown or can be coupled to an input
of a preamplifier stage (not shown) prior to stage 214.
Feedback network 252 can include video signal black
and white level adjusting circuits, as well as frequency
selective feedhack to provide peaking at one or more
selected video signal frequencies. The amount of AC
and DC feedback can be varied -to adjust the circuit gain
and operating point, respectively.
FIGURE 3 shows a low current, low power consump-
tion video output stage 314 employing complementary con-
ductivity type transistors 326 and 332. A video output
stage of this type is described in detail in the afore-
mentioned Technical Communications Bulletin published
... .. _
by Mullard Limited.
Briefly, bias for PNP transistor 332 is provided
by resistors 340, 343 and 376 in conjunction with an
operating supply voltage (+240 volts). A resistor
351 is coupled from the operating supply voltage to the
junction of respective collector resistors 371 and 372
of transistors 332 and 326. Input video signals are applied
the the
to / base of NPN transistor 326 and AC coupled to / base
of PNP transistor 332 via a capacitor 375, so that tran-
sistors 326 and 332 operate in antiphase or push-pull rela-
tion. Push-pull output signals from transistors 326 and 332
appear at the junction of resistors 371 and 372 and are
coupled through a-resistor 360 to an output terminal. A
capacitor 378 serves to provide sufficient AC gain for
transistor 332. Feedback network 352 is similar to feed-
back network 252 of stage 214 (FIGUR~ 2).
Transistors 326 and 332 each exhibit a nonlinear
18 -
-
~ ~3~6~ RCA 71,111
conduction characteristic in a region of low current
conduction at or near cut-off. A bandpass llmiting low
pass Eilter network Rl, Cl ls similarly arranged and per-
~orms the same function as the corresponding network
of FIGURE 2. A diode D1 arranged in the emitter circuit
of transistor 326 serves to inhibit cut-off of transistor
326 in the presence of negative-going input signals which
tend to cause transistor 326 to operate at or near cut-off,
as discussed in connection with the circuit of FIGURE 1.
Likewise, an additional diode D2 serves to inhibit operation
of PNP transistor 332 at or near cut-off in response to
positive-going input signal transi~ions which tend to cause
such operation.
FIGURE 4 shows a Class A video output stage in
5 accordance with the present invention. Input signals
the
are coupled to ~ base of a transistor 426 and amplified
output signals appear across a collector load resistor 473.
A degenerative direct current feedback network 452 is
coupled from the output to the input of the video output ~.
stage as mentioned in connection with FIGURES 1-3. Stage
414 also includes a low pass filter network Rl, Cl and
a degenerative feedback network Dl, R2, all arranged
as shown, to inhibit operation of transistor 426 at or
near cut-off in the presence of rapid negative-going
input signal transitions.
It is noted that a Class A video output stage
is less susceptible to operation at or near cut-off in
response to applied input signals, compared to low current,
low power consumption stages of the types shown in
FIGURES 1-3, since a Class A stage typically operates at
- 19 -
~ 3~6~ RCA 71,111
a higher quiescent current level. The generation of RF
harmonics can be suppressed by increasing the quiescent
curren-t level, although tllis undesirably increases power
consump-tion and operating -temperature. These and other
problems in a Class A stage can be alleviated in accordance
with an arrangement including networks Rl, Cl and D1, R2
as shown in FIGURE 4.
Each of FIGURES 5, 6 and 7 illustrates a modifica-
tion to the emitter circuit of the input amplifier transis-
tor (e.g., transistor 26 of FIGURE l~ of a video output10
stage in accordance with the present invention. The
emitter circuit of a common emitter transistor 526 of
FIGURE 5 includes a diode Dl and an additional resistor R3
the
coupled in series between / emitter of transistor 526
and a point of reference potential VR. The role of diode D
has been discussed previously. In this example, resistor
R2 is omitted (i.e., the value o R2 is infinite~. Resistor
R3 serves as an emitter degeneration resistor to improve
the gain stability of the video stage. Resistor R3 may
2~ be required when, for example, input signals (Vi~ are sup-
plied to / base of transistor 526 from a preceding pre
amplifier stage having significant gain, and direct
current degenerative feedback is provided from an output
of the video output stage to an input of the preamplifier,
thereby developing relatively high feedback loop gaint
Additional combinations of resistor R3, resistor
R2 and diode Dl in the emitter circuit of an input amplifier
transistor of a video outpu-t stage are shown in FIGURES
6 and 7.
Various transistors exhibiting a small collector-
- 20 -
~ 6~ RCA 71,111
base feedback capacitance (e.g., less than 2.5 picofarads~
which have been ~ound suitable for the load transistor
(e.g., transistor 32 in FIGURE 1~ :include the types
BFR 88, BF 391 or RCA types RCP lllC and BF 458. Suitable
types for common base transistor 28 of FIGURE 1 include
RCA types RCP lllC and BF 458. A suitable type for the
small signal transistor (e.g., transistor 26 of FIGU~E 1
is the type BC 147. A suitable type for diodes D
and D2 is the type BAX 13 or IN 914.
In video output stages of the type described,
values of resistor Rl can be chosen from between zero ohms
and 3.3 kilohms, for example. Similarly, values for
capacitor Cl can be chosen from between zero (when the
base capacitance of the input amplifier transistor alone
is sufficient to provide effective filtering~ and 56 pf, -
for example. Factors determinative of the values of
resistance and capacitance ultimately chosen include the
type of input amplifier transistor used, the quiescent current
of the transistors, and stray circuit capacitances. These
factors likewise apply to the selection of the values of
resistors R2 and R3. The value of resistor R2 can be
about 150 ohms or greater; the value of resistor R3 can
be from zero ohms to about 100 ohms, for example.
Although the invention has been disclosed in
terms of particular circuit embodiments, it should be
recognized that other arrangements can be devised by those
skilled in the art without departing from the scope of the
invention.
For example, transistors of opposite conductivity
3~ type from that shown can be employed, in which event the
- 21 -
~ ~3~ RCA 71,111
I conduction polarity of diodes Dl and D2 should be reversed.
Also, diodes Dl and D2 can comprise a transistor base-emitter
junction, a dlode-connected transistor, or similar device.
- 22 -
:, :
