Note: Descriptions are shown in the official language in which they were submitted.
~13534~
This invention relates to current sources and in partic-
ular to such current sources in which the magnitude of the output
current of the source is determined by the magnitude of a control
voltage.
Conventional current sources which are designed to pro-
duce an output current whose magnitude is a function of an input
signal to the source, are frequently provided with a negative feed-
back loop whose function it is to keep the output current as nearly
as possible to the required magnitude.
In practice such current sources may comprise a multi-
stage amplifier, with the input signal being applied to an input
of a first stage and the current output being provided by an output
stage. As long as the requirements of accuracy of transconductance
are not too stringent, the output stage frequently consists of a
single transistor, with the feedback signal being derived from the
emitter circuit of the transistor. ~owever, when a high degree of
accuracy is required, it has heretofore been necessary, in general,
to use so-called "Darlington" or other multiple transistors in the
output stage, which tend to increase the cost of the current
source, particularly in application involving high voltages.
It is an object of the present invention to provide an
accurate current source which does not require the use of multiple
transistors.
In accordance with the present invention, there is pro-
vided a current source arrangement in which the magnitude of current
flowing in an output path of a transistor amplifier stage is
arranged to be controlled by an input signal applied to an input
of the arrangement, said input signal being applied to the
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1~.353~9
transistor amplifier stage by way of a differential amplifier,
comprising means to derive a negative feedback signal whose magni-
tude is dependent upon that of said current in said output path,
means to apply said negative feedback signal to an input of said
differential amplifier, means to derive a further feedback signal
whose magnitude is dependent upon the value of input current to
said transistor amplifier stage, and means to apply said further
feedback signal as a positive feedback signal to an input of said
differential amplifier wh.ereby to reduce the dependence of the
current flowing in said output path on the gain of the transistor
amplifier stage.
In accordance with. the present invention, there is also
provided a current source arrangement comprising a differential
amplifier having inverting and non-inverting inputs and an output,
and a transistor having an input electrode, an output electrode,
and a control electrode, the input electrode being connected by
way of a resistive element to one pole of power supply means and
the output electrode being connected by way of a load circuit to
the other pole of said power supply means, the output of the dif-
ferential amplifier heing electrically coupled to the controlelectrode of the transistor, means to apply a control signal to
one input of the differential amplifier, feedback means to apply
to another input of the differential amplifier a first feedback
signal dependent upon the voltage developed in operation across
said resistive element, means to derive a second feedback signal
dependent on the current flow to the control electrode of the
transistor, and means to apply said second feedback signal as a
positive feedback signal to said one input of the differential
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~135349
amplifier.
According to one aspect of the present invention, in acurrent source arrangement in which the magnitude of current flow
in an output path of an amplifier stage is arranged to be dependent
on the magnitude of an input signal applied to an amplification
stage preceding said amplifier stage, there are provided also at
least two feedback paths from said amplifier stage to said preced-
ing amplification stage, one of which feedback paths serves to
inject into the amplification stage a negative feedback signal
whose magnitude is dependent on the current flow in said output
path~ another one of the feedback paths being arranged to transmit
a signal dependent on a current flowing in an input of said
amplifier stage such as to reduce the dependence of the current
flow in said output path on the
- 2b -
113S34~
.
~3--
gain of said amplifier stageO
In accordance with another aspect of the present
i~vention in a current source-arrangement in which the magni-
~ude of current flowing in an output path of a transistor ampli-
fier stage is arranged to be controlled by a control signal
applied to an input of the arran~ement 7 said co~trol si~nal
being applied to the transistor amplifier stage by way of a
differential amplifier to an input of which is also applied a
negative feedback signal whose magnitude is dependent upon that
10 of said current in said output path, there are provided means to
provide a further feedback signal to an input of said differen-
tial amplifier whose value is dependent upon the input current to
said transistor amplifier stage such as to reduce the dependence
of the current flowing in said output path on the gain o~ the
transistor amplifier stage.
~ he control signal and the said further feedback signal
are conveniently applied to a non-inverting input of the differ-
ential amplifier, and the negative feedbac~ signal to an inver-
ting input thereof. ~he negative feedback signal may be derived
20 from the voltage drop across a resistive element in the emitter
circuit of a~ output transistor in the transistor amplifier
stage. ~he said further feedback signal is conveniently derived
from a first resistive element in the control signal input path
of the transistor amplifier and applied to the non-inverting
input via a second resisti~e element, with the control signal
being applied to the same input of the differential amplifier
via a third resistive element.
In accordance with a further aspect of the present
invention a current source arrangement comprises a differential
30 amplifier havin~ inverting and non-inverting inputs and an out-
put, and a transistor having an input electrode, an output
electrode, and a control electrode; the input electrode being
connected by way of a resistive element to one pole of power
supply means and the output electrode being connected by way of
a load circuit to the other pole of said power supply means, t~e
output of the differential amplifier being electrically coupled
to the control electrode of the transistor, means to apply a
control signal to one input of the differential amplifier~
: 1~353~9
--4--
eedback means to apply to another input of the differential
a~iplifier a signal dependent u~on the voltage developed in
Gperations across said resistive element, and further means
for deriving a signal dependent on the curren~ flow through
the control electrode of the transistor, said further means
bein~ electrically connected to ~aid on~ input o~ the di~fer-
ential amplifier.
Preferably~ said differential amplifier is a high gain
amplifier having substantially linear amplification characteri-
stics, of the kind known as operational amplifiers.
Said ~urther means for deriving a signal dependent onthe current flow through the control electrode of the transistor
conveniently comprises a first resis~ive element connected be-
tween the output of the dif~erential amplifier and the control
electrode of the transistor, thereby also electrically coupling
said output to said control electrode, a second resistive ele-
ment by means of which the output of the differential amplifier
is connected to said one input thereof, and a tnird resistive
element connecting said one input to a source of said control
signal whereby said control signal is applied to said one input.
~ he control signal applied in operation of~the current
source to said one input may be a DC signal voltage derived from
a resistance divider network.
Said control signal may also include an A.C~ component
superimposed on the D.C. signal in a known manner.
A current source arrange~ent in accordance with the
present invention is suitable for use with electronic line units
for telephone systems.
A current source arrangement in accordance with the
present invention will now be described by way of example with
reference to the accompanying drawing, of which:
~ igure 1 shows the current source arrangement sche-
matically;
~ igure 2 shows the current source arrangement diagram-
matically; and
~ igure 3 shows, in diagrammatic form, a circuit incor-
porating two complimentary current source arrangements in push-
pull modeO
. ~ .
~L1353
_5_
Referring first to ~igure 17 a current source inaccord~nce with ~ne present invention comprises a high gain
linear am~lification stage provided by an operational amplifier
1~ and an amplifier stage in the form of a ~unction transistor 2,
whose control electrode, or base, 8 is electrically coupled
wi~h th~ output o~ ~he am~ icr lo ~he input el~ctrod~, o~
emitter 7, of the transistor 2 is connected to the inverting
input of the amplifier 1 and also~ via a close tolerance resistor
B ~ to a voltage rail 9. Connected to the non-inverting input
10 of the amplifier 1 is a voltage source 11 supplying a co~trol
signal voltage to the amplifier and hence to the base 8 of the
transistor 2, the value of the control voltage determining
the current flow at the output electrode, or collector 6, of
the transistor 2. Additionally control means 4 and 5 are conn-
ected to the non-inverting input and the output respectively of
the amplifier 1, arranged to derive a signal dependent on base
current of the transistor 2 and to apply a signal to the non-
inverting input of the amplifier 1 such that the dependence of
the collector current on the gain of the transistor is reduced.
20 A load circuit (not shown) to be driven by this current source
arrangement is connected in use to the collector 6 of the tran-
sistor 20
A signal indicative of the base current is derived by
means 5 and is fed by mea~s of the control means 4 into the non-
inverting,input of the amplifier, any deviation of this signal
from its design value as determined by the control voltage
causing a variation'in the signal applied to the non-inverting
input. A consequent change in current flow t'nrough the emitter
and hence the resistor 3 results in a change of the voltage
applied to the emitter and thus compensates for this signal
variation thereby bringing the collector current nearer to the
intended valueO
~ he operation of a particular embodiment of this con-
stant current source will become apparent from the description
below with reference to ~igure 2 of the drawings, in which parts
easily identifiable as being equivalent to components of ~igure
1 carry the same referenceO
_ .. . .. .. ... .. . . . .
~5349
~ eferring now to ~igure 2, the output of the operat-
ional amplifier l is coupled by means of a resistor 16 to the
ba~ 8 OL~ the tra~sistor 2~ ~Jhose emitter 7 is connected to the
voltage rail 9 by wa~ of a resistor 3 and to the i~verting
in ut of the amplifier 1 through a resistor 14 which serves as
a feedback path and can also be arranged to e~ualize the source
impedances to the two inputs o~ the amplifier. A resistance
divider network comprising resistors 18 and 19 connected across
the voltage rails 9 and 10~ from which power is also supplied
10 to the amplifier lg provides the biassing D.C. control vo~ta~e
onto which may be superimposed an A.C. control voltage in the
form of an A.C. signal applied to the terminal 20 of the capaci-
tor 210 ~he biassing voltage derived from the resistance net-
work 18, 19 and~ where applicable, the A.C. signal is applied
to the non-inverting input of the amplifier 1 through a resistor
17, with a resistor 15 connecting this input to the output of
the amplifiex. ~he influence of resistors 14, 18, and 19 on
the A.~. performance of the current source is negligible. The
accuracy of the current source can be made dependent solel~J on
20 resistor matching, thus obviating the need to adjust each indivi-
dual source in dependence on the individual performance charac-
teristics of the transistor used, as the following calculation
shows, in which:-
Vin , the voltage obtained from the resistance divider network
Vref= the voltage at the non-inverting input of the amplifier
V0 = the voltage at the output of the amplifier
Vb = the voltage ~etween base and emitter of the transistor
Ib = the base current of the transistor
Ie = the emitter current of the transistor
3 Ic ~ the collector current of the transistor
re ' the dynamic emitter-base resistance of the transistor
R3 = the resistance value of resistor 3, etcO
Summing currents at the non-inverting input of the
operational amplifier, one obvains:
Vin ~ Vref ~VO - Vref ~ o (1)
Rl~ Rlg
,
1.
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~135349
--7--
Rearranging this expression gives:
Vref = Vi 15 , -- ~ VO P~l~ (2)
R17 ~ R15R17 + R15
3ut; since current flow th~ough the resistor R3 will tend to
stabilise at a value such that the volta~e at the emitter
electrode 7 of the transistor 2 is virtually equal to Vref:
V = V~ef + Vb ~ IbR16
Defining
VbDc ~ Vb ~ Iere (4a)
Vb ~ref R e , (4b)
writing re/R = k and substituting for Vb from equation (4b)
in equation (3) yields:
VO = Vref (1 ~ k) ~ VbDc + IbR16 (5)
Using this equation (5) in equation (2) gives
V = Vin R15 ~ VbDc R17 ~ IbR16R17
ref l5 - kR17 R15 - kR17 -R15 - kR17 (6)
Xowever,
I ~ Vref Ib ' (7)
and by substitution from equation (6)
I = VinR15 ~ VbDcR17 ~ IbR16R17 Ib (8)
R3(~15 kR17) R3(R15-~R17~ 3( 15 17)
In order to make the collector curre~t Ic independent of the
base current Ib, and hence of the current amplification ~,
where
c (8a)
it is necessary to ensure that
R3(R-15-kR17) - 9 . (9)
,,
.... . . . . . ..
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1~35349
~ . .
--8--
or given k is small~ to a first order that
R~
6 ~ ~ (10)
~ 15 R17Thus under D.C. conditions the accuracy of the current source
subject to th~ restrictions (9) a~d ~10) abov~ is indaed
dependent only on resistor matc`ning and independen~ of the
cu~rent gain ~ of the transistor. ~ny deviation of Ic from
thepredicted value is caused by variations in VbDc and re-
sistive tolerances which result in i~perfect cancellation of
the base current Ib and hence in an error in the transconduct-
ance dIc of the circuit, i~e~ the variation of collector
d Vi~
current Ic in dependence on Vin.
~he error in the quiescent current under D.C. con-
ditions is the same as that in the A.C. case dealt with below,
except for the additional error due to VbDc variations. As
this error is su~stantially independent of the absolute value
of the collector current Icg only the differential of Ic with
respect to VbDc needs to be calculated. ~hus, frOm equation (8)
dIc _ Rl
dVbDc Rl~-k~17) R~ '~
which on account of equation (10) yields
dIC
dVbDc
and therefore
dIC = dVbDc ( 11 )
R16 .. I
i~e. the effect of variation in VbDc as defined by e~uation l~
(4a), on the collector current Ic is inversely proportional
to the resistance value of resistor 16.
~ lthough being sufficient for a calculation of the
accuracy of the constant current source under D.C. conditions,
': " ,`~ ~. ' :
~, :
-, ~
~ 53
_9_
the first order approxi~ation expressed in equation (9) above
is not precise enough to permit an assessment of the accuracy
of the current source under A.C. condi-tions~ A good approxi-
mation is, however, possible by using a sensitivit~ analysis,
for a given resistor R3, involving the transconductance
d c
m dVin
and the current gain ~ of the transistor~
Rearranging equation (8), using the relationship (8a)
between Ic and Ib, and defining a resistor ratio X as
X = ¦ 1 - R16R17
R3(R15 - kR17)
gives
. gm R3 r 15 1 x r 1 (12)
l R15 - kR17 ~ L ~ +
~he sensitivity factor S for a fractional change
m caused by a fractional chan~e A ~ is
gm
defined by ~ gm = S 4 ~ , providing (13)
gm
a measure for the dependence between the two variables.
In the limit of infinitesimal change.s dgm and d~
expression (13) becomes
dgm = S d~ . I
gm ~ (14).l
iOe~ gm x dg (15
Substit~ting for gm from e~uation (12) gives
gB x dgm = X . (16)
m d~ -X ~
.
: ~
1135349
--10--
It follows that the dependence f gm on ~ is only
slight by vir~ue o~ X being a very small number approaching 0
in the case of ideal resistor. matching. Ideally9 the ~
~epandence o~ the transconductance is removed co~pletely but
for practical values of X there still is a very slight
dependence on account of the inite curr0nt gain.
~ xpressions similar to (16) can be derived for the
sensitivity of the current source to changes of the trans-
conductance gm with respect to R3 and X, ~iz:
3 dgm ~ -1 (17)
gm dX
and
X x dgm = -X ~18)
gm dX X ~ ~
Equation (17) shows that variations in gm are direct-
ly propor~ional to variations in R3~ underlining the neeessity
for a very low tolerance resistor ~.
The ef~ect of variations in the resistor ratio on
gm is very low as shown by (18), the physical explanation being
that the tolerance of resistor ratio X proportionally effects
the base current, but the base current itself is only a small
20 proportion of the total collector current.
~ he performance of the circuit shown in ~i~ure 2
has been evaluated in practical tests, with the resistor ratio
X being realized to an accuracy of 0.03%O Measurements
carried out showed no discernible difference over the audio
frequency range between the predicted and the actual trans-
conductanee.
Using di~ferent transistors 1 with eurrent gains
between 20 and 100, worst possible resistor tolerances of
0.1% and a perfeet operational amplifier, the error between
3o predicted and measured performance is still less than ~ 0.2%
under ~.C. conditions~
~ ypical VariatiOnS in VbDc for a power transistor
. are I 100 mV resuiting in a change of the eolleetor ourre~t
~ ..' ' :
.
1~35345~ .
~11--
o approximately ~ 5G~A~ i.e. ~ 002% of the quiescent current.
he near-independence of the output current with res-
ec~ to the gain of the transistor as achieved in the fore-
going current source arrangement allows easy matching of two
or more sucn current source arrangements.
Althou$n a current so~rce i~ accordance ~ith the
invention will give, for the same applied DC control signal
voltage, the same current output over a large range of tran-
sistor gains, the value at which the source output stabilizes
10 nay vary slightly between different sources, for instance on
account of resistor tolerances. These variations will, in general,
be small enough to be o~ no conse~uence when replacing one source
with another. ~owever, if two current sources are simply
connected in series to work in a push-pull mode7 e.g~ to drive
a telephone subscriber's line~ even the slightest mismatch
between the current outputs of the two sources may present
problems in that, as each of the current sources attempts to
keep the common output current to a value appropriate to its
own stable state, one or the other of the sources may saturate
~0 and cease to operate in the above described manner. A circuit
arrangement designed to overcome this problem is shown in ~igure
3.
~ hus, each of the two current sources making up the
circuit comprises an operational amplifier 1, ll, whose output
is connected, by way of a resistor 16, 161 to the base of the
transistor 2,21 respectively, with transistor 2 being an n-p-n
transistor, and transistor 21 being a nominall~ matched, comple-
mentary p-n-p transistor. ~he use of a matched pair of comple-
mentary transistors 2,21 provides for the impedance presented
3 by each current source to the respective line of the two wire
line 31 to bee~ual, since thereby the collectors 6 and 61 of
the transistors 2 and 21 are connected respectively to the
positive line (+) and the negative line (-) of the two wire
line 31. ~he otner end of the two wire line 31 is connected to
a load circuit (not shown) such as e~g. a subscriber's instru-
ment
,~
~ .
11;~5349
-12-
~ he feed back paths of each of the t~o current
s~u-ces~ vhat is the n~ga~ive ~eed back path via resistors 1~
ar~d i4 , and the further ~eed back path including resistors 15
and 16, and 151 and 1~1 respec~ively, are identical in arrange-
rent and function to the corresponding feed back paths of the
circuit sho~n in ~igure 20
~ he power sup ly to the arrangement is by means o~
voltage rail 29, carr~ing a negative voltage of suitable magni-
tude~ and grounded voltage rail 30.
~lso connected between the voltage rails 29 and 30
is a resistor divider networX com~rising resisto~s 22, 23, 2~9
25, all having the same value7 and eoual resistors 19 and 191.
~ his resistor network determines the average voltage
o~ the two wire line~ i.e. the mean of the voltages on the posi-
tive and the negative line of the two wire line 31.
If the two current sources of the arrangement are
perfectly matched~ the average voltage of the line pair lies
half way between the voltage levels at rails 29 and 30. Ifg
however, the t~o current sources are imperfectly matched, that
is to say that they stabilize individually at different current
levels of collector current for the same applied control input
voltage, the average voltage of the two wire line ~1 moves away
from this half way point, in a direction so as to decrease the
voltage between the collector and the corresponding current
rail of that transistor which draws the higher current and
increase the corresponding voltage at the other circuit. This
- shi~t of the average voltage at the two wire line, also termed
common mode shift causes an e~ual and opposite change, with
respect to the nearest current rail~ of the input voltage levels
Vin and Vinl, the change bein~ such that the current flow
through that transistor~ which initially drew the higher curr-
ent, is reduced and the current ~low through the other tran-
sistor is increased. ~his common mode shi~t is arrested when
both transistors dra~ the same current, with the average vol-
tage stabilizing at the new value. ~esistors 22 to 25 thus form,
in conjunction with resistors 19 and 191, a third ~eed back
path which ensures that the collector currents of the two tran-
;
: ~ ~
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:1135349
--13--
sis~ors 2 and 21 a re equal~ i.e~ that lc - ICl.
A diffarential voltage change, on the other hand, which
causes t~e two lines of the two wire line 31 to move indivi-
dually away from the average voltage in opposite directions,
leaves the average voltage unchanged~ and thus will produce no
cha~e in ~he levels of Vin and ~in IL trAerefore an anti-
phase AC signal is applied to the terminals 20 and 201S and
the output currents Ic and ICl vary~ in antiphase, in accordance
w_th that signal, no ~eed back effect will be produced. Pro-
10 vided also, that the resistors 22 to 24 are approximately equalto twice the appropriate value of resistor 18 of ~igure 2 in
order to leave the above calculations unchanged, the third feed
back path does not~ therefore, interfere with the ~C operation
of the arrangement.
In a similar way, the curren~s Ic and ICl are not
affected by any differential signal produced within the two
wire line~ such as may be produced in a subscriber's instrumen~.
~ s aforesaid, the collector currents Ic and ICl are
largely unaffected b~ differential signals generated witnin the
two wire line circuit, but respond only to either a common
mode voltage shift, due to e.g. imperfect matching of the two
current sources or an as~Jmmetry in the line circuit on account
of lea'~age currents, or to &ntiphase signals applied to the
terminals 20 and 201. ~herefore, by roviding a further circuit
(not shown) which is unaffected by variations in the collector
currents, but de~ects differential signals which are generated
within the two wire line circuit, the present arrangement may
be incorporated in, and form part of an electronic nybrid cir-
cuit. ~he detection of such line generated signals may be
~0 achieved in the following way. When signals are sent to the
subscriber's instrument, that is when antiphase signals are
applied to the terminals 20 and 201, then the emmiter and coll-
ector vo]tages generated oy a given transistor are out of phase
with each other; and by suitable addition these voltages may be
made to cancel each other~ &nd conse~uently no output signal is
provided by said furtner circuit. Differential signals whicn
are generated within the line circuit do, however, produce in-
phase collector and emmiter voltages at each of the =ransistors,
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:
113534'13
--14--
resu ving in an outpuv signal at the said further circuit.
he arrangement including a said further circuit tnus provides
~or a separation oi incoming and outging signals as is required
~or an eiectronic h~brid circuit such as may be used in the
conversion from two wire to four wire transmission and vice
versa.
Resistors 26 and 27 7 and non-linear devices 28 &nd 29
on the positive and the negative line respectively form part
of an overvoltage or lightning protection arrangement. ~he
devices 28 and 29 may e.g. be non-linear resistors, or zener
diodes. In the constant current sources described above,
known means for providing the DC control voltages, other than
resistor divider networks ma~ of course be employed, and &ny
other modificatio~s of the current sources above, which are
obvious to those skilled in the art are included in the scope
of the present inventionO
.,
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