Note: Descriptions are shown in the official language in which they were submitted.
254
1 -l- RCA 69,827
HIGH FREQUENCY FERRORESONANT POWER SUPPLY FOR
A DEFLECTION AND HIGH VOLTAGE CIRCUIT
Thls invention relates to a high frequency
ferroresonant power su~ply for a deflection and hiah voltage
circuit.
In deflection and high voltage power supplies for
television receivers, the B+ supply voltage for the deflec-
10 tion circuitry and the high voltage ultor accelerating
potential typically are derived in two different manners.
The B+ voltage is derived from the AC line mains which is
rectified and filtered; whereas the ultor accelerating
potential is derived from rectified flyback pulses obtained
15 from a horizontal output or flyback transformer. With such
an arrangement, two relatively independent and costly
power supplies must be used.
To regulate the high voltage, either the high
voltage itself is regulated directly or the B+ voltage is
20 regulated typically through relatively complex electronic
series switching, or shunt regulators. Such circuits are
relatively costly and subject to failures which require
additional protective circuitry to disable the television
receiver under abnormal increases in high voltage.
Many television receivers include circuitry to
maintain a constant raster width with varying ultor beam
current. This can be accomplished by altering the B+
raster voltage so that it tracks the changing ultor voltage
in such a way that the raster width and thus the picture
30 size remains constant with changing ultor voltage. Typically,
the B+ voltage change is accomplished by inclusion of a
series resistor conductively coupled to the flyback trans-
former primary winding or by use of additional B+ regulator
control circuitry which senses beam current variations and
35 correspondingly changes the B+ voltage. In the former
approach, power may be unnecessarily dissipated in the
series resistor, while in the latter approach additional
circuit complexity and cost may be incurred.
Z54
1 -2- R~A 69,827
Some ~+ regulators employ a 60 Hz AC line mains
regulating transformer, such as a 60Hz ferroresonant trans-
5 former, to provide a regulated B+ voltage. Because opera-
tion is at the low frequency of 60Hz, a relatively large
and heavy transformer must be used. Furthermore, the high
voltage is then independently supplied by means of a
relatively large flyback transformer designed to accommodate
10 relatively large power flows.
Other television receiver regulator circuits,dis-
cussed in the prior art, regulate the high voltage by pro-
viding a flyback transformer which itself is operated in
the ferroresonant mode. Flyback pulses are coupled to the
15 flyback primary winding. The ultor high voltage winding
is then tuned to the desired frequency. Because the B+
supply is derived from a separate source such as the AC
line mains supply, a separate regulator circuit must be pro-
vided if the B+ voltage is also to be regulated. If the B~
20 voltage is unregulated, other circuitry may be required to
maintain a constant raster width.
In many typical flyback derived high voltage cir-
cuits, the high voltage provides a peak voltage substantially
less than the required ultor potential in order to reduce
25 the number of turns required of the high voltage winding.
A high voltage multiplier then steps up the voltage to the
required level. Since the design of many voltage multipliers
requires that both polarities of the AC voltage be used, it
is desirable for the multiplier to be driven by an AC voltage
30 with the magnitudes of the positive and negative polarity
portions close to being equal. If one polarity is much smaller
than the other, the capacitors and diodes active during this
polarity contribute very little voltage. This is thesituation
when using the flyback pulse as a source for a voltage
35 multiplier circuit. A sextupler multiplier using six diodes
and six capacitors is required to obtain a three times
multiplication when a low duty cycle flyback pulse is
applied to a voltage multiplier.
ll~V2S4
1 -3- RCA 69,827
In aeeordanee with a preferred embodiment of the
invention, a ferroresonant power supply for a defleetion and
high voltage circuit in a television receiver comprises a
5 source of alternating eurrent voltage. A ferroresonant
transformer has a magnetic core, a first winding coupled to
the source of alternating current voltage, and a high voltage
winding wound around a core portion of the magnetic core and
eoupled to a high voltage terminal for developing a high
10 voltage. A second winding is wound around the core portion
of the magnetic core and is eoupled to a scan supply voltage
terminal for developing a sean supply voltage. Means provides
sufficient capacitance to at least one winding of the
ferroresonant transformer for generating eireulating eurrents
15 for saturating the eore portions under the high voltage and
seeond windings during eaeh eycle of the alternating current
voltage. This provides a regulated high voltage and regulated
scan supply voltage. A deflection switch is coupled to a
deflection winding for generating trace and retrace intervals
20 each defleetion cycle. A source of trace voltage is coupled
to the defleetion winding for developing seanning current in
the defleetion winding. First means eouples the regulated
sean supply voltage to the source of trace voltage. An ultor
terminal provides an ultor accelerating potential. High
25 voltage means is eoupled to the high voltage terminal and to
the ultor terminal for developing the ultor accelerating
potential from the regulated high voltage.
IN- THE DRAWING:
FIGURE l illustrates an electrical schematie diagram
30 of a high frequeney ferroresonant power supply for a
defleetion and high voltage eireuit embodying the invention;
FIGURE 2 illustrates a high frequeney ferroresonant
transformer eore and winding strueture used in the eireuit
of FIGURE l;
FIGURE 3 illustrates a cross-section of the trans-
former of FIGURE 2 along the line 3-3;
FIGURE 4 illustrates a different core arrangement
for the transformer of FIGURE 2; and
FIGURE 5 illustrates waveforms associated with
40 the eireuit of FIGURE l.
259~
1 -3a- RCA 69,827
In FIGURE 1, AC line mains voltage of illustra-
5 tively 120 VAC, 60Hz, is coupled to terminals 21 and 22 and
then to input terminals 23 and 24 of a full-wave bridge
rectifier 25. A current limiting resistor 30 is coupled
between terminals 21 and 23. A DC voltacle of illustra-
tively +150VDC is developed at a terminal 26 and is filtered
10 by a capacitor 27 coupled across terminal 26 and a terminal
28 which is the common ground current return terminal not
isolated from the AC line mains supply. A resistor 29 is
coupled to the +150 volt supply terminal 26. A low DC
voltage supply of illustratively +20DVC is developed at
15 the cathode (terminal 32)of a zener diode 33 that is
~14~254
1 -4- RCA 69,827
coupled to resistor 29.
A high frequency square-wave power oscillator 35
5 comprises a sinewave oscillator 36, a push-pull squaring
stage 37 and a power output stage 38. Sinewave oscillator
36 is self-oscillating and comprises a transistor 39 with
a collector electrode coupled to a resonant LC tank circuit
45 comprising a capacitor 40 and a winding 41a of a coupling
10 transformer 41. Collector voltage for transistor 39 is
obtained from the +20 volt supply that is coupled to a tap
terminal 48 of winding 41a through a resistor 42. A bypass
capacitor 43 is coupled to tap terminal 48.Resistor 42 reduces
the 20V to 17V and capacitor 43 aids in removing the riPPle
15 from the rectified 60Hz input voltage. AC feedback for main-
taining oscillator 36 in a self-oscillating mode is pro-
vided by a capacitor 44 coupled between tank circuit 45 and
the base electrode of transistor 39. DC bias for the base
electrode is provided by voltage dividing resistors 46 and
20 47 coupled to resistor 42.
Oscillator 36 developes a high frequency sinusoidal
voltage in winding 41a. The resonant frequency of tank cir-
cuit 45 is selected, for example, to be close to the hori-
zontal deflection frequency, l/TH, of approximately 15.75
25 kilohertz. Horizontal retrace pulses 67 obtained from a
utility horizontal flyback transfoxmer 68 is coupled to
a synchronizing input terminal 69 of oscillator 36. Retrace
pulses 67 are AC coupled from terminal 69 to the emitter
of transistor 39 through a capacitor 70 and a resistor 71
30 of voltage dividing resistors 71 and 72. Retrace pulses 67
synchronize the frequency of oscillator 36 to the horizontal
deflection frequency by turning off transistor 39 within
the horizontal retrace interval.
The high frequency sinusoidal voltage in winding
35 41a of oscillator 36 is coupled by means of a winding 41b
of transformer 41 to the bases of push-pull transistors 49
and 50 respectively through resistors 51 and 52. A center
tap of winding 41b is grounded. Squaring stage 37 converts
the sinusoidal voltage developed by oscillator 36 into a
~14~1~S4
1 -5- RCA 69,827
square-wave voltage of the same frequency. The square-wave
voltage is more suitable than a sinewave for driving power
5 output stage 38.
The high frequency square-wave voltage developed
by squaring stage 37 is coupled from a winding 53a of a
coupling transformer 53 to the bases of push-pull power
output transistors 54 and 55 through a winding 53b of
10 transformer 53 and through respective resistors 56 and 57.
A resistor 58 and a capacitor 59 coupled in parallel are
coupled between a center tap of winding 53b and the common
junction point of the emitters of transistors 54 and 55.
Resistor 58 and capacitor 59 function to provide a negative
15 bias voltage at the bases of the power output transistors.
A diode 60 is coupled across the collector-emitter
electrodes of transistor 54, with the cathode of diode 60
coupled to the collector of transistor 54. Similarly a
diode 61 is coupled across the collector-emitter electrodes
20 of transistor 55 with the cathode of diode 61 coupled to
the collector of transistor 55. Diodes 60 and 61 function
to limit the peak voltage of undesirable voltage spikes
which might damage the transistors.
Power output stage 38 provides a high frequency
25 alternating square wave voltage 64 at its output terminals
62 and 63, the collector electrodes respectively of tran-
sistors 54 and 55. Voltage 64 functions as a source of
unregulated energy or as an excitation voltage for a high
frequency ferroresonant transformer 65. An input or primary
30 winding 65a is coupled across output terminals 62 and 63 of
power output stage 38. The supply voltage for power output
stage 38 is obtained from the unregulated +150 volts DC at
terminal 26 which is coupled to a center tap terminal 66
of primary winding 65a.
High frequency ferroresonant transformer 65
comprises primary winding 65a, a low voltage secondary wind-
ing 65b, a high voltage secondary winding 65c, and a
magnetic core 165. As illustrated in FIGURE 2, magnetic
2S4
1 -6- RCA 69,827
core 165 comprises two core portion 165a and 165b. Core
portion 165a is formed as a C-shaped member. Core portion
5 165b is formed as a relatively thin rectangular slab of
màgnetic material with a relatively large surface area to
volume ratio.
As illustrated in FIGURE 2, primary winding 65a is
wound around the center section of C-shaped core portion 165a-
10 ~ow voltaqe secondary winding 65b is wound around slab coreportion 165b. High voltage winding 65c is concentrically
wound around low voltage winding 65b. Each of the secondary
windings 65b and 65c may be layer wound around cylindrical
coil forms 265b and 265c respectively. Other suitable
15 winding arrangements may be substituted, as, for example,
high voltage winding 65c being layer wound directly over
low voltage winding 65b. Segmented pi windings for low and
high voltage windings may also be used. Alternatively,as
illustrated in FIGURE 4, core 165 may be a rectangular core
20 formed of 2 C-shaped cores 765 and 865 butted together along
their legs with the legs 765a and 865a of the two C-shaped
cores being of reduced cross-sectional area. Low voltage
winding 65b and high voltage winding 65c, not illustrated in
FIGURE 4, are then concentrically wound around legs 765a and
25 865a as in FIGURE 2, with input or primary winding 65a being
wound around the opposite leg as shown in FIGURE 2.
As illustrated in FIGURE 1, a conductor of low
voltaqe secondary winding 65b is coupled to a terminal 101 and
another conductor is coupled to a ground current return
30 reference terminal 102 that is conductively isolated from
the AC line mains supply. This terminal may be at earthground
potential. Low voltage secondary winding 65b is coupled to a
scan supply voltage terminal 301 through a half-wave recti-
fier 401. The high frequency alternating current voltage
35 developed by low voltage winding 65b across terminals 101
and 102 is half-wave rectified by rectifier 401 and
filtered by a capacitor 501. A B+ scan supply voltage of
illustratively ~120 volts DC is produced at scan supply
114~ZS~
1 -7- RCA 69,827
voltage terminal 301.
Other tap conductors of low voltage secondarywinding
5 65b are brought out from the winding and are coupled to
respective rectifiers 403~405 to provide low DC voltages of
+30 volts, +72 volts and +210 volts at respective terminals
303-305. Filter capacitors 503-505 are respectively coupled
to the cathodes of diodes 403-405.
A horizontal deflection circuit 73 comprises a
conventional horizontal oscillator and driver circuit 74, a
deflection trace switch 76 comprising a damper diode 77 and
a horizontal output transistor 78, a horizontal retrace
capacitor 79, and a series coupled arrangement of a hori-
15 zontal deflection winding 80 and a trace capacitor 81. The
voltage Vt across trace capacitor 81 functions as a source
of trace voltage for horizontal deflection winding 80.
During each horizontal trace interval, trace switch 76 is
conducting and couples the trace voltage Vt across horizontal
20 deflection winding 80, thereby developing the required
horizontal sawtooth scanning current in the deflection wind-
ing.
To obtain the trace voltage Vt, trace capacitor 81
is coupled to the B+ scan supply voltage termina' 301
25 through the primary winding 68a of utility flyback trans-
former 68. Thus the average or DC value of trace voltage
Vt substantially equals the B+ scan supply voltage of +120
volts DC.
During the horizontal retrace interval, with
30 trace switch 76 cut off, horizontal deflection winding 80
and horizontal retrace capacitor 79 resonate for one-half
cycle of oscillation. The horizontal retrace pulses
developed in primary winding 68a of utility flyback trans-
former 68 are transformer coupled to fly~back secondary
35 windings 68b and 68c. Terminals 82 and 83 of secondary
winding 68b couple utility retrace pulses to such cir-
cuitry as the blanking and horizontal sync circuits.
Secondary winding 68c functions as the source of retrace
pulses 67 used to synchronize oscillator 36 of the high
ZS4
1 -~- RCA 68,827
frequency square-wave power oscillator 35.
In many conventional television receiver power
5 supply circuits, the ultor accelerating potential is de-
rived from rectified retrace pulses. In the circuit of FIGURE
1, however, it is the high frequency alternating current high
voltage developed across high voltage secondary winding
65e that generates the ultor voltage. This alternating
10 eurrent high voltage developed aeross terminals 106 and 107 is
reetified and multiplied by a voltage multiplier cireuit 84
eomprising three diodes 85-87 and three eapaeitors 88-90.
The cathode of diode 87 is coupled to an ultor terminal U,
at which terminal an ultor accelerating potential of illus-
15 tratively +27 kilovolts DC is developed. An intermediate DChigh voltage developed at the eathode of diode 85 may serve
as a foeus voltage for the focus electrode of a television
receiver cathode ray tube.
With ferroresonant transformer 65 providing both
20 a high voltage in high voltage secondary winding 65c and
a low voltage in secondary winding 65b, both the ultor
aceelerating potential and the B+ scan supply voltages are
regulated without the necessity of relatively complex and
failure-prone electronic regulator circuitry. To regulate the
25 voltages aeross secondary windings 65b and 65c, a resonating
eapaeitor 91 may be eoupled to low voltage seeondary winding
65b at terminal lOl,as illustrated in FIGURE 1, or to another
winding wound around thin slab eore portion 165b. The
value of capacitor 91 is selected such that capacitor 91 and
30 low voltage sesondary winding 65b resonate near the frequeney
of the exeitation souree, that is, near the 15.75 kHz
frequency of the high frequency alternating eurrent voltage
64. Under certain circumstances, as described below,
capacitor 91 may be omitted to reduce cost. If suffieient
35 winding eapaeitanee exists in the high voltage seeondary
winding, an additional eapaeitor will be unneeessary.
The eireulating resonant current flowing in
winding 65b and capacitor 91 aids in magnetieally saturating
Z59~
1 -9- RCA 69,827
the core under both low voltage winding 65b and high voltage
winding 65c during each half-cycle of the circulating current
5 oscillation. By so saturating the core, the induced voltage
in both secondary windings 65b and 65c are thereby regulated.
As illustrated in FIGU~E 5a, the voltage V65a
which appears across primary winding 65a of the high
frequency ferroresonant transformer 65 is a symmetrical
10 square wave voltage of frequency 15.75 kHz with a periodof TH
= 63.5 microsecond. The high voltage across secondary winding
65c is also a relatively symmetrical square wave voltage
V65c as illustrated in FIGURE 5b. The square-wave developed
by low voltage winding 65b across terminals 101 and 102 is
15 illustrated in FIGURE 5c, with the flat topped portion
occurring during conduction of rectifier 401.The input current
i65a to primary winding 65a (from terminal 26) is illustrated
in FIGURE 5d and is a relatively constant current except near
the switching instants of transistors 54 and 55, near the
20 leading and trailing edges of waveform V65a of FIGURE 5a.
The resonant or circulating current ic flowing in
capacitor 91 and low voltage secondary winding 65b between
terminals 101 and 102, illustrated in FIGURE 5e, aids in
magnetically saturating the core portion 165b under both high
25 voltage secondary winding 65c and low voltage secondarYwinding
65b. Saturation will occur near the peak portions 94 and 95
of the circulating current waveform ic.
To provide for saturation of core portion 165b
while the core portion 165a under primary winding 65a
30 remains unsaturated, the cross-sectional area of slab 165b of
FIGURE 2 is made smaller than the cross-sectional area of C-
shaped core 165a. As illustrated in FIGURE 3 by the sectional
view taken along the line 3-3 of FIGURE 2, the cross-section-
al area a=(t)~-(w), the product of the thickness t and
35 width w of slab 165b, is relatively much smaller than the
cross-sectional area A=wf, the product of the sides wand f of
C-shaped core 165a. For the value listed below, the ratio a/A
equals approximately 0.19.
It may be desirable to limit the temperature
`-` 114C~Z54
1 -10- RCA 69,827
increase in the saturatinq core portion 165b under low voltage
~econdary winding 65b and high voltage secondary winding 65c
5 after energization of the circuit of FIGURE l.The saturation
flux density BSat of the core material decreases with in-
creasing temperature. Because,in a ferroresonant trans-
former,the voltages developed across secondary windings 65b
and 65c is a function of BSat, it may be desirable to limit
10 the temperature rise by providing a secondary winding and
core structure with increased cooling capability. This
temperature rise may occur due to increased core losses at
high frequency operation and due to relatively large I2R
losses from such factors as the relatively large high
15 frequency saturating circulating current.
As illustrated in FIGURES 2 and 3 core portion
165b comprises a thin slab of thickness t, width w, and
length 1. With typical values listed below,the ratio of ~
surface area to volw~le of slab 165b is relatively larqe, 40
20 to 1, for example, thereby permitting increased cooling of
the thin slab.
Furthermore, the inner diameter D of cylindrical
coil form 265b is also larger than the thickness t of slab
165b, thereby permitting windings 65b and 65c to be loosely
25 wound around slab core portion 165b with a relatively
large air space between the windings and the core. Con-
vective cooling of the core is thereby enhanced. Such a
core configuration is described in U.S. patent
~o. 4,242,245, issued 14 April 1981, to F.S. Wendt.
, . . . .
If the increase in core temperature is of relatively
little concern, a conventional core geometry for core
portion 165b may be used,such as a square or circular cross-
35 section core with the windings 65b and 55c more tightlywound around the core. As illustrated in FIGURE 5b, the high voltage V65c
across high voltage winding 65c is a relatively symmetrical
square wave with the magnitude of positive portion 92 approxi-
ZS~
1 -11- RCA 69,827
mately equal to the magnitude of negative portion 93.With a
peak-to-peak voltage swing of 18 kilovolts for V65c,for example,
5 only three rectifiers and two or three capacitors are required
in high voltage multiplier circuit 84 of FIGURE 1 to obtain
an ultor voltage of +27 kilovolts, for example. Diodes 85
and 87 rectify the positive portions of V65c, whereas diode
86 rectifies the negative portion. Thus, the ultor voltage
10 equals approximately twice the positive magnitude of V65c
added to the negative magnitude.Capacitor 90 may be omitted
if the conductive coatings on the kinescope provide sufficient
filtering capacitance. Conventional high voltage supplies which
rectify positive retrace pulses of the same magnitude as that
15 of positive portion 92 of FIGURE 5b may require five or six
diodes and associated capacitors in the multiplier arrange-
ment to obtain the same ultor voltage.
The described high frequency ferroresonant trans-
former system embodying the invention provides a regulated
20 B+ scan supply voltage and thus a regulated trace voltage
Vt, and also provides a regulated high voltage. With such
an arrangement, the design criteria for horizontal deflec-
tion circuit 73 may be considerably relaxed. For example,
with flyback transformer 68 no longer required to transfer
25 load energy to the ultor, the flyback transformer may be
considerably reduced in size, as relatively little load
current flows in the flyback transformer. With little DC
current flowing through horizontal output transistor 78, its
size, current and voltage ratings and heat sinking re-
30 quirements are reduced. With suitable circuit redesign, a
low impedance deflection winding 80 may be used, which
would then require a much lower voltage peak of 200V,
illustratively, instead of the relatively large horizontal
retrace pulse of l,OOOV typically being developed.
With the high frequency ferroresonant transformer
65 supplying both the B+ scan supply voltage and the ultor
high voltage, relatively good picture width stability may
be obtained without the use of electronic control circuitry
4U
- ~\
2~4
1 -12- RCA 69,827
or discrete series resistances. As beam loading of the
ultor terminal U increases, the ultor accelerating
5 potential decreases. The increased DC portion of the load
current flowing in high voltage wind:Lng 65c acts to de-
magnetize somewhat core portion 165b and to shift the
operating point of that core portion away from greater
saturation slightly towards the knee of the transformer B-H
10 hysteresis loop, thereby decreasing the high voltage
somewhat. However, with low voltage windlng 65b and high
voltage winding 65c sharing a common saturating core portion,
increased load current flowing in winding 65c will also
decrease the B+ scan voltage, thereby providing substantial
15 raster width regulation.
Alternatively explained, the existing leakage
inductances in the transformer 65 provide an increased
voltage drop with increased video beam loading causing both
the high voltage and the B+ voltage to decrease.The leakage
20 inductance between the high and low voltage secondary windings
65b and 65c,and the degree and location of core saturation
are adjusted for providing raster width regulation.
A relatively large number of winding turns is
required to develop the relatively high voltage across high
25 voltage secondary winding 65c. By proper choice of such factors
as winding configuration, layer separation and conductor wire
size, the interwinding stray or distributed capacitance may be
sufficilently large to enable the high voltage windinq 65c to
resonate so as to saturate core portion 165b,and thus requlate
30 the voltages in both the high and low voltage secondarywindings.
Such distributed resonating capacitance is illustrated in
FIGURE 1 by a capacitor 665 coupled across high voltage
winding 65c, although in actuality the total capacitance
is distributed along the winding turns.
With the distributedcapacitance 665 andhigh voltage
winding 65c providing the circulating current for saturating
core portion 165b, capacitor 91 is no longer required.
AginG or failure of discrete components that may cause an
114~3Z54
1 -13- RCA 69,827
increase of the high voltage is eliminated. Furthermore,
using a ferroresonant transformer to provide the high voltage
5provides intrinsic high voltage protection capability, as
changes in a winding inductance or in the capacitance value
of a resonant capacitance will typically result in a loss
of ferroresonant operation and a decrease of the winding
voltage.
With ferroresonant transformer 65 providing both
the high voltage and the B+ scan supply voltage, a relatively
large rise in core temperature after initial circuit energi-
zation may be tolerated without substantially affecting
raster width. Because the high voltage seconday winding 65c
15and low voltage winding 65b are wound around a common core
portion 165b, a decrease in BSat with an increase in core
temperature decreases both the ultor and the B+ voltages,
thereby providing a substantial degree of raster width
r~gulation.
Typical values for a high frequency ferroresonant
transformer 65 as illustrated in FIGURES 1-3 and using the
distributed capacitance 665 associated with high voltage
winding 65c, are as follows:
CORE 165: C-shaped core portion 165a with a cross-
25sectional axea of 0.360 square inch (2.32 cm2), outer leg
length of 2.0 inch (5.1 cm) and center section length of
2.8 inch (7.1 cm); thin slab core portion 165b of thickness
t = 0.110 inch (2.79 mm), width w = 0.610 inch (15.5 mm),
length 1 = 2.8 inch (7.1 cm), cross-sectional area of 0.067
30square inch (43.2 mm2); core material is a ferrite with
BSat of around 4000 gauss at 25C, such as Ferroxcube 3E2A
from Ferroxcube Corporation, Saugerties, New York, or such
as RCA 540 from RCA Corporation, Indianapolis, Indiana.
PRIMARY WINDING 65a: 30/40 Litz nylon wrap
35insulated enameled copper wire, layer wound with four layers,
center tapped, bifilar wound, 200 turns total, with no
insulating layers between winding layers; winding length
of 1.55 inch (3.94 cm).
LOW VOLTAGE WINDING 65b: cylindrical coil form
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1 -14- RCA 69,827
265b with an inner diameter D = 0.715 inch (18.2 mm),
outer diameter of 0.850 inch (21.6 mm); and a length of
1.675 inch (42.55 mm). Winding 65b with 25/38 Litz nylon
5wrap insulated enameled copper wire, bifilar, layer wound,
with 190 turns total with four layers of about 48 turns in
each layer; a fifth layer of four turns provides a cathode
ray tube filament voltage of approximately 6.3 volts, 900
miliampere; winding length of 1.675 inch (42.55 mm).
HIGH VOLTAGE WINDING 65c: cylindrical coil form
265c with an inner diameter of 1.150 inch (29.21 mm), an
annular thickness of 0.060 inch ~1.52 mm), and a length of
1.050 inch (26.67 mm). Winding 65c of #38 gauge (0.1007 mm)
enameled copper wire, layer wound with 32 layers with 147
15turns in first 31 layers and 43 turns in last layer, each
layer separated from the other by a 0.002 inch (0.05 mm)
to 0.004 inch (0.10 mm) mylar insulator; total number of
winding turns equals 4600; winding length of 0.75 inch
(19 mm).
