Note: Descriptions are shown in the official language in which they were submitted.
3~
LARGE DYNAMIC RANGE MULTIPLIER FOR A
MAXIMAL-RATIO PREDETECTION DIVERSITY COMBINER
Background of the Invention
This invelltion relates to diversity combining systems
and, more particularly, to a large dynamic range mult.iplier
for maximal-ratio divers.ity comb.iners.
The need for space d.iversity combining arises in moblle
radio systems because the rad.io-frequency (RF) s.ignal pa-th
between a mobile transmitter and a base receiver is general-
ly not l.ine of sight, but instead consists of many reflected
and scattered RF signal paths having varying amplitudes and
phases. Furthermore, in mobile radio systems operating at
relatively high frequencies, for example at 800 MHz, deep,
rapid fading, commonly referred to as Rayleigh fad.ing, must
be contended with. By utilizing an antenna array having
space diversity, the foregoing effects may be substantially
reduced. According to space diversity, antennas of an an-
tenna array are spaced at predetermined distances from one
another, for example, at a distance of at least one-quarter
wavelength from one another. The probability that deep
fades will occur simultaneously at all antennas of a space-
diversity antenna array will be extremely low. Thus, a
composite signal formed by coherently combining each of the
RF signals from a space-diversi-ty antenna array w.ill theo
retically have a s.ignal level at least as high as the
strongest RF signal received by the antenna array.
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One practical prior art technique of coherently combin-
ing the antenna RF signals from a space-diversity antenna
array is known as "equal-gain predetect.ion diversity combln-
ing". Exemplary equal-gain predetection diversity combiners
are those described in an art.icle by D. Brennan entitled,
"Linear Diversity Comblning Techniques", publ.ished in IRE
Proceedings, June 1959, at pp. 1075 to 1101 and in U.S.
patent no. 3,471,788 to W.S. B.ickford et al. In these prior
art comb.iners, the antenna slgnals are converted to inter-
mediate frequency (IF) signals which are then cophased wi-th
one another and thereafter linearly combined to provide a
composite IF signal. For example, the IF signals developed
from each antenna RF signal may be phase allgned w.ith a
locally generated signal of a reference frequency, or may be
phase aligned to a selected one of the IF signals, or may be
phase aligned with respect to the compos.ite IF s.ignal. Once
the IF signals from each antenna RF signal are cophased with
one another, they may then be linearly added by appropriate
circuitry to prov.ide a coherent composite IF signal which is
the vector sum of the individual IF signals.
In order to cophase each IF signal, prior art equal-
gain predetection diversity combiners include circuitry,
commonly referred to as a "branch", for dividing the IF sig-
nal into first and second portions, mixing the first portion
with a reference signal to provide a firs-t product signal
that has a phase equal to the difference in phase between
the first portion and the reference signal, and mixing the
first product signal with the second portion of the IF s.ig-
nal to provide a second product signal that will be cophased
with the reference signal. S.ince the second product signals
of each branch are cophased with one another, they may then
be linearly added by appropriate circuitry to provide the
composite coherent IF signal. If the second portion of the
IF signal and the first product signal were each amplified
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l,inearly so that the magnitude of each would be proportional
to the .input IF signal, the magnitude of the second product
signal would theoretically be proport,ional to the square of
the magnitude of the ,input IF signal. However, ,in the con-
ceptual design of the prior art equal-gain comb.iner, the
f.irst product signal is ampl.itude l.im.ited prior to the input
to the second stage of mixing. Consequently, the second
product s.ignal will be directly proportional to the magni-
tude of the input IF signal rather than to .its square.
In such equal-gain combining systems, it is necessary
that all of the antenna ~F signals must have substantially
the same mean signal level due to the fact that the first
product signal is amplltude l.imited prior to the second mix-
.ing. ~eca~se the first product signal is amplitude limited,
the second product signal from the second stage of mix,ing
will not be proportional to the square of the magnitude of
the input IF signal. Thus, if IF signals rece,ived by all
branches do not have substantially the same mean signal
level, the composite IF signal may be significantly degraded
in signal-to-noise ratio s.ince a weak signal received by one
branch w,ill be weighted substantially equally with a strong
signal received by another branch.
The foregoing inadequacy of prior art equal-gain pre-
detection diversity combiners may be improved by utilizing
the prior art combining technique known as "maximal-ratio
predetection combining". In such maximal-ratio predetection
combining systems, signals are not limited prior to the sec-
ond mixing in each of the branches. It is desired that all
signals are proportionally related to the input IF signal so
that the magnitude of the second product signal will be pro-
portional to the square of the magnitude of the input IF
signal. As a result, branches receiving strong signals will
receive more emphasis than branches receiving weak signals.
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Since Rayleigh fading experienced in 800 MHz systems
may c~use instantaneous amplitude variations between the RF
signals received at different antennas of a space-diversity
antenn~ array that are in excess of 40 decibels (db~, the
linear dynamic range of the branch circuitry must accommo-
date IF signals ha~ing at least a 40 db dynamic range ~nd
second product signals having amplitude variations in excess
of 80 db, which is twice the dynamic range of the input IF
signals due to the squaring by the second mixing operation.
Thus, the particular circuit implementation for providing
the second mixing operation must provide substantially
idealized multiplication over an extremely large dynamic
range.
A prior art product multiplier capable of providing the
.desired performance over an output dynamic range in excess
of 80 db has not been practically achieved in the past.
Commercially available line~r inte~rated-circuit balanced
mixers such as the Motorola~ C15g6 have been utilized as the
second mixer in maximal ratio combiners designed for mili-
tary applications~ These integrated circuit mixers consist
of a quad differential amplifier with cross-coupled outputs
to provide full-wave balanced multiplication of the two
input signals. Each differential pair is powered by a con-
stant current source. Such a mixer will not accommodate
input signals having a dynamic range in excess of 30 db,
since its linear output dynamic range is only 50 to 60 db.
A doubly~balanced FET mixer, such as that described by
Highleyman and Jacob in the article~ "An Analog Multiplier
Using Two Field Effect Transistors", IRE Transactions on
Communications Systems, Vol. CS-10, pp. 311-317, September,
1962, may also be used for the second mixing operation with
some expected impr~vement in dynamic range, but without any
appreciable reduction in complexity or cost~ :
Accordingly, it is an object of the present invention
to provide an improved low-cost, large dynamic range product
,~
multiplier for a maximal-ratio predetection diversity com-
biner that coherently combines a plurality of amplitude and
phase varying input signals of substantially the same fre-
quency.
It is another object of -the present invention to pro-
vide an improved low-cost, large dynamic range product mul-
tiplier for a ~aximal-ratlo predetectlon divers.ity combiner
su.itable for use in a d.iversity recelver for coherently com-
b.ining RF signals havlng a dynamic range in excess of forty
decibels (40 dB).
Summary of the Invention
According to the .invent.ion, there is provided a maxl-
mal-ratio predetection diversity combiner for coherently
combining over a predetermined dynamic range a plurality of
input signals of substantially the same frequency that have
unknown and varying phases and magnitudes with respect to
one another. The plurality of input signals are co-phased
with each other within the combiner so that these signals
are phase-coherent with each other prior to being linearly
summed together at the output of the combiner. Furthermore,
the amplitude of each input signal is, in effect, squared
within the combiner prior to the l.inear summation to give
greater emphasis to the siynals having the larger magni-
tudes.
For each input signal, the maximal-ratio predetection
combiner includes: circuitry for dividing the input signal
into first and second portions; a first mixer for multiply-
ing the first portion of the lnput slgnal with a reference
signal that is produced within the combiner to provide a
first product signal having a phase that is the difference
between the phase of the input signal and the reference sig-
nal; a filter network for prov.iding a variable phase shift
,
to the first product signal with the phase shift being a function
of the frequency of the first product signal; and a second
mixer for multiplying the second portion of the input signal
with the phase-shifted first product signal to provide a second
product signal that is substantially independent of the phase
of the origianl input signal. The second product sig~als, which
are each cophased with the reference signal, may then be linearly
combined by combining circuitry to provide a composite coherent
IF signal.
The second mixer of the maximal-ratio diversity combiner
preferably is a field-effect transistor devic~ that propor-
tionally multiplies, over the predetermined dynamic range of the
combiner, the second portion of the input signal and the first
product signal to provide the second product signal. Thus, the magni-
tude of the second product slgnal is proportional to the square o
the magnitude o~ the respective input signal. Therefore, since
strong input signals receive more emphasis than weak input
signals, the signaL-to-noise degradation from the weak input
signals is greatly reduced. Since a single FET device provides
essentially ideal multiplication, the second product signal is
proportional to the square of the respective input signal over
the entire dynamic range of the input signal, which, in some
applications, may be in excess of forty decibels (40 db) for
the input signal.
More particularly, there is provided:
A maximal-ratio predetection diversity combiner for
coherently combining a plurality of input signals each having
a linear dynamic range exceeding forty decibels (40 db), having
substantially the same predetermined frequency and further
having unknown and varying phases and magnitudes with respect
to one another, said maximal ratio predetection diversity
combiner comprising:
means for generating a reference signal having a pre-
determined reference frequency for each input signal;
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means for dividing each input signal into first and second
portions;
first means for multiplying the first portion of each
input signal with the reference signal to provide a first
product signal having a phase that is the difference between
the phase of the input signal and the reference signal;
means for providing a variable phase shift to said first
product signal, said phase shift being a function of the
frequency of said first product signal;
second means for multiplying the second portion of each
input signal and the corresponding phase-shifted first product
signal to provide a second product signal that is substantially
co-phased with the reference signal and substantially independent
of the phase of the input signal, said second multiplying means
being comprised of only a single field-effect transistor (F~
said FET being predeterminedly biased for multiplying the
second portion of the input signal and the first product signal
to provide the second product signal, such that the magnitude of
the second product signal is proportional to the product of the
~o magnitudes of the second portion of the input signal and the
first product signal over substantially twice the dynamic range
of the corresponding input signal; and
means for intercoupling the second product signals developed
from each input signal to provide a phase coherent composite
signal.
There is also provided:
A diversity receiving system comprising antenna array means
having a plurality of substantially independent antennas for
receiving a radio signal of a predetermined frequency, each
antenna providing an input signal, the input signals from the
antennas having unknown and varying phases and magnitudes with
respect to one another; converting means coupled to the input
signals from the antennas for converting the frequency of the in-
put signals to an intermediate frequency; intermediate frequency
amplifying and filtering means coupled to the converted input
signals from the converting means for filtering the converted
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input signals; and maximal-ratio predetection combining means
coupled to the filtered input signals from the intermediate
frequency amplifyling and filtering means for coherently combining
the filtered input signals to provide a coherent composite signal,
said combining means including:
means for generating a reference signal having a predeter-
mined reference frequency for each filtered input signal;
means for dividing each filtered input signal into first
and second portions;
first means for multiplying the first portion of each
filtered input signal with the reference signal to provide a
first product signal having a phase that is the difference
between the phase of the filtered input signal and the reference
signal;
lS means for providing a variable phase shift to said first
product signal, said phase shift being a function of the
frequency o~ said first product signal;
second means for multiplying the second portion of each
filtered input signal and the corresponding phase-shifted first
product signal to provide a second product signal that is
substantially co-phased with the reference signal and sub-
stantially independent of the phase of the input signal, said
second multiplying means being comprised of only a single field-
effect transistor (FET), said FET being perdeterminedly biased
for multiplying the second portion of the filtered input signal
and the first product signal to provide the second product
signal, such that the magnitude of the second product signal
is proportional to the product of the magnitudes of the second
portion of the filtered input signal and the first product
signal over substantially twice the dynamic range of the
corresponding input signal; and
means for intercoupling the second product signals
developed from each filtered input signal to provide a phase
coherent compoiste signal.
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Brief Description of the Drawings
.
Fig. 1 is a block diagram of a diversity receiving system
that may advantageously utilize the present invention.
Fig. 2 is a block diagram of a maximal-ratio predetection
diversity combiner which may advantageously utilize the present
invention.
Fig. 3, including Figs. 3A and 3B taken together, is a
detailed circuit diagram of a portion of the circuitry in
each branch of Fig. 2, together with a portion of the circuitry
of Fig. 2 common to all of the branches.
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Detailed Description of the Preferred Embodiment
In Fig. 1, there is illustrated a diversity receiver
including six directional sector antennas in an antenna array
100, conversion stages 110-115, IF selectivity stages 120-125
and a maximal-ratio predetection diversity combiner 130. The
signal received by each directional sector antenna is tuned
to the same radio frequency by the conversion stages 110-115.
The diversity receiver of Fig. 1 may be advantageously utilized
at the base station of a mobile radio system, such as the
system described in the Federal Communications Commission filing
by American Radio Telephone Service, Inc., of Baltimore,
Maryland, entitled "An Application For A Developmental Cellular
Mobile and Portable Radio Telephone System In The Washington-
Baltimore Northern Virginia Area", filed on February 14, 1977,
and in copending application, Serial No. 345,723 by Frank J.
Cerny, Jr., and James J. Mikulski, entitled "Instantaneously
Acquiring Sector Antenna Combining System", filed on February
15, 1980 and assi~ned to the same assignee as the instant
application. Prior to the aforementioned copending application,
~0 radio-telephone systems have typically utilized omnidirectional
antennas, as shown in U.S. Patent No. 3,471,788, instead of
high-gain directional sector antennas as shown in this copending
application, for providing omnidirectional coverage. By
utilizing a maximal-ratio predetection diversity combiner 130
with a directional antenna array 100 comprised of a plurality
of directional gain antennas whose patterns are spatially
distributed to provide an omnidirectional pattern, the effective
coverage area of the antenna array may be substantially increased
while providing an omnidirectional receiving pattern. However,
systems utilizing antenna arrays
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comprised of either d,irect.ional or omnidirectional antennas
may advantageously utilize the present invention.
The signals received by each antenna of the antenna
array 100 are converted to an .intermediate frequency by con-
version stages 110-115, filtered by IF stages 120-125, and
applied to max.imal ratio predetect.ion combiner 130. The
combiner 130 continuously cophases the branch IF signals
received from each antenna of the antenna array 100 and
thereafter linearly adds these cophased branch IF s.ignals
together to provide a compos.ite IF signal. ~n combining the
branch IF signals, it is desirable that signal-to-no.ise
rat.io degradations which may be .introduced by low-level s.ig-
nals or deep nulls received by one or more of the sector
antennas of the antenna array 100, be avo.ided. Thus, the
entire diversity rece,iver should be capable of linearly
accommodating the expected dynamic range of each sector
antenna signal from the antenna array 100. The expected
dynam,ic range should be linearly accommodated not only by
the conversion stages 110-115 and the IE' stages 120-125 but
also by the maximal-ratio predetection diversity combiner
130. It has been found that, in order to provide optimum
suppression of the noise pops due to Rayle.igh fading, a
l.inear dynamic range of at least 40 db should be maintained
throughout the diversity receiver of Fig. 1. Thus, in order
to prov.ide a linear dynamic range for the maximal-ratio pre-
detection diversity combiner 130, it is necessary that the
second multiplier (230 of Fig. 2) provlde essentially ideal
multiplication so that the multiplier product signal is pro-
port.ional to the product of the magnitudes of the multiplier
input signals over the entire dynamic range of each multi-
plier input signal. Prior art combiners have failed to pro-
vide such proportional multiplication for signals having a
linear dynamic range in excess of 40 db.
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In accordance with the present invention, it has been
found that a FET device will provide essentially ideal
multiplication over the wide dynamic range encountered in
maximal ratio predetection combiners. The FET device is
preferably biased such that the qulescent gate-to-source
voltage is approximately one-half of the gate pinchoff volt-
age. The -two signals to be multiplied are applied to the
gate and source terminals of the FET device, respectively.
By filtering the FET drain current, id~ at the d~ifference
frequency, wl-w2, the ~ollowing equation is obtained:
id = (IDSs/vp2) Vl V2 CS(Wl-W2)t
where IDSs = the steady state drain saturation
current;
Vp = the gate pinchoff voltage;
Vl = the magnitude of the input signal at
radian frequency wl; and
V2 = the magnitude of the input signal at
radian frequency w2.
The development of the foregoing equation is described
in detail in my prior U.S. patent no. 3,716,730.
According to the present invention, a FET device will
provide substantially ideal multiplication as predicted by
the foregoing equation as long as the FET input signal lev-
els are properly ratioed. For example, a Siliconix U-310
FET iS capable of providing a product signal having a dynam-
ic range in excess of 130 db. For the U-310 FET, one deci-
bel of output compression (indicating the limit of linear
product multiplication) occurs when the gate signal level is
-3 dbm or the source signal level is +9 dbm. Assuming that
a receiver requires an input signal level of -112 dbm to
provide a 20 db quieting signal, the threshold levels for
the gate signal may be set at -62 dbm and the source signal
at -50 dbm (total of -112 dbm). Thus, the signals at the
gate and source terminals of the FET may each vary over a
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dynamic range of 59 db (e.g. for the gate, -3dbm -[-62dbm] =
59db) without producing output compression. Furtherm~re, if
the signal-to-noise ratio of the receiver IF circuitry is 7
db for a 20 db quleting signal, the siynals at the gate and
source terminals may each vary over a dynamic range of 66 db
(i.e. 59db + 7db = 66db). Thus, the FET product signal has
a dynamic range that .is 132 db, twice the dynamic range of
the s.ignals at the gate and source terminals.
In Fig. 2, there .is .illustrated a more deta.iled block
d.iagram of the diversity receiver of Fig. 1, as shown in the
aforementioned copending application. The diversity rece.iv-
er of Fig. 2 shows only three of the six branches of the
diversity receiver of Fig. 1, although any number of branch-
es may be utilized in practicing the present invention.
Branches 200, 201 and 202 are comprised of substantially
identical c.ircu.itry, each branch provid.ing a product signal
that .is both phase coherent with the other branch product
signals and proport.ional to the square of the magnitude of
the signal received by its respective sector antenna.
In the diversity receiver of Fig. 2, the frequency of
local oscillator 208 determines which radio channel the
diversity receiver is tuned to. The RF signal received by
each branch sector antenna 220 is combined by mixer 221 with
the signal from local oscillator 208 to provide an IF signal
at 21.4 MHz. The IF signal from m.ixer 221 is then applied
to IF bandpass filter 222, which may be a monolithic band-
pass f.ilter of conventional design similar to that described
in U.S. patent no. 3,716,808. The filtered IF signal from
filter 222 is then applied to IF amplif.ier 223. The output
from IF amplifier 223 is split and fed forward via two paths
to mixer 230. The first portion of the IF signal is linear-
ly amplif.ied by IF ampl.ifier 229 and thereafter appl.ied to
mixer 230. IF amplifier 229 linearly ampl.ifies the first
portion of the IF signal to provide a signal level
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that is within the input dynamic range of mixer 230. The
second portion of the IF signal is appl.ied to mixer 225
together with -the 1.72 MHz composite IF signal which is fed
back v.ia amplifier 206 and fllter 224. By feed.ing ~ack the
S IF slgnal, the IF strlp of the dlverslty recelver forms a
closed feedback loop that is regenerative on noise. Thus,
the randomly varylng phase of each branch IF slgnal relatlve
to the compos.ite IF signal is added into the closed loop via
mixer 225/ and then subtracted out at mixer 230. By th.is
process, the random phase variations are removed from each
branch IF signal in relation to the composite IF signal.
The result is that each branch IF output signal is cophased
to the composite IF signal. Alternatively, in other ar-
rangements, the branch IF signals need not be cophased with
the composite IF signal, but may be cophased to a locally
generated reference signal.
Referring back to branch 200, the component of the out-
put signal from mlxer 225 at the dlfference frequency of
19.68 MHz has a relatlve phase which ls the dlfference be-
tween the phase of the branch IF slgnal at 21.4 MHz and the
composlte IF signal at 1.72 MHz. This resultant output slg-
nal is llnearly amplified by second IF amplifler 226 and
applled to bandpass filter 227 to provide a variable phase
shift to the resultant signal. Filter 227 may be a two-pole
crystal fllter havlng a center frequency of 19.68 MHz and
passband bandwidth of 2 KHz. The phase shift, provlded by
fllter 227, is a function of the absolute frequency of the
resultant signal. The slgnal from filter 227 ls linearly
ampl.ified by thlrd IF ampllfler 228 to provlde a signal
level that is within the input dynamlc range of mixer 230;
thls ampllfied signal is applied to the second input of
mixer 230. Mixer 230 multiplles the ampllfied 19.68 MHz
difference product signal from ampl.ifier 228 wlth the ampll-
fied 21.4 MHz IF signal from ampllfler 229 to provlde a
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resultan-t output product signal having a 1.72 MHz difference
frequency that is cophased with the compos.ite IF signal,
thus being substantially free of the random phase variations
of the input IF s.ignals. The phase d.ifference result.ing
from mixer 225 .is subtrac-ted from the phase of the ampl.ified
21.4 MHz IF s.ignal from amplifier 229 to produce ~he 1.72
MHz difference product s.ignal from m.ixer 230, which is co-
phased wlth the compos.ite IF signal and free from the random
phase variations of the branch IF signal. The resultant
output signal from mixer 230 is proportional to the square
of the level of the input IF signal to that branch. The
resultant output signals from the mixer 230 of each branch
are linearly added together to form one composite IF s.ignal
at 1.72 MHz. This composite IF signal is the output s.ignal
from the maximal ratio predetection diversity combiner; it
.is fed to a 1.72 MHz IF bandpass filter 204 and fourth IF
amplifier 205. The composite IF signal from amplif.ier 205
may then be applied to a conventional demodulator that is
appropriate for recovering the method of information modula-
tion being utilized within the system. The composite IF
signal from amplifier 205 is further amplified and then
amplitude limited by fifth IF amplifier 206 to provide a
high-level amplitude-limited composite IF signal which is
applied to each mixer 225 through each filter 224. Filter
224 may be either a bandpass filter or a low-pass filter
having an operating fre~uency of 1.72 M~z. Automatic gain
control is applied to all branches of the combiner by con-
trolling the gain of each IF amplifier 223 with an AGC con-
trol voltage from AGC circuitry 207. This control voltage
may be obtained by rectifying, amplifying, and low-pass fil-
tering a portion of the composite IF signal from the output
of mixer 230.
Figs. 3A and 3B taken together illustrate in detail
the corresponding blocks of the circuitry of Fig. 2. Each
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branch 200, 201 and 202 ~f Fig. 2 contains essentially the
same circuitry that is illustrated in Figs. 3A and 3B. In
Fig. 3A, the branch IF signal applied to FET device 310 is
the ~ignal provided by IF filter 222 of Fig. 2. Each desig-
nated portion of Figs. 3A and 3B corresp~nds to the block of
Fig. 2 identified in parentheses after each designation.
Referring to Fig. 3A, the branch IF signal, having a
nominal frequency of 21.4 MHz, is amplified by FET device
310, which may be a Motorola~ N~04, before application to
the IF amplifier and A~C stage consisting of am~ ier 311,
which may be a Motorola~ C1350P. The output of ~his IF
amplifier is then divided into two portions, one portion
being applied to FET device 313, and the other portion being
applied ~o amplifier 315. FET device 313, which may be a
Motorola~N4416, is biased to operate as a mixer~ The FET
device 313 mixes the signal from the this IF amplifier with
a feedback signal derived from the composite IF signal. The
output of FET device 313 is then transf~rmer-coupled to
amplifier 314, which may be a Motorol ~ MC1350P. Amplifier
314 is biased to provide linear amplification to signals
from FET device 313. The output ~f amplifier 314 is then
filtered by a two-pole monolithic crystal filter 316, which
has a center frequency of 19.6B MHz and a passband of 2 KHz.
The monolithic crystal filter 316 provides for both n3rrow-
band filtering and phase-shifting that is ~ ~unction of the
frequency of the signal from amplifier 314. The phase-
shifted output from monolithic crystal filter~316 is then
applied to amplifier 317, which may be an RCA~ A3~86.
Amplifier 317 likewise provides linear amplifica~ion ~o sig-
3Q nals from the mon~lithic crystal filter 316. The output of
amplifier 317 is transfcrmer-coupled to the source terminalS
of ~ET device 31B. The second portion of the branch IF sig-
nal from IF a~plifier 311 is coupled to amplifier 315, wh~ch
also provides linear amplification. The output fr~m ampli
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fier 315, which may be a Motorola~MC1350P, is ~ransformer-
coupled to the gate terminal ~ of FET device 318.
In accordance with the present inventi~n, FET device
318 proportionally multiplies the signals from amplifiers
317 and 315 to provide the 1.72 MHz difference pr~duct sig-
nal at its drain teI~unal D. The FET device 318 is preferably a large ~cwer FET
device, such as the Siliconi ~U-310. According to another
feature of the present invention, the product signals ~rom
the FET devices 318 of each branch may be combined simply by
interconnecting the respective FET drain terminals. The
drain terminals of FET devices 318 may be paralleled since
the drain of a FET is essentially a constant-current source
when the load conductance is at least an order of magnitude
larger than the output conductance of the FET device; the
paralleling of the drain terminals of the FET devices of
each branch does not degrade either the gain or th2 dynamic
range of the individual FET devices. According to yet
another feature of the present invention, additional ~ET
devices 3181 may be paralleled with the FET device 318 in each
branch to provide an increased dynamic range frsm the paral-
leled FET devices. Thus, nei~her the intercoupling of the
FET devices nor the parallelinq of ~dditional FET devices
requires any additional circuitry. In addition, the connec-
tions from the amplifiers 315 and 317 to the FET device 318'
2~ may be reversed, such that the amplifier 317 is coupled to
the gate terminal and amplifier 315 is coupled to the source
terminal indicated on Fiq. 3B by "G" and "S!! in parentheses.
The additional circuitry shown in Fig 3B is primarily
circuitry that is common to all of the branches of the
diversity combiner. The AGC circuitry includes transistors
319 and 320, which are capacitively coupled to the drain
terminals of the branch FET devices 318 for providing an AGC
voltage therefrom. The transistors 319 and 320 may be any
suitable silic~n transistors, ~uch as the Motorola MPS6517
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and Motorc>l(~MPS6513, respectively. The composite IF signal
is coupled to amplifier 322 and ~hereafter to limiter 321
for providing a feedback signal to FET device 313 of each
branch of the diversity combiner. The feedback signal from
limiter 321 is applied via a low-pass filter 323 to the FET
device 313 of Fig. 3A. Amplifier 322 ~ay be a Motorola(~)
MC1350P, and limiter 321 may be an RC 3086.
In summary, a maximal~ratio predetection diversity com-
biner has been described hereinabove which provides a com-
posite IF signal having a dynamic range in excess of 100 db.
The large dynamic range of the diversity combiner has been
achieved by utilizing an active FET device to proportionally
multiply linear IF signals over their entire dynamic range.
The circuit configuration for achieving this result is very
simple, not requiring a multitude of current sources, dif-
ferential pair arrangements or balanced circuit arrange-
ments. Thus, the cost of the inventive diversity combiner
has been substantially reduced. The diversity combiner of
` the present invention may be expanded to accommodate a large
number of branches simply by paralleling additional branches
with the drain terminal of the branch FET devices 318 of
Fig. 3B.
..
