Note: Descriptions are shown in the official language in which they were submitted.
Description
High Efficiency, Light Weight
Audio Amnlifier and Power Su~lv
.. ~ . .
Technical Field
This invention relates to methods for audio
signal amplification and to audio amplifier circuitry
and power supplies therefor.
Background Art
Solid state circuit components have brought
incredible reduction in the size, weight and cost
of audio amplifier circuitry and have also achieved
increased fidelity in sound reproduction as compared
~ with vacuum tube technology of a prior-generation.
In an attempt to exploit to the limit the potential
of solid state circuitry, audio engineers have striven
to provide the user with increased power ratings
while simultaneously achieving decreased distortion
levels. Their efforts have met with resounding
success, but have produced some undesirable side
effects primarily in the areas of increased weight,
cost and power consumption. For example, a-commer-
cially available state-of-the art 400 watt ampli-
fier typically weighs anywhere from 16 kg to over
38 kg depending upon the particular design and choice
of materials. Such amplifiers normally employ costly
components necessitated by the peak loads which they
must carry, and generate significant amounts of heat
which must be dissipated to avoid component damage.
With regard to the transformer weight problem
an obvious approach would be to reduce the number
of windings and/or the gauge of the wire making up
the transformer coils. However, reduction in the
number of windings also reduces the inductance in
the primary coil, thereby increasing idling currents
through the coil and contributing to both heat gen-
eration and increased power consumption. The conven-
tional method for achieving low idling currents in
the primary has been to use a large number of wind-
ings. This approach also requires a large number
of windings in the secondary to keep the voltage
in the secondary at the proper level. The other
obvious alternative for weight reduction, i.e. reduc-
tion in the wire gauge, is not an acceptable solu-
tion since the internal resistance of each coil would
be increased, leading to excessive heat generation
and power loss upon high power demands being placed
on the transformer. ~onventional wisdom has thus
taught the necessity of increasing the size and
weight of the transformer whenever a transformer
powered amplifier is redesigned for increased power
rating.
An alternative approach for reducing ~he overall
weight, size and cos~ of audio amplifiers has been
to reduce the total input power requirement without
decreasing output power capability~ Such increases
in amplifier efficiency permit the use of less costly,
lower weight power supplies, and can be achieved
by reducing the power dissipation which normally
attends the conventional use of output transistors
in the output stage of the amplifier. When such
power dissipation decreases are achieved, additional
weight and cost savings are realized beyond those
realized in the power supply since the weight~ size
and cost of the heat sinks normally required b~ the
output transistors in the amplifier ma~ also be
4~4
reduced.
U.S. Patent No. 3~426,290 to Jensen is represen-
tative of one known approach for increasing amplifier
efficiency by keeping the voltage supplied to the ..
output transistor of the amplifier very close to
the output voltage level, thereby permit~ing opera-
tion of the output transistor in a condition which
is at all times only slightly out of saturation.
When operated in this condition, the actual voltage
drop across the output transistor will be maintained
quite low and the power dissipated by the transistor
(equal to voltage across the transistor X current
through the transistor) will be correspondingly
reduced. A rather complex regulator is employed
in the Jensen circuit to maintain the desired voltage
supply to the output transistor wherein energy is
stored in an inductive capacitive circuit by means
of a switching transistor operated at high speed
in response to a control signal derived from the
audio input signal. By operating the switching
transistor in full "on" or full "off" condition to
maintain the desired voltage supply to the amplifier
output transistor, energy consumption by the combined
regulator and output transistor is reduced over that
which would be consumed by an output transistor
operated with a conventional fixed supply voltage~
While producing a decided advantage in amplifier
efficiency, the Jensen circuit is only truly effec-
tive if the switching transistor is operated at high
frequencies, which can in turn cause transient in
terference distortion in the amplifier output signal.
~.S. Patent No. 4,054,843 to Hanada discloses a
i similar circuit to that disclosed in Jensen.
An alternative approach to achiev;ng improved
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amplifier efficiency is disclosed in the patent to
Dryden (3,319,175) which discloses a stepped voltage
supply operated in response to the voltage level
of the amplifier output whereby the minlmum voltage
from the available power supply voltages sufficient
to achieve the desired amplification is applied
across the power amplifying elementO While useful
for the purposes disclosed, Dryden employs only a
single transistor as the power amplifying element
for each polarity of the output voltage and thus
the entire difference between the load voltage and
the connected supply voltage appears across the
output transistor. Significant power losses will
thus occur unless a large number of discrete supply
voltages are provided by the power supply circuitry.
Each such discrete voltage requires a separate ampli-
tude comparator and associated switching device thusadding significantly to the cost of the power supply.
Still another approach disclosed in the prior
art is illustrated in U.S. Patent No. 3,622,899 to
Elsenberg. In this patent a low power dissipation
amplifier circuit is disclosed including plural
transistors coupled in series to a load terminal
wherein the transistors are energized by respective
voltage sources having different magnitudes and
wherein the transistors are biased to operate as
amplifiers in sequence in response to an input signal
of increasing magnitude. This type of circuit causes
each output transistor to be driven into saturation
as the next higher voltage output transistor is
brought into operation, causing substantially the
entire voltage drop in the amplifier output stage
I (that is the difference between the supply voltage
and the load or output voltage) to appear across
only a single output transistor at any one time.
This arrangement of circuitry requires output tran-
sistors having substantial power ratings unless a
relatively ~arge number of output transistors and
discrete supply voltages are provided. Either ap-
proach will add to amplifier cost. The patents to
Woehner (3,772,606) and to Sampei et al. (3,961,280)
disclose circuit arrangements similar to that de-
scribed above with reference to the Elsenberg patent.
~he patent to Schade, Jr. (3,887,878) discloses
a transistor series amplifier wherein plural series
connected transistors in the output stage are biased
to share the total voltage drop in the output stage
to permit use of lower cost components. ~owever,
this patent fails to disclose a technique for re-
ducing the total power dissipation in such transis-
tors.
Still other techniques for reducing the cost
of amplifier power supplies have been disclosed in
the prior art. For example in the U.S. patent to
Munch, Jr. (3,542,953) a technique is disclosed
wherein a single power supply may serve two Class
B amplifier circuits designed to amplify the same
audio signal by phase inverting the input to-one
amplifier to cause the amplifiers to draw peak cur~
rent from the power supply in alternation. Munch,
Jr. does not, however, suggest how such a technique
can be employed in a system employing dual amplifiers
(such as in a sterophonic system) for amplifying
two separate signals.
None of the prior art systems discussed above
addre~sses directly the problem of reducing power
I supply weight and costs by modifying the supply
itself in a manner to employ less costly lighter,
weight components while maintaining the power supply
capabilities required by the amplifier circuit.
The patent to Chun (3,466,527) discloses a
.~ circuit for reducing the cost and size of a trans-
former based voltage supply circuit including a duty
cycle controlled switch in the A.C~ power supply
circuit of the transformer primary. The switch
functions to regulate output voltage from the sec-
ondary. However, the lower cost and weight capabil-
ity achieved by the concepts disclosed in Chun arederived by operating the duty cycle controlled switch
over only a quarter cycle volt-time integral and
do not in any way suggest how such a circuit design
could be employed in an audio amplifier circuit in
lS a manner to obtain power supply weight and cost
reductions based on the characteristics of the in-
coming audio signals.
Disclosure of the Invention
It is a general object of this invention to
overcome the difficiencies of the prior art by pro-
viding an amplifier circuit and power supply of signi-
ficantly reduced weight and cost achieved by simul-
taneously increasing the efficiency of the ampli-
fier output stage and modulating the energization
of the amplifier power supply in response to a charac-
teristic of the signal being amplified.
Another object of this invention is to provide
a transformer based power supply for an ampliier
in which the transformer primary is energized by
a pulsed supply which is duty cycle modulated par-
tially in response to the signal being ampliied.
The duty cycle is controlled in a manner to insure
that adequate power is available in the power supply
--7--
output energized by the transformer secondary while
at the same time minimizing the time during each
cycle when idling currents are flowing through the
primary winding of the transformer.
The above objects are achieved in one specific
embodiment by providing means to cause current pulses
to be transmitted to the primary winding of the trans-
former and circuit means responsive to the signal
being amplified in a manner to control the power
of the pulses transmitted to the primary winding.
This is done to increase and decrease power of the
pulses as the signal increases and decreases in
amplitude, so that the power means provides to the
amplifier means power of a magnitude related to power
requirements of the amplifier means to provide an
output corresponding to the signal. In one embodi-
ment, the pulses may be formed by a pulse generator
and the circuit means may include switch means op-
eratively connected to the primary winding of the
transformer to interrupt current to the primary
winding. There are other means to make the switch
means non-conductive at periodic intervals to cause
the current pulses to flow through the transormer.
The circuit means further comprises modulating
means to control the power of the current pulses
to the transformer primary winding. This is desir-
ably done by controlling the duration of open periods
of the switch means in a manner that pulses of shorter
duration pass through the primary winding when the
amplitude of the signal is relatively small, and
pulses of greater ~uration are delivered through
the primary winding of the transformer when the
- amplitude of the signal is relatively large.
The pulse generator based embodiment may also
include comparator means responsive to a first value
related to amplitude of the signal and to a second
value related to power then ava;lable from the trans-
former for the amplifier means. The comparator is
arranged to provide a control signai of a value
related to a difference ~etween the first and second
values `to provide an output of a magnitude corre-
sponding generally to the amplitude of the signal.
The first value is a voltage, the magnitude of which
increases and decreases with the amplitude of the
signal. The second value is a voltage rela~ed to
the voltage at an output end of said transformer.
Desirably, there is an absolute value detector ar-
ranged to receive the signal and to produce a direct
current output having a magnitude related to the
amplitude of the signal. The absolute value detector
means is arranyed to transmit its direct current
output to the comparator as said ~irst value. Prefer-
ably, there is also a non linear peak detector ar-
ranged to receive the direct current voltage fromthe absolute value detector and produce an output
voltage for the comparator. The output voltage of
the non-linear peak detector corresponds to the
output from the absolute value detector. The non-
linear peak detector is more responsive to relativelyrapid variations in amplitude in the signal and less
responsive to relatively small variations in ampli
tude of the signal. There is another comparing means
operatively connected to two ends of the secondary
winding of the transformer to produce a summing value
related to a difference in absolute magnitude of
the voltages at the ends of the transformer secondar~
i winding. This other comparing means is arranged
to produce a second control voltage related to the
summing value. The comparator means and the other
comparing means are arranged to transmit a control
output which is responsive to relative magnitude
of the first control cueput signal from the compara-
tor and the second control signal from the othercomparing means. The output is arranged to control
power of current pulses in the primary winding of
the transformer.
The pulse generator employed in the pulse gen-
erator embodiment generates control pulses at a pulse
frequency which is sufficient to cause each pulse
cycle to have a duration no greater than the minimum
response period required of the amplifier~ The
pulses are transmitted to switch means to turn the
lS switch means "on" and "off" at the pulse frequency
to cause current pulses to flow through the primary
winding of the transformer. In this manner, the power
output of the transformer may be rendered more respon-
sive to major amplitude variations of the signal~
In an alternative embodiment of the transformer
based power supply of this invention, the primary
of the transformer may be connected with a source
of alternating current combined with a switch means
arranged to cyclically interrupt the alternating
current in response to a control signal produced
by control means operatively connected to the switch
means to cause the switch means to be conductive
at selected portions of current cycles from the power
input terminal means. The control means causes the
switch means to be conductive for shorter periods
of time for lower power requirements of the amplify-
ing apparatus, and to be conductive for longer periods
of time for higher power requirements of the ampli-
Ey i ng appar at u s . Des i r ably, I he con tr ol me sns cau se s
~L6;~
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the switch means to be conductive during a latter
portion of each current pulse from said power input
terminal. In one embodiment, there is a rectifying
. means operatively connected with the power input
terminal means and the primary winding to cause only
positive current pulses to be directed to the primary
winding. The switch means comprises voltage respon-
sive switch means which becomes conductive at pre-
determined voltage levels of the current pulses.
In the-preferred form of this embodiment, the switch
means comprises a silicon controlled rectifier con-
nected in series between the power input termina~
means and the primary winding of the transformer.
In another embodiment of the transformer based
power supply, the power input terminal means is con-
nected to the primary winding to cause alternating
current to be delivered to the primary winding.
The switch means is a voltage responsive means to
cause the switch means to be conductive at predeter-
mined voltage levels during latter portions of thecurrent pulses. In the preferred form of this addi-
tional embodiment of the power supply, the switch
means comprises a first triac connected in series
with the primary winding. Operation of the--first
triac is governed by circuitry which responds to
both the magnitude of the signal being amplified
and the value of the voltage output from the trans-
former secondary. The triac is caused to fire during
precisely defined portions of the duty cycle, thereby
regulating the amount of current flow through the
transformer primary and the corresponding transfer
i of energy to the transformer secondary.
I In another form of the transformer based power
I supply, the triac switching means discussed above
4a~3~
further includes cut off circuitry for shutting off
current flow through the primary of the transformer
before the normal waveform of the alternating current
supply has returned to zero. Such cut off circuitry
may include a second triac which operates to com-
mutate the first triac into a non-conductive state~
In the method of the present invention/ a series
of current pulses are directed through a primary
winding of a transformer to cause voltage pulses
to be imposed on a secondary winding of the trans-
former. These voltage pulses are in turn transmitted
to power input terminals of an amplifier means.
The method further comprises controlling power of
the pulses transmitted to the primary winding to
increase and decrease power of the pulses as the
signal increases and decreases in amplitude. In
this manner, power delivered to the amplifier means
is matched to the power requirements of the ampli-
fier means, thus providing an output corresponding
to the signals being amplified. Preferably, this
is done by controlling the duration of the pulses
delivered to the primary winding. A control signal
is generated by comparing a first value related to
amplitude of the signal and a second value r-elated
to power then available from the transformer.
With respect to one embodiment of the method,
power is supplied to the amplifying apparatus at
a relatively constant voltage. In one form of this
method, there is transmitted a series of first direct
current pulses through a primary winding of a trans-
former. The primary winding of the transformer is
caused to be conductive at pre-selected time periods
! during latter portions of each pulse, so as to cause
i current pulses to flow through the primary winding
and cause a ma~netic field to build up around the
transformer. While the magnetic field is building
up, substantial current flow in the secondary wind-
ing of the transformer is prevented. The secondary
winding of the transformer is caused to bë conduc-
tive after each first current pulse in the primary
winding, so as to cause second current pulses to
flow in the secondary winding. These second current
pulses are directed to power output terminal means.
The first current pulses are controlled in response
to power requirements at the output terminal means
in a manner to cause first current pulses of greater
power to flow during periods of higher power require-
ments at the output terminal means, and to cause
first current pulses of lower power to flow during
periods of lower power requirements of the output
terminal means. Power of the first current pulses
is controlled by utilizing a silicon controlled recti-
fier in series with the primary winding. The silicon
controlled rectifier is made conductive at higher
voltage levels when greater power is needed, and
is made conductive at lower voltage levels when less
power is needed.
In accordance with a second form of the-method,
power at a substantially constant voltage is supplied
by means of alternating current pulses transmitted
to the primary winding of the transformer. The pri-
mary winding of the transformer is caused to be con-
ductive during latter portions of each half cycle
of the pulses, so that positive and negative current
pulses flow through the primary winding to cause
corresponding positive and negative current pulses
! to flow through the secondary winding of the trans-
i former. This form of the method further comprises
~6~34
-13-
rectifying the positive and negative current pulses
from the transformer to transmit positive current
to a positive output terminal and to transmit nega-
tive current pu~ses to a negative output terminal.
Further, the periods of conductivity for the primary
winding are controlled in a manner such that the
primary winding becomes conductive at higher and
lower voltage levels of the alternating current
pulses delivered to the transformer, thereby causing
current pulses of greater or lesser power to be
transmitted throuyh the primary winding, in response
to a power requirement of the output terminals.
A triac may be utilized by connecting it in series
with the pr;mary winding. When used, the triac is
caused to be conductive at higher voltage levels
in response to greater power requirements at the
output terminals, and to become conductive at lower
voltage levels during periods of lower power require-
ments at the output terminals.
Still another object of this invention is to
provide a high efficiency audio amplifier including
series arranged transistors connected to respective
stepped voltage levels wherein the transistors are
controlled in a manner to more evenly distr-ibute
the voltage drop over the interconnected transistors,
thus reducing power rating requirements and resulting
in less distortion in the amplifier output.
In one embodiment of the audio amplifier, a
plurality of series connected output transistors
are connected in emitter follower configuration to
the output of the amplifîer. The transistor closest
; to the output is connected through its base to signal
input means. The remaining transistors are controlled
through their base electrodes by transistor control
~6~4
means designed to cause the series connected transis-
tors to be non-conauctive under conditions where
said amplifying apparatus is amplifying a signal
having an amplitude below a predetermined magnitude.
` 5 The transistor control means causes the second tran-
sistor to become conductive under circumstances where
the input signal is of a higher magnitude so that
current at a higher voltage is delivered to the
output terminal, with the result that current flows
from the higher voltage point to the second and first
transistors to the output terminal. In the preferred
form, the transistor control means is responsive
to output voltage from the first transistor to the
load terminal. Further, the transistor control means
is operatively connected to a base electrode of the
second transistor in a manner that control current
to the base electrode of the second transistor begins
to flow~when the output voltage reaches a predeter-
mined voltage level, so as to cause the s~econd tran-
sistor to be conductive and cause current to flowfrom the higher voltage point through said first
and second transistors to the output terminal.
Desirably, the transistor control means is further
characterized in that the~first control means sup- .
plies base current to the base electrode of the second
transistor according to a functional relationship
of the output voltage at the output terminal. This
is done in a manner that voltage of the current sup-
plied to the base electrode of the second transistor
varies as a function of magnitude of the output volt-
age, with the voltage of the base current to the
second transistor being a voltage level intermediate
j the voltage at the higher voJtage point and the out-
put voltage, with the result that voltage drop across
4~3~
~15-
the second and first transistors is shared between
the second and first transistors.
Specifically, the transistor control means com-
prises a control transistor having a first main cur-
rent carrying electrode connected to the base elec-
trode of the second transistor, and a second main
current carrying electrode connected to voltage divid-
ing means connected between said output terminal
and a related higher voltage source. The base elec-
trode of the control transistor is connected to ajunction point between a pair of voltage dividing
resistors, which in turn are connected between higher
and lower voltage sources.
There is diode means inter-connecting the lower
voltage point with the second electrode of the first
transistor, so that when the signal voltage is at
a level higher than a voltage level of the lower
voltage point, the lower voltage point is blocked
off from the higher voltage point.
In the preferred configuration, there are two
sets of transistors arranged in push pull relation-
ship, with at least two transistors being in each
set. One set of transistors conducts during positive
portions of the signal voltage, and the other set
conducts during negative cycle portions of the signal
voltage. The first set is connected to greater and
lesser positive voltage points on the secondary wind-
ing of the transformer, while the other set of tran-
sistors is connected to greater and lesser negative
voltage points on the secondary winding of the tran-
sistor. The first and second control means functions
in substantially the same manner as indicated pre-
¦ viously herein.
; In the method of the present invention, a siynal
4~
-16-
is amplified by directing the signal to a base elec-
trode of a first transistor, having a first main
current carrying electrode connected to a load ter-
minal, and a second main current carrying electrode
connected to a lower voltage source~ Current is
caused to flow from the lower voltage source through
the first transistor during periods when amplitude
of the signal is within a lower predetermined range.
During periods when the signal is within a higher
predetermined range, a control current is directed
to a base electrode of a second transistor. This
second transistor has a ~irst main current carrying
electrode connected to the second main current carry-
ing electrode of the first transistor, and a second
main current carrying electrode connected to a higher
voltage source.
Said method is further characterized in that
the base current directed to the base electrode of
the second transistor is at a voltage level inter-
mediate a voltage at the higher voltage terminal
- and voltage at the load terminal. Thus, voltage
drop across the second and first transistors is shared
by the second and first transistors.
In still another embodiment of the subject inven-
~5 tion, the audio amplifier includes a primary outputtransistor the base of which is responsive to the
audio signal and the emitter collector circuit of
which is connected between a first supply voltage
and the amplifier output. Second and third tran-
sistors have their emitters connected through diodesto the collector of the first transistor and have
their colle~ctors connected, respectively, to second
I and third supply voltages which are hiyher than the
, first supply voltage. In this embodiment control
~6~t34
-17-
circuit means are provided responsive to the ampli-
fier output to cause the second and third transistors
to sequentially commence conducting in response to
the output voltage from the amplifier exceeding the
first and second suppiy voltages, respectively.
By this arrangement the voltage drop within the
output stage of the amplifier occurs in one of the
three ways as follows:
(a) solely across the first transistor,
(b) solely across the first and second transistors,
or
(c) solely across the first and third transistors.
An additional feature of this invention is to
provide a transformer based power supply circuit
for, an audio amplifier in which the transformer primary
coil is energized by an A.C. current duty cycle modu-
lated by a solid state switch which is, in turn,
controlled by a phase shift network wherein the amount.
of phase shift is a function of the power supply
output voltage and of the audio signal being ampli-
fied~ The phase shift network may be connected by
means of a light photon coupled communication link
to control circuitry responsive to the output voltage
of the power supply and a signal which tracks the
audio signal being amplified. By this arrangement,
the duty cycle of the A.C. signal applied to the
transformer primary may he modulated to cause.the
amount of current flowing through the primary to
be adjusted to be just sufficien~ to satisfy the
power needs of the amplifier, thereby substantially
reducing primary coil idling currents. In a pre-
ferred embodiment, the power supply circuit may be
j equipped with an automatic shut down capability in
I . response to any one of the following conditions:
-18-
over current or over voltage in the arnplifier output
or a direct current fault in the amplifier circuitry~
In yet another embodiment of this invention,
the duty cycle controlled power supply circuit is
desirably equipped with a transformer having a sec-
ondary to primary turns ratio below 1.0 with a primary
induction above 30 millihenries and a coil wire gauye
diameter above no. 18 when the transformer i5 used
to produce approximately 1000~ watts at a maximum
+ 75 volt D.C. output from conventional 117-125
volt, 60 cycle alternating currentO
Still other and more specific objects of this
invention will become apparent from a consideration
of the following Brief Descr.iption of the Drawings
and Best Mode for Carrying Out the Invention.
Brief Description of the Drawinqs
Figure 1 is a schematic diagram illustrating
the basic concepts of the~present invention;
Figure 2 is a block diagram of one embodiment
of a power supply and amplifier designed in accor-
dance with the present invention;
Figure 3 is a circuit diagra~. of the power
supply and amplifier illustrated in Figure -2; -
Figure 4 is a modification of the switching
circuitry for controlling the operation of the power
supply illustrated in Figure 3;
Figure 5 is a circuit diagram of a stepped voltage
power supply constructed in accordance with the present
invention,
Figure 6A, 6B and 6C illustrate various current
cycle waveforms flowing through the primary winding
¦ of the power supply of Fiyure 5;
i Figure 7A is an alternative embodiment of a
g~
--19--
stepped voltage power supply constructed in accor-
dance with the present invention;
Figure 7B is a~first modification of the switch-
ing circuitry for controlling the operation of the
stepped voltage power supply of Figure 7A;
Figure 7C is a second modification of the switch-
ing circuitry for controlling the operation of the
stepped voltage power supply of Figure 7A;
Figure 8A through 8C illustrate various current
cyle waveforms flowing through the primary winding
of the power supply of Figure 7A;
Figure 9A through 9C illustrate various current
cycle waveforms flowing through the primary winding
of a power supply switched in accordance with the
circuitry of Figures 7B or 7C;
Figure lOA is a circuit diagram of a conven-
tional power supply and audio amplifier;
Figure lOB is a diagram of typical conduction
periods of a conventional transformer secondary of
the type illustrated in Figure lOA;
Figure lOC is a diagram of typical conduction
periods of a transformer secondary constructed in
accordance with the present invention;
. Figure lOD is a diagram of the peak load con- -
duction periods of a transformer secondary constructed
in accordance with the present invention;
Figure 11 is an alternative embodiment of-a
push-pull amplifier constructed in accordance with
the present invention and adapted to receive stepped
voltages from the power supply of the present in-
vention;
Figures 12A, 12B and 12C are diagrams o the
voltage drops across the push-pull transistors em-
ployed in the amplifier of Figure 11;
-20-
Figure 13 is another embodiment of an amplifier
constructed in accordance with the present invention;
Figure 14 is a diagram of the voltage output
of the amplifier illustrated in Figure 13;
Figures 15A and 15B are circuit diagrams of
a preferred embodiment of the left channel of an
amplifier constructed in accordance with the present
invention;
Figure 16 illustrates a portion of the right
channel of an amplifier constructed in accordance
with the present invention; and
Figure 17 is a circuit diagram of a power supply
for the preferred embodiment of the present invention,
wherein various safety control features have been
incorporated into the circuit.
Best Mode For Carrying Out the Invention
Referring to Figure 1, a highly schematic diagram
of an amplifier circuit 2 and power supply circuit
4 designed in accordance with the subject invention
is illustrated. Electrical energy is supplied to
the system by means of an oscillating power supply
6 which may take the form of a pulse generator or
a source of commercially available current-such as
conventional 117-125 volt, 60 cycle alternating
current. The primary coil 8 of a specially designed,
light weight transformer 10 is connected to the
oscillating power supply 6 by means of a duty cycle
control circuit 12 designed to modulate the amount
of energy suppiied to primary coil 8 in at least
partial dependence upon a characteristic of the audio
signal which is to be amplified by the system for
I conversion into sound by speaker 14. The audio
~ signal characteristic may be supplied directly over
4~
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the audio input signal conductor 16 or may be pro-
vided through a feedback line 18 connected with the
audio amplifier output 20. As will be explained
in greater aetail hereinbelow, the provision of a
duty cycle control 12 in the supply circuit to the
primary coil 8 results in the possibility of employ-
ing a much lighter weight transformer than has beenthought to be necessary heretofor.
- The transformer secondary coil 22 is connected
to a rectifier and amplifier power supply 24 designed
to provide a direct current supply voltage to ampli-
~ fier 2 which may take the form of a voltage which
varies widely in amplitude in dep~ndence upon ampli-
tude changes in the input audio signal or the form
of a relatively constant output voltage. If the
later type of amplifier power supply is employed,
a voltage-level feed back line 26 maybe employed
to cause the duty cycle control 12 to adjust the
amount of power supplied to the primary coil 8 in
order to assist in maintaining constant the output
voltage from amplifier power supply circuit 24.
As will also be described in greater detail
below, amplifier 2 has been specially designed to
permit the use of low cost, low power rated-trans
istor components. When designed in accordance with
this invention, the amplifier may be operated in
a manner to minimize the amount of heat which must
be dissipated by the heat sinks in the output stage
of the amplifier. This operation permits further
reduction in the weight of the amplifier and power
system by permitting the use of smaller, lighter
weight heat sinks than have previously been required
by amplifiers providing comparable power output.
In particular the amplifier, which may be of the
,,. ., . ~
-22-
Class B push-pull type, includes at least a pair
of output transistors 30 and 32 for amplifying the
audio signal supplied on line 16 to the respective
base electrodes thereof. In one embodiment of this
invention amplifier 2 includes additional transistors
34 and 36 connected in series with transistors 30
and 32, respectively, to provide higher absolute
supply voltage levels to the amplifier output 20
when necessary~ When these higher voltages are
unnecessary, transistors 30 and 32 remain non-con-
ductive as determined by supply voltage control
means, 38 and 40, respectively~ When transistors
34 and 36 are non-conductive, voltage is supplied
to the collectors of transistors 30 and 32 through
diode5 42 and 44, respectively, which are in turn
connected to lower voltage level taps 46 and 48,
respectively, of amplifier power supply 24. To
further reduce the power rating requirements for
transistors 30 through 36~ control means 38 and 40
are designed to cause the voltage drop across the
transistor pairs 30~ 34 and 32, 36 to be equally
shared wherever transistors 34 or 36 are conductive,
respectively. The manner by which this is achieved
and the beneficial results that flow therefrom will
be discussed in greater detail below.
~ eference is now made to Figure 2 which is a
block diagram illustrating one embodiment of the
~present invention. There is a speaker 100 which
is driven by an amplifier 102, which in turn derives
its power from a transformer 104, having primary
and secondary windings 104a and 104b, respectively.
The speaker 100 and amplifier 102 are or may be o~
I conventional design. As shown herein, the amplifier
, 102 comprises a pair of transistors 106 and 108
.
-23-
having signal input terminals 110 and 112, respec-
tively, through which the audio signal is fed into
the amplifier 102. Positive portions of the audio
signal cause the transistor 106 to be conductive
` 5 while negative portions of the audio signal cause
the transistor 108 to be conductive in a manner such
that an output current corresponding to the audio
signal is supplied to the speaker 100. In the par-
ticular embodiment shown herein, the amplifier 102
is arranged to operate at maximum power output when
plus 80 volts is applied to the transistor 106 and
minus 80 volts is applied to the transistor 108.
For an appreciation of the significance of the
present invnetion, attention is now directed to the
transformer 104. In a conventional power amplifier
(e.g. a 400 watt amplifier), the power transformer
would weigh at least 9 kg. The reason for this is
as follows. Current flow in the primary winding
is equal to the voltage input multiplied by the time
the voltage is applied, divided by the inductance
of the transformer. For a time varying input volt-
age, such as a 60Hz, 117 volt house current, analysis
reveals that in order to keep the current in the
primary winding (i.e. the magnetizing current-) within -
a proper value, the inductance must be made rather
large. This requires that a large, heavy transformer
be used in a conventional high power amplifier.
In the embodiment of Figure 2, a power trans-
former 104 can be made of a size which is a very
small fraction of the size of a transformer of a
conventional amplifier of comparable power output.
In this particular embodiment, the transformer 104
weighs only about ~2 kg, or 1/40th the weight of
the transformer of typical prior art amplifiers of
-24-
the same power ratiny. The reason for this phen-
omenal reduction in transformer size is that the
present invention permits the inductance of the trans-
former 104 of the present invention to be made rela-
tively small. When a voltage i5 applied across theprimary winding 104a of the transformer, the mag-
nitizing current climbs rapidly. Within several
microseconds the current will have reached a high
value of approximately twenty amps or so, and at
this time an electronically controlled switch 116
is opened. At this point in time, an amo~nt of
energy is stored in the magnetic field surrounding
the primary winding. This stored energy can be
considered to be analogous to the energy stored in
the electric field of a capacitor. The opening of
the switch 116 causes the field to begin collapsing,
- which causes the energy to be transEerred to the
secondary winding to deliver the energy to the amp-
lifier 102. By turning the switch 116 on and off
at a relatively high frequency (i.e. 20 KHz), 20,000
voltage pulses are delivered each second to the
amplifier 102.
The power delivered by the transformex 104 is
controlled by controlling the time within which the
switch 116 is open for each current pulse. This
is accomplished by tracking the audio signal which
is to be amplified by the amplifier 102 and then
comparing this tracking signal to the voltage imposed
across the amplifier 102. This produces a control
signal which is u~ilized to control the duration
of each current pulse delivered to the primary wind-
I ing of the transformer 104. In other words, on the
¦ assumption that the switch 116 is being opened and
! closed at a frequency of 20 KHz, the duration of
~6~
~5
each period would be 50 microseconds. During thosetime periods where the power requirements o~ ampli-
fier 102 are low, during each 50 microsecond periodthe switch 116 would be open for a relatively large.,
' 5 fraction of that time (e.g. 25 to 35 microseconds).
When the power requirements of the amplifier 102
are relatively high, the switch 116 would be open
in each period for a much shorter duration.
The audio signal which is to be amplified is
directed into an absolute value detector 118. This
signal can have both positive and negative portions.
The absolute value detector 118 provides an output
when the negative portions of the audio signal become
positive while maintaining these negative portions
at the same magnitude relative to a zero line. The
output of the absolute value detector 118 causes
the negative portions of the audio signal to be in~
verted.
The output of the absolute value detector 118
is then directed to a non-linear peak detector 120
which is characterized in that it has a rapid re-
sponse time for rapidly varying large signals, is
less responsive to more slowly varying signals, and
is virtually unresponsive to small signals~varying
2S about any arbitrary average level.
The output of the non-linear peak detector 120
is fed into a comparator 122. There is also a power
output feedback 124 which is responsive to the volt-
age impressed across the power input terminals of
the amplifier 102. This power output feed back 124
,transmits to the comparator 122 a voltage generally
proportional to the voltage at the power input ter
minals of the amplifier 102. The comparator 122
then "compares" the input from the non-linear peak
-26-
detector 120 and the input from the power output
feedback 124 to produce a control signal generally
proportional to the difference between the two in-
puts.
This control signal generally corresponds to
the increment of increase or decrease in the dif-
ference between the two input signals and is used
to control the duration of the regularly timed cur-
rent pulses in the primary winding 104a of the trans-
former 104.
There is a pulse generator 126 which functions
to generate a pulsed wave of a constant voltage,where the gaps between the pulses are of approxi-
mately the same duration as the pulses themselves.
The pulses are of a same frequency as the desired
current pulses for the transformer 104. In the
particular embodiment described herein, where the
frequency of the current pulses in the transformer
are 20 KHz, the output from the pulse generator 126
would be of the same frequency.
The output from the pulse generator 126 is di-
rected to a square wave to triangular wave converter
128. This converts the square wave form from pulse
generator 126 to a wave form where each pulsé has
the configuration of an isosceles triangle, where
during the duration of each pulse, the voltage climbs
at a substantially constant rate to a peak at the
middle of the pulse, and then declines at a constant
rate through the latter half of the pulse.
The output from the square wave to triangular
wave converter 128 is transmitted to a ramp time
modulator 130, ana this ramp time modulator 130 also
- receives the control signal from the comparator 122.
The modulator 130 in effect "chops off n the upper
L6f~
-27-
portion of the triangular wave form produced by the
square wave to triangular wave converter 128.
The output of the ramp time modulator 130 is
a constant voltage pulse signal having the same fre-
quency as that of the pulse generator 126. The dura-
tion of each pulse is directly proportional to the
duration of the "unchopped" bottom portion of the
triangular wave form from converter 128. Thus~ it
can be appreciated that the duration of the pulses
from modulator 130 are proportional to the magnitude
of the control signal from comparator 122.
The pulses from ramp time modulator 130 are
used to open and close the switch 116 in a manner
that the switch 116 is closed during the duration of
each of the puls~s from ramp time modulator 130.
Thus a pulse of relatively short duration produces
a corresponding current pulse of a relatively small
magnitude, since the current has such a very short
time period to build up or l'ramp up". It can be
appreciated that as the voltage pulses from modula-
tor 130 increase in duration, the magnitude of the
current pulses in the transformer primary winding
104a increase, correspondingly, in a manner that a
pulse of longest duration from modulator 13-O pro- -
duces a current pulse of the largest magnitude in
the transformer primary winding.
Power may be suppIied to the primary by means
of a conventional plug 132 and bridge rectifier 134
for converting regular 110-120 volt, 60 cycle current
to direct current. The bridge rectifier 134 is con-
nected to the upper terminal 136 of the transformer
primary winding 104a. The lower terminal 138 of
the transformer primary winding 104a is connected
to the aforementioned switch 116, which in turn is
~6~
-28-
connected to ground. When ~he switch 116 is conduc-
tive, direct current flows through the primary 104a.
The secondary winding 104b of the transformer
104 is center tapped at 140 to ground. The upper
terminal 142 of the secondary winding 104b is connected
to an upper positive output terminal 144 through
a diode 146 which permits only positive current to
be directed to the output terminal 144. In a similar
manner, the lower terminal 148 of the secondary wind-
ing 104b is connected to a lower negative output
terminal 150, through a second diode 152 which permits
only negative current pulses to be transmitted to
the output terminal 150. There are a pair of shunt
connected capacitors 154 and 156, one of which is
connected to the positive output terminal 144 at
a location between that terminal 144 and diode 146,
and the other capacitor i56 being connected to the
negative output terminal 150 at a location between
that terminal 150 and diode 152. The other plates
of the two capacitors 154 and 156 are both connected
to ground.
In view of the foregoing description, the op-
eration of the present invention may be understood
in relation to the amplification of a typica-l audio
signal, such as the audio signal of a musical com-
position. This signal will be made up of some lower
frequency oscillations (fundamental tones) on which
are imposed any number of higher frequency oscilla-
tions (overtones), with the amplitude of these os-
cillations varying over a wide range (e.g. from the
sound generated by full orchestra to the quiet sound
; of a single woodwind instrument playing a melodic
theme). With regard to the amplitude variations
in the signal, although these amplitude variations
34
might seem to the listener to be in some cases very
abrupt, in actuality the very sharp amplitude changes
of any great magnitude would generally take place
in a time period no less than about one thousandth .
of a second. For example, the rise time associated
with the noise generated by a very sharp percussion,
such as that generated by slapping two wood blocks
together is generally greater than a thousandth of
a second.
Assume that an audio signal is to be amplified
by the circuit of Figure ~. This signal would be
directed into the two signal input terminals 110
and 112 of the amplifier 102, and would also be di-
rected to the absolute value detector 118. As indi-
cated previously, this audio signal is converted
by the absolute value detector 118 to a direct cur-
rent wave form which in turn is transmitted to the
non-linear peak detector 120 to provide a ~smoothed"
signal to the comparator 122.
Since upon initial start up, there is no voltage
generated at the output terminals 144 and 150, the
feedback signal provided by the power output feed-
back 124 would be zero or substantially zero. Accord-
ingly, the comparator 122 would generate a-rather
strong output signal to the ramp time modulator 130.
The ramp time modulator 130 would in turn transmit
pulses of the desired ~requency to the electronic
switch 116~ with these pulses being of the maximum
duration. In other words, the switch 116 would con-
tinue to turn "on" and "off" at the same frequency,
but the duration of the "on" periods would be at
a maximum. Accordingly, the current pulses passing
! through the primary winding 104a would ramp up to
maximum amperage and thus deliver full power to the
~6~4!t84
-30-
output terminals 144 and 1500 Within a very short
period of time, (i.e. about two hundred microseconds)
the voltages applied to the power input terminals
158 and 160 of amplifier.l02 would build up to the
proper operating level.
At this time, the power output feedback 124,
would transmit to the comparator 122 an output signal
related to the voltage level at the power input ter-
minals of the amplifier 102. ThereaEter, the compara~
lQ tor 1~2 would continue to provide to the ramp time
modulator 130 a control signal related to the power
requirements of the amplifier 102. In other words,
when the comparator 122 receives an input signal
from the non-linear peak detector 120 which indicates
that the amplitude of the audio signal is increasing,
there will be a greater disparity between this audio-
related signal and the existing signal from the power
output feedback 124 so that the voltage of the control
signal to the ramp time modulator 130 increases.
This will in turn cause the current pulses through
the primary winding 104a to increase in duration
to deliver more power to the amplifier 102 and thus
raise the voltages supplied at the output terminals
144 and 1500 On the other hand when the amplitude
of the audio signal declines, the comparator 122
would detect that the difference in the signal from
the non-linear peak detector 120 and that from the
power output feedback 124 is smaller, so the control
signal transmitted by the comparator 122 to the modu-
lator 130 would be of a lower voltage. This wouldin turn shorten the duration of the current pulses
through the primary winding 104a, thus delivering
I less power to the amplifier 102. From the above
I description it can readily be appreciated that com-
-31-
parator 122 will in effect "track" the audio signal
to maintain the voltage impressed upon the power
input terminal 158 and 160 of the amplifier 102 so
that these voltage levels are varied in such a manner
that they remain only moderately above the power
requirements of the amplifier 102. In actual prac-
tice, there would generally be a voltage drop across
each of the transistors 106 and 108 of approximately
five volts. In the particular embodiment shown herein,
the operating components of the invention are selected
so that the maximum voltage which would be applied
acxoss the terminal 158 and 160 of the amplifier
102 would be plus 80 volts and minus 80 volts.
As a further advantageous feature of the present
invention, attention i5 directed to the two shunt
connected capacitors 154 and 156. Inherent in the
operation of the present invention is the feature
that at any particular time, only a relatively small
amount of power need be stored in-the two capacitors
154 and 156 to respond to rapid increases in the
power requirements of the amplifier 102. The reason
for this is that the power pulses through the primary
winding 104a are of such a high frequency, and the
response time to increase the duration (and--thus
increase the power) of these current pulses in the
transformer ]4 can occur in such a very short time,
that the transformer 14 can respond in a matter of
a fraction of a millisecond to begin delivering full
power to the amplifier 102. Thus, these capacitors
154 and 156 can be made with reasonably small capa-
city, thus creating a further savings in both weigh
and expense.
! Another desirable feature of the present inven-
tion is the efficiency achieved. Let it be assumed
34
-32-
that the circuit constants have been arranged to
cause a constant voltage of five volts across each
of the amplifier transistors 106 and 108~ Let it
further be assumed that the res;stance of the load
(i.e. the speaker 100) is eight ohms. Then let us
examine three situations:
1. Where the voltage applied to the amplifier
102 is plus 25 and minus 25 volts,
2. Where the applied voltage to the ampliier
is plus 35 and minus 35 volts, and
3. Where the applied voltage is plus 45 and
minus 45 volts.
The power output is equal to the voltage squared
divided by the resistance. For the first situation
(where the applied voltage îs plus 25 and minus 25
volts), there would be a drop of five volts across
each transistor 106 and 108 when each is conducting,
and a drop of twenty volts across the eight ohm load.
The actual power output would then be twenty squared
divided by eight, which gives fifty watts. The loss
at the transistors 106 and 108 would be 12 1/2 watts.
Therefore out of the total of 62 1/2 watts used (50
watts and 12 1/2 watts) fifty watts are actually
utilized in the speaker, for an efficiency of 80%.
In the second situation (where the voltage ap-
plied across the amplifier 102 is between plus 35
and minus 35 volts), the power delivered to the speaker
would be equal to thirty squared divided by eight
for 112 watts. The power dissipated in the transis-
tors 106 and 108 would be 18.75 watts, which gives
an efficiency of 85.7%.
In the third situation (where the voltage applied
to the amplifier 102 i~ plus 45 and minus 45 volts),
i 200 watts are actually utilized in the speaker 100
-33-
and only 25 volts dissipated in the transistors 106
and 108, for an efficiency of 88.8%.
In a conventional amplifier where the f~ll vol-
tage of plus 80 volts and minus 80 volts is impressed
across the amplifier power input termianls àt all
times, the efficiencies for the three situations
outlined above would be 25%, 37% and 50% respectively.
Since the efficiency of the apparatus of the present
invention is quite high, a relatively small amount
of energy is dissipated in the transistors 106 and
108. For this reason the heat sinks for these transis-
tors 106 and 108 can be made relatively small. It
has been found that with these various weight savings,
a power amplifier can be built according to the present
invention with a power rating of four hundred watts,
and with a total apparatus weight of only
5.5 kg.
With reference to Figure 3, the circuitry o
the embodiment of the present invention illustrated
in Figure 2 will now be described in more detail.
The plug 132 is adapted to be connected to a conven-
tional wall socket which produces 110 to 120 volts
at 60 Hz. The plug 132 connects to the bridge recti-
fier made up of four diodes D101, D103, D105-, and
D107 arranged to transmit positive D~Co current to
the primary winding 104a. A small capacitor 162
(fifty to two thousand microfarads) is connected
between the output of the bridge rectifier 134 and
ground to prevent the D.C. output from the bridge
rectifier 134 from going to zero.
The absolute value detector 118 has two input
terminals 164 and 166 which are arranged to receive
a signal input from a stereo player unit. The signals
! transmitted to terminal 164 and 166 are in turn trans-
-34-
mitted to an operational amplifier 168 through a
set of four diodes D109, Dlll, D113 and D115. These
diodes D109-D115 are so arranged that whenever there
is a relatively s~rong signal at one input 164 and
166 and a relatively weak signal at the other input
164 and 166, the highest value will be transmitted
to the operational amplifier 168. The highest nega-
tive value will pass through the diodes D109 and
D113 while the highest positive value will pass through
the diodes Dlll and D115. As indicated previously,
the output from the operational amplifier 168 is
an output signal where the negative portions of the
audio signal have been made positive.
This signal passes through a diode D117 of the
non-linear peak detector 120. Signals of large value
are within the operating envelope of the diodes Dll9
and D121 and thus are passed immediately with the
capacitor 170 providlng a more constant output.
Smaller signal variations are largely blocked by
resistor R101 and diodes Dll9 and D121.
The output from the non-linear peak detector
passes through a resistor R103 to one input terminal
of an operational amplifier 172 of the comparator
122. The input to the other terminal of the-opera-
tional amplifier is from the power output feedback
124. Power output feedback 124 comprises two re-
sistors R105 and R107 connected in series to the
positive input terminal 158 of amplifier 102 and
a thixd resistor R109 connected between ground and
a location between the resistors R105 and R107.
These resistors R105-R109 step down the voltage from
the level at the terminal lS8 to less than 15 volts,
which is a voltage which the operational amplifier
172 is able to handle. The operational amplifier
has a resistor Rlll to provide a feedback voltage.
- The output from the operational amplifier 172
is transmitted through a resistor R113 to the ramp
time modulator 130. A aiode D123 is connected be-
tween the location of resistor R113 and the modula-
tor 130 to pass signals from the amplifier 172 above
a certain value to ground.
The pulse generator 126 may be provided as one
of a number of commercially available pulse genera
torsr such as a Fairchild US 78540. As shown herein,
this pulse generator comprises an operational ampli-
fier 174 having a pair of Lesistors R115 and R117
provided as feedback loops. One input terminal 176
of the operational amplifier is connected to one
plate of a capacitor 178, with the other plate of
the capacitor 178 being connected to ground. The
other terminal of the operational amplifier 176 is
connected through a resistor Rll9 to ground.
The square wave to triangular wave converter
128 comprises a resistor R121j which receives the
voltage pulses from pulse generator 126. The resistor
R121 is connected to one plate of a capacitor 180,
the other plate of which is connectea to ground.
Capacitor 180 converts the square-wave pulsed signal
from generator 126 to a triangular shaped waveO
The triangular shaped wave is transmitted through
one input terminal of an amplifier 182 which simply
amplifies the triangular wave and transmits it to
the ramp time modulator 130. The ampli~ier 182 has
a feedback resistor R123 connected to the other input
terminal thereof, and also a resistor R125 connected
between the other input terminal and ground.
The ramp time modulator 130 may be a differen-
tial amplifier of conventional design such as the
-36-
TL0074 from Texas Instruments Corp. As shown herein,
this modulator 130 comprises two transistors Q101
and Q103 which compare the two voltages directed
to the two bases of the transistors Q101 and Q103;
the transistor having the lower input will conduct~
The emitters of the two transistors Q101 and Q103
are connected to a positive voltage source through
the resistor R127. Thus, for purposes of illustra-
tion, let it be assumed that the waveform from tri-
angular wave converter 128 is imposed upon the transis-
tor Q101 and the control signal from operational
amplifier 172 in comparator 122 is imposed on tran~
sistor Q103. In this situation, those portions of
the triangular shaped wave which are above the control
signal voltage will cause transistor Q103 to conduct.
The collector of transistor Q103 is connected through
a resistor R129 to a negative voltage terminal, with
the resistance of R127 being substantially greater
than the resistance of R129. Consequently, when
transistor Q103 is non-conducting, the point 184
located between transistor Q103 and resistor R129
goes negative to cause a transistor Q105 to conduct,
which in turn causes the transistor Q107 to be con-
ductive and the transistor Q109 to be non-conductive.
Thus, the output from the ramp time modulator 130
is a series of pulses, with the duration of each
pulse coinciding with the portion of the triangular
shaped wave from converter 128 which is below the
voltage control signal.
The switch 116 comprises a first transistor
Qlll, the output of which is connected to the bases
o~ two parallel connected transistors ~113 and Q115.
! Thus, when transistor Qlll becomes conductive, it
causes the two transistors Q113 and Q115 to conduct,
thus permitting a current pulse to pass through the
primary winding 104a of the transormer 104. When
transistor Qlll becomes non-conductive in response
to the termination of the pulse from ramp time mod-
ulator 130, transistors Q113 and Q115 also become
non-conductive and switch 116 opens. The magnetic
field created in primary winding 104a by the current
pulse thereafter collapses, causing a transfer of
energy between primary winding 104a and secondary
winding 104b. In this manner, power is delivered
to amplifier 102.
A modification of the circuitry of Figure 3
is shown in Figure 4. In Figure 4, there is shown
only those portions of the circuitry of Figure 3
which are necessary for proper orientation o the
components shown in Figure 4 to the other components
of the present invention.
The reason for th~ modification of Figure 4
is that in some circumstances the voltage at the
negative input terminal 160 of amplifier 102 may
have an absolute magnitude that is too low relative
to ~he voltage at the positive terminal 158. In
this situation, it would be desirable to override
the signal generated by the operational amplifier
172 of the comparator 122 and provide a correction
signal of greater magnitude to the ramp time modu~
lator 130. This is accomplished in the circuitry
of Figure 4 in the following manner. The power input
terminals 158 and 160 o~ amplifier 102 are connected
each through a related resistor R401 and R403 to
a summing junction 400. The output from the summing
unction 400 is a voltage which is the algebraic
sum of the voltages at the amplifier input termina1s
158 and 160. Thus, if the absolute magnitude of
-3B-
the negative voltage is greater than the absolute
magnitude of the positive voltage (minus 40 volts
and plus 30 volts), the output at the junction 400
would be a negative value (e.g. minus ten volts).
On the contrary, if the positive voltage at the
terminal 158 has an absolute magnitude greater than
that of the negative voltage at the terminal 160,
the output at junction 400 will be positive.
The summing junction 400 has a first connection
through a diode D401 and resistor R405 to the nega-
tive terminal of an operational amplifier 402~ The
summing junction 400 has a second connection through
another diode D403 to the positive terminal of opera-
tional amplifier 402. The diode D401 permits only
negative current to pass therethrough, while the
diode D403 permits only posit;ve current to pass
therethrough. At a location between the diode D403
and operational amplifier 402, there is a resistor
R407 connected to ground. There is also a ~esistor
R409 to provide feedback to the negative terminal
of the operational amplifier. The output of the
operational amplifier 402 leads through a diode D405
which will only pass positive current to the rPsistor
R113. There is also provided a diode D407 which
passes positive signals from the comparator opera~
tional amplifier 172 to the resistor R113O
When the voltages at the terminals 158 and 160
are equal, the output from the summing junction 400
is zero. In those circumstances where the negative
voltage at the terminal 160 is greater than the positive
voltage at terminal 158, there would be a negative
output from the summing junction 400 which would
be transmitted through the diode D401 to the ampli-
fier 402, which would then produce an output signal
116'~4~34
-39-
through the diode D405. If the difference between
the voltages at terminal 158 and 160 is sufficiently
large, this signal will override the signal from
the operational amplifier 172 (which is directed
through diode D407) to cause ramp time modulator
` 130 to increase the power of the current pulses and
thus correct the difference between the absolute-
magnitudes of the voltages at terminals lS8 and 160.
In those circumstances where the positive voltage
at the terminal 158 is higher than the negative voltage
at the terminal 160, the output from the summing
junction 400 will be positive, and thus a positive
voltage will be transmitted through diode D403 to
the amplifier 402, which in turn will produce an
output signal. However, since the voltage at terminal
158 is already relatively high, in most circumstances
the signal from the amplifier 172 would be suffi~
ciently large to overcome the signal from the opera-
tional amplifier 402 and provide a control signal
2~ of sufficient strength to cause ramp time modula-
tor 130 to increase the duration of the power pulses
to the transformer 104 and thus correct the disparity
between the absolute maynitude of the voltages at
terminals 158 and 160.
As indicated previously herein, when the ampli-
fier of the present invention is operating as an
audio amplifier, the major amplitude variations in
the signals to be amplified would occur in a time
period no less than approximately 1/1000 of a second.
Therefore, the apparatus should be able to respond
within this time period to change its power output
from a low level tv a relatively high level. For
this reason, the frequency of the control pulses,
which in turn controls the frequency of the current
-40-
pulses through the transformer primary winding 104a,
should be at least a thousand cycles per second,
and pxeferably at least two thousand cycles per second.
However, there are further advantages in operating
the current pulses at a much higher frequency, in
a range of fifteen to twenty-five thousand cycles
per second (desirably about twenty thousand cycles
per second). First, this permits the size of the
transformer to be made quite small. Also, the fre-
quency is at a sufficiently high level that it wouldnot produce any undesired sounds in the normal audio
range, which is normally below twenty thousand cycles
per second. Further, this frequency is n~t so high
as to be beyond the capacity of the switching circuits
used.
The basic concept of utilizing power output
feedback in an audio amplifier transformer to control
the energy transfer between the primary and secondary
windings of the transformer in response to the magni-
tude of the audio input signal can be effectively
employed in a transformer which produces a fixed
voltàge output across the secondary. Moreover, when
a fixed output transformer is involved, the power
output feedback can also be used to assist in main- -
taining the voltage output constant. The Figure
5 embodiment of the present invention illustrates
such a system. The main power transformer 500 (i.e.
magnetic field coil) has primary and secondary wind-
ings 500a and 500b, respectively.
For reasons more fully explained in connection
with Figure 11 below, the secondary 500b is tapped
to provide plural positive and negative voltages
i stepped in 25 vvlt increments from 25 to 75 volts.
I The positive terminals are respectively designated
-41-
El through E3, while the negative terminals are
respectively designated E4 and E6.
The opposite ends of primary winding 500a are
connected through current rectifying diodes to the
two terminals of a conventional power source, indi-
cated at 502, which can be a wall socket providing
current at 60 Hertz and 120 volts. Two leads 504
and 506 from power source 502 are respectively con-
nected through related rectifying diodes D501 and
D503, and then in series with a silicon controlled
rectifier 508, to the upper end 510 of transformer
~ primary winding 500a. The leads 504 and 506 are
also respectively connected through related second
rectifying diodes D505 and D507 to the lower end
512 of primary winding 500a~ An examination of the
diodes D501, D503, D505 and D507 makes it readily
apparent that these four diodes are arranged in a
rectifying bridge so that on each half cycle from
the power source, a positive voltage is directed
to the silicon controlled rectifier 508 and thence
to the upper end 510 of the transformer primary wind-
ing 500a.
Silicon controlled rectifier 508 is controlled
by an audio input signal as will be explained-more
fully hereinbelow. The audio input signal is directed
into an input terminal 514, and thence to a junction
point 516. The resistors R501 and R503 are connected
between junction 516 and a suitable voltage source
(e.g. a 75 volt source) to provide a base voltage
level of, for example, 0.7volts. This voltage is
developed by diode D509 connected from the junction
of R503 and R501 to ground. The voltage is in turn
directed to an operational amplif;er 518. Feedback
to the opera~ional amplifier is obtained ~rom the
-42-
junction 520 of a diode D515 and capacitor 522 in
the transformer seconaary circuit. The point 520
is connected through two voltage dividing resistors
R505 and RS07 to ground. At a junction point 526
between the two resistors R505 and R507 there is
a feedback connection to the operational amplifier
518. The output of operational amplifier 518 is
directed to a suitable control apparatus indicated
at 528. This control apparatus 523 is connected
to the control terminal of silicon controlled recti- ¦
fier 508 in a manner such that at higher outpu~ levels
from operational amplifier 518, the rectifier 508
is caused to conduct at higher voltage levels on
the latter portion of each half cycle. In like manner,
when the output of the operational amplifier 518
is lower, the silicon controlled rectifier 508 is
caused to fire at lower voltaye levels.
The center of secondary winding 500b is tapped
at 530 to ground. The upper half of the winding
500b is tapped at two intermediate locations 532
and 534 to respectively provide the positive 25 volt
and 50 volt output for power terminals El and E2,
while the upper terminal 524 of winding 500b provides
the positive 75 volt output for terminal E3~ In
like manner, the bottom half of secondary winding
500b is tapped at intermediate locations 536 and
538 to provide the intermediate voltage levels of
minus 25 and minus 50 volts at terminals E4 and E5,
respectively, while the lower end 540 of the secondary
winding provides the minus 75 volt output for terminal
E6.
~ The three points 532, 534 and 524 are each con
! nected through respective blocking diodes D511, D513
I and D515 to their respective output terminals. A
~3~6~
-43-
first capacitor 542 i5 provided between ground and
the lower voltage output terminal (i.e. positive
25 volt terminal). A second capacitor 544 is con-
. nected between the positive 25 volt output terminal
and the positive 50 volt output terminal, and thethird capacitor 522 is in like manner connected
between the positive 50 volt output terminal and
the positive 75 volt output terminal. Capacitors
542, 544 and 522 have sufficient capacitance to
compensate for any abrupt power demand from the
related output terminal so as to maintain the output
voltage terminals at nearly constant voltage level.
The lower half of secondary winding 500b is
connected to its negative output terminals E4, E5,
E6 through three blocking diodes D517, D519 and D521.
Capacitors 546, 548 and 550 are respectively connected
between the negative output terminals in substan-
tially the same manner as the corresponding components
for the upper half of the primary winding. However
the blocking diodes D517 through D521 are reversed
so as to permit only negative current to pass to
the output terminals E4 through E6.
In the operation of the power supply of Figure
5, the output voltage in the winding 500b is-reg~
ulated entirely by the circuitry controlling the
silicon controlled rectifier 508. The SCR control
circuitry 508 is in turn governed by the audio input
signal in a manner such that when the input signal
is of a greater amplitude, the silicon controlled
rectifier 508 is caused to conduct for greater portions
of each power half cycle to transmit more current
to primary winding 500a. This mode of operation
! can best be illustrated with reference to Figures
6A, 6B and 6C.
-44-
In Figure 6At there is a representation of the
voltage delivered from power source 502 throuyh the
two diodes D501 and D503 to the silicon controlled
rectifier 508. It can be seen that because of the
action of the diodes D501 through D507, a positive
sine wave voltage pulse is delivered to silicon con-
trolled rectifier 508 on each half cycle. Let it
be assumed that the audio input signal to the amplifier
- is of a relatively low amplitude, so that the power
requirement of the amplifying circuit is quite low.
In this condition, the silicon controlled rectifier
508 is caused to conduct only at the very end of
each half cycle. The point of conduction of each
half cycle is illustrated at 600, and rectifier 508
remains conductive until the current has reached
zero, at point 602. Thus, it can be seen that the
current is being delivered to the primary winding
500a in rather short increments of time, and at a
lower voltage level.
As the audio input signal reaches a greater
amplitude, silicon controlled rectifier 508 is caused
to conduct at a higher voltage level for the latter
part of each half cycle, as illustrated in Figure
6B. The point of conduction occurs at 604, and each
cut off point is indicated at 606. It can be seen
that not only is the voltage higher, but also the
time increment of each current pulse is longer, so
that greater power is delivered to the primary wind-
ing 500a.
Finally, in Figure 6C, there is shown the situa-
tion where the audio input signal is at a maximum
amplitude, thus Ieadiny to a demand for maximum power
from the transformer. In this situation, silicon
controlled rectifier 508 is caused to conduct near
-45-
the peak voltage at the beginning of the latter half
of each half cycle, as illustrated at 608 in Figure
6C, with the cut off point being indicated at 610.
Thus, it can be seen that current is being delivered
at yet a higher voltage and for a longer duration
during each half cycle.
As the current builds up in the primary winding
500a during the latter part of each half cycle of
current, no current flows in the secondary winding
500b, because of the arrangement of the blocking
diodes D511 through D521. However, at the end of
each half cycle of current through the primary winding
500a, after the current is shut off, the field around
500a collapses so as to create a voltage drop across
the secondary winding 500b and cause current to flow
through the secondary winding to supply power to
the six capacitors 522 and 542 through 550 and the
six output terminals El through E6.
As indicated previously, when the amplitude
of the audio input signal is at higher levels t the
power requirements on the transformer 500 are greater.
During these periods, current will flow through the
primary winding 500a for longer increments of time
to store more energy in the magnetic field of primary
winding 500a. When the current is shut off in the
primary winding at the end of each half cycle, the
magnetic field in the primary winding collapses to
induce a flyback voltage across the secondary winc ng
50Qb. The diodes DSll through D521 are arranged
so that current flows through the secondary winding
to charge the capacitors 542, 544r 522, 546, 548
and 550 to maintain the voltage at the power terminals
i El through E6 at the proper level.
Figure 7A illustrates another embodiment of
116Z4~4
--46--
a stepped voltage power supply constructed in accor-
dance with the present invention. As in the Figure
5 embodiment, there is a power source such as wall
plug 702 including two leads 704 and 706. Lead 704
is connected to a triac 708. The opposite side of
the triac 708 connec~s to the ~pper end 710 of primary
winding 700a in transformer 700. The other lead
706 of power source 702 connects to the lower end
712 of primary winding 700a.
The triac 708 serves a function similar to that
of the silicon control rectifier 508, except that
triac 708 conducts on both positive and negative
half cycles of the current from power source 702.
Switching circuitry controls the triac 708 in a manner
such that it conducts during the latter portion of
each half cycle for longer or shorter periods of
time, depending upon the power requirements of the
amplifier. This switching circuitry is substantially
the same as the circuitry which switches silicon
controlled rectifier 508 in the Figure 5 embodiment
of the~invention, and thus the circuit components ¦~
will not be described in further detail.
The secondary winding 700b of transformer 700
is tapped to ground at its center point 714._ The
upper half of secondary winding 700b is tapped at
two intermediate locations 716 and 718 to provide
the positive 25 volt and 50 volt outputs for the
power terminals El and E2. A connection 720 to the
upper end of winding 700b provides the positive 75
volt output for power terminal E3. In like manner,
the bottom half o~ the secondary winding is tapped
j at three equally spaced locations, 722, 724 and 726
to respectively provide the negative 25, 50 and 75
volt outputs E4, E5 and E6.
34
-47-
The positive and negative 75 volt leads 720
and 726 are attached at opposite ends to a first
bridge rectifier, generally indicated at 728. The
positive output of bridge rectifier 728 is connected
by lead 730 to the positive 75 volt power terminal
E3, and the negative output of the bridge rectifier
728 is connected to the negative 75 volt power output
E6 through lead 732. The positive and negative 50
volt leads 718 and 724 are connected to the opposi~e
ends of a second bridge rectifier 734. The positive
output terminal of bridge rectifier 734 is connected
by lead 736 to the positive 50 volt power terminal
E2, while the negative output from the bridge recti-
fier 734 is connected~through lead 738 to the nega-
tive 50 volt power terminal E5. Finally, the posi-
tive and negative 25 volt leads 716 and 722 are con-
nected to opposite ends of a third bridge rectifier
740. The output leads 742 and 744 of bridge recti-
fier 740 are respectively attached to the positive
and negative 25 volt power terminals El and E4.
As in the Figure 5 embodiment, six~capacitors,
designated 746 through 756, are provided between
the power output terminals El through E6 to compen-
sate for any abrupt power demand on the related
output terminal so as to maintain the output voltage
terminals at nearly constant voltage ~evel.
To describe the operation of the embodiment
of Figure 7A, reference is made to Figures 8A, 8B
and 8C. The current through ~rimary winding 700a
is not rectified and is thus an alternating current.
The triac 708 is caused to conduct in the latter
half of each half cycle, whether it be a positive
I or negative half cycle. When the power requirements
; of the amplifier are low, the control apparatus acts
-48-
to cause triac 708 to conduct for only a very short
time period at the end of each half cycleO This
is illustrated in Figure 8A, where the point of con-
duction of each half cycle is illustrated a~ 800.
Triac 708 remains conductive until the voltage has
reached zero at point 802. Thus, it can be seen
that current is being delivered to the primary wind-
ing in rather short increments of time, and at a
lower voltage level.
As the control signal reaches a greater ampli-
tude, the triac 708 is caused to conduct at a higher
voltage level for the latter part of each half cycle,
as illustrated in Figure 8B, where the point of con-
duction is indicated at 804, and each cutoff point
is indicated at 8060 It can be seen that not only
is the voltage higher, but also the ~ime increment
of each current pulse is lGnger, so greater power
is delivered to primary winding 700a.
Fina~ly, in Figure 8C, there is shown the situa-
tion where the input signal is at a maximum ampli-
tude, thus providing for maximum power requirements.
In this situation, triac 708 is caused to conduct
near the peak voltage at the beginning of the latter
half of each cycle, as illustrated at 808, with the
cutoff point being ind;cated at 810.
Current flows in the secondary winding 700b simul-
taneously with the flow of current in 700a, with
the current in 700b also being an alternating cur-
rent. With regard to the flow of current through
the two 75 volt leads 720 and 726, since this cur-
rent flows through the rectifying bridge 728, the
output to the power terminal E3 is always positive,
while the output to the terminal E6 is always nega-
tive. In like manner, current is directed from the
-49-
intermediate terminals 716, 718, 722 and 724 through
the two bridge rectifiers 734 and 740 to provide
positive current to the output terminals E2 and El
at the positive 50 and 25 volt levels, and to provide
negative current to the power output terminals E5
and E4 at the`negative 50 and 25 volt levels, re-
spectively.
It has been found that by using the power supply
circuitry of the present invention, the transformer
can be made relatively small and still provide ade-
quate power. For example, the transformer in the
present invention can be made from 1/4 to 1/10 the
size of the transformer in a conventional audio amp-
lifier of comparable power rating, with one hundred
and seventy five windings in the primary and two
hundred in the secondary.
It is sometimes desirable to adjust the cutoff
point of current flow through transformer primary
700a in order to more precisely control the charac-
teristics of the energy transfer across the trans-
former windings during each half cycle of current
from the power source. For example, shutoff of the
primary current prior to the zero crossover voltage
point in the current waveform eliminates idling cur-
rents in the primary winding during the remainingportion of the waveform half cycle. Accordingly,
Figures 7B and 7C illustrate two modifications to
the switching circuitry of Figure 7A, both of which
modifications enable the transformer primary winding
700a to receive current during more narrowly defined
portions of the power cycle~
In Figure 7B, a second Triac 758 is connected
¦ in parallel wi~h Triac 708. A control apparatus
~ as previously described in connection with Figure
~6~ 4
-50-
5 controls the operation of Triacs 708 and 758 re-
spectively via leads 760 and 762. A capacitor 764
is connected in series with Triac 758 and acts to
periodically shunt Triac 708. The circuit of Figure
7B operates as foiiows. In response to an audio
input signal, output from the control apparatus
causes Triac 708 to conduct at some point during
each positive and negative half cycle of the current
from power source 702. At a later predetermined
time the output from the control apparatus causes
Triac 758 to conduct, whereupon current is diverted
from Triac 708 and begins to flow through capacitor
764. Triac 708 shuts off but current continues to
pass through Triac 758 and capacitor 764 to trans-
former primary 700a until the voltage buildup incapacitor 764 reaches a level sufficient to turn
Triac 758 off, thus ending current flow through the
transformer primary. Capacitor 764 is very small
in order to limit the conductive state of Triac 758
to a short period of time.
Figure 7C illustrates a second modification
o~ the embodiment of Figure 7A where the single triac
708 is replaced by a switching apparatus indicated
generally at 766. A pair of GTO SCR's are connected
in parallel with one another. One GTO, designated
GTOb, conducts on positive half cycles and a second
GTO, designated GTOa, conducts on negative half
cycles. Blocking diodes are provided at D701 and
D703. Each GTO becomes conductive at a predetermined
voltage, as provided by the control mans, and becomes
non-conductive, as provided by the control means,
within a predetermined control time period, prefer--
ably one millisecond.
Figures ~A, 9B and 9C illustrate the manner
~16~ 34
-51-
of switching common to both the Figure 7B and 7C
modifications. At low power requirements the cur-
rent flow begins at 900 and ends at 902 near the
. latter part of the last part of each half cycle.
At intermediate power requirements, the on-off switch-
ing occurs earlier in the latter half of each half
cycle, illustrated in Figure 9B at 904 and 906.
At peak power requirements, the switching occurs
near the peak of each half cycle, illustrated in
Figure 9C at 908 and 910. With this arrangement,
the transformer can be made yet smaller.
For purposes of clearly demonstrating the advan-
tages of the various duty cycle controlled power
supply embodiments discussed above, reference is
made to Figure lOA, which illustrates the basic con-
figuration of a conventional amplifier power supply.
Conventional 117-125v 60 cycle AC is supplied from
PS to the primary winding lOOOa of a transformer
1000. The secondary winding lOOOb of transformer
1000 has its upper end connected through a diode
D1001 to the upper terminal of the amplifîer 1002.
The lower terminal of the secondary winding lOOOb
is connected through a second diode D1003 to the
lower terminal of amplifier 1002. Upper and--lower
capacitors 1004 and 1006 respectively maintain the
voltage imposed upon the amplifier 1002 at a substan-
tially constant value. Normally, the supply voltage
has a peak voltage input of approximately 169 volts.
Let it be assumed that the input voltage imposed
at the upper terminal of the amplifier 1002 is de-
signed to be plus 75 volts and the voltage at the
lower terminal is minus 75 volts. The center of
the secondary winding lOOOb is normally tapped to
ground.
An audio sound typically has peak power require-
ments of relat;vely short duration and average power
requirements of longer d~ration which are possibly
1/20th of the peak power requirement. Thus, most
of the time the amplifier is operating at only l/lOth
to l/20th of full power. To understand the implica-
tion of this fact, reference is made to Figure lOB
illustrating the sine wave o~ incoming voltage sup-
plied to the primary of a conventional audio ampli-
fier transformer connected to receive conventional117 to 125v alternating current. The turns ratio
of the primary and secondary winding of the conven-
tional transformer is such that with the primary
conducting at least some current throughout the
entire sine wave of the incoming voltage, the peak
voltage generated in the secondary is just slightly
larger than the plus 75 and minus 75 volt level
required by a conventional audio amplifier. When
the amplifying component of the amplifier is demand-
ing only average power, current flows in the sec-
ondary winding for only a very short period of time
at the very peak of the sine wave of the input volt-
age. This time period is indicated at 1008 in Figure
lOB. When there are peak power requirements,--there
is an immediate drain on the conventional storage
capacitors 1004 and 1006 of the power supply to lower
their voltage levels slightly, and the result is
that the secondary winding is conducting for a longer
time period~ so that the conducting portion of the
sine wave of Figure lOB is broadened out to, for
example, lines indicated at lOlOa and lOlOb. It
should be noted that since the two lines lOlOa and
lOlOb are spaced further apart, the voltage produced
in the secondary winding lOOOa is moderately down
-53-
from the peak voltage delivered at 1008.
In designing a transformer suitable for use
in a conventional amplifier power system as described
above, careful consideration must be given to acco-
modating the idling current in the primaryn Idlingcurrent is the current which flows in the primary
when no current is flowing in the secondary. In
a transformer having a small number of windings in
the primary and thus a small inductance, the primary
idling current may become large enough to cause the
transformer to heat up to an undesired extent. This
fact dictates the use of a primary coil having a
large number of windings.
A suitable audio amplifier transformer of con-
ventional design must also be capable of accomodatinga relatively large current flow through the primary
and secondary coils in order to handle peak power
demands. Thus the wire forming the coils must be
of sufficient diameter to allow the transformer to
deliver high current at peak loads, without too much
internal resistance. The result is a very large,
heavy transformer having a large number of windings
- to keep the inductance in the primary sufficiently
high, and relatively thick wire to keep the-resistance
relative low in spite of the rather long length of
wire in the transformer.
In contrast with a conventional power supply
transformer, a transformer desi~ned for use in the
duty cycle controlled power supply of the present
invention will normally be ormed with a higher sec-
ondary to primary turns ratio than is conventional
in non-duty cycle controlled transformers employed
! in commercial amplifier power supply circuits. With
' such a turns ratio, the point on the sine wave input
~.~.6~
-54-
to the primary at which current would stop flowing
in the secondary can be made to occur well down the
back slope as illustrated at point 1012 of Figure
lOC. Without duty cycle control the power supply
equipped with such a transformer would be supplying
voltage substantially above the desired plus 75 and
minus 75 volt levels normally required by conven-
tional audio amplifiers. With duty cycle control,
no current will flow in the primary of ~he trans- --
former when the duty cycle switching element is open
except for very small leakage currents allowed by
the solid state switching element when in the open
condition. These leakage currents can be ignored
for purposes of this discussion. The switching
element remains in a non-conducting position until
the voltage in the primary drops to a point 1014
just above the 1012 level. Then, current flows in
the secondary between points 1014 and 10120 If the
switching element is a self-commutating SCR, the
switching ele~ent will remain in a conducting state
down to point 1016 but no current will be flowi~g
in the secondary from point 1012 t~ point 1016 since
the diodes connecting the storage capacitor's in
the power supply to the transformer secondar-y will
become back biased.
Causing current flow in the secondary of a duty
cycle controlled power supply during the time between
points 1014 and 1012 of Figure lOC would appear at
first observation to be a little more inefficient
than causing secondary current flow in a conventional
amplifier as represented by 1008 in Figure lOB.
This is because the capacitors 1004 and 1006 only
! want to accept current at the 75 volt level. Thus,
there are some resistance losses which occur in the
-55-
.
duty cycle controlled transformer, these being repre-
sented by the shaded triangle between points 1012,
1014 and lOlB in Figure lOC. However, a duty cycle
controlled transformer can get by with many fewer
turns (only a small fraction of the turns in a conven-
tional transformer), so that the length of wire in
the transformer is reduced. This reduces the internal
resistance of the transformer proportionately.
Let it now be assumed that the smaller duty
cycle controlled transformer is operated at peak
power requirements as is illustrated in Figur~ lOD.
In this situation the switching element moves the
"switch on" point further up the sine wave, which
would be at a maximum at approximately point 1020
near the peak of the sine wave. Let it further be
assumed that the capacitors 1004 and 1006 are suffi-
ciently large so that they are maintaining voltages
of plus 75 and minus 75 volts pretty much at that
level. The voltage generated in the secondary at
point 1020 would be substantially higher than the
plus 75 volt level (possibly 90 volts), ignoring
losses in the transformer. Therefore, the voltage
difference represents the losses in the transformer
itself. These losses are represented in the shaded
triangle between points 1012, 1020 and 1022 in Figure
lOD. Due to the fact that peak power is rarely re-
quired by audio amplifiers for more than a very short
time, the somewhat greater losses represented by
Figure lOD can be tolerated in order to obtain the
compensat;ng gains of cutting off idling currents
during the first part of each cycle of the conven-
tional A.C. input sine wave. When the switching
element in the primary coil includes means for turn-
ing off the current flow in the primary coil prior
~6~B4
-56-
to return of the waveform to a zero voltage (such
as is illustrated in Figures 9A through 9C), even
greater reduction in idling current losses can be
achieved.
In summary, most of the time an amplifier power
supply is operating in the low power mode, as shown
in Figure lOC~ Accordingly,~the smaller duty cycle
controlled transformer of the present invention can
operate with about the same efficiency as the much
larger prior art transformer. This is due in part
to the number of primary and secondary turns being
cut down substantially, thereby permitting the trans-
former wire to be much shorter and thus offering
less resistance in the transformer itself. This
lessened resistance makes the existence of the idling
current in the latter portion of each half cycle
more tolerable. When higher power levels are re-
quired, there is potential for greater inefficiency.
However, this is offset by the low internal resist-
ance of the transformer, and in any case it is pos~sible to live with these higher inefficiencies for
a short period of time, since they will not be large
enough to overheat the transformer.
The following table includes the resul~s-of ~ -
testing several different transformer designs wherein
the transformers were tested by connecting the 50
volt output terminals of the secondary to two 150
watt light bulbs. The output of the secondary wind-
ing was maintained at 300 watts. Temperature was
measured at the top center portion of the transformer.
-57-
TABLE I
Turns Turns Resist- Temp C
Trans~ in RatioPrimary ance after
former Pri- Sec/ Wire Induct- Primary/ 21
Number marY Prim~ Size ance Sec. Min.
9 149 0.6~ #18~#17 58.6 485/733 31
8 131 0.82 #18&~17 52~7 400/78g 45
7 113 1.03 #18 32.2 335/812 62
1 90 1.14 #18 33.7 330/88~ 54
4 113 1.14 #18 32.3 330/993 67
6 113 1.14 #1~ 33.7 330/88~ 71
3 141 1.14 #20 58.1 538/1560 75
2 113 1.14 ~19 ~3 452/1470 84
177 1.14 #20 97 800/2340 107
The results of these tests indicate that a
preferable duty cycle transformer designed for opera-
tion in a power supply built in accordance with this
invention would be a transformer having a secondary
to primary turns ratio below 1.0 with a primary in-
ductance above 30 millihenries and a coil wire gauge
diameter above no. 18 when the transformer is used
to produce a maximum + 75 volt D.C. outpu~ from con-
ventional 117-125 volt, 60 cycle alternatin~ current.
Figure 11 depicts an amplifying apparatus 1100
designed to utilize the stepped voltage power- supply
illustrated in Figures 5 and 7A-7C. A signal voltage
is provided at 1102, and the output of the amplifier
is connected through a load (shown herein as a speaker
1104) to ground. The apparatus employs two sets
of transistors arranged in push-pull relationship,
with each set being connected in series. The first
- set QllOl, Q1103 and Q1105 are NPN transistors, and
these are used to amplify pos;tive portions of the
input signal. The other set of transistors Q1107,
i 35 QllO9 and Qllll are PNP transistors and are used
~6~
-58-
to amplify negative portions of~the input signal.
In the following description, the overall operaticn
of the first set of transistors QllOl, Q1103, Q1105
will be described in detail, with the understanding
that the same description would apply to the opera-
tion of Q1107, QllO9 and Qllll with respect to the
negative portions of the signal.
It will be noted that the emitter electrode
llQ6 of QllOl is connected to a power output terminal
1108 of the load 1104, and the collector electrode
1110 of QllOl is connected through a diode DllOl
to a D.C. voltage source El having a magnitude of
plus 25 volts. The emitter electrode 1112 of the
second transistor Q1103 is connected to the collector
`15 electrode lllO of QllO1, and the collector electrode
1114 of the transistor Q1103 is connected through
a second diode D1103 to an intermediate D.C. voltage
source E2 having a magnitude of plus 50 voltsO
Finally, the third transistor Q1105 has its emitter
electrode 1116 connected to the collector electrode
1114 of Q1103, and its collector electrode lllB is
connected directly to a higher D.C. voltage source
E3, which is shown as having a magnitude of plus
75 volts. --
As discussed previously herein, under the topic
heading "Background of the Invention'l, various ar-
rangements of series coupled transistors, with the
stepped voltage sources of increasing magnitude are
shown in the prior art. It is believed that a better
understanding of the operating features of the pre-
sent invention will be achieved by preceding a de-
tailed description of the present invention with
i a general discussion of the general mode of operation
of prior art devices using an arrangement of series
-59-
connected transistors with stepped voltage sources.
To disc~ss generally the prior art modes of
operation, when the signal voltage is relatlvely
small (e.g. below 25 volts) only the first transistor
QllOl would be conductive, and all of the power would
be derived from the 25 volt power source E1. The
obvious advantage is that there is less voltage drop
across the transistor QllOl and thus an increase
in efficiency.
When the signal voltage closely approaches the
value of the first voltage level, in prior art de
vices the signal voltage is then applied in some
manner to the base of the transistor Q1103 to make
it conductive, so that the power is derived from
the 50 volt source E2, with the 25 volt source being
blocked out by the diode DllOl. While the signal
is fluctuating between the 25 volt and 50 volt level,
substantially all or at least the major portion of
the voltage drop is taken across the second transis-
tor Q1103.
In like manner, when the signal voltage rises
above the 50 volt level, the voltage signal is ap-
plied to the base of the transistor Q1105 to make
it conductive and thus derive power from the--75 volt
power source E3. Also, with the signal voltage fluc-
tuating between the 50 and 75 volt level, substan-
tially all, or at least the major portion o~, the
voltage drop is taken across the third transistor
Q1105. Thus, with regard to the prior art devices,
each of the transistors must be made with the capability
of withstanding the voltage drop imposed across the
transistor at the current levels existing at th~
i various voltage levels.
~ Reference is again made to Figure 11. To discuss
-60-
.
specifically the present invention, it will be noted
that the signal input terminal 1102 is connected
through an operational amplifier 1120 to a biasing
transistor Q1113. The collector electrode of transistor
Q1113 is connected through a resistor RllOl to a plus
75 volt source. The base electrode 1122 of the trans-
istor QllOl is connected at a junction point between
the resistor RllOl and transistor Q1113 to provide
a forward bias to the transistor QllOl. The base
electrode 1124 of the second transistor Q1103 is
connected to a first switch and control means, indi-
cated at 1126, and the base electrode 1128 of the
third transistor Q1105 is connected to a second switch
and control means, indicated generally at 1130.
lS The second set of transi5tors Q1107, QllO9 and
Qllll are similarly connected. Thus, the biasing
transistor ~1113 is connected in series~ with a re-
sistor R1103 to a minus 75 volt source, with the
base electrode 1132 of transistor Q1107 connected
to a terminal between transistor Q1113 and resistor
R1103. the respective base electrodes 1134 and 1136
of transistors QllO9 and Qllll are respectively
connected to a third switch and control means 1138
and a fourth switch and control means 1140.--Nega-
tively stepped voltage sources E4, E5 and E6 areprovided in the same manner as the sources El, E2
and E3.
As shown herein, the input from 1102 is through
an operational amplifier 1120 to the base 1142 of
biasing transistor Q1113~ There is a feedback from
the output junction 1144 between the transistors
QllOl and Q1107 back through the resistors R1105
! - and R1107 to ground. From the junction 1146 inter-
! mediate transistors RllOS and R1107, there is a
~6
-61-
feedback connection back to operational amplifier
1120. The resistors RllO9 and Rllll provide an
initial biasing voltage to the transistors QllOl
and Q1107, respectively.
The general function of each of the switch and
control means 1126, 1130, 1138 and 1140 is to cause
a related transistor to become conductive at the
appropriate time and then to apportion the voltage
drop across the related transistor so as to minimize
the power which must be dissipated by any transistor
at any particular time. The manner in which this
is accomplished can best be described with reference
to the graphs of Figures 12A, 12B and 12C.
In Figure 12~ the voltage drop across the first
transistor QllOl is plotted against the output volt-
age. Let it be assumed that the signal voltage has
climbed to a low level of five volts. This voltage
is applied to transistor QllOl to cause it to become
conductive to transmit current from the postive 25
volt source El through transistor QllOl and to the
output terminal 1108. Thus the voltage at output
terminal 1108 will be approximately 5 volts, and
the voltage drop across transistor QllOl will be
approximately 20 volts. As the signal current increases -
to a value closer to the 25 volt level, the voltage
level at output terminal 1108 increases, while the
voltage drop across transistor QllOl decreases.
When the signal voltage comes within one or
two volts of the 25 volt level, the first switch
and control means 1126 becomes operative and directs
current to the base electrode 1124 of transistor
Q1103 at a voltage level intermediate the level of
the output voltage and the value of the positive
fifty volt source E2. The graphs of Figure 12A and
-62-
12B illustrate this relationship in a somewhat idealized
manner, where the first switch and control means
1126 functions to apply a voltage to base electrode
1124 consistently mid-way between the output voltage
and the plus 50 volt level at E~, so that the voltage
drop across the two transistors QllOl and Q1103 remain
substantially equal for all output voltages between
25 and 50 volts. In the actual embodiment shown
herein, the apportioning of the voltage drop across
transistors QllOl and Q1103 would depart moderately
from this idealized situation~
When the signal voltage comes very close to - ~
the 50 volt level, then the second switch and control
means 1130 makes the third transistor Q1105 become
conductive and also transmits base current to the
base electrode 1128 of Q1105 at a sufficiently high
voltage so that only a portion of the total voltage
drop is across the transistor Q1105. In like manner,
the first switch and control means 1126 continues
to supply current to the base electrode 1124 of transis~
tor Q1103 so that the voltage drop across Q1103 is
within its apportioned share of the total voltage
drop across the three transistors Q1105, Q1103 and
QllOl.
`25 Again, the somewhat idealized situation is shown
in Figure 12A~ 12B and 12C, in that with the output
voltage between 50 and 75 volts, the voltage drop
is equally apportioned among all three transistors.
In actual practice, the apportioning would not be
that precise.
The manner in which the four switch and control
means 1126, 1130, 1138 and 1140 operate will now
! be described. Since each of the four switch and
i control means are substantially identical,~only the
-63-
first means 1126 will be described in detail.
In the first switch and control means 1126,
there is a control transistor Q1115 having its col-.
lector electrode 1148 connected to the base electrode
1124 of the second power transistor Q1103. The base
electrode 1150 of transistor Q1115 is attached to
a junction point 1152 between two voltage dividing
resistors R1113 and R1115. The other end of the
resistor R1113 is connected to a positive 75 volt
terminal, while the other end of resistor R1115 is
connected to ground.
The emitter electrode 1154 of the transistor
Q1115 is connected through a resistor R1117 to a
juncture point 1156 between two voltage dividing re-
sistors R1119 and R1121. The other end of resistor
R1121 is connected to a positive 75 volt source,
while the other end of resistor Rlll9 is connected
to the main output line 1158 leading to the output
terminal 1108. A capacitor 1160 is connected in
parallel with the resistor Rlll9 to alleviate rapid
voltage changes across the resistors Rll9 and R1121.
As discussed previously herein, it is aesirable
to have transistor Q1103 become conductive when the
signal voltage (and consequently the output._voltage
which should be substantially identical to the signal
voltage) reaches a level just below the 25 volt levelO
It is also desirable to have the current supplied
to the base electrode 1124 of the transistor Q1103
at a voltage level approximately intermediate the
output voltage and the next stepped voltage in the
power source, which is the 50 volt power source E2.
Accordingly, when the output voltage reaches approxi--
I mately the 25 volt level, it is desired that the
~ base electrode 1124 of transistor Q1103 have a cur-
-6~-
rent delivered thereto at a voltage approximately
midway between 25 and 50 volts (e.g. appro~imately
37 1/2 volts)~
The value of the resistances R1113 and R1115
are selected so that when little or no base current
is flowing to the base electrode 1150 of the transis-
tor QlllS the voltage at junction point 1152 is ap-
proximately 37,5 volts. The values of the two resis-
tors Rlll9 and R1112 are so selected that when the
output voltage comes within one or two volts of the
voltage of ~he lowest power terminal (i.e. 25 volts)
the voltage at junction 1156 is approximately 3~.2
volts so as to apply a forward bias to the emitter
electrode 1154 of the transistor Q1115 to cause tran-
sistor Q1115 to conduct and transmit base currentto the base electrode 1124 of the transistor Q1103.
Since the collector electrode 1114 of transistor
Q1103 tends to follow the voltage of base electrode
1124 within a fraction of a volt, the immediate ef-
fect would be to bring the voltage at the emitterelectrode 1112 of Q1103 to approximately 37~5 volts.
Thus, with an output voltage of approximately 2~
volts, the voltage drop across transistor Q1103 would
be approximately 12.5 volts, and the voltage-drop
across transistor QllOl would be 12.5 volts, thereby
causing the power dissipated to be shared equally
by QllOl and Q1103.
As the signal voltage increases in the 25 volt
to 50 volt range, the voltage at junction point 1156
likewise increases so as to tend to drive the voltage
at the emitter electrode 1154 of transistor Q1115
upwardly. This causes an increase in current to
the base electrode 1150 of Q1115, thus raising the
voltage of junction point 1152 to a level closer
-65-
to the voltage of emitter electrode 1154 and also
causing transistor Q1115 to be more conductive so
that greater current is supplied to the base electrode
1124 of transistor Q1103 at yet a higher voltage.
The effect of this is to raise the voltage at emitter
electrode 1112 of transistor Q1103 even higher (i.e.
closer to the 50 volt level). Thus, as the output
voltage increases from the 25 volt level toward the
50 volt level, the voltage drop across transistor
Q1103 diminishes to apportion the voltage drop ~e-
tween the transistors Q1103 and QllOl.
By the time the signal voltage reaches the
second power source increment level (i.e. the 50
volt level), substantially all of the voltage drop
is across the load, and there is very little voltage
drop across the two transistors QllOl and Ql1030
At this time, the second switch and control means
1130 becomes operative to cause the third transistor-
Q1105 to become conductive. Because this is accom-
plished in substantially the same manner as in thefirst switch and control means 1126, the operation
of the means 1130 will be summarized only briefly.
It can be seen that there is a control tran-
sistor Q1117 having a collector electrode 1148a --
connected to the base electrode 1128 of transistor
Q1105. There are a pair of voltage dividing resis-
tors R1113a and R1115a producing a voltage level
of approximately 62.5 volts at juncture point 1152a.
Also, the two voltage dividing resistors R1119a and
R1121a are arranged such that when the output voltage
reaches a level just below the 50 volt level, the
voltage at junction point 1156a is approximately
! - 63.2 volts.
i Thus, when the output voltage comes quite close
~.~..6~
-66-
to the 50 volt level, a forward bias is applied be-
tween the emitter electrode 1154a of transistor Q1117
and base electrode llSOa to cause the transistor
Q1117 to conduct, thçreby supplying base current
to the base electrode ii28 of transistor Q1105 to
cause transistor Q1105 to conduct. As soon as Q1105
becomes conductive, the voltage of the emitter elec-
trode 1116 of Q1105 rises to a level close to that
of base electrode 1128 of Q1105 (i.e. approximately
63.2 volts). This causes the diode D1103 to block
off the 50 volt power source so that all of the power
is derived from the plus 75 volt power source.
When the output voltage is slightly above 50
volts, the voltage at which current is being delivered
through transistor Q1115 to the base electrode 1124
of transistor Q1103 is intermediate between the output
voltage and the voltage of the current to the base
electrode 1128 of transistor Q1105. Thus, the volt-
age drop from the 75 volt source to the level just
above the 50 volts being delivered to output terminal
1108 is apportioned between the three transistors
QllOl, Q1103 and Q1105. As the output voltage in-
creases further toward the 75 volt level, the voltages
at the ]unction points 1156 and 1156a increase propor~. -
tionately to raise the voltage o~ the currents respec-
tivley delivered to the base electrodes 1124 and
1128 of transistors Q1103 and Q1105, thus increasing
the voltage level at the respective emitter electrodes
1112 and 1116 of transistors Q1103 and Q1105. Conse-
quently, the voltage drop across the three transis-
tors QllOl, Q1103 and Q1105 continues to be appor-
tioned between the three transistors. As indicated
! previously, the apportioning illustrated in the graphs
of Figures 12A, 12B and 12C is somewhat idealized~
-67-
and the actual voltage drops will depart somewhat
from the precisely equal apportionment.
The operation of the third and fourth switch
and control means 1138 and 1140 is substantially
the same as that of the first and second switch con-
trol means 1126 and 1130 respectively, except that
the means 1138 and 1140 operate on the negative por-
tions of the input signal. Accordingly, the operation
of the means 1138 and 1140 will not be described
in detail.
It is sufficient to note that transistor of
switch control means 1138 is designated Qlll9 while
the control transistor of switch control means 1140
is designated Q1121. The control transistors Qlll9
and Q1121 operate in substantially the same manner
as corresponding transistors Q1115 and Q1117 to make
the power transistors QllO9 and Qllll, respectively,
conductive at the proper negative voltage levels.
- Transistors Qlll9 and Q1121 also control the voltage
level of the emitter electrodes of transistors QllO9
and Qllll to apportion the voltage drop across the
three transistors Q1107, QllO9 and Qllll.
Referring now to Figure 13, an alternative ar-
rangement of the output stage transistors and tran-
sistor control means for use in an audio amplifierof the type illustrated in Figure 11 is disclosed.
In particular, those components in Figure 13 which
are identical with the components of Figure 11 have
been identified by the same reference numerals. The
voltage dividing resistors R1115 and R1113 are se-
lected so that the voltage at 1152 will be 37 1/2
volts when the voltage at 1156 reaches approximately
38 volts as described above, thereby causing tran-
sistor Q1115 to conduct. This in turn causes the
~68-
emitter of transistor Q1103 to jump up to the 37
1/2 volt level, raising the input voltage to transis-
tor QllOl to 37 1/2 volts. Diode DllOl now operates
to block off the 25 volt power source. As the output
signal on line 1158 climbs toward 50 volts, the volt-
age at 1156 also climbs upwardly toward the 50 volt
level. By the time the audio signal reaches the
50 volt level, the voltage at 1156 will also have
reached 50 volts, thereby increasing the voltage
supplied to collector 1110 of Q1101 to 50 volts.
The operation of the Figure 13 circuit is iden-
tical to that of Figure 11 up to this point. How-
ever, as the input signal voltage rises still further
above 50 volts, transistor control 1130 operates
to switch transistor Q1105 on, thus causing the
potential appearing at emitter 1116 of Q1105 to be
applied directly to the collector of transistor QllOl
through diode D1303. Due to the bias at 1156a at
this time, the potential applied to the collector
of QllOl will be 67 1/2 volts. This will have the
effect of back biasing diode D1301 to cause voltage
from source E3 to be applied directly to transistor
Q1101 through transistor Q1105.
Figure 14 is a graph representing the operation
of the circuit of Figure 13, wherein line 1401 repre-
sents the output voltage on line 1158 of Figure 11
and line 1402 represents the voltage applied to the
collector 1110 of transistor Q1101.
A preferred embodiment of the left and right
channels`of a stereo amplifier constructed in accor-
dance with the present invention is illustrated in
Figures 15A, 15B and 16. Referring first to the
I left channel circuitry 1500 illustrated in Figures
15A and 15B, a left channel input signal is received
- ~9 -
at terminal 1502 and preconditioned in a high fre-
quency filter 1504 which rolls the audio signal off
beyond 20KHz. This filter acts to prevent transient
in~er-modulation distortion.. Upon leaving filter
1504, the input signal enters operational amplifier
1506 and thereafter passes to transistors Q1501 and
Q1503, which split the input signal into positive
and negative halves. The positive half of the signal
is fed to the upper half of the left channel amplifier
and the negative half of the signal is fed to the
lower half of the left channel amplifier. Because
the upper and lower halves of the left channel ampli-
fier are symmetrical, only the upper half will be
described in detail.
The output of transistor Q1501 is level shifted
upwardly through the action of resistors R1512 and
R1513 to the base of transistor Q1505. The signal
appearing at the output collector of transistor Q1505
is transmitted to transistor Q1509. At very low
output power requirements, the emitter current from
transistor Q1509 flows through series arranged diodes,
indicated at 1508, to the base of output transistor
Q1513, whereupon transistor Q1513 begins conducting.
Current from 25 volt power source 1512 then-passes
through output transistor Q1513 to an output inductor
1510 and on into the loudspeaker.
When output voltages above approximately 25
volts are required, the amplifier output current
is derived from the 50 volt power source 1514 through
output tran~sistor Q1517. Similarly, when output
voltages above 50 volts are required, transistor
Q1509 drives output transistor Q1521 to derive current
! from the 75 volt power soruce 1516. Switching circuit
1518, including transistors Q1525 and Q1527, acts
~3h6 2~34
-70-
to apportion the voltage drop across the power transis-
tors Q1513, Q1517 and Q1521.
The left channel amplifier includes an over-
current protection circuit 1520. In the event of
a short circuit across the output transistors, heavy
currents are drawn through the amplifier, thereby
creating a voltage drop across the emitter xesistor
R1571 of output transistor Q1513. This voltage drop
in turn switches over-current protection transistor
Q1533 on, and the current which normally flows through
transistor Q1505 to the base of transistor Q1509
is instead diverted to flow through the collector
of over-current protection transistor Q1533. Thus
deprived of its drive current, transistor Q1509 will
not turn on and the output transistors will not con-
duct. Consequently, the high power dissipations
otherwise occurring under short circuit conditions
are prevented.
Cross-over distortion is minimized by the action
of transistors Q1537 and Q1539 in cross-over preven-
tion circuit 1522. Transistors Q1537 and Q1539~
together with the 1-2-3-4 series arranged diodes
1524, resistors R1520 and R1521, and capacitor C1513,
form a bias network which develops a slight~forward
voltage drop between the bases of transistor Q1509
in the upper half of the left channel amplifier and
transistor Q1511 in the lower half of the amplifier.
This forward voltage drop places transistors Q1509
and Q1511 on the verge of conducting. When an audio
signal is received by the ampliier, Q1509 and Q1511
will conduct immediately and without discontinuity
in the amplifier output waveform, thereby resulting
in very low distortion of the audio signal.
For purposes of convenience t the values of all
-71-
the capacitors and resistors employed in the circuit
of Figures 15A and 15B are listed in Table II below.
Figure 16 illustrates the input portion of the
circuitry for the right channel ampli~ier. The right
channel amplifier includes a network for shifting
the phase of the incoming audio signal by 180 in
order to better utilize the amplifier power supply.
In all other respects, the right channel amplifier
circuitry is identical to that of the left channel
amplifier illustrated in Figures 15A and 15B.
Statistical analysis of stereo broadcasts indi-
cates that the vast majority of audio signals as-
sociated with one channel of such broadcasts are
in phase with the audio signals of the other channel.
Prior art high fidelity amplifiers generally process
incoming stereo signals without any modifications
of the phase between the channels~ operating in what
is known as single end~d fashion~ The components
of a stereo amplifier operating in single ended
fashion, however, tend to drain additional energy
from the power supply. When the amplifier output
voltage is high, the positive side of the power
supply furnishes power to both channels but the
negative side of the power supply does no wo~k.
When the amplifier output voltage is low, the nega-
tive side of the power supply furnishes power to
the amplifier but the psoitive side is not working.
Greater efficiency can be obtained from the
amplifier if both sides of the power supply work
continuously. In such situations, the power supply
is said to be operated in bridge. Power can be de-
livered in the bridge mode to a two channel stereo
I amplifier by inverting the incoming signals in one
-72-
of the amplifier channels and thereafter processing
both channels in an out-of-phase fashion. As a
result of the change in phase relationship between
the otherwise generally in-phaSe stereo signals,
positive power will always be required by one of
the two amplifier channels while the remaining channel
during any given power cycle will require negative
power. Thus~ regardless of the value of the ampli-
fier output voltage, the posi$ive and negative excur-
sions of the power supply during each power cycle
will both be employed. The greater power available
at the amplifier output due to the fact that the
power supply is being utilized more efficiently can
increase the output power by about 15-20%.
Again referring to Figure 16, the right channel
' amplifier is indicated generally at 1600. The audio
input signal to the right channel is received at
terminal 1601 and fed to inverting network 1602.
The inverting network, which consists of capacitors
C1601 and C1603 in combination with resistors R1601
R1603 and R1605, drives the inverting terminal o
operational amplifier 1604. The values of the net-
work components are listed in Table III below. Driving
the inverting terminals of operational ampli-fier
1604 produces an output signal which is 180 out
of phase with the input signal. As p'reviously dis-
cussed, the majority of audio signals in each channel
of a stereo broadcast are in phase. Consequently,
the use of the inverting network generally results
in a 180 phase difference between the operation
of the left channel and the operation of the right
channel of the amplifier.
Figure 17 is a preferred embodiment of the power
source for the left and right channel'amplifiers
.6~
-73-
illustrated respectively in Figures 15A, 15B and
16. Referring now to Figure 17, when switch 1700
is closed, current begins to flow from A.C. power
line 1702 through a phase shift network 1704 to Diac
1706 and Triac 1708. Triac 1708 turns on, permitt-
ing current flow through the primary 1710a of trans-
former 1710. The magnetic field in primary 1710a
builds up, transferring energy to transformer secondary
17]0b and then to electrolytic energy storage capa-
lQ citor banks 1716, 1718 and 1720. Storage capacitor
bank 1716 is designed to maintain a constant 25 volt
output at the 25 volt power source, capacitor bank
1718 is designed to maintain a constant 50 volt output
at the 50 volt power source, and capacitor bank 1720
is designed to maintain a constant 75 volt output
at the 75 volt power source. The capacitors in the
capacitor banks become fully charged within the first
100 milliseconds after the power supply is turned
on.
When the voltages at the three power supplies
respectively reach their preferred voltage levels
of 25, 50 and 75 volts, control transistor Q1701
is forced into conduction and the emitter current
from Q1701 flows through an LED diode 1712. --In
response to the emitter current, LED 1712 emits a
red light which shines on photoresistor 1714 and
lowers the photo-resistance value thereof. The
lowering of the resistance value of photoresistor
1714 in turn acts to shunt some of the current flow-
ing through phase shift network 1704, subsequentlyshifting the phase of the A.C. line signal and caus-
ing the Diac 1706 and Triac 1708 to fire at a later
point on the incoming A.C. line sine wave.
Changes in the firing point of the Diac and
-74-
Triac create variations in the conduction angles
and corresponding variations in the amplifier out-
put voltage. 5uch variations provide a means for
tracking the audio signal whenever the audio signal
5 frequency is below the repetition rate of the power
supply line, e.g., below a frequency of 120 Hertz
(2x60 ~ertz). The incoming audio signal is summed
at the junction of resistors R1765 and R1767, and
is low passed by the time constant of the parallel
combination of resistors R1765, R1767 and capacitor
C1733. The resulting signal is t'nen rectified by
diode D1709 to form a D.C. voltage which is propor-
tional to the output of the power amp:Lifier. This
proportional D.C. voltage is applied to capacitor
C1735, from whence it is fed to control trans;stor
Q1701. ~ontrol transistor Q1701 thereafter controls
the operation o~ LED 1712 to vary the time constant
of phase shift network 1704 as discussed above,
generating greater amplifier output voltages under
high signal conditions and lesser output voltages
under lower signal conditions. The output of the
power supply thus effectively tracks incoming audio
signals having frequencies in the low audio range~
This tracking ability makes it possible to urther
reduce the cost, size and weight of the amplifier
unit.
Automatic shut down of the Figure 17 power sup-
ply as a result of over-current conditions is achieved
through the use of operational amplifier i722 and
transistors Q1703 and Q1705. If a fault condition
results in over-current delivery to the audio ampli-
fier, an over-currerlt trip signal from the circuit
of Figure 15A is fed to the base of transistor Q1707.
Transistor Q1707 switches on, causing the input port
-75-
of operational amplifier 1722 to go high. The output
of operational amplifier 1722 likewise goes high,
switching transistors Q1703 and Q1705 on. The emitter
of transistor Ql.705 is connected to the 25 volt
source and the collector of Q1705 is connected to
capacitors C1723 and C1725. When transistor Q1705
turns on, the charge from the 25 volt supply is
transferred to capacitors C1723 and C1725. Current
then flows through LED 1712, causing the LED to shine
brightly on photoresistor 1714. The resistance of
photoresistor 1714 is accordingly lowered to a value
sufficient to shunt virtually all of the current
from the phase shift network 1704~ thereby shutting
the power supply down.
When the power supply is shut down, LED 1712
is maintained in a lighted condition by the charge
stored ~n capacitors C1723 and C1725. After a short
period of time ~i.e., somewhere between 1/2 and 1
minute) the charge on capacitors C1723 and C1725
dissipates through the LED and the LED begins to
~o dark again. The resistance of photoresistor 1714
consequently begins to rise and the power supply
cuts back on. If the fault condition has been re-
moved, the power supply remains on and the audio
amplifier operates as before. If~ however, the fault
still exists, the over-current trip line activates
transistor Q1707 and the power shut-off sequence
is repeated.
An over-voltage trip network is indicated at
1724. The audio signal from the output of the audio
amplifier drives the network, including resistors
R1751, R1753, R1755, R1757 and R1759, capacitor C1731,
and diodes D1701 and D1703, to form a D.C. signal
which represents the time-average half-way rectified
76-
audio signal. Note that diodes D1701 and D1703 serve
as OR gates as well as rectifiers. The output at
the jUnGtiOn of resistors R1751 and R1753 (e.g.,
the time-average audio voltage) charges capacitor
C1731. Capacitor C1731 is chosen such that a value
representative of a pre-selected over-voltage value
will cause capacitor C1731 to trip operational ampli-
fier 1722, thereafter turning on transistors Q1703
and Q1705 to shut down the power supply in a manner
analogous to that which occurs during over-current
conditions.
If for any reason ~i.e., amplifier failure or the
dropping of a tone arm~ a D.C. component should appear
at the amplifier output~ a D.C. voltage will appear
at the junction-of resistors R1761 and Rl7630 This
voltage is carried through D.C. fault trip line 1726
to operational amplifier 1722, causing the operational
amplifier to trip. When the D.C~ component is posi
tive, diode D1705 will conduct into the positive
port of operational amplifier 1722 and the operational
amplifier will go high. When the D.C. component
is negative, diode D1707 will conduct into the nega-
tive or inverting port of operational amplifier 1722
and the output oE the operational amplifier-w-ill
also go high. In both cases, the power supply will
be shut down following the switching of transistors
Q1703 and Q1705 and the energization of LED 1712.
Tables listing the values of the various resis-
tors and capacitors illustrated in Figures 15A, 15B,
16 and 17 follow. As previously discussed7 Table
II contains listings for the circuitry of Figures
15A and 15B. Table III includes the components of
I the inverting network of Figure 16, while Table IV
discloses preferred values for the resistors and
capacitors of Figure 17.
~L~6~
--77--
TABLE II
VALUES OF THE RESISTOF<S AND CAPACITORS USED IN
THE LEFT CHANNEL AMPLIFIER OF E'IGURES 15A AND 15B
:
Resistors
5 R1501 - 15k52 R1526 - 39k Q R1551 -22kQ
R1502 - 2k~ R1527 - 100 n R1552 -18k n
R1503 - 6.2kQ R1528 - 100 Q R1553 -4.7k fl
R1504 -390Q R1529 - lkQ R1554 - 39Q
R1505 - 2.7Q R1530 - 5.6k Q R1555 -27kSL
10 R1506 -- 9.lkQ R1531 -- 120 ~ R1556 -- 4.7Q
R15t)7 - 1.5kQ R1532 - lkS~. R1557 - 3.3kQ
R1508 - 1.5kQ R1533 - 5.6kQ R1558 -2.2kQ
R1509 - 9.1kQ R1534 - 1205L R1559 -3.3k Q
R1510 - 1.5kQ R1535 - 1.5kQ R1560 -33 n
15 R1511 - 1.5kQ R1536 - 1.5k Q R1551 -2.7k Q
R1512 - 4.7kQ R1537 - 2.4kQ R1562 -2.7kQ
R1513 -910SL R1538 - 22kn R1563 - lUk n
R1514 - 47~ R1539 - 22k5L R1564 - 220 5L
R1515 - lkQ R1540 - 18k~ R1565 - 220Q
20 R1516 - 4.7kQ R1541 - 4.7k~L - R1566 - 6.2 S~
R1517 -910 Q R15~12 - 3!3Q - R 567 - 220 Q
R1518 - 47Q R1543 - 27kQ R1568 -220 Q
R1519 -- lkQ R1544 - 4.7kQ R1569 - 56Q
R1520 - 12kQ R1545 - 3.3kQ R1570 -6251
25 R1521 - 5.bkQ R1546 - 2.2kQ R1571 - .lQ
R1522 - 5kQ R1547 - 3.3kSL R1572 - .15
R1523 - 10Q R1548 - 33Q R1573 -2~7Q
R1524 - 10Q R1549 - 2.4kSL R1574 -2.7Q
R1525 - 12k Q R1550 - 22k Q
-78
TABLE II (Continued)
Capacitors
C1501 - 200 pf C1517 - .01 f C1531 ~ pf
'~ C1503 - .001 f ~1519 - ~01 f C1533 - .039 f
5 C1505 - 470 uf C1521 - 22 u~ C1535 - .039 f
C1507 - 100 pf C1523 - .0033 f C1537 - .1 f
C1509 - 100 pf C1525 - 22 uf C1539 - .1 f
C1511 - 200 pf C1527 - .0033 f C1541 - .33
C1513 - 4.7 uf C1529 - 180 pf C1543 - .33
10 C1515 - 10 pf
. _ . . _
TABLE III
VALUES OF THE RESISTORS AND CAPACITOR5 USED IN
THE INVERTING NETWORK OF THE RI&HT
- CHANNEL AMPLIFIER OF FIGURE 16
_
15 Resistors Capacitors
R1601 - 3k Q C1601 - 200 pf
R1603 - 12k ~ C1603 - 33 pf
R1605 - 15k Q -
--79--
TABLE IV
VALUES OF THE RESISTORS AMD CAPACITORS USED
IN THE POWER SUPPLY OF FIGURE 17
Resistors
-
R1701 - 1.2kQR1725 - 9.1 megQ R1747 - 2 megQ
R1703 - 150kQ R1727 - 510k Q R1749 - 1.6 meg5
R1705 - 180kQ R1729 - 5.6k Q R1751 - l99k Q
R1707 -- 27kQ R1731 -- 6.8k52 R1753 -- 3~3kQ
R1709 - 27kQ R1733 - 22k~ R1755 - 3.3kQ
R1711 - 150k S~ R1735 - 1 Q R1757 - 4-7k~
R1713 - 1.5kQ R1737 - 3.6k Q R1759 - 4.7ksL
R1715 - 390 Q ~ R1739 - 20k ~L R1761 - 3.3kQ
R1717 680n R1741 - 100kQ R1763 - 6.8kQ
R1719 - 200Q R1743 - 200kQ R1765 - 30kQ
R1721 - 680Q R1745 - 20k Q R1767 - 60kQ
R1723 - 10k Q
,
Capacitors
C1701 - .lf C1713 - 2200u~ C1725 - 2200uf
C1703 - .013f C1715 - 2200uf C1727 - 22u~
C1705 - .01~ C1717 - 2200uf C1729 - ~7uf
C1707 - 2200uf C1719 - 3000uf C1731 - .033f
C1709 - 2200u~ C1721 - 3000uf C1733 - 2.2uf
C1711 - 2200uf C1723 - 2200u~ C1735 - 2.2uf
~6~
-80-
Industrial ApplicabilitY
The above disclosed amplifier circuitry and
methods are of particular economic importance in
the field of audio amplifiers. The concepts disclosed
herein permit dramatic reduction in the weight and
cost of providing suitable power supplies and power
transistors for high ~idelity audio amplifiers and
are particularly well adapted to stereophonic systems.
As an example of the dramatic weigh~ reductions
possible through implementation of the concepts of
this invention, a completed commercial embodiment
of a power supply and amplifier rated at 400 watts
weighs only 4 kg. In contrast, the lightest prior
art commercial amplifier of comparlable power rating
weighs approximately 16 kg.
