Note: Descriptions are shown in the official language in which they were submitted.
1 OVERLOAD PROTECTOR
BACKGROUND OF THE INVENTION
The resent invention relates to overload
protectors for preventing excessive radio-frequency
energy, applied to an input, from reaching an output.
There are many applications in which a delicate
input circuit must be protected from excessive voltages,
current or power. For example, certain field effect
transistors and other devices cannot tolerate excessive
voltage at their inputs. Other examples include the
input to certain receivers. For example, in aircraft
weather radar systems, the input to a mixer can be
damaged by excessive power emanating from nearby radio
sources or reflected power originating from the radar
itself. Radar systems are especially susceptible to
overloading because they incorporate an antenna serving
the dual function of transmitting relatively high energy
pulses and receiving very faint signals of the same
form. In the event such an antenna is damaged or broken
off, it is likely that these high energy pulses will not
be safely transmitted but will be coupled directly into
the input of the radar receiver.
Such radar systems have been very difficult to
protect. Known methods of protecting the radar have
included rotating the weather radar antenna so it would
not receive damaging signals from reflections or from
other nearby, operating radar systems until the aircraft
has left the heavily trafficked area. One unsuccessful
method for protecting the radar system is to turn off
its power. However, even when power has been removed,
the radar front end and its sensitive components are
still exposed to receipt of damaging energy from nearly
high frequency sources.
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I 9
1 A known technique for protecting a sensitive
input is by shunting the input with one or more stages
of pin diodes. A relatively large, radio frequency
signal placed across the pin diodes will forward bias
them. The forward biasing will persist because of the
capacitance of the diode. This approach is inherently
limited since this power mutt be absorbed by the pin
diode which must -therefore have a high power rating.
Consequently, the pin diode tends to be rather slow and
will allow substantial power to reach the protected
circuit before the diode becomes effective. Moreover,
pin diodes do not provide a perfect short but will only
reduce the dynamic shunting impedance across the input
of the protected circuit.
With high frequency circuits it is often
desirable and practical to take advantage of the
relatively short wavelengths of signals propagating
through a circuit. For example, a -transmission line may
have one end shorted or open but depending upon the
effective electrical length of the transmission line,
the other end can appear as either an open or short
circuit. Similarly, depending upon the spacing of ports
on a transmission line, either complete or no coupling
will occur between ports. This phenomena is used in
circulators, directional couplers and hybrid couplers.
These various effects can be produced with wave guides
cables, micro strips, strip lines and through known
equivalent circuits that simulate the effect of a
transmission line.
Accordingly, there is need for a device for
protecting a delicate circuit by interrupting a higher
energy power flow more quickly and more completely than
has been possible with systems of the prior art.
I
1 SUMMARY OF THE INVENTION
In accordance with the illustrative embodiment
demonstrating features and advantages of the present
invention, there is provided an overload protector for
safely transferring signals from an input to an output.
The protector has a power-dissipating element, a
detection means and a diversion means. The detection
means it coupled to the input or providing a bias
current in response to a signal at the input in excess
of a predetermined magnitude. The diversion means is
coupled to the input, the output, the power-dissipating
element and the detection means. The diversion means
can receive the bias current and, in response, redirect
power at the input from the output to the
power-dissipating element.
In one embodiment, protection is provided to a
high frequency detection system having an antenna, a
high frequency power source and a power-dissipating
element. The detection system also has a phased means
having at least a first, a second and a third port. The
first and second ports are connected to the antenna and
power source, respectively. The first port is phased to
communicate with the second and third ports. The second
and third ports, however, are phased to prevent
communication between them. Also included is a
processing means for responding to signals having a
predetermined pattern to produce a detected signal. The
detection system also has a protector means coupled to
the power-dissipating element, the third port and the
processing means for diverting signals issuing at the
third port from the processing means to the
power-dissipating element. Thus, the processing means
is protected from excessive signals.
An embodiment of a protector according to the
principles of the present invention can selectively
transfer power from an input to an output. The
:~Z~8~
1 protector employs a diversion transmission means coupled
between the input and a power-di~sipating element for
conveying power there between. A reflex means of the
protector is coupled to the diversion transmission means
for reflecting power thereon away from the
power-dissipating means. The protector includes an
operative means for altering the extent of reflection
provided by the reflex means. Thus, the protector can
divert power from the output to the power-dissipating
10 element-
By employing devices of the foregoing type, a
highly effective protector is achieved that can quickly
and completely divert power from a protected circuit to
power-dissipating component. In a preferred embodiment,
the input is coupled through a directional coupler to a
node marking the start of two quarter wavelength
branches. One branch extending toward the protected
circuit is shunted by a first limiter diode, the other
line terminating in a power-dissipating resistor. A
controllable stub connected to this power-dissipating
resistor has another shunting limiter diode connected at
a spacing of one quarter wavelength from the
power-dissipating resistor. Preferably, these diodes
are forward biased by a Skeptic diode detector driven
by the directional coupler. Thus, an excessive signal
can effectively reconfigure the circuit to detour the
damaging power.
The protector can be extremely fast since these
Skeptic diodes can be designed to respond almost
immediately to the excessive incoming power by producing
a rectified current. The system, which is a passive
apparatus, can be designed to switch off in five to
twenty nanoseconds or better depending upon the
components chosen and the input power to be switched
off. In one constructed embodiment the protector was
rated for handling 1.5 Kilowatts with a 16 microsecond
~2~119
l pulse width (duty cycle of .003) and a 1.2 microsecond
recovery time.
By switching the power to an external load, the
protector can be constructed from micro strips which
might otherwise be damaged. Thus, notwithstanding
unintended loading due to mismatching, a greatly
improved power handling capability, about one order of
magnitude greater, is achieved because the redirecting
of power keeps real power at the diode an order of
magnitude lower than the maximum rating of the diode.
BRIEF DESCRIPTION OF THE DRAWINGS
The above brief description, as well as other
objects, features and advantages of the present
invention, will be more fully appreciated by reference
to the following detailed description of a presently
preferred but nonetheless illustrative embodiment in
accordance with the present invention when taken in
conjunction with the accompanying drawings, wherein:
Fig. l is a schematic block diagram of a high
frequency detection system including an overload
protector according to the principles of the present
invention;
Fig. 2 is a more detailed schematic of the
overload protector of Fig. l;
Fig. 3 is plan view of a micro strip version of
the circuit of Fig. 2;
Fig. PA is a simplified, partial, equivalent
circuit diagram illustrating some of the micro strip
transmission lines ox Fig. 3 when power it being
conveyed from input to output; and
Fig. 4B is a circuit diagram similar to that of
Fig. PA but showing conditions existing when power is
being conveyed from the input to a power-dissipating
element.
1 DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT
In Fig. 1 a high frequency detection system, such
as an aircraft weather radar system, operating at 9.35
GHz, is illustrated. It will be understood that this
environment is exemplary and various other sensitive
systems operating at other frequencies can be protected
instead. In this specification the term radio-frequency
means frequencies above audio and below infrared. The
illustrated radar system includes antenna 10 and
magnetron 12 connected to a first port 14 and second
port 16, respectively, of a phased means, shown herein
as wave guide circulator 18. Circulator 18 is phased so
that antenna 10 can communicate with either magnetron 12
or third port 20 but -third port 20 and magnetron 12 (a
high frequency power source) do not intercommunicate.
Wave guide port 20 is connected through wave guide to
coaxial cable transition 21 to input Jo of an overload
protector comprising protection means 24 and
power-dissipating element 26. Transition 21 may be a
commercially available device in which a wave guide
terminates with an internal conductive probe acting as
the center line of a coaxial cable. The output Jo of
protector 24 connects to the input of a processing
means, shown herein as the front end or mixer of a radar
receiver. Mixer 28 and its associated circuitry may be
the conventional radar circuit found in an aircraft
weather radar system, although other sensitive circuits
may be protected instead.
Referring to Fig 2, the previously mentioned
overload protector is illustrated in a more detailed
schematic diagram showing power-dissipating element 26
as an external grounded load. Element 26 is connected
to connector Jo which may be an OSM-type of connector.
Input Jo (an OSM-type female connector) connects to one
port of directional coupler 30 whose corresponding port
Lo l19
1 connects to node A. The two other corresponding ports
of directional coupler 30 are separately connected to a
matching 50 ohm resistor 32 and to the input of a
matching network 34. Matching network 34 is designed to
correct impedance mismatches that might otherwise exist
between the circuitry served by it. The input Jo is
designed to work into a 50 ohm characteristic impedance.
Directional coupler 30 is designed to have a -33dB
coupling between input Jo and the input of matching
network 34. Directional couplers such as coupler 30 are
known per so. See for example, Members of the Staff of
the Radar School, MUTT., Principles of Radar,
McGraw ill Book Co., Inch (1952). pp. 834-39; Dr. Max
Fogies, Modern Microelectronics, Research and Education
Association, New York, New York (1972) pp. 222-25. As
explained in those references, directional couplers can
be Fabricated from wave guides, cables, strip lines or
micro strips and the same phenomena can be produced with
an equivalent circuit composed of inductors and
capacitors. In this embodiment, directional coupler 30
is designed to afford duplex communication between input
Jo and node A.
The output of matching network 34 connects to a
detection means (also referred to as an operative mean)
in the form of a pair of unidirectional conducting
devices 36 and 38. Devices 36 and 38 are preferably
Skeptic diodes (for example type DMJ, manufactured ho
Alpha Industries of Woburn, Maws.) having both of their
anodes connected to the output of matching network 34.
A low past filter means includes a shunting
storage capacitor 39 connecting between ground (that is,
a reference potential and the junction of test point TO
and the cathodes of diodes 36, 38. The filter means
includes an inductor or chose 40 connecting between test
point TO and node C. Also, 68 ohm carbon composition,
direct-current return resistor 43 it shown connected
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between ground and test point TO.
Configured in this fashion, a sufficiently large
signal at input Jo can forward bias diodes 36 and 38 which then
act as detectors for producing a voltage across capacitor 39 and
a bias current Is through inductor 40 to node C.
Node C is coupled through beam lead capacitor 42
(10 pi) to the anodes of parallel limiter diodes 44 and 46 whose
cathodes are grounded. For the specified operating frequency
diode 44 may have a recovery time of 20 nanoseconds and diode
46, 10 nanoseconds. Diodes 44 and 46 may be pin diodes, type
numbers COLA 3132-02 and COLA 3131-01 respectively, by Alpha
Industries of Woburn, Mass. A direct current return is provided
by inductor 48 which connects between ground and the junction
of the anodes of limiter diodes 44 and 46 and the output connector
Jo.
The balance of the circuitry of Fig. 2 is herein
referred to as a diversion means. Line A-C between nodes A and
C is referred to herein as a main (or as a third) transmission
means. Line A-C is preferably a quarter wavelength transmission
line which may be formed from micro strips, although embodiments
employing wave guides, cables or other equivalent circuits are
possible. A variable impedance is provided by a semiconductor,
limiting diode 50, herein referred to as a cancellation (or as
an interrupt) means. Limiting diode 50 has its anode connected
to node C and its cathode grounded. In one preferred embodiment,
diode 50 is a pin diode, type COLA 3133-03 manufactured by Alpha
Industries, Woburn, Mass., having a recovery time of 50 Nina
seconds. Another quarter wavelength line segment between nodes
A and B, line segment A-B, is similar to line A-C and is referred
to as a diversion transmission means. Node B is coupled through
capacitor 52 (a capacitor identical to capacitor
..~,~
8 -
mob/
~23~
1 42) and OHM connectors I to 50 ohm matching termination
26. Line B-D, a second transmission means connected
between nodes B and D, is another quarter line similar
to the two other lines, line A-C and line By A
S shunting means (also referred to as a reloading or
reflex means) is shown as a shunting diode 54 with its
anode connected to node D and its cathode grounded.
Diode 54 is a semiconductor providing variable impedance
(that is, an impedance varying diode) and may be
identical to previously mentioned diode 50. A fourth
transmission means, line D-E, is connected between node
D and node E and is serially connected with line B-D to
form a line stub. In a referred embodiment line D-E is
effectively one half wavelength long.
Referring to Fig. 3, a practical embodiment of
the circuit of Fig. 2 is illustrated as a micro strip
circuit. It will be understood that this circuit could
be fabricated with discrete components where the various
transmission lines are synthesized by an equivalent
circuit, especially for lower frequencies. Alterna-
lively, the circuit could be made with wave guide
although the latter would be substantially more
difficult to fabricate. The illustrated circuit employs
a four-walled aluminum frame 60 onto which are mounted
the three previously mentioned OSM-type connectors Jo,
Jo and Jo'; The outer conductive cowls of connectors
Jo, Jo and Jo are screw mounted to aluminum frame 60.
The interior circuitry is mounted on a micro strip
board comprising an aluminum ground plane I
electrically and physically connected to frame 60 and
having the same outside dimensions as it. On the side
of frame 60 opposite plane 62 an identically sized
aluminum cover plate (not shown) is attached by screws
to the frame. Ground plane 62 has laminated to its
inside face a low loss dielectric material, preferably
composed of polytetrafluoroethylene which is .010 inch
1 thick. The various conductive strips illustrated upon
dielectric material 64 are metal laminations which may
be photo chemically etched into the pattern shown.
Micro strip board material can be obtained from Rogers
Co., of Chandler, Arizona, the dielectric material being
referred to as RT/Duroid 5880. In this schematic
components previously described in Fig. 2 bear identical
reference numerals.
The majority of the illustrated strip lines are
dimensioned to provide a 50 ohm characteristic
impedance. Specifically, the strip lines aligned
between connectors Jo and Jo, the previously mentioned
lines A-B, B-D, D-E, as well as the line between
connector Jo' and node B, are all designed to have a 50
ohm characteristic impedance. The width of these 50 ohm
strip lines it .031 inch. Similarly designed is the
strip line AYE running between elements AYE and 32 in
the directional coupler.
The directional coupler 30 includes a strip AYE
which is spaced about .035 inch from strip 30B for
approximately 0.225 inch. Directional coupler 30 is
designed to operate at an input frequency of 9.3S GHz
with -33dB coupling from input Jo to element AYE. The
right end yin thy view) of strip AYE is terminated ho
previously mentioned chip nest ion 32 which connect to
grounded pad 66. Pad 66 is a metal lamination resting
atop dielectric material 64 hut having a slot cut
there through and reaching the aluminum ground plane 62.
This typical slot is approximately 0.13 inch long and
0.031 inch wide with rounded end. The slot is
connected to ground plane I by soldering pad 66 to the
ground plane 62.
Previously mentioned matching network 34 is shown
herein a shunting capacitive element AYE, a widened
metallic pad for capacitively shunting signals to the
underlying ground plane 62. A strip line also .~31
if
1 inch wide) then reaches from shunting capacitor AYE to
previously mentioned Skeptic diodes 36, 38, shown
herein as a parallel combination, hermetically sealed
into a common package by the manufacturer. The strip
34C between component AYE and 36 is shunted about
approximately two thirds of the way towards component 36
by strip line inductor 34B which connects to grounding
pad 68, a pad again having a soldering slot for
connecting to ground plane 62. Strip 3~B is a
quarter wave direct-current return acting as a
radio-fre~uency choke. Components AYE and 34C provide
an impedance matching network so that the 50 ohm strip
line from strip AYE is matched to the lower impedance
presented by Skeptic diodes 36, 38. The dimensions and
thus the values of elements AYE and 34C are selected
according to the impedance at diodes 36, 38.
A shunting storage capacitor 39 is formed by the
area of pad 39 which is approximately .04 square inch.
Previously mentioned direct-current return resistor 43
connects between pad 39 and the illustrated (typical)
slot in grounded pad 70. A strip inductor, .007 inch
wide and .242 inch long connects between pad 70 and the
strip line running between connector Jo and diode I at
a point nearer to the diode. Diode 46 is generally in
the shape of a cubical chip waving terminals formed on
opposing faces. Its cathode face is solder-connected to
ground plane 62 exposed through the slot AYE cut through
dielectric material I Area 72 is grounded to prevent
bypass currents. The anode of diode 46 is connected to
the micro strip on either side of slot AYE by a 99-99%
pure gold ribbon, .005 inch by .0025 inch. A gap in the
micro strip between diodes I and 50 is spanned by beam
lead capacitor 42. Previously mentioned diodes 44 and
50 are situated in slots AYE and AYE, respectively,
(similar to slot AYE so that each of their cathodes
connect to ground plane 62. Again, their anodes connect
I
1 to a gold ribbon spanning the slows AYE and AYE.
The anode of diode 50 connects to previously
mentioned node C. The micro strip between nodes C and A
is the previously mentioned quarter wavelength trays-
mission strip and is, in -this embodiment, approximately
0.190 winch long, in view of the operating frequency of
9.35 GHz. Of approximately the same length is the
perpendicular micro strip line running from node A to
node B. Soldered between node B and strip 74 is beam
lead capacitor 52. Strip 74 leads to load connector
Jo'. The strip between node B and node D is similar in
length to line A-B. At node D limiter diode 54 is
soldered within slot AYE atop ground plane 62. Again,
the anode of diode 54 connects to a gold ribbon spanning
either side of slot AYE. A folded micro strip between
node D and E is approximately twice the length of strip
B-D and is open at node E.
To facilitate an understanding of the principles
associated with the foregoing apparatus, the operation
of the equipment of Figs. 1, 2 and 3 will be explained
using the simplified schematic of Figs. PA and 4B. In
Fig. 1, circulator 18 operates such that high frequency
power from magnetron 12 is transferred to antenna 10 in
short bursts without coupling a significant signal into
port 20. Signal reflected ho targets eventually cause
a return to be received by antenna 10 and coupled
through circulator 18 into port 20. After passing
through transition 21 the return signal is coupled to
input connector Jo of protector 24.
The relatively small signal appearing at
connector Jo does not produce a sufficient signal to
charge capacitor 39 fig. 2). Accordingly, any current
IT through inductor 40 is negligible. Thus diode 54 and
50 are not forward biased and remain essentially a very
high impedance Jan open circuit). As a result, the open
circuit at node still appears as an open circuit at
13
1 node D. One quarter wavelength therefrom at node B this
open circuit appears like a short across load 26. This
short at node B causes the line By to appear like an
open circuit from node A. Since there are no other
S diode or other components shunting the energy in the
micro strip connection between connectors Jo and Jo,
signals are conveyed without reflection between those
connectors. An equivalent circuit of the micro strip
under these conditions is shown in Fig. PA, wherein node
B is shown grounded to produce what appears to be an
open circuit when viewed from node A.
Referring again to Fig. 1, we now assume that the
magnetron pulse applied to antenna 10 is reflected back
into port 14 due to a nearby obstruction or due to
damage to antenna 10. This pulse is therefore at a
relatively high power level. Alternatively, a nearby
radar signal, a likely happening at a crowded airport,
can be directed into antenna 10 to produce an excessive
signal at port 14. Consequently, an excessive signal it
conveyed from port 20 to connector Jo. This pulse may
typically rise at the rate of 10 watts per nanosecond.
Accordingly, a significant amount of energy is
coupled from the directional coupler 30 (Fig. 2) through
matching network 34 to detecting diodes I 3B. The
high speed rectification provided by them causes a rapid
charging o capacitor 39. Consequently, a bias current
IBM eventually reaching about 60 ma flows through
inductor 40 and forward biases diodes 54 and 50. This
current dramatically reduces their dynamic impedance and
presents an effective short circuit from their anode to
cathodes. These effective short circuits cause the
grounding of the micro strips as illustrated in Fig. 4B.
Nodes C and D have been grounded by their respective
limiter diodes as illustrated in Figs. 2 an 4B. Since
the line A-C, having its nod C grounded, it a quarter
wavelength long, the effect of line A-C as seen from
~Z~8~1~
1 node A is that of an open circuit. Similarly, line B-D
has its node D grounded so that the line appears from
node B as an open circuit. Consequently, there is an
undisturbed signal path from input connector Jo -through
S line A-B and connector Jo to power-dissipating element
26. Therefore, the excessive power on connector Jo is
dissipated externally. Moreover, the effective open
circuit presented by line A-C provides excellent
isolation to keep destructive power from ever reaching
the protected circuits.
It will be appreciated that as the input power
rises, should any of it leak past line A-C, additional
protection is provided by diodes 44 and 46 (Fig. 2).
Such leaked power can forward bias diodes 44 and 46.
These diodes have a certain amount of capacitance so
that they effectively remain forward biased to shunt
power so that any signal reaching output connector Jo is
relatively small. Significantly, diode 44 and
especially diode 46, can be selected to have a very fast
response since these diodes need not dissipate much
energy.
When the excessive signal ceases, all of the
diodes can return to a relatively non-conducting state.
For example, diodes 44 and 46 can be discharged through
inductive choke 48. Similarly, capacitor 39 a well as
diodes 50 and 54 can be discharged through resistor 43
which is effectively connected in parallel across them.
With the just described embodiment, the bias
current IT can be generated rather quickly (5 to 20
nanoseconds). However, the return to normal operation
is designed to take somewhat longer, up to 1.2
microseconds, and is a function of the pulse width and
the power level applied. This recovery time is limited
by the time constant established by resistor 43. Also,
another limiting factor is the recovery time associated
with diodes 44, 46, 50 and 54, especially the latter
g
1 two. Once these elements discharge the system is then
in a condition to operate us originally described with
power flowing from input Jo to output Jo, essentially no
power being conveyed to power-dissipating element 26.
It is to be appreciated that various
modifications may be implemented with respect to the
above described preferred embodiment. For example, the
power rating of the foregoing system can be changed by
the expedient of specifying diodes with a different
power rating or by placing more or fewer diodes in
parallel to change the effective power rating
Additionally, it is possible to change the size of the
micro strip by simultaneously changing the dielectric
constant of the underlying nonconductive material.
Also, while a directional coupler is shown driving the
detector for producing bias current, alternate coupling
techniques can be employed, including an ohmic
connection. Also, in many cases a quarter or half
wavelength line can be increased by multiples of half
wavelengths without changing the effect of the system.
Also, while a 50 ohm characteristic impedance is
disclosed, clearly, in alternate embodiments, other
impedances can be employed. Furthermore, while a
micro strip configuration has been shown, discrete hard
I wired components, waveguid~ systems, strip line systems
or coaxial cable systems can be employed depending upon
the required power handling capability, reliability,
weight and size limitations, etc. Additionally, the
values of components and the specific components
selected can be changed depending upon the required
frequency, band width, power handling capability,
temperature stability, accuracy, leakage requirement,
interference immunity, etc. Also, wile the protector
has been shown guarding the input to a radar receiver,
any system for receiving an oscillating signal can be
protected by the foregoing circuit. In addition, short
~Z~8:~9
16
1 circuits provided by the illustrated diodes can be
accomplished by other devices including transistors, or
other fast switching devices. It will be further
appreciated that while in some instances a diode
produces a shorting effect resulting in an open circuit,
the length of awn associated transmission line can be
altered so that an open/shorted diode can produce either
an open or short circuit.
Obviously, many modifications and variations of
the present invention are possible in light of the above
teachings. It is, therefore, to be understood that
within the scope of the appended claims, the invention
may be practiced otherwise than as specifically
described.
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