Note: Descriptions are shown in the official language in which they were submitted.
lZ'~6688
TITLE OF THE IN~ENTION
METHOD OF DIGITAL SIGNAL TRANSMISSION
BACKGROUND OF THE INVENTION
1. Field of the invention:
The pr~sent invention relate~ to a method of
transmitting a digital signal along a multipath fading
transmission line such as a radio transmission in an ur~an
area.
2. Description of the Prior Art:
In recent years, digital signal processing
techniques have spread to the field of mobile communications
so as to meet increasing demands for communication privacy,
intelligent communication, a~d matching with peripheral
communication networks. In urban areas where such demands
are most intensive, the quality of communication is
considerably lowered by multipath propagati~n due to
reflection and diffraction caused by buildings and
surrounding geographical features. In digital transmission,
when the propo~ation delay time difference between waves
constituting the multipath is increased to such an extent
that it is no longer negligible with respect to the length
of a time slot, the bit error rate characteristics are
highly degraded by waveform distortions and follow-up
failures in synchronous systems.
FIG. 1 is a phase transition waveform diagram showing
an example of phase transition waveform of a transmission
signal transmitted by a digital signal transmission method
according to the present invention;
FIGS. 2 through 12 are phase transition waveform dia-
grams showing examples of phase transition waveforms of trans-
mission signals transmitted by a digital signal transmission
method according to the present invention;
FIG. 13 is a phase transition waveform diagram showing a
specific example of phase transition waveform of a transmission
signal transmit~ed by a digital signal transmission method
according to the present invention;
FIGS. 14 through 18 are phase transition waveform
lZ7~i688
diagrams of other examples of phase transition waveforms in
a time slot of FIG. 1;
FIG. 19 is a block diagram of a generator circuit
for a transmission signal according to a first embodiment of
the present invention;
FIG. 20 is a block diagram of a circuit
arrangement of a quadrature modulator 1905 as shown in FIG.
19;
FIG. 21 is a block diagram of a circuit
arrangement of a differential coding circuit 1902 as shown
in FIG. 19;
FIG~ 22 is a block diagram of a circuit
arrangement of a wveform generator circuit 1904 as shown in
FIG. 19;
FIG. 23, is a block diagram of another circuit
arrangement of the waveform generator circuit 1904 as shown
in FIG. 19;
FIG. 24 is a block diagram of a differential
detector of a binary-phase system;
FIG. 25 is a block diagram of a differential
detector of a quadrature-phase system;
FIG. 26 is a block diagram of a differential
detector of an octal-phase system;
FIG. 27 is a diagram explaining a detected output
signal in the presence of a two-wave multipath propagation
in a~digital signal transmission method of the present
invention;
FIG. 28 is a vector diagram illustrating the
combined phase of a D- and U-waves in order to determine the
detected output in FIG. 27;
FIG. 29 is a block diagram of a diversity model in
the presence of multipath propagation in a digital signal
~ transmission method of the present invention;
12'7~6~8
FIG . 3 0 is a diagram explanating a detected output
signal in the presence o~ a two-wave multipath propagation
in a digital signal transmission method of the present
invention;
FIGS. 31 through 35 are graphs showing average bit
error rate characteristics according to the present
invention in the presence of two-wave Rayleigh fading
FIG . 36 is a block diagram of a transmitter
circuit according to a second embodiment of the present
invention;
FIG. 37 is a phase transition waveform diagram
showing the phase transition of a transmission signal
transmitted by a first conventional digital signal
transmission method;
FIG. 38 is a diagram explaining a detected output
signal in the presence of a two-wave multipath propagation
in the first conventional digital signal transmission
method;
FIG. 39 is a vector diagram illustrating the
combined phase of a D- and U-waves in order to determine the
detected output in FIG. 38;
FIG. 40 is a phase transition waveform diagram
showing the phase transition of a transmission signal
transmitted by a second conventional digital signal
transmission method;
~ - FIG. 41 is a block diagram of a detector in the
second conventional digital signal transmission method;
FIG. 42 is a diagram explaining a detected output
signal in the presence of a two-wave multipath in the second
conventional digital signal transmission method; and
FIG. 43 is a vector diagram illustrating the
combined phase of a D- and U-waves in order to determine the
detected output in FIG. 42.
668~3
A first conventional digital signal transmiscion
method will be described, by way of example, with
reference to FIGS. 37 through 39~
FIG. 37 is a phase transition waveform diagram
showing the phase transition of a signal transmitted by the
first conventional digital transmission method. Designated
at T is a time slot interval which is a minimum unit for
transmitting one data symbol. When the data is a 1, a 180
phase transition occurs, and when the data is a 0, no phase
transition takes place. This modulation technique is called
differential coding BPSK (Binary Phase Shift Key~ng).
Such a transmitted signal can be detected, for
example, by differential detection (delay line detection)
using a delay line with a delay of one time slot. As a
typical example of multipath propagation, consideration will
be given as to how a detected output signal behaves in the
presence of a two-wave multipath transmission line having a
propagation delay time difference which is not negligible as
compared with the time slot interval. A wave arriving
earlier will be called a D~wave, and a delayed wave will be
called a U-wave.
FIG. 38 is a diagram explaining how the detected
output signal behaves when the transmitted signal as shown
in FIG. 37 is subjected to the differential detection in the
presence of the two-wave multipath propagation. FIG. 38(a)
shows a phase transition of the D-wave. A phase transition
of the U-wave which arrives with a propagation delay time
difference from the D-wave is illustrated in FIG. 38(b).
The detected output at a particular time is the vector inner
~ product of the combined phase of the two waves at that time
-- 4 --
~Z~66~3
'- . ..
waves at that time and the combined phase of the two waves
in a preceding time slot. For e~ample, the detected output
in a region B in FIG 38(c) is the vector inner product of
the combined phase of the two waves at a time B' and that
at a time B.
FIG. 39 shows the combined phase of the D- and
U-waves in order to determine the detected output at each
of the times A through C. The ratio of the amplitudes of
the D- and U-waves is indicated by p, and the phase
difference between them by ~. For example, the absolute
value of the detected output at the time B is the inner
¦ product of vectors OB' and OB in FIG. 39, i.e., the square
of the line segment OB. Therefore, the detected outputs at
¦ the respective times A through C in ~IG. 38~c) are given,
using the cosine theorem, as follows:
A ... indefinite
B ... an~l + p2 + 2p cos ~)
C ... indefinite
i whe~e an (an = + 1) is a data sequence being transmitted.
¦ In the regions A and C~ the detected outputs are
indefinitedepending on the data valuec in the preceding and
subsequent time slots. After the differential detection, a
low-pass filter is normally placed in order to remove
1 harmonics and undesired noise components Therefore, the
I waveform o a final detected output signal is as shown by
I the dotted line in FIG. 38(c), which is produced by
filtering the solid-line waveform of ~IG. 38(c), and
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constitutes a portion of an e~e p~ttern. If p is close to 1
and a is about 180, the detected output in the region B
which is an effective detected c~ltput is substantially zero.
Thus, the eye is closed, and the bit error rate
characteristics are degraded At this time, since the
ineffective detected outputs in the regions A and C are much
larger than the effective detected output in the region B,
the eye is largely fluctuated in the direction of the time
base, making a reproducing clock unable to follow it and
thereby resulting in a greater degration of the bit error
rate (See, for example, "Bit Error Rate Characteristics in
Rayleigh Fading Having a Propagation Delay Time Difference"
by Onoe et al, Papers of Tech. Group on Commun. Syst., IECE
JAPAN, CS81-168, 1982, or "Analysis of Instantaneous Bit
Errors due to Multiwave Propagation and An Error Generating
Mechanism based on A Bit Synchronous System" by Takai et al,
Papers of Tech. Group on Commun. Syst., IECE JAPAN,
CS83-158, 1984).
In order to reduce the degradation of the eyes
pattern and the fluctuation of the eye causing the
degradation of the bit error rate characteristics, there has
been proposed a method such that the phase transition
waveform of a transmitted signal is designed so as to
produce a plurality of kinds of detected outputs and the
problems are improved by a diversity effect produced by
combining the plurality of kinds of detected outputs. One
example of such a second conventional digital signal
transmission method will be described below with reference
to FIGS. 40 through 43.
-- 6 --
~276688
FIG. 40 shows the phase transition of a signal
transmitted by the second conventional digital signal
transmission method. T indicates one time slot for data.
When the data is a 1, the phase is rotated twice in one
d~rection, each by 90 in every half time slot. When the
data is a 0, the phase is rotated twice in a different
direction from the above, each by 90 in every half time
slot. This modulation technique is called DSK (Double Shift
Keying).
Such a transmitted signal can be detected by a
differential detector having a delay line with a delay of
half time slot as shown in FIG. 41. Denoted in FIG. 41 at
4101 is an input terminal, 4102 a multiplier, 4103 a
half-time-slot (T/2) delay element, 4104 a low-pass filter,
and 4105 an output terminal. Unlike the first conventional
example, the carrier is rotated by 90 in phase by the
half-time-slot delay element 4103. As with the first
conventional example, consideration will ~e given as to how
a detected output sisnal behaves in the presence of a
two-wave multipath transmission line having a propagation
delay time difference t which is not negligible as compared
with the time slot.
FIG. 4~ is a diagram explanating how the detected
output signal behaves when the transmitted signal as shown
in FIG. 40 is subjected to the differential detection in the
presence of the two-wave multipath propagation. FIG. 42(a)
shows a phase transition of the D-wave. A phase transition
of the U-wave which arrives with the propagation
~Z7~6~38
delay time difference T from the D-wave is illustrated in
FIG: 42tb). The detected output at a particular time is
the vector inner product of the combined phase of the two
waves at that time and a phase attained by rotating
the combined phase of the two waves in a
~eding half time slot by 90.Eor example, the detected output
in a region B in FIG. 42(c) is the vector inner product of
the combined phase of the two waves at a time B and a phase
attained by rotating the phase at a time B' by 90.
FIG. 43 illustrates the combined phase of the D-
and V-waves in order to determine the detected output at
each of the times A through E. The ratio of the amplitudes
of the D- and ~-waves is indicated by p, and the phase
difference therebetween by a . For example, the absolute
value of the detected output at the time B is the square of
the line segment O~ in view of the fact that the vector OB is
pe~ ciær to thevector OB' in FIG. 43. Therefore, the
detected outputs at the respective times A through E in
FIG: 42(c) are given, using the cosine theorem, as follows:
A ... indefinite
... an~l + p2 ~ 2p co~ a1
C ... an~1 + p2 + 2anP sin a)
D ... an(1 + p2 + 2p cos a)
E ... indefinite
where an ~an ~ + 1) is a data sequence being transmitted.
In the regions A and E, the detected outputs are
indefinite dep~ing on the data values in the preceding and
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subsequent time slots. A~tually, the cut-off frequency of
the low-pass filter 4104 is selected to be low enough to
prevent intersymbol interference. Therefore, the output
signal that has passed through the low-pass filter 4104 is
produced by filtering the'solid-line waveform of'FIG.
42tc), and constitutes a portion of an eye pattern as shown
by the dotted line in FIG. 42(c). Since the regions B,
D and the region ~ produce complementary detected outputs
as described above, the eye will not be closed.
Furthermore, inasmuch as at least one of these effective
detected outputs does not become smaller than the
ineffective detected output in the region A or E, any
fluctuation of the eye Ln the direction of the time base i9
reduced,'and any degradation of the bit error rate due to a
follow-up failure of a re~x~w~ing clock is small.
With the second conventional digital signal
transmlssion method, as described above, the bit error rate
characteristics are largely improved in a mult~path fading
tra~smLssion line by a kind of di~ersity effect by
combining the mutually different outputs in the regions B,
D and the re~ion C, making high-speed digital
tran3mission possible (See, for example, "A Modulation
Technique Suffering from Less ~iming Fluetuation of Eye
Patterns in Multipath Fading" by S. Ariyavisitakul et al,
Papers of Tech. ~x~p on ~un. Syst., l~ J~PA~, CS84-67, 1984).
Because the second conventional digital signal
transmission method has two phase transitionc per time
_ g _
~27~i88
slot, however, it occupies a frequency bandwidth about ~wic~
that of the ordinary phase modulation, and hence is poor in
frequency utilization efficiency. This method is, in
principle, capable of only binary transmission, and cannot
reduce the bandwidth through multivalued transmission- The
bandwidth could be reduced to a certain extent by reducin
the amount of phase transition or smoothing the phase
transition more than a stepped pattern, but the bit error
rate characteristics would be highly degraded ("A Further
Study of Anti-Multipath Modulation Technique DSK ~
Analysis of Generalized DSK Modulation and Considerations
for a Narrow-Band Scheme" by S. Ariyavisitakul et al, Papers
of Tech. Group on Commun. Syst., IECE JAPAN, CS85-108,
1985).
Furthermore, according to the second conventional
digital signal transmission method, ~he bit error rate
characteristics can no longer be improved in principle when
the delay time difference exceeds 0.5 in terms of t/T which
is normalized with the time slot interval T. This is
because in a region in which t/T is 0.5 or more, the regions
B and D disappear, and the diversity effect which would be
produced by combining the two kinds of detected outputs is
no longer obtained.
SUMMARY OF THE INVENTION
It is an object of the present invention to
provide superior bit error rate characteristics in the
presence of multipath fading and to improve frequency
utilization efficiency through multi-phase transmission, or
to improve bit error rate characteristics with respect to a
multipath transmission line having a greater delay time
difference for allowing high speed transmission of a digital
signal.
~ In order to achieve the object, the present
invention employs, in a transmission apparatus for
transmitting digital data, a transmission signal having a
lZ76688
phase transition waveform in each of the tlmc slot~s of data,
the time-slot phase transition waveform having var~ing
change rate of phase or a phase jump, a time-sl~t phase
transition waveform in any desired time s]ot an~ a time-slot
phase transition waveform in a time slot which comes in
prescribed time slots subsequent to the de~ired time slot
being identical to each other in shape irrespective of
information to be transmitted, the information to be
transmitted being present in a phase difference betwcen the
same positions of the phase transition waveforms in the time
slots that are spaced apart from each other by the
prescribed time slots.
By using the above transmitted signal and
effecting a differential detection employing a delay element
capable of delaying the signal for the prescribed time
slots, different detected outputs according to multipath
propagation are produced. Through a kind of diversity
effect produced by combining these outputs with a low-pass
filter, bit error rate characteristics are highly improved
in the presence of multipath fading for allowing high-speed
transmission of a digital signal. By making the phase
difference transmitting the information multiphase, a
multivalued transmission is rendered possible with ease for
increasing frequency utilization efficiency without
degrading the characteristics with respect to the multipath
propagation. Dependent on the kinds of phase transition
waveforms in the time slots, improvement can be also attained
for a delay wave with t/T > 0.5.
The above and other objects, features and
advantages of the present invention will become more
apparent from the following description taken in
conjunction with the accompanying drawings in which preferred
embodiments of the present invention are shown by way of
illustrative example.
12766~38
DESCRIPTION OF THE PREFERRED EMBODIMENTS
A digital signal transmission method according to
an embodiment of the present invention will hereinafter be
described with reference to the drawings.
FIG. 1 is a phase transition waveform diagram
showing an example of a phase transition waveform of a
transmission signal transmitted by a digital signal
transmission method according to the present invention:
A time-slot phase transition wavefonm ~(t) (0 < t < T) in
a time slot of data is different from a conventional
phase modulation method of a fixed value in that its
differential coefficient is variable, or it has a phase
jump. FIG. 1 shows an example in which there is a phase
jump. The time-slot phase transition waveform ~(t)
has a stepped pattern with a phase jump, indicated by
, between a front halE portion Tl and a rear half portion
T2. 'rhe phase transition waveforms in a first time slot
and a ~n ~ l)th time slot which are spaced apart from each
other by n time slots are identical to each other in shape,
and.are shifted by ~ according to the.l-n~ormation to be
transmitted. Stated otherwise, n-time-910t differential
coding is effected. For example, when a binary-phase
qystem with 0 and ~ as 9 is employed, information of one
bit per time slot.can be transmitted, and when a quadrature-
phase system with 0, ~/2, ~, 3 ~/2 as 3 is employed,
information of two bits per time slot can be transmitted.
is generally indicated as follows:
~ 2m (m = 2P, p - 1, 2, 3...) ...~1)
where i is the data value which is converted into a Gray
code to be transmitted, and 0 < i < m, i ~ Integer.
Therefore, if the phase transition waveform in the first
time slot is ~(t~, then the phase transition waveform in
the ~n + l)th time slot i~ expressed as ~(t - nT) I ~.
~Z766~38
Assuming that a phase shift bearing information
is indicated as a phase shift ~a(t) from the absolute
phase, the phase shift ~a(t) is a stepped function having a
fixed value in each time slot, and can be expressed, using
a data vzlue sequence idq produced by the n-time-slot
differential coding of a data sequence iq ~q ~ Integer) that
has been converted to a Gray code to be transmitted, as
follows:
a~
~t) = q~ ~ idq 2m {U~t-qT) - U(t - ~q-l)T)}
I l (t > O)
U(t) -
0 (t < O) ...~2)
There may be a plurality of kinds of t~Y~slot phase
transition waveforms ~(t). For the n-time-
slot diferential coding, n kin~s of time-slot phase transition
waveforms ~l(t), ..., ~n(t) are available.
If it is assumed that
~ r(t) - O (t < O, t > T, 1 ~ r ~ n) ...(3)
then the general formula of~the phase transition waveform
~(t) of a transmission signal in the digital signal
transmission method of the present invention can ~e
expressed, using the equation ~2), as follows:
~t) ~ r (t - (qn+r-l)T) + ~ (t)
q=-~ r=l
_ ~ n~ ~r (t - (qn+r-l)T)
r=l
~ 2
+ q=_x idq ' -m {U(t-qT) - U~t-(q-l)T)~ (4)
The phase transition waveform of the transmission signal
according to the present invention resides in the first term
of the equation (4), with the second term being the same as
that of the conventional differential coding phase modulation.
The time-splot phase transition waveforms ~l(t), ~2(t),
~n(t) may include those which are
` 127~688
identical to each other. In a special case, all of them
may be identical to each other. At any rate, the ti~e-slot phase
transition waveforms ~t~ that are spaced from each other
by the n time slots need to be identical to each other.
~he value of n may be 1, and in such a case, the t~me-slot phase
transition waveforms ~(t) are of one
kind, and the phase transitiOn wave Frms in all of the time
slots are of identical shape. Where the time slot
pnase transition waveforms ~t) are of one kind, the phase
transition waveform ~(t) of the transmission signal is
expressea, using the equation (2), as follows:
~(t) = ~ ~(t-qT) + a (t)
q
co .
= ~ ~tt-qT)
q=_~
~ q~ ~ idq- 2m {U(t-qT) - U(t-(q-l)T)}
FIG. 1 shows the stepped tlme-slot
phase transition waveform ~t) as described above.
A duty ratio ~l/T indicating the position of a phase jump
may be of any desired value. The duty ratio may of course
be SO %, that is, the phase jump ~ may be located centrally
in the tirne slot. The phase jump ~ may be of any desired
magnitude and may be leading or lagging in phase. In a
special case, the phase jump ~ may be ~. As described
later on, as the phase jump ~ is larger, i.e., it is closer
to ~, the characteristics under multipath fading are
improved, but the envelope with a limited band varies to a - `
large extent.
As described above, there may be a plurality of
kinds of time-slot phase transition waveforms ~(t)o~IG~ 2 shows a case in
which there are a plurality of kinds of phase jumpC ~. FIG. 3 shows a case
in which there are a plurality of kinds of duty ratios. FIG. 4 illustrates
a case in which there are a plurality of k~s of ccmb~tions of phase jumps
.. . ~ . ~ . .
I4 - -
~27668E~
and duty ratios. FIG. 5 shcws a case in which the transition dire~tions
of p~ase jumps ~ are alternately leading and lagging. FIG. G illustrate~
a case Ln which there are a plurality of kinds of masnitudes of
phase jumps ~, and FIGS. 7 an~ 8 show cases in which there are . a
plurality of kinds of duty ratios, with the distance n between oorresponding
time slots being of an even numker and in~icated by 2n'. FIG. 9 shows a
case in which the transition directions of phase jumps ~ constitute a quasi-
random binary seguence having a period equal to the di~tance n between
oorresponding time slots~ FIG. 10 shows a case in which there are fur~her
a pl~x~ity of k~s of magnitu~es of phase jumps ~, and FIGS. 11 and 12
show cases Ln which there are a plurality of kinds o duty ratios.
FXG. 13 is a phase transition waveform diagram
showing the phase transition waveform of a transmission
signal transmitted by a digital signal transmission method
of the present invention, in which the time-slot
phase transition waveforms ~(t) are of one kind and of a
stepped pattern with the duty ratio being 50 % and the
phase jump ~ - ~, and in which n = 1, i.e., one time-510t
dlfferential coding, and multiphase number m - 4, so that
two bits can be transmitted per time slot.
The time-slot phase transition waveforms
~(t) may have a plurality of steps as shown in FIG. 14.
The time-slot phase transition waveforms ~(t) may
be of a triangular shape as shown in FIG. 15, instead of
the stepped pattern. The gradient of straight lines
constituting the phase transition waveform corresponds to a
frequen~y shift from the carrier frequency. Therefore, for
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the e~ample of FIG. 15, one time slot is composed of two
fre~uencies. The gradient of straight 1ines cons~ituting
the time-slot phase transition waveform may be of
various ~inds as shown in FIG. 16, and may have a p`nase
jump as sho~n in FIG. 17. Fur~herm~ore, in orcer to reduce
the transmission spectral wid~h, the phase transitlon
waveform may be of a pattern p.ocuced by smoothins any one
of the above-mentioned ~aveforms. For e.ca~ple, it may be
of a raised cosine waveform or a Gaussian waverorm as shown
in FIG. 18.
A method of producing the transmission signal as
described above will hereinafter be descrioed with
reference to embodiments of t~e invention.
FIG. 19 is a block diagram of a generator circuit
for a transmission signal according to a first embodiment
of the present invention. Designated in FIG. 19 at 1901 is a
data input terminal; element 1902 is d difrecential coding circuit,
element 1903 is an oscillator; element 1904 is a waveform yenerator circuit;
ëlernent 1905 is a quadratucelnodulator, and element 1906 is a transmission
signal output terminal. Digital data to be trarl~mitted is supplied
from the data input terminal 1901 and subjected to
differential coding in the differential coding circuit
1902. The waveform generator circuit 1904 generates
modulation signals in I- and Q-phases according to the
differential coded data- The oscillator 1903 produces a
~ carrier, which is modulated by the modulation signals in
the I- and Q-phases in the quadrature modulator 1905 into a
i8~
transmission signal that is outputted from the transmission
signal output terminal 1906.
FIG. 20 shows a circuit arrangement of the
quadrature modulator 1905 shown in FIG. 19. Denoted in FIG.
20 at 2001 is a 90 phase shifter; elements 2002 and 2003
are balanced modulators, and element 2004 is a combiner.
The carrier signal supplied from the oscillator 1903 is
modulated by the I-phase modulation signal from the waveform
generator circuit 1904, using the balanced modulator 2002,
into an I-phase modulation signal from the waveform
generator circuit 1904, using the balanced modulator 2002,
into an I-phase modulated signal. The carrier signal is
also phase-shifted by 90 by the 90 phase shifter 2001, and
then modulated by the Q-phase modulation signal from the
waveform generator circuit 1904, usiny the balanced
modulator 2003, into a Q-phase modulated signal. The I- and
Q-phase modulated signals thus produced are combined by the
combiner 2004 into a modulated transmission signal, which is
outputted from the transmission signal output terminal 1906.
FIG. 21 illustrates a circuit arrangement cf the
diff~rential coding circuit 1902 shown in FIG. 19. Denoted
at 2101 and 2104 are Gray code converter circuits; element
2102 is an adder, and element 2103 a delay element.
When the multiphase number is m (m = 2, 4, 8 ... ),
a p-bit parallel data value sequence indicated by the
equation (1) is applied to the Gray code converter circuit
2101. The data value sequence iq that has been converted
~ into a Gray code is applied to the adder
- 17 -
lZ76688
2102 in which it is added (MOD m) to data which has been
produced by delaying the output of ~he alder ~02 by the
delay element 2103 for the n time slots, i.e., n clock
ulses. The output from the adder 2102 is converted by the
Gray code converter circuit 2104. Therefore, the p-bit
input parallel data value sequence is converted into a Gray
code to produce a p-bit parallel data value sequence idg
subjected to n time-slot differential coding.
FIG. 22 is illustrative of a circuit arrangement
of the waveform generator circuit 1904 with respect to a
quad-rature-phase system in which the phase transition
waveform ~(t) is indicated by the equation (5). Designated
at 2201 is an I-phase data input terminal; element 2202 is a
data clock output terminal; element 2203 is a Q-phase data
input terminal; elements 2204 and 2206 are shift registers;
element 2205 is a binary counter; element 2207 is a ROM
(Read-Only Memory); element 2208 is a clock generator;
elements 2209 and 2210 are D/A converters; elements 2211 and
2212 are low-pass filters; element 2213 is an I-phase
modulation output terminal, and element 2204 is a Q-phase
modulation output terminal. For the ~uadrature-phase
system, the output idg from the differential coding circuit
1902 is a 2-bit parallel data with its most- and
lesat-significant bits being supplied from the I- and
Q-phase data input terminals 2201 and 2203, respectively.
The supplied data sequences are delayed respectively by the
shift registers 2204 and 2206 to provide modulating data in
the present time slot and modulating data in time slots
subsequent to and preceding in the present time slot. In
FIG. 22, the shift registers 2204 and 2206 produce
- 18 -
12766~38
at Qd the modulating data in the present time slot, and at
Qe through Qg and Qa ~h~gh Qc the mo~ating da~a in the ~hxe time
slots ~ubsequent to and preceding the present time slot.
The I- and Q-phase modulating waveforms are written in the
ROM 2207 according to the mcdulating data. In FIG. 22, each
time slot i8 composed of 16 ~ampling point~. Addresse~ A7
through A17 of the ROM 2207 are used as select signals for
determining which modulating data is to be det~cted, and are
supplied with the mXh~ating data in the three time ~lot~
sub~equent to and preceding the present time ~lot. To
addres~es A0 through A3 of the ROM 2207, there is ~pplied a
signal, as a modulating-waveform readout signal, which is
produced by Ere~uency-dividing a reference clock slgnal
from the clo~k generator 2208 with the binary counter 2205.
The ROM 2207 produces outputs X0 through X7 and Y0 through
Y7 which are converted into analo~ signals by the D/A
c~nverter~ 2209 And 2210' and the low-pa8s filters ~211, and 2212
which remove olded components, the analog signals serving
as the I- and Q-phase modulating signals. For modulation in a
re nultl-pbase system such a3 an octal-phase system, it 18 neCe~sary
~o-ha~e a~ many shift registers as the number o~ p in
the equatlon (1) and corresponding ROM addresses.
The modulatLng waveform in each time slot which is
written in the ROM 2207 will be described below.
Basically, the I- and Q-phase modulating waveforms MI(t),
MQtt) may be obtained by the following equations from the
pha~e transition waveform ~t) of the transmission signal
-- 19 --
~27668~3
which is derived by the equation (5) from the data value
sequence idq subjected to the differential codingand to be
transmitted:
. MI(t) = cos ~tt) -'
MQ(t) = sin ~(t) ...(6)
Since these waveforms are wideo2nd signals as they are,
they are limit~?d in band width by a oand~ itin~ filt~ havinq
an impulse response h(t). The equations (6) are then
modified as follows:
+to
MI(t) = Jcos ~(t ~ h(~)dl
...(7)
r~to
MQ(t) =) sin ~t -1) h~T)d~ '
Various band-limiting filters capable of passing lower
frequencies, such as of the cosine-square type or the
Gaussian type, may be employed. Such various filters have
different impulse responses h~t). As an example, the
impulse response h~t) of a cosine-square type filter having,
a cut-off angular frequency ~o and a roll-off coefficient y
is given by: ' ~
' h(t) = ~ sin ~ot cosY ~o t ...(8)
~ ~ot 1-(2y ~o t/~ )2
The cut-off angular frequency ~o should be selected to be
about the same as the modulation rate which is a reciprocal
of the time slot interval T. - :
In the R0.~ 2207 shown in FIG. 22, there are
written the I- and Q-phase modulating waveforms MI(t) and MQ(t)
,
:, . : '.' .
- 20 - .
1;2766~1~
for one time slot according to the equations (7). The
range of integration (-to, to) in the equations (7) is
selected to be about the same as the spread of the im?ulse
response h(t), and is equal to the three time slots
subsequent to and preceding the pzesent time slot in the
arranqement OL FIG. 22. The mcdula-~Lng dat~-in the chree
time slots subsequent to and preceaing the present .ime
slot are required in order to calculate the phase
transition waveform ~(t) from the equation (5). Therefore,
the waveform dat~ fo~ one tim~ slot is calculated by the ~uations (7) with
respect to all of the ~lo~ul~ting datd patte~ls in the ~e-ent ti.ne slot & the
three time slots ~ sequent to and preceding the present ti~e slot, and is
written in the RCL~ 2207. A modulating waveform is selected
by the addresses ~4 through A17 of the ROM 2207 which are
the modulating data in the present time slot and the three
time slots subsequent to and preceding the present time
slot.
This is substantially the case with a plurality
of kinds of phase transition waveforms ~(t) in the time
slots just like the time-slot phase transition waveform ~(t) is
indicated by the equation (4). The I- and Q-phase
modulating waveforms MI~t), MI(t) for one time slot, as
calculated by the equations (7), may be written in the ROM.
In determining ~(t) in the equations (7) from the equation
14), it is necessary to find the value of r (1 < r < n) in the
time-slot phase transition wavefon~ ~r(t) in the present time
slot. Therefore, the waveform datas to be written in the
~276~
ROM are calculated with respect to not only the modulating
data patterns but also representative of which one is the
time-slot phase transition waveform ~r(t) in the present
time slot.
Accordingly, the circuit arranfgement of the
waveform generator 1904 shown in FIG. 19 must be modified as
shown in FIG. 23. Denoted in FIG. 23 at 2201 is an I-phase
data input terminal; element 2202 is a data clock output
terminal; element 2203 is a Q-phase data input terminal;
elements 2204 and 2206 are shift registers; element 2205 is
a binary counter; element 2208 is a clock generator; element
2209 and 2210 are D/A converters; elements 2~11 and 2212 are
low-pass filtersi element 2213 is an I-phase modulation
output terminal, and element 2214 is a Q-phase modulation
output terminal. These components are the same as those
shown in FIG. 22. The circuit arrangement of FIG. 23
differs from that of FIG. 22 in that a binary counter 2301
indicating the present value or r is added, and addresses
A18 and A19 are added to a ROM 2302 for selecting a waveform
according the value of r. The binary counter 2301 has a
period n, which is 4 in the arrangement of FIG. 23.
A method of detecting a transmission signal in the
digital signal transmission method of the present invention,
as described above, will be described belo~.
In the digital signal transmission method of
present invention, a signal is detected by a differential
detector having a delay line for n time slots. This detect-
ing method is the same as the conventional method, and
described in detail, for example, in "Data Transmission", by
William R. Bennet ~ James R. Davey, McGraw-Hill Book Co. New
York, 1965. The detecting method will be described briefly
hereinbelow.
-
127~i688
FIG.24 illustrates a circuit arrangement of adifferential detector of a binary-phase system. Designated
in FIG. 24 at 2401 is an input terminal, 2402 a multiplier,
2403 a low-pass filter, 2404 an n-time-slot (nT) delay
element, and 2405 an output terminal. The signal is delayed
for the n time slots by the n-time-slot delay element 2404.
Unlike the second conventional arrangement, the carrier
phase remains identical at the input and output. The
low-pass filter 2403 not only removes a component having a
frequency which is twice as high as that of the carrier
produced by the multiplier 2402, but also serves to combine
a plurality of detected outputs. The low-pass filter 2403
is preferably a so-called Nyquist filter naving a cut-off
frequency that is half the symbol transmission rate, i.e.,
(1/2)T, and also having attenuation characteristics asymmetrical
with respect to the cut-off frequency.
FIG. 25 shows a circuit arrangement of a dif-
ferential detector of a quadrature-phase system. Denoted in
FIG. 25 at 2501 in an input terminal; elements 2502 and 2506
are multipliers; element 2503 is a -45 phase shifter;
element 2505 is a +45 phase shifter; element 2504 is an
n-time-slot (nTJ delay element; elements 2507 and 2508 are
low-pass ~ilters; element 2509 is an output terminal A, and
element 2510 is an element
- 23 -
~2~6~i88
output terminal B. The circuit arrangement of FIG. 25
differs fro~ that of FIG. 24 in that time -45 phase shifter
2503 and the +45 phase shifter 2505 are employed to effect
differential detection with respect te two mutually
perpendicular axes for demodulating 2-bit parallel data.
The other operation is the same as that of the circuit
arrangement of FIG. 24.
FIG. 26 shows a circuit arrangement of a
differential detector of an octal-phase system. Denoted in
FIG. 26 at 2601 is an input terminal; elements 2602 through
2605 are multipliers; element 2606 is an n-time-slot (nT)
phase shifter; element 2607 is a -22.5 phase shifter;
element 2608 is a +22.5 phase shifter; element 2609 is a
+67.5 phase shifter; element 2610 is a -67.5 phase
shifter; element 2611 through 2614 are low-pass filters;
element 2615 is a comparator; element 2616 is an output
terminal B. The phase shi~ters 2607 through 2610 serve to
effect differential detection with respect to three axes
which are 45 displaced for demodulating 3-bit parallel
data. The comparator 2615 detects whether or not both
inputs applied thereto are of teh same polarity.
~ The manner in which the digital signal
transmission method of the present invention exhibits good
bit error rate characteristics in the presence of multipath
fading will be described below.
FIG. 27 is a diagram in which a transmission
system of a binary-phase system having stepped time-slot
phase transition waveforms as shown in FIG. 1 is
~Z766~3~
employed as a transmission signal to be transmitted by the
digital signal transmission method of the invention, and
explains a detected output signal produced by detecting
such a transmission signal with the differential det~ctor
of FIG. 24. FIG. 27(a) shows the manner of D-wave phase
transition of a time slot and a time slot
which is n time slots subsequent to the former time slot. The
t~æ-slot phase transition waveforms in both o th~ time slots
are of an identical stepped pattern as described above.
The phase transision of the U-wave which comes with a time
delay equal to the propagation delay time difference ~ that
~ s not negligible as compared with the time slot interval T
is as shown ~n FIG. 27~b). The detected output at a
particular time is the vector inner product of the combined
phase of two waves at that time and the combined phase of
the two waves in a time ~hich is the n time slots
prior to the above time~ For example, the detected output
in a region B in FIG. 27~c) is the vector inner product of
the combined phase of the two waves at a time B' and that
at a time B.
FIG. 28 shows the combined phase of the D- and
U-waves in order to determine the detected output at each
of the times A through E. The ratio of the amplitudes of
the ~- and U-waves is indicated by p, and the phase of the
carrier of the U-wave as seen from the carrier of the
D-wave is indicated by a. The phase axis of FIGS. 27~a)
and 27(b) is such that the leading direction is positive.
lZ76~88
Based on FIG. 28, when the waveform is not dis.o ed bv
the low-2ass filter 2403, or the cut-off freauenct is
sufficiently high as compar-d with the ~ata t 2nsmission
rate, the detected outputs at the res~ec_iv2 times A
through E in FIG. 27(c) are as follows:
~egions A, E
inde inite
Regions B, D
1 + p2 + 2P cos ~ ... (9)
Region C
1 + p2 + 2 Pcos(~
In the regions A and E, the detected outputs are rendered
ind~finite depending on the data values in the preceding and
subsequent time slots. Actually, the cut-off frequency of
the low-pass filter 24~3 is selected to be low enough to
prevent intersymbol interference. Therefore, the output
signal that has passed through the low-pass filter 2403 is
proauced ~y filtering the solid-line waveform of FIG.
27(c), and constitutes a portion of an eye pattern as shown
by the dotted line in FIG. 27(c). Since the regions B,
D and the regions C produce complementary detected outputs
and will not be simultaneously eliminated with respect to
any value ofp or ~, the eye will not be closed.
Furthermore, inasmuch as at least one of these effective
detected outputs does not become smaller than the
ineffective detected output in the region A or E, any
fluctuation of the eye in the direction of the time base is
~766~8
reduced, and any degradation of the bit error rate,due to a
follow-up failure of a re~x~ucing clock i~ small.
Detected outputs under two-wave multipath propa~ation, includln~ a
multiphase system, will be considered. Assuming that a
transmission data sequence is an tan = + 1), a multiphase
number i~ m (m - 2, 4, 8, ...), and complex m~ltiplicative noises
indicating fading of the D- and U-waves are sl(t~' ant ~2(t)
the detected outputs in the regions B, C, D ca~ be
expressed as follows tthe complex envelopes of the D- and
U-waves are expressed by sl~v, s2-v where v (¦v¦ - i) is
the complex envelope of a transmitted wave):
Regions B, D
an5in~ ~/m) ~lsl ~ s212)
Region C ... ~10)
an~in~ ~/m~ ~Isl exp~ 52l2~
It should be noted in the equations ~10) that the carrier
phase of the D-wave in the detected output of the region C
is shifted by ~.
A~ described above, the principle of the present
invention for improving a digital signal transmis-
~ion in the presence of multipath fading can be said to
be a kind of diversity effect by combining different
detected outputs. Therefore, an appropriate diversity
model can be considered for evaluating bit error rate
characteristics. Now, a bit error rate in the presence of
two-wave multipath fading which meets the condition of a
delay time difference T with the both regions B, D and
~2~7~688
region C being present will be evaluated.
A model of a maximum ratio combiner 2901 supplied
with, as two branch inputs, signals Ul~v~ u2-v tv is the
complex envelope of a modulated signal, ¦v¦ = 1)
accompanied by fading expressed by complex mNltiplicative noises
ul(t), u2(t), and a differential detector 2902, as shown in FIG. 29, wdll be
concldered. In the maximum ratio combiner 2901, the sum of S/N
ratios of the inputs is equal to the S/N ratio of the
output ~where N is noise power). Therefore, at this time,
the complex envelope of the received signal at the input of
tne differential detector 2902 is indicated by
. _
~¦ul¦2 + ¦u2¦2 v, and the detected output from the
differential detector 2902 is given by:
an(¦ull2 1 lu212) ...~11)
When the two detected outputs in thè equation (10) are
combined equally, the first and second terms
of the equation ~11) correspond to the two detected outputs
of the e~uation (10), and are expressed by:
Ul ' ~sin( ~m)-(sl + s2) ...~12)
ul ~ ~sin( ~/m)-tsl exp~ s2)
Thus, lt will be understood that FIG. 29 shows a diversity
model employing stepped time-slot phase transition wave-
forms in the digital signal transmission method of the present
invention in the presence of two-wave multipath fading.
The equations (12) indicate that the diversity effect,
strictly speaking, has, at diversity branches, a combined
wave of coming waves having the delay time di~ference T and
- 28 -
~2~;688
a combined wave of those coming waves with the carrie~
phase difference ~etween them being shifted by ~.
The diversity characteristics employing the
maximum ratio combiner can analytically be evaluat~d.
Assuming that R is a complex variance matri~ < ~ > (i, j
= 1, 2) of {ui/~N } (i = 1, 2), the average bit error rate
Pe when the S/N ratio is large is expressed by the
following equation, as desc~ibed in "Communication Sys,ems
and Techniques", by M. Schwartz and W. R. Bennett & S.
Stein, McGraw-Hill Book Company, New York, 1966:
2-ae~ R ...(13)
Assuming that the fading of the D-wave and the fading of
the U-wave are independent of each other an~ the average D~U ratio is
OdB, det R is calculated as follows:
det R = ~`f sin( ~/m)-sin~ /2)}2 ... (l~)
Y = S/N ratio
Therefore, the average bit error rate Pe is:
_ l _ ... (15)
2 {y sin( ~/m) sin(~ /2)}2
Thus, from the viewpoint of the bit error rate, the optimum
value of ~ is ~, and the average bit error rate is degraded
twice at ~/2 and seven times at ~/4. With respect to the
effect on the multiphase number m, the average bit error
rate for q~ture-phase syste~ is degraded ~ice that for binary-phase sys~
and the average bit error rate for octal-phase syste~ is degraded
seven times that for bin2ry-phase system. According to the paper
"A Further Study of Anti-l~ultipath Modulation Technique DSX
- 29 -
. . .; .
¢~
i~766~38
~ Analysis of Generalized DSX Modulation and Considera-
tions for a Narrow sand Scneme" by S. Ariyavisitakul et al,
Papers of ~ h. Grou? cn C~n. Syst., IEC~ ~P~, CS35-103, 193,, th.e a~-er2ce
bit ~rror rate Pe of 9-DS~ w`nich is calculated bv th~ same
method and is a generalization of DSX discussed as th~ second
conventicn l dicit~l sicnal ~rans~ission m~thod is e.~ressed by:
1 ...(16)
Pe
3S:~ 2- {y sin a - sin( ~/2)}2
Comparison of the equations (15), (16) shows that the
former is different from the latte~ in that the parameter
which dc~inates the diversity effect is separated from the
amoun~ of phase transition ~/m(a) for delivering
information. Therefore, by selecting ~ to be of an optimum
value ~, thR digital signal transmission method of the present
invention can produce better characteristics than those of
the conventional digital signal transmission methods,
particularly DS~, i.e., the second conventional
arrangement. For e~ample, the quadrature-phase digital signal
tra~smission method with ~ = ~ of this in~ention has bit
error rate characteris.ics equal to those of ~/2-DS~ which is the
secor.d conventional digi~ signal transmission method, and
has frequency utilization efficiency about twice that of
the conventional method since the method of this invention
is capable of quadrature-phase transmission while the
conventional method is capable of only binar~y-phase t~ansmiS-
sion. Thus, the digital signal transmission method of
this invention has better bit error rate characteristics
- 30 -
.... . . ....
127~
and it is capable of multiphase transmission.
The aforesaid analysis has been made with
refer2nce to two-r~ave multi?a_~ faaing, bu. ca~ easily be-
eYtendec to multiwaves. For e.Yam~le, when three waves ar-
involved, there are three detec.ed out?uts, and the model
shown in EIG. 29 has three branches, resulting in much
be~-er characteristics. Diversity branches are also
inc-eased whe~ a pluralit~ o. waverorms are employed as t~-slot
phase t-ansition waveforms ~(t).
While the ~res~nt ir.v~ntion has ~e~n des~ribe~ above~th
ste?ped wavefor~s e.~?loyed as the ti~e-slot phase transition
wavefor~.s ~t), it is easily e:~tended to
any desired dif~erPn. waveforms.
FIG. 30 is ~ diagram illustfative of a detected output in the
presenc~ of two-wavelnultipath ~ropayation, as with FIG. 27, with
respect to any time-slot phase transition waveforms
~(t)- Like the exm2?1e of FIG. 27, the detected output is
roughly classified into three regions F, G, H, the regions
A, E corresponding to the regions F, H, respectively. In
these regions, the output is dn ineffective detected output and has a
polarity which does not necessarily coincide with the
value of data transmitted. The regions B, C, D correspond
to the region G. In this region, t~e output is an effective detected
output and has a polarity WhiCh necessarily coincides with the
,
. -- 31 --
` . . ` . . ` :.. ~?: ,
~2 ;t668~ -
value of data transmitted. Although there is no de~ini,e
divisions in the region G, different types of detecled
outputs appear as indicated by the solid line in FIG.
30(c). The wavefo-m shown by the solid line in FIG. 30(c)
is filtered by a low-pass filter to form a portion o- an
eye pa'_ern as indic~ted by the dot~ed line in FIG. 30 ( c) .
The detected out?ut in the resion G can be
expressed, with z used as a parameter, similarly to the
ecuation (10), as follows:
Resion G
. an sin( ~/m)
( 15l ex?{; II)(Z)} + 5~ e~:?{i 'r~(Z ~ 'r)~ 12)
= an sin( ~/m)
;(¦5l e~p[j { ~(z) - ~(z -.)~] + s2l2)
where T < z < T ...~17)
Thus, inso~ar as the time-slot phase transition ~Javeforms
~(t), which meet the condition:
~ (z) - ~(z - T), cons,. (T < z < T) ...(18)
the equation (17) is not constant, and it can be seen that
bit error rate characteristics in the presence of multipath
fading can be improved by the kind of diversity effect
produced by combining different detected outputs. The
conditio~ of the equation (18) indicates that the time-slot phase
transition waveforms ~(t) have a varying rate of
change of phase or a phase jump.
When the raised cosine waveform or Gaussian waveform as
shown in ~IG. 1~ is employed, the spectrum utill2ation
- 32 -
~ 27~68~3
efficiency is good stated otherwise, greater protection
against band limitations i9 obtained.
Examples of the average bit error rate
characteristics in the presence of two-wave Rayleigh
fading having a delay time difference will be described
with reference to typical examples of the digital signal
transmission method of this invention.
FIG. 31 is a graph in which the.time-slot
phase transition waveforms are of one type in the form of
a stepped pattern having one phase jump ~ ) at the
center of the time slots as shown in FIG. 1, the graph
~howing average bit error rate characteristics for the quad-
rature-phase system with respect to the S/N ratio. The
average bit error rate characteris~ics o the con~entional
digital signal transmission method, QPSK (Quadrature Phase
Shift Reying), are al~o shown in the same graph for
comparison. As shown in FIG. 31, in QPSK, errors are
produced that cannot be reduced even by increasing
the S/N ratio. According to the digital signal
transmis6ion method of this invention, no such phenomenon
occur~, and the bit error rate characteristic~ are highly
improved.
FIG. 32 similarly shows an average bit error rate
with re~pect to the delay time difference T. In the range
ofO~T/T < 0.5, the average bit error rate is improved to a
great extent. At T~T = 0 or T~T 2 0.5, no improvement is
achleved, and the characteristics are closer to those of
- 33 ,~
lZ~66~38
.
QPS~ because the region C or the resions s, D dis~?~e~r in
FIG. 17, losing the diversity effect.
The bit error rate can be improved wi~h respec~
to a la-se value o. ~/T by s~iftinc the posi~~on o ~.~e
phase 3ump ~ from the center of the time slcts. ~IG. 33
shows an average bit error rzte with res?ec- tO ~he aelav
tim_ di ~erence T when the phase jum? ~ is pos .ioned at an
in.ernallv dividea point in the ra,io 1:2. Tne bit error
rate can be im?roved thereby up to the range of about T/T <
0.7. ~ DSX discussed as the second conventional method,
it is im?ossible in principle to im?rove the bit error rate
at ~/T > 0.5.
FIGS. 34 and 35 show averaqe bit error rate
characteristics for the quadr2ture-phase system with respect to
the delay time difference T with the time-slot
phase transition waveforms being triangular and raised
cosine waveforms as shown in FIGS. 15 and 18. The bit
error rates are similarly improved up to about ~/T = 0.7.
In each of FIGS. 34 and 35, the maximum value of phase
transition in the time slots is ~, and the characteristics
are degraded below this value and are not virtually
improved above this value.
In this embodiment, as describnd bove, the bit error
rate characteristics can be improved to a large extent in
the presence of multipath fading by employing a
transmission signal having a phase transition waveform ineach of
time slots of data, the time-slot phase transitio:- wav~form having
.. ..
- 34 -
: ..
~ ~:76688
a va~ying rate of change of phase or a phase jump, the
time-slot phase transition waveform in any desired time slot
and the time-slot phase transition waveform in a time slot
which comes prescribed time slots subsequent to the desired
time slot being identical to each otherin shape irrespective
of information to be transmitted, the information to be
transmitted being present in a phase difference between same
positions of the phase transition waveforms in the time
slots that are spaced apart from each other by the pr-
escribed time slots. The spectrum utilization efficiency is
increased by multiphaes transmission as compared with DSK
described above as the second conventional method. By
selecting time-slot phase transition waveforms, bit error
rate characteristics can be improved even with respect to
multipath fading having a larger delay time difference.
A second em~odiment of the present invention will
hereinafter be described with respect to FIG. 36.
FIG. 36 shows a transmitter circuit in a digital
signal transmission method according to a second embodiment
of the present invention. Denoted in FIG.36 at 1901 is a
data~input terminal; element 3603 is a transmission signal
generator circuit; the data input terminal 1901 and the
transmission signal generator circuit 3603 are identical to
those in the first embodiment shown in FIG. 19; elements
3610 through 3612 are first through kth antennas of a k
system; elements 3607 through 3609 are level adjusters of
the k system, and elements 3604 througn 3606 are first
through (k-1) delay elements of a k-1
- 35 -
~ ~6688
system. The level adjusters 3607 through 3609 may also
have an amplification capability. As a detecting method at
the receiving end, n t~me-slot differential detec-
tion as shown in FIGS. 24 through 26 according to the
first embodiment is employed.
The digital signal transmission method according
to the aforesaid arrangement will hereinafter be described
with reference to FIGS. 27, 32 through 36, and the equation
(18).
FIGS. 32 through 35 show bit error rate
characteristics, as described above~ for the transmission
signal which is produced by the transmission signal
generator circuit 3603, propagated through two-wave
Raileigh fading paths having the delay time difference 1,
and then received and detected. Now, it is assumed that
the delay time difference T of the propagation paths, i.e.,
delay spread, is smaller than the time slot interval T. This
condition is applied to the case where the delay spread is
small as in a building, or the transmission rate is low.
When T/T is close to 0, e.g., where the time-slot
- phase transition waveforms ~(t) are of a stepped pattern,
the region C in FIG. 27 is reduced. Where ~(t) is of a
general waveform, the lefthand side of the equation t18)
varies to a smaller extent as z varies. This results in
elimination of the diversity effect which would be produced
by combining different detected outputs as described with
reference to the first embodiment. As a consequence, as
.:,,
. : ,.. .
_ ~ ~6 ~ _
~ 276~i~38
T/T approaches 0 in FIGS. 32 through 35, the bit error rate
characteristics are less improved. But, by giving, at the
transmitting end, a delay in the range o~ from 0 to 0.5 or
from 0 to 0.7, which is the improving range for ./T, the
bit error rate characteristics can be im?roved.
The delay elements 3604 throuch 3606 shown in
FIG. 36 serve to give such a delay in the transmitti~g end.
The delay elements should be designed to set, to Tm/T, the
time difference rm between a wave arriving first and a
wave arriving last at the receiving end, including a
delay due to the difference in paths from the respective
antennas, so that the delay will not exceed 0.5 or 0.7,
which is the maximum improving range for r~T that is
determined by the time-slot phase transition waveforms
~(t). The level adjusters 3607 through ~609 serve to
substantially equalize, at the receiving point, the average
levels of waves accompanied by fading and coming from the
respective antennas. The first through kth antennas are
required to be located at spaced intervals or be composed - `
of antennas having different planes of polarization such
... ~
- that the fadings of the respective paths from the antennas
- .;
to the receivi`ng point are mutually uncorrelated. One
simplest and most effective arrangement is the case in -
which k = 2. In this case, the time difference Tm between ~
waves arriving from two antennas is preferably selected to -
be 0.2 to 0.4 in terms of Tm/T, which is the best point of
an bit error rate determined by the time-slot phase transition
.... .
... . - . - . ~ . - ~ ~ _ 3 7 _ . . ; . ~
1~7~688 -
waveform ~(t).
With the second embodiment of the present
invention, as described above, one transmission signal is
transmitted with a time delay from different antannas for
thereby obtaining a diversity effect even when T/T is
small, so that the bit error rate characteristics can be
improved. The diversi~y is highly effective in reducing
the size of a receiver and making it portable since only
one antenna is required on the reciever.
.. ... ..
:, . .. ;
. .
~i~..r
. .
