Note: Descriptions are shown in the official language in which they were submitted.
20~6~44
BACKGROUND OF THE INVENTION
The present invention relates generally to a video
signal processing system for processing a wide bandwidth
video signal into a reduced bandwidth signal suitable for
transmission and/or recording via a narrow bandwidth signal
medium whereby the information content of the wide bandwidth
video signal is retained in the reduced bandwidth signal and
the reduced bandwidth signal is compatible with conventional
narrow bandwidth reception apparatus, and for receiving
and/or reproducing and processing the transmitted reduced
bandwidth signal for recovering therefrom the information
content of the original wide bandwidth signal.
The present invention relates more particularly to a
signal processing system applicable to a narrow bandwidth
format video cassette recorder (VCR) for converting a wide
bandwidth input video signal to a reduced bandwidth video
signal containing the information content of the input wide
bandwidth video signal within the reduced bandwidth whereby
the reduced bandwidth video signal may be recorded and
reproduced conventionally by such narrow bandwidth format
VCR, and for processing the reproduced narrow bandwidth video
signal to recover the information content of the wide
bandwidth video signal therefrom whereby a wide bandwidth
video signal may be reconstructed for yielding improved video
bandwidth of the reproduced signal comparable to the full
bandwidth of the input video signal, while maintaining
backward compatibility of the recorded reduced bandwidth
video signal for playing back video cassettes recorded by
this improved video signal processing system on available
conventional narrow bandwidth format VCRs.
Conventional consumer type VCRs record video information
onto video tape cassettes in one of several formats. The
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X056'744
well-known VHS format system uses a relatively narrow
bandwidth format and produces degraded picture quality in
comparison to standard broadcast video chiefly because the
recorded VHS format video signal has insufficient horizontal
resolution. An enhanced VHS format type recording system,
popularly called Super VHS or s-VHS, produces enhanced
picture quality by recording a wider bandwidth video signal
on the video tape cassette using a higher FM carrier
frequency for the luminance information, thus yielding
improved picture resolution. Such a format requires a higher
FM carrier frequency, higher quality tape in the cassette and
higher quality recording and playback mechanisms, heads and
circuitry. However, the S-VHS format is not backward
compatible with standard VHS format VCRs. That is, although
an S-VHS format VCR can reproduce (play back) cassettes
recorded on either S-VHS format or standard VHS format VCRs,
a standard VHS format VCR cannot play back cassettes recorded .
on S-VHS format VCRs.
It has long been a goal of video engineers to increase
the amount of information transmitted through a given
narrowband channel, such as an NTSC signal channel, which is
limited to a nominal 4.2 Mhz of useful bandwidth. Because
the frame and line rates (temporal and vertical resolution)
are usually fixed, restricting the bandwidth translates into
restricting the horizontal resolution. In some cases, the
nominal bandwidth of the channel is limited to 3 MHz or even
2.5 MHz, resulting in an image with insufficient horizontal
resolution.
It has long been recognized that in scanned television
systems, the signal energy is concentrated spectrally in the
spatio-temporal domain at periodic intervals according to the
scanning frequencies, and the video spectrum has so-called
'holes' , that is, gaps between these discrete signal areas in
which the signal energy is very small, such gaps also
occurring at regular intervals. The NTSC composite (i.e.
'colorplexed') color video system represents a system which
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2056'44
uses one of these 'holes' to carry the color information. In
the NTSC system, the signal, i.e., chrominance or 'chroma'
signal containing the color information is transmitted
encoded onto the baseband video as color difference or
mixture signal quadrature, i.e. two-phase, amplitude
modulation of a suppressed nominally 3.58 MHz sub-carrier
(i.e. AM sidebands of a pair of suppressed subcarriers in
phase quadrature) which carrier's frequency (3.579545 MHz) is
very carefully selected (227.5 times the horizontal scanning
frequency of 15.734 KHz) so that a minimum disturbance occurs
when a color video signal is displayed on a black and white
receiver. Specifically, the NTSC color subcarrier frequency
is interleaved horizontally, vertically, and temporally with
the luminance signal spectrum to minimize crosstalk and
intermodulation between the luminance and chrominance
components of the composite video signal.
It was recognized at around the time of the adoption of
the NTSC colorplexed system that such frequency spectrum
'holes' could also be used to transmit additional horizontal
information to increase the horizontal resolution of the
reproduced image. In such systems, the higher frequency
horizontal information is interleaved spectrally with the
lower frequency horizontal information in a similar manner as
the chrominance information is in the NTSC color system. An
article entitled "REDUCTION OF TELEVISION BANDWIDTH BY
FREQUENCY INTERLACE" by E. A. Howson et al. in J. Brit. I. R.
E., February, 1960, pp. 127-136 contains a description of
such a system which utilized analog signal processing
techniques. This described system, however, could not
accurately reproduce the full bandwidth image in its original
form because it was unable to completely remove the artifacts
resulting from the frequency interleaving, which manifested
themselves as annoying dot crawl patterns.
Sampled data digital video signal processing techniques
were later developed using sub-Nyquist sampling (sometimes
termed subsampling) to address the problem. These techniques
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2056'44
involved replacing every odd sample in a first video line
with a zero-valued sample, and then on the next line,
replacing every even sample with a zero-valued sample. On
alternate frames, the patterns are reversed.
German Patent Publication No. 82100286.2 entitled
"Verfahren zum Ubertagen von Fernsehsignalen uber einen
genormten bandbreitebegrenzten Ubertragunskanal and Anordnung
zum Durchfuhren des Verfahrens," January 1, 1982 by
Professor Wendland et al. describes principles of offset
subsampling and bandwidth compression as applied to advanced
television systems. This patent publication also describes
techniques for implementing television systems in accordance
with the described principles.
Theoretically, the Howson et al. frequency folding
technique and the sub-Nyquist sampling technique are
equivalent, when the folding carrier frequency 'ff' is one
half the sampling frequency 'f,'. But, although
theoretically equivalent, the later sampled data digital
systems provided improved reconstruction of the received
image because of the existence of line and frame combing
techniques, which had not been developed at the time of the
Howson et al. system. The sub-Nyquist sampling techniques,
however, were developed for totally sampled data digital
systems as data reduction (i.e. compression) techniques in
digital systems, and the signals generated by these sampling
techniques were not generally intended to be passed through
a narrowband analog channel.
In an article entitled "DEVELOPMENT OF HDTV RECEIVING
EQUIPMENT BASED ON BAND COMPRESSION TECHNIQUE (MUSE)", by
Kojima et al. in IEEE Transactions on Consumer Electronics,
Vol. CE-32, No. 4, Nov. 1986, pp. 759-768, another data
compression scheme is described which achieves bandwidth
compression by sampling each pixel once every other frame.
This scheme works well for non-moving images.' For moving
images, a motion vector is developed, and the actual rate of
sampling of each pixel is adaptively varied in response to
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2056'44
the motion vector so that a sample of the pixel is
transmitted every other frame on the average, but more often
when that pixel is representing a moving image.
U.S. patent 4,831,463 issued May 16, 1989 to Faroudja
describes apparatus for processing a video signal having a
predetermined bandwidth in order to pass the video
information through a limited bandwidth channel, such as
magnetic tape. In the apparatus described in this patent, a
video signal pre-processor includes a comb filter to produce
spectral holes, such as described above, between spectrally
active areas in the video signal spectrum. A folding circuit
folds the baseband video luminance signal about a
predetermined folding frequency selected so that aliases of
the baseband luminance signal are placed in the spectral
holes previously made in the video signal. A low pass filter
then filters the resulting folded video signal so that its
bandwidth is about one-half (1/2) the bandwidth of the
original video signal. The resulting signal may then be
transmitted through the limited bandwidth channel.
The Faroudja '463 patent further describes a post-
processor which receives the folded signal from the limited
bandwidth channel. The post-processor includes an unfolding
circuit which. unfolds the received signal about a
predetermined unfolding frequency. A comb filter then
processes the unfolded signal to remove the alias components
resulting from the unfolding process. The signal produced by
this comb filter closely approximates the original video
signal in terms of the bandwidth and information content.~y
It is interesting to note that the Howson et al. article
discussed two bandwidth reduction techniques for video
luminance signals by frequency interlacing or interleaving.
In the first technique discussed, the video luminance signal
spectrum is divided into two equal half-bands (i.e. band
split at frequency 'f') and the upper half-band (i.e. the
high band luminance from frequency 'f' to frequency '2f') is
used to modulate a sub-carrier which has its frequency set to
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~U56~44
be near the upper frequency limit of the normal video band
(i. e. '2f'). The lower sideband of the modulator output is
selected and combined with the original lower half-band,
whereby the resultant frequency-interlaced signal after
combination contains all of the original luminance signal
information but in one-half the bandwidth of the original
signal, thus being suitable for transmission over a reduced
bandwidth channel.
In the second technique discussed by Howson et al.,
instead of dividing the main video luminance signal into two
half-bands and modulating the '2f' sub-carrier with the high
frequency half-band only, the entire main video (i. e.
baseband) luminance signal is used to modulate the '2f' sub
carrier. The lower sideband of the modulator output signal
contains the required interleaved signal in correct frequency
relationship with the main baseband video signal. If the
modulator output is added to the main signal and the
resultant added signal is passed through a low-pass filter
having its cutoff frequency at approximately one-half the
sub-carrier frequency, the low-pass filtered output signal
consists of the correct composite reduced bandwidth signal
(with the sub-carrier suppressed). Howson et al. teach that
this second technique avoids the need for using complementary
low-pass and band-pass filters as required by the first
technique employing band-splitting, and Howson et al. adopted
this second technique in a described experimental apparatus,
although the summary abstract appearing in the Howson et al.
article somewhat misleadingly implies that the first
technique using band-splitting was employed.
The folding/unfolding system described in the Faroudja
X463 patent is similar in principle to the second technique
described and adopted by Howson et al., in that the folding
modulation sub-carrier frequency in the Howson et al. system
is selected to be an odd multiple of one-half the line scan
frequency while the frequency of the folding heterodyne
oscillator/mixer or of the sub-Nyquist sampling clock applied
- 6 -
2056'44
to the multiplier used as the folding modulator in Faroudja
'463 is selected to be an odd harmonic of one-half the line
scan rate or a harmonic of an odd multiple of the line and
frame scan rate, and in both systems the folding modulation
is performed on the baseband luminance signal and both
systems thus necessarily require low-pass filtering after
folding to remove frequencies greater than one-half the
folding frequency from the folded signal.
Both the Howson et al. article and the Faroudja '463
patent describe folding systems which, if incorporated into
an improved VCR, could not produce cassettes which could be
played back on conventional VCRs without introducing
unacceptable artifacts into the displayed image, that is,
recordings of such a folded video signal would lack 'backward
playback compatibility' to existing VCRs. This is primarily
due to the amplitude of the folded high frequency components
present within the spectrum of the low frequency components
on the previously recorded cassette. The magnitude of the
folded high frequency components is sufficiently high as to
introduce intolerable artifacts and degradation (dot crawl,
twinkling, line flicker, etc. ) into an image display produced
from a video signal from which the folded high frequency
components were not properly removed.
Reference is also made to an article by T. S. Robson
entitled "A Compatible High Fidelity Television Standard for
Satellite Broadcasting", appearing in "Tomorrow's
Television", SMPTE, (1982) at pages 218-236. This article
proposed an extended definition 'MAC' component type video
signal system including filtering high frequency diagonal
information from conventional video signals to make available
certain gaps in the signal spectrum, and using a 3.-D sampling
process to deliberately alias useful high frequency
information into these gaps, followed by truncation of the
original signal spectrum. A post filter/interpolator would
be used to restore the folded energy to its correct high
frequency location, thereby recreating the original spectrum.
2056'44
This article also describes the necessity for pre- and post-
filtering to avoid aliasing in the displayed image. On a
conventional display device without 3-D post filtering,
however, the high frequency alias product will be present,
resulting in image impairment.
It is desirable that an improved video recording system
be able to record wider bandwidth video signals on a standard
quality cassette than those recordable by conventional narrow
bandwidth VCRs, while still maintaining backward
compatibility with conventional narrow bandwidth VCRs, and
not require especially high quality magnetic tape or record
and playback mechanisms. That is, it is desirable that
standard quality narrow bandwidth media video cassettes may
be recordable with wider bandwidth, higher frequency video
information using the improved system and be able to be
compatibly played back by conventional narrow bandwidth VCRs
without producing noticeable visual artifacts in the
reproduced image, even if the conventional VCR may not be
able to reproduce the full bandwidth signal recorded on such
a cassette.
Howson et al. were not concerned with backward
compatibility of the interleaved signal, but instead
suggested including a pre-emphasis filter for boosting the
interleaved high frequency components of the folded luminance
signal in order to minimize the effects of crosstalk from the
low frequency luminance components during the transmission of
the folded signal through the channel and sub-carrier
interference at the receiver. If a video cassette recorded
by a VHS format VCR modified to include the system taught by
Howson et al. were played back on a standard VHS format VCR,
the interference of the pre-emphasized high frequency
components which would not be removed would produce an even
more objectionable image than that produced by the Faroudja
system.
The Faroudja '463 patent does not include any teaching
of compatibility with pre-existing recording media and
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~056'~44
apparatus, other than a mention that such is desirable and is
one object of that patent's invention. There is no teaching
whatsoever in the Faroudja '463 patent of any apparatus or
process for achieving backward or 'downward' compatibility
with existing playback apparatus. As described above, a
recording made by a system according to the Faroudja '463
patent's teaching is not backward compatible with existing
playback apparatus because of the high level of the folded
luminance high frequencies in the luminance low frequencies.
Thus, the need has remained for improving the video
resolution obtainable by the present limited bandwidth video
recording and playback techniques, media and mechanisms in a
manner which retains backward playback compatibility with
existing VCRs and VCPs.
It is an object of the present invention to provide a
video signal processing system by which an input wide
bandwidth full resolution video signal may be transmitted or
recorded as a limited bandwidth video signal as by broadcast
or video recording via a limited bandwidth medium, while
still retaining in the limited bandwidth signal essentially
the information content of the input wide bandwidth signal in
such a form that the limited bandwidth signal may be
compatibly received or reproduced by conventional narrow
bandwidth apparatus to produce a video image without
objectionable artifacts.
It is also an object of the present invention to provide
a video signal processing system capable of receiving or
reproducing such a transmitted or recorded limited bandwidth
video signal and reconstructing therefrom a wide bandwidth
video output signal corresponding to and containing the
information of the input wide bandwidth video signal so as to
produce a full resolution video image therefrom.
It is further an object of the present invention to
provide a video signal processing system applicable to
conventional narrow bandwidth video recording and playback
apparatus for improving the resolution of video recordings
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made and reproduced thereby without requiring higher quality
recording and playback mechanisms or recording media.
Still further objects of the present invention include
providing a video recording apparatus capable of recording an
input wide bandwidth video signal within a limited bandwidth
such that the recorded limited bandwidth signal may be
compatibly reproducible by conventional narrow bandwidth
playback apparatus, and providing a video playback apparatus
capable of reproducing such a recorded limited bandwidth
signal to recover therefrom a wide bandwidth playback signal
corresponding to the input wide bandwidth video signal.
Additional objects of the present invention include
providing video signal processing techniques for encoding an
input wide bandwidth signal into a limited bandwidth video
signal containing in encoded form essentially the information
of the input wide bandwidth video signal within a
substantially reduced bandwidth and compatibly reproducible
by conventional narrow bandwidth playback apparatus for
producing a video image without objectionable artifacts, and
for decoding such a limited bandwidth video signal to produce
therefrom a wide bandwidth video signal corresponding to the
input wide bandwidth video signal for producing a video image
having the same resolution as the input wide bandwidth video
signal.
In accordance with one aspect of the present invention,
there is provided a video signal processing system by which
it is possible to bandwidth-reduce an input wide bandwidth
video signal into a narrow bandwidth video signal, by
suitably processing the input wide bandwidth video signal to
convert it into a narrow bandwidth video signal while
retaining the video information content of the input wide
bandwidth video signal, the resultant converted narrow
bandwidth video signal being recordable and reproducible via
a narrow bandwidth recording medium.
- 10 -
2056'44
In accordance with another aspect of the present
invention there is provided a video recording system capable
of converting an input wide bandwidth video signal into a
narrow bandwidth video signal while retaining the video
information content of the input wide bandwidth video signal,
and recording the converted narrow bandwidth video signal
onto a narrow bandwidth recording medium, which recorded
narrow bandwidth video signal also retains therein the video
information content of the original input wide bandwidth
video signal, such that the thusly recorded narrow bandwidth
video signal may be compatibly reproduced by conventional
narrow bandwidth reproducing devices to provide a reproduced
compatible narrow bandwidth video image of comparable quality
to that of conventional narrow bandwidth video recordings.
In accordance with a further aspect of the present
invention, there is provided a video recording and
reproducing system capable of converting an input wide
bandwidth video signal to a narrow bandwidth video signal
retaining the video information content of the input wide
bandwidth video signal by an adaptive process controlled by
a control signal derived from the input video signal,
recording the converted narrow bandwidth video signal onto a
narrow bandwidth recording medium, which thusly recorded
narrow bandwidth video signal also retains encoded compatibly
therein the video information content of the original input
wide bandwidth video signal as well as the control signal,
whereby the recorded narrow bandwidth video signal may be
compatibly reproduced by conventional narrow bandwidth
reproducing devices to provide a compatible narrow bandwidth
video image of comparable quality to that of conventional
narrow bandwidth video recordings and without objectionable
artifacts, and which inventive system is also capable of
reproducing the recorded narrow bandwidth video signal from
the recording medium, recovering the control signal therefrom
and processing the reproduced narrow bandwidth signal in
accordance with the recovered control signal to reconstruct
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the reproduced narrow bandwidth video signal into a wide
bandwidth video signal corresponding to the original input
wide bandwidth video signal.
In accordance with the present invention, a full
y bandwidth input video signal is passed through an encoder
which generates an encoded reduced bandwidth video signal
having a limited bandwidth low frequency luminance component
with a de-emphasized amplitude high frequency luminance
component folded into it. Encoded video signals produced by
the encoder according to the present invention can be output
to a narrow bandwidth video medium or channel, e.g. be
recorded on a conventional narrow bandwidth magnetic medium
such as a VHS format video cassette.
When played back by a VCR embodied in accordance with
the present invention, the reproduced narrow bandwidth
encoded video signal is passed through a decoder wherein the
folded de-emphasized amplitude high frequency luminance
component may be recovered by unfolding and amplitude re
emphasis processing and the full-bandwidth video signal
thereby restored, while when played back by a conventional
narrow bandwidth VCR lacking any capability for unfolding and
recovery of the recorded folded high frequency luminance
component, the reproduced folded high frequency luminance
component is at a sufficiently low amplitude level, by virtue
of the advantageous de-emphasis performed thereon during the
encoding processing of the present invention, that no
objectionable interference will result in the reproduced
narrow bandwidth video image, thus providing backward
playback compatibility of the encoded recording with
conventional narrow bandwidth VCRs.
The video recording and reproducing system in accordance
with the present invention is embodied in part by an encoder
for receiving and suitably processing a wide bandwidth
baseband video input signal which includes a baseband
luminance component having low and high frequency luminance
(luma) components and a chrominance (chroma) component, to
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2056'744
produce therefrom an encoded band-limited luminance signal in
which the high frequency luminance components of the input
wide bandwidth video signal are advantageously folded into
the spectral bandwidth of the low frequency luminance
components, along with an encoded chrominance-plus-motion
signal containing the chrominance component of the input wide
bandwidth video signal together with a motion representative
signal derived from the input video signal and utilized in a
motion adaptive processing of the input video signal in the
encoder, thereby producing an encoded video signal having its
bandwidth limited or compressed to a narrow bandwidth
relative the bandwidth of the original input video signal
while still retaining the high frequency luminance detail
information of the original input video signal. The encoded
luminance and chrominance-plus-motion signals from the
encoder, which are suitable for a narrow bandwidth video
channel or medium, may thus advantageously be recorded onto
a narrow bandwidth capacity videotape as by conventional
narrow bandwidth format VCR recording apparatus.
The encoding process in accordance with the present
invention carefully controls the magnitude of the high
frequency luminance component adaptively during the folding
operation to an appropriate level in order to ensure that the
folded high frequency component in the band-limited signal
will not generate objectionable noise, interference or
artifacts in the displayed video image when the recorded
encoded band-limited signal is reproduced by a conventional
narrow bandwidth format VCR or VCP, thereby assuring backward
playback compatibility of improved recordings processed in
accordance with the present invention with conventional
narrow bandwidth format VCRs and VCPs.
In contrast to prior bandwidth folding techniques such
as that proposed by Howson et al. in which the high frequency
components of the luminance signal are boosted in amplitude
by a pre-emphasis process or that described both by Howson et
al. and by the Faroudja '463 patent, in which the baseband
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2056'744
luma signal is folded without any amplitude adjustment, the
encoding processing in accordance with the present invention
splits the baseband luminance signal into high and low
;
frequency components; as described but disfavored by Howson
et al., and then advantageously adaptively de-emphasizes the
high frequency components of the luminance signal during the
folding operation with the degree of de-emphasis being
governed by the energy measured in the high band luminance,
so that the folded high frequency luminance component in the
encoded band-limited luminance signal will not generate
objectionable interference or noise in a displayed video
image when the encoded band-limited video signal is
reproduced by a conventional narrow video bandwidth
reproduction device, for thereby providing backward
compatibility of the encoded band-limited video signal with
existing narrow video bandwidth reproduction devices.
De-emphasis of the high frequency luminance components
of the folded band-limited video signal can be performed in
various ways, for example in a fixed manner. However, the
present inventors have found it to be particularly
advantageous to apply an adaptive de-emphasis to the high
frequency luminance components whereby those high frequency
luminance components in the folded band-limited signal which
will most tend to generate objectionable noise or
interference in the displayed video image on reproduction by
a conventional narrow video bandwidth reproduction device,
for example high amplitude - high frequency luminance
components, will be de-emphasized to a greater extent, while
those high frequency luminance components in the folded band-
limited signal which have less tendency to cause noise or
interference on narrow bandwidth reproduction and which might
tend to suffer noise degradation in the displayed wide-
bandwidth video image on reproduction by a wide bandwidth
reproduction system if greatly de-emphasized, for example low
amplitude - high frequency luminance components, will be de-
emphasized to a lesser extent or passed unaltered. Such an
- 14 -
2US6'~44
adaptive de-emphasis technique has the advantage that it
provides backward compatibility to the encoded band-limited
video signal while also preserving the image quality of the
reproduced wide-bandwidth video signal. Thus, the high luma
de-emphasis during the folding operation in the encoder is
preferably performed adaptively in accordance with the
present invention.
Furthermore, in advantageous contrast to the prior
baseband folding systems described in the Howson et al.
article and the Faroudja '463 patent which require low pass
filtering of the luma signal after folding because the
folding process, performed on the baseband luminance signal,
results in unwanted upper sideband components, in the
encoding process according to the present invention the
luminance signal can be band-split prior to de-emphasis
processing and folding, the folding operation performed
directly upon the high frequency luminance component signal
by a sampling technique whereby the folded de-emphasized high
band luminance signal, after being band-shifted to the low
band frequency spectrum, is then simply combined with the low
band luminance signal to produce the band-limited folded
luminance signal, without requiring any lowpass filtering of
the folded luminance signal. Alternatively, the present
invention may implement folding in the baseband in
combination with de-emphasis of the luminance high
frequencies.
It has previously been known to apply different
processing, such as different types of filtering, to a video
signal depending upon some factor such as, for example, the
degree of motion in the video image, for instance where one
type of filtering is more appropriate for portions of the
video image signal in which there is no motion or only a
small amount of motion while another different type of
filtering is more appropriate for portions o~f the video image
signal exhibiting a greater degree of motion. A motion
signal may be generated by analyzing the video image signal
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2056'44
to determine the degree of motion occurring in the video
image, and then this motion signal may be used to control a
switch which appropriately applies the video image signal to
different filters depending upon the degree of motion in the
video image. The motion signal may for example control a
soft switch at the outputs of different filters so that
amounts of the filtered signals from the different filters
provided to an output may be controlled depending upon the
motion content of the respective portions of the video image.
In accordance with a further aspect of the present
invention, the encoder also generates a motion representative
signal derived from the input wide bandwidth video signal and
utilizes this motion representative signal for effecting a
motion-adaptive spatio-temporal filtering of the video signal
during the encoding processing. So that this generated
motion representative signal can advantageously be also
utilized in the decoding process for effecting a
corresponding motion adaptive processing of the reproduced
video signal, the motion representative signal is itself
suitably encoded into the band-limited video signal in such
a way that it may be recovered on reproduction and utilized
in reconstructing the wide bandwidth video signal from the
encoded band-limited video signal. However, at the same
time this encoding of the motion signal onto the encoded
band-limited video signal must not give rise to noise or
interference when reproducing the encoded band-limited video
signal by a conventional narrow bandwidth format reproduction
device. Therefore, in accordance with a further aspect of
the present invention, the motion signal may advantageously
be encoded onto a vacant portion of the video signal spectrum
which will not affect narrow bandwidth reproduction, for
backward compatibility with existing narrow bandwidth
reproduction devices. In a practical implementation of the
inventive system for the conventional VHS format, this is
advantageously accomplished in accordance with a further
aspect of the present invention by suitably encoding the
- 16 -
~~'~44
motion signal onto vacant portions of the VHS format color-
under signal in such a way that the encoded band-limited
video recording may be compatibly reproduced by a
conventional VHS format VCR without interference, yet permit
an improved VCR according to the invention to extract the
encoded motion signal by a decoding process on the playback
side.
The video recording and reproducing system in accordance
with the present invention is also embodied by a decoder for
processing the encoded band-limited video to unfold and
recover the high frequency luminance components encoded in
the band-limited video signal and to restore the wide
bandwidth of the original input video signal by a bandwidth-
expansion reconstructive processing which includes unfolding
of the folded signal and performing a motion-adaptive
filtering of the unfolded signal.
In accordance with another aspect of the present
invention, the decoder performs a re-emphasis processing of
the recovered unfolded high frequency luminance components to
restore them to their original amplitude and then combines
them with the recovered low frequency luminance components so
that a wide bandwidth video signal can be reconstructed
therefrom whereby a wide bandwidth video image may be
reproduced from the reconstructed wide bandwidth video
signal. Advantageously, where the de-emphasis processing
during encoding is performed adaptively, the re-emphasis
processing is also performed adaptively, preferably with a
transfer function which is the inverse of that employed
during the encoding side de-emphasis processing, in order to
realize a more faithful reconstruction of the amplitude
relationship of the high frequency luminance components in
the unfolded wide bandwidth video signal.
The decoder also provides for recovering the encoded
motion signal from the encoded band-limited video signal, so
that the recovered motion signal may be utilized in the
decoding processing. In accordance with a further aspect of
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2056744
the invention, when implemented for use with the conventional
VHS format, in the decoder the encoded motion signal,
previously combined together with the chrominance component
of the video signal into a chrominance-plus-motion signal
during encoding, is separated from the chrominance component
during the decoding process.
These and other features and advantages of the present
inventive system will be described in greater detail in the
following detailed description taken together with the
accompanying drawings.
BRIEF DESCRIPTION OF THE DRAWINGS
FIGURE 1 is a block diagram of the record section of a
video recorder implemented in accordance with the present
embodiment;
FIGURE 2 is a block diagram of an encoder which is a
part of the record section of FIG. 1;
FIGURE 3 is a block diagram showing in more detail an
adaptive luminance signal separation portion, a motion signal
generating portion and a chrominance signal separation
portion of the encoder illustrated in FIG. 2;
FIGURE 4 is a block diagram showing an implementation of
the adaptive luminance separation and motion signal
generation portions of the encoder illustrated in FIG. 3;
FIGURE 5 is a detailed block diagram of a soft switch of
an adaptive luminance filtering section of the encoder
illustrated in FIG. 2;
FIGURE 6a is a block diagram of a folding circuit which
is a part of the encoder of FIG. 2; FIGURE 6b is a more
detailed block diagram of a control signal generator of an
adaptive de-emphasis circuit which forms part of the folding
circuit; FIGURE 6c is a graph showing a gain/de-emphasis
transfer function of the control signal generator of FIG. 6b,
including a noise coring function; FIGURE 6d is a detailed
block diagram of a folding modulator in the folding circuit;
FIGURE 6e is a plot of the vertical versus horizontal
- 18 -
2J~~744
frequency characteristics of a folding modulation performed
in the folding circuit; FIGURE 6f is a plot of the vertical
versus temporal frequency characteristics of the folding
modulation; FIGURE 6g is a block diagram of an alternative
implementation of the folding circuit; FIGURE 6h is a block
diagram of a further alternative implementation of the
folding circuit; FIGURE 6i shows an input signal; FIGURE 6j
shows the input signal of FIG. 6i after folding by sub-
Nyquist sampling; FIGURE 6k shows the input signal of FIG. 6i
after folding with a fixed de-emphasis of the high frequency
component; FIGURE 61 shows the input signal of FIG. 6i after
folding with an adaptive de-emphasis of the high frequency
component in accordance with an embodiment; FIGURE 6m shows
the input signal of FIG. 6i after folding with an adaptive
de-emphasis of the high frequency component including a noise
coring operation; FIGURE 6n is a block showing a further
alternative implementation of a folding circuit in accordance
with an embodiment; and~FIGURE 6o is a more detailed block
diagram of an alternative band-splitting filter shown in FIG.
6n;
FIGURE 7a is a detailed block diagram of a
chrominance/motion signal combining circuit of FIG. 2;
FIGURE 7b is a diagram showing the relationship between the
NTSC color carrier and the motion signal prior to encoding;
FIGURE 7c is a diagram showing the relationship between a VHS
format color-under carrier and the encoded motion signal in
an even track (channel); FIGURE 7d is a diagram showing the
relationship between a VHS format color-under carrier and the
encoded motion signal in an odd track (channel); FIGURE 7e is
a diagram illustrating phase opposition cancellation of the
reproduced encoded motion signal in a conventional VCR; and
FIGURE 7f is a block diagram of a conventional chroma comb
filter;
FIGURE 8 is a block diagram of a playback section of a
video recorder implemented in accordance with the present
embodiment;
- 19 -
~u5~744
FIGURE 9 is a more detailed block diagram of a decoder
in the playback section illustrated in FIG. 8;
FIGURE l0a is a block diagram of an unfolding circuit of
the decoder illustrated in FIG. 9; FIGURE lOb is a detailed
block diagram of an adaptive re-emphasis circuit forming a
part of the unfolding circuit illustrated in FIG. 10a; FIGURE
lOc shows a re-emphasis/gain transfer function of a control
signal generator of the adaptive re-emphasis circuit,
including a coring function; FIGURE lOd shows a folded
signal; FIGURE l0e shows an unfolded signal; FIGURE lOf shows
the unfolded signal of FIG. lOf with spatial filtering; and
FIGURE lOg is a block diagram of a soft switch of a spatio-
temporal post-filter circuit of the unfolding circuit in FIG.
10a;
FIGURE lla is a detailed block diagram of a
chrominance/motion signal separation circuit of the decoder
of Fig. 9; FIGURE llb shows the coefficients of a digitally
implemented quadrant selective filter forming part of the
chrominance/motion signal separation circuit; FIGURE lic is
a three-dimensional plot of the spatial frequency response of
the quadrant selective filter; FIGURE iid is a two
dimensional plot taken across the horizontal midplane of the
three-dimensional plot in FIG. iic; and FIGURE lle is a
selectivity plot of the horizontal frequency response of the
quadrant selective filter;
DETAILED DESCRIPTION
The system of the present invention may be implemented
using analog and/or digital signal processing techniques.
For sake of example, an implementation of the system
will be described below using i
digital signal processing. However, given the description
herein, one of ordinary skill in the art will understand that
the invention may be practiced using analog techniques and
how such may be implemented.
In the Figures, equalizing delays have been omitted for
- 20 -
2056"44
the purposes of simplicity. One skilled in the art of video
signal processor design will appreciate the need for such
delays to properly time-align pixels subject to different
delays on different processing paths due to the differing
processing performed in those paths. One skilled in the art
would understand where such delays would be necessary and how
long each of the delays would have to be, and such delays
will not be described or discussed below.
In addition, in the Figures, various filters are used
for filtering in the horizontal, vertical, and temporal
directions, having both high pass and low pass response
characteristics. One skilled in the art of video signal
processor design will appreciate that some of such filters
may be constructed as known comb filter designs, and would
understand how to properly select the delay periods of the
respective delay lines, the number of taps and the weighting
of the taps. Consequently, the detailed design of such comb
filters will not be discussed below, unless such a design
detail is important for other reasons. Further, where A/D
and D/A converters are shown or described in the present
disclosure, one skilled in the art would understand the
desirability of preceding or following such converters with
anti-aliasing or sampling clock rejection lowpass filters,
respectively, and how this could be implemented, and such
will not be further described in detail below.
Also, in the Figures and in the following detailed
description, various embodiments constructed in accordance
with the present invention are directed to an NTSC composite
video baseband signal. One skilled in the art would
understand how to modify the embodiments in order to process
a PAL video signal, a SECAM video signal or a video signal
according to any other standard. Such embodiments could
still be constructed in accordance with the teaching of the
present invention.
Figure 1 is a block diagram of a portion of a record
section of a video signal recorder according to the present
- 21 -
~a~~~~44
embodiment. In Figure 1, an input terminal 5 is coupled to a
source (not shown) of a video signal, for example, an NTSC
composite video signal. Input terminal 5 is coupled to an
input terminal of an encoder 10. A first output terminal of
encoder 10 is coupled to an input terminal of a luminance
(LUMA) record circuit 20, similar to those found in
conventional VCRs. An output terminal of the luminance
record circuit 20 is coupled to a record head 40 in a
standard tape transport mechanism, similar to those found in
conventional VCRs. A second output terminal of encoder 10 is
coupled to an input terminal of a chrominance (CHROMA) record
circuit 30, similar to those found in conventional VCRs. An
output terminal of the chrominance record circuit 30 is also
coupled to the record head 40.~ The record head records the
signals supplied to it by the luma and chroma record circuits
20, 30 onto magnetic tape (not shown) in a standard video
cassette.
In operation, the encoder 10 takes a standard full
bandwidth composite NTSC video signal and generates an
encoded luminance signal L,, which has the same reduced
bandwidth as a standard luminance signal produced by a
conventional narrow bandwidth VCR, but with a de-emphasized
high frequency component folded into its spectrum. The
encoded luminance signal L, thus contains all the luminance
information from the full-bandwidth NTSC input signal, within
the reduced bandwidth which can be recorded on the magnetic
media of the videotape cassette, thus allowing standard
quality cassettes and record and playback mechanisms to be
used. In addition, the reduced amplitude of the folded high
frequency luminance component in the encoded recording will
not cause objectionable artifacts in a displayed image if the
recorded cassette is subsequently played back on a standard
narrow bandwidth VCR. The luminance record circuit 20
records the encoded luminance signal L~ in exactly the same
manner as a limited bandwidth (i.e. narrow band) luminance
- 22 -
2oss~44
signal is conventionally recorded in a standard VCR. In a
conventional VHS format VCR recording circuit, for example,
the separated NTSC luminance signal is frequency modulated
onto a luminance carrier which can vary in frequency between
approximately 3.4 - 4.4 MHz (~0.1 MHz) and, after filtering
to remove sidebands below 1.2 MHz, occupies a band of
frequencies around 1.2 - 7 MHz.
The encoder 10 also produces a composite chrominance
plus-motion signal C+Mr to be supplied to the chrominance
record circuit 30. This composite signal includes the
standard chrominance information signal of the input NTSC
composite video signal as one component, and a motion
representative signal as another component.
As is described more fully below, in the record side
processing performed by the encoder 10 of Figure 1, a motion
representative signal M is developed by analysis of the image
motion in the input video signal and is advantageously
utilized as a control signal for motion-adaptive processing
of the video luminance components of the input video signal
prior to folding. On reproduction, use of the same motion
representative signal M utilized during the record side
luminance processing can significantly facilitate luminance
processing during reconstruction, so the motion
representative signal M is additionally processed in encoder
10 to advantageously combine it with the video chrominance
component signal C to provide the composite chrominance-plus
motion signal C+Mr to the chroma record circuit 30 for
recording. Details of the encoding and decoding of the
motion representative signal M will be more fully described
later.
The chrominance record circuit 30 records the
chrominance-plus-motion signal C+Mr in exactly the same manner
as the conventional chrominance signal is recorded in a
standard VCR. In a VHS format VCR for example, the 3.58 MHz
NTSC chrominance sub-carrier frequency is heterodyned down
- 23 -
X056744
(i.e. down-converted) to about 629 KHz to provide a color-
under carrier. In accordance with an implementation of the
encoder of the present embodiment for the :standard VHS VCR
format which will be more fully described later, the
composite chrominance-plus-motion signal C+Mr is modulated on
a color-under carrier and supplied to the record head 40
together with the luminance signal L, to be conventionally
recorded by the record head 40 onto the video tape in the
cassette. It will be understood by those skilled in the art
l0 in connection with the chrominance encoding and decoding in
accordance with one embodiment that the NTSC chroma carrier is
a two-line sequence, phase alternating by 180° every other
line, whereas the color-under chroma carrier is a four-line
sequence advancing or retarding in phase by 90° per line,
with the phase advancing or retarding on alternate tracks.
Figure 2 is a more detailed block diagram of the encoder
10 illustrated in Figure 1. In Figure 2, an input terminal
105 corresponds to input terminal 5 in Figure 1. For
facilitating the signal processing, digital signal processing
techniques may advantageously be implemented, and so input
terminal 105 is coupled to an input terminal of an analog-to-
digital converter (A/D) 102 which produces a digitized
(quantized) composite video output signal V. An output
terminal of A/D 102 is coupled to respective input terminals
of an adaptive luminance signal separator 104, a motion
signal generator 106 and a chrominance signal separator 114.
An output terminal of the adaptive luminance signal separator
104 is coupled to an input terminal of a folding circuit 108.
As will be more fully described below, folding circuit 108
performs band-splitting of the separated luminance signal
from luminance separator 104 into low and high frequency
luminance components, performs an adaptive de-emphasis
processing of the high frequency luminance component,
modulates a folding carrier with the thusly de-emphasized
high frequency luminance component to fold the adaptively de-
- 24 -
2056'44
emphasized high frequency luminance component into the low
frequency luminance component spectrum, and adds the folded
adaptively de-emphasized high frequency luminance component
into the low frequency luminance component to thereby provide
a bandwidth-limited folded luminance signal Lf.
An output terminal of the folding circuit 108 is coupled
to an input terminal of a digital-to-analog converter (D/A)
110 by which the digital folded luminance signal Lf is
converted to an analog signal L~ suitable for conventional
analog recording. An output terminal of D/A 110 is coupled
to a first output terminal 115 of encoder 10 to which is
supplied the analog luminance signal L~ to be recorded.
Output terminal 115 is coupled to the input terminal of the
luminance record circuit 20 of Figure 1.
A motion representative signal output terminal of the
motion signal generator 106 is coupled to a control input
terminal of the adaptive luminance signal separator 104 and
a motion signal input terminal of chrominance/motion signal
combining circuit 116. A separated chrominance signal output
terminal of the chrominance signal separator 114 is coupled
to a chrominance signal input terminal of a
chrominance/motion signal combining circuit 116. The
composite digital chrominance-plus-motion signal C+M output
by chrominance/motion signal combining circuit 116 is coupled
to an input terminal of a second digital-to-analog converter
(D/A) 118 which outputs the analog chrominance-plus-motion
signal C+M~. An output terminal of D/A 118 is coupled to a
second output terminal 125 of the encoder 10. Output
terminal 125 couples the analog chrominance-plus-motion
signal C+Mr to the input terminal of the chrominance record
circuit 30 of Figure 1 for recording onto the magnetic tape
in a videocassette.
In operation, the encoder 10 of Figure 2 first converts
the composite video signal input at input terminal 105 to a
sampled data multi-bit digital composite video signal V using
- 25 -
.~ ,-- ~- ~ n
r
l~._i~~Lg~
A/D 102. For an NTSC signal having a nominal bandwidth of
approximately DC - 4.2 MHz for example, the sampling
frequency may be selected to be about 10 MHz. Digital
composite video signal V is supplied to the adaptive
luminance separator 104, which extracts the luminance
component L therefrom and performs a motion-adaptive spatio-
temporai filtering of the separated luminance signal; to the
motion signal generator 106, which derives a motion
representative signal M (hereafter simply referred to as
'motion signal M') therefrom for controlling the motion-
adaptive filtering on the encoder side and also on the
decoder side; and to a chrominance signal separator 114,
which extracts the chrominance component C therefrom.
The extracted luminance signal L is further processed by
the folding circuit 108. This circuit folds the adaptively
de-emphasized high frequency component of the luminance
signal L back into the bandwidth of the lower frequency
luminance component so that all the information in the full
bandwidth baseband luminance signal L is contained in a
folded luminance signal Lf having a reduced bandwidth of about
e.g. 2.5 MHz. The adaptive folding circuit 108 will be
described in more detail below. The folded luminance signal
Lf is converted to an analog signal L, in D/A 110. This
signal is in a form which can be conventionally recorded on
a video cassette by luminance recording circuitry 20 and
record head 40 in Figure 1.
The extracted motion signal M and the extracted
chrominance component signal C are combined into a single
composite chrominance-plus-motion signal C+M in the
chrominance/motion signal combining circuit 116. A
chrominance/auxiliary signal combining circuit, which may be
used as the chrominan~e/motion signal combining circuit 116,
is described in more detail in commonly assigned Canadian
patent application serial No. 2,053,488-5 filed
October 15, 1991. Moreover, for a particularly advantageous
implementation of the present invention in the case of
- 26 -
2056'44
adapting it for compatibility with conventional VHS format
video cassette recording and reproduction, use may be made of
the chrominance component processing, recording and
reproduction conventions according to the standard VHS format
by compatibly encoding the motion signal into the VHS
format's recorded chrominance component signal, as will be
more fully described later.
The chrominance-plus-motion signal C+M is converted into
an analog signal C+R,. by D/A 118. This signal is in a form
which can be recorded on a video cassette by standard
chrominance recording circuitry 30 and record head 40 in
Figure 1.
As is known in the video signal processing art, frame
comb low pass filtering (temporal low pass filtering) may be
used to extract the luminance component from a composite
video signal with no loss of spatial resolution. However, in
the presence of motion in the video image, undesirable
artifacts are introduced into the frame comb extracted
luminance signal. Line comb low pass filtering (vertical
comb low pass filtering or spatial low pass filtering) may
also be used to extract the luminance component, even in the
presence of motion. However, the luminance component
extracted by line combing has decreased spatial (diagonal)
resolution. It is preferable to extract the luminance signal
using frame comb filtering in order to preserve spatial
resolution, unless there is motion in an area of the image,
in which case it is preferable to use line comb filtering in
that area.
Figure 3 is a more detailed block diagram of a portion
of encoder 10 illustrated in Figure 2. In Figure 3, an input
terminal 205 is coupled to the output terminal of the A/D
converter 102 of Figure 2. Input terminal 205 is coupled to
respective input terminals of a vertical high pass filter
(VHPF) 202, a temporal high pass filter (THPF) 204, a
horizontal band pass filter (HBPF) 206 and to respective
minuend input terminals of subtractors 208 and 210. An
- 27 -
2~'~44
output terminal of the VHPF 202 is coupled to an input
terminal of a horizontal high pass filter (HHPF) 212 which
may, for example, have a cutoff frequency selected at 1.7
MHz. An output terminal of horizontal HPF 212 is coupled to
a subtrahend input terminal of subtractor 208. An output
terminal of subtractor 208 is coupled to an input terminal of
a horizontal low pass filter (HLPF) 209. An output terminal
of HLPF 209 is coupled to a first data input terminal of a
soft switch 214. An output terminal of soft switch 214 is
coupled to an output terminal 215. Output terminal 215 is
coupled to the input terminal of the folding circuit 108 of
Figure 2.
An output terminal of THPF 204 is coupled to an input
terminal of a horizontal high pass filter (HHPF) 216 and to
a minuend input terminal of a subtractor 218. An output
terminal of HHPF 216 is coupled to respective subtrahend
input terminals of subtractors 210 and 218. An output
terminal of subtractor 210 is coupled to a second data input
terminal of soft switch 214.
An output terminal of subtractor 218 is coupled to an
input of a signal magnitude detector (rectifier) 220. An
output terminal of magnitude detector 220 is coupled to an
input terminal of a signal spreader 222. An output terminal
of signal spreader 222 is coupled to an output terminal 225
and to a control input terminal of soft switch 214. Output
terminal 225 is coupled to the motion signal input terminal
of chrominance/motion signal combining circuit 116 of Figure
2.
An output terminal of HBPF 206 is coupled to an input
terminal of an anti-crosstalk processor 224 for processing
the chrominance component prior to combining with the motion
signal. An output terminal of anti-crosstalk processor 224
is coupled to an input terminal of a chrominance signal
modulator 226 forming part of the chrominance/motion signal
combining circuit 116 of Fig. 2 as will be described in
detail later.
- 28 -
2056' 44
In operation, the horizontally and vertically high pass
filtered signal HV,,p produced by the serially coupled VHPF 202
and HHPF 212 contains all the chrominance information present
in the composite video signal V in addition to all the
spatial detail information. This signal HVhp is subtracted
from the composite video signal V by differencing in
subtractor 208 to produce a diagonally lowpass-filtered
signal HV,P containing only the luminance information. The
output signal HV,p from subtractor 208 is applied to HLPF 209
which may e.g. have a cutoff frequency selected at 3.3 MHz,
for removing the horizontal frequency spectrum components
above 3.3 MHz from HV,P in order to avoid aliasing noise in
the spatially reconstructed luminance signal during playback
processing (as will be described later in reference to Fig.
lOf), thereby providing at the output of HLPF 209 a spatially
derived luminance signal Ls. The spatially derived luminance
signal LS produced by HLPF 209 therefore contains only
luminance information, but has reduced diagonal resolution.
Temporally and horizontally high pass filtered signal
HT~p, produced by the serially coupled THPF 204 and HHPF 216,
also contains all the chrominance information present in the
composite video signal V, in addition to most of the temporal
detail information. This signal HThp is subtracted from the
composite video signal V by differencing in subtractor 210 to
produce a temporally derived luminance signal LT. The
temporally derived luminance signal LT produced by subtractor
210 therefore contains only luminance information at full
spatial resolution, but has reduced temporal resolution.
The temporally high pass filtered signal T,,P from THPF
204 contains motion information at horizontal low frequencies
and chrominance information at high horizontal frequencies.
Thus, the output signal HT,,P from HHPF 216 is subtracted by
differencing in subtractor 218 from the temporally high pass
filtered signal T,,P to derive a horizontally lowpass-filtered,
temporally highpass-filtered signal H,pT,,p which is a bipolar
- 29 -
2056'44
motion-representative signal. The signal HiPThP varies in
magnitude as a function of both the magnitude of the motion
in the image (that is, the greater the degree of motion in
the image, the greater the signal magnitude) and the contrast
between the moving and still portions of the image. This
signal H,pTeP has greatest magnitude at the edges of an object
having high contrast with respect to the background against
which it is moving. Where the background and the moving
object are close in intensity, the motion-representative
signal H,pT,,P has a low magnitude. In addition, quickly moving
objects with soft edges also produce a low magnitude motion
signal. Finally, even with quickly moving, high contrast
objects, the motion-representative signal H,pT,,p is usually
only strong within several pixels of the moving object's
edge.
In order to minimize the effect of these variations in
the derived motion-representative signal H,pThp, magnitude
detector 220 detects the magnitude of the motion-
representative signal H,pT,,p from the subtractor 218 and
produces a single bit signal indicating either the presence
or absence of motion for that pixel. A known magnitude
detector 220 may include a multiplexes having a control input
terminal responsive to a sign bit of the applied motion-
representative signal H,PT,,P. The motion-representative signal
H,pTbp would be coupled to a f first input terminal of the
multiplexes and an input terminal of an arithmetic negator
circuit. An output terminal of the arithmetic negator
circuit would be coupled to a second input terminal of the
multiplexes. The output terminal of the multiplexes produces
the magnitude (absolute value) of the motion- representative
signal H,pTeP. If the sign bit is a logic ' 0' , indicating, for
example, that the motion-representative signal value is
positive, then the multiplexes couples the first input
terminal, carrying the motion-representative signal H,PT,,P to
the output terminal. If the sign bit is a logic '1',
- 30 -
~~~~ 14~
indicating that the motion-representative signal value is
negative, then the multiplexer couples the second input
terminal, carrying the arithmetic negative of the motion-
representative signal H,pT,,P (which would be a positive valued
signal) from the negator to the output terminal.
This magnitude signal is then supplied to a known
comparator circuit. The comparator circuit compares the
magnitude signal to a predetermined threshold value. If the
magnitude signal exceeds the threshold value, then the
comparator circuit produces an output signal which is a logic
'1' signal. If the magnitude signal is less than the
threshold value, then the comparator circuit produces an
output signal which is a logic ' 0' signal . The output of
this comparator is a single bit motion-representative signal
which is a logic '1' in the presence of motion, and a logic
'0' otherwise.
This single bit motion-representative signal is spread
vertically and horizontally by signal spreader 222 to
generate the spread motion signal M. Optionally, the signal
may be spread temporally, vertically and horizontally by
signal spreader 222. Apparatus for spreading such a single
bit motion representative-signal is described in detail in
co-pending commonly assigned U.S. Patent No. 5,083,203 dated
January 21, 1992. The spread motion signal M produced by
signal spreader 222 is a multi-bit digital signal
whose value gradually decreases from a maximum value in
moving areas (as indicated by the single-bit bi-level signal
having a logic '1' value) to a minimum (zero) value in the
still region area around the moving area in the vertical and
horizontal directions (and optionally, temporally). This
spread motion signal M is used in the encoder 10 for
adaptively processing the video signal V as described below.
The motion signal M is also compatibly encoded so as to be
recordable and reproducible, to be recovered and utilized by
a decoder as will be described in detail later.
As described above, in the absence of image motion, the
- 31 -
2i'I~6'74~
luminance signal L is preferably the temporally derived
luminance signal L.~., but in the presence of image motion, the
luminance signal L is preferably the spatially derived
luminance signal LS. Soft switch 214 will continuously vary
the proportion of the two input signals LT and LS which can
be coupled to the luminance signal L output terminal 215 in
response to the value of the motion signal M. If the value
of the motion signal M is zero, or nearly zero, indicating no
motion or a low level of motion, then the soft switch
produces an output signal L which is composed completely of
the temporally derived luminance input signal L.L. If the
value of the motion signal M is at a maximum, or nearly
maximum, indicating a high level of motion, then the soft
switch 214 produces an output signal L which is composed
completely of the spatially derived luminance signal LS. At
intermediate values of the motion signal M, the output signal
contains some proportion of each of the input signals L.I. and
LS. The operation of soft switch 214 will be described in
more detail below.
The NTSC chrominance component is extracted from the
composite video signal V in a known manner using the
horizontal BPF 206. The separated chrominance component
signal (modulated on the 3.58 MHz NTSC chroma subcarrier)
from horizontal BPF 206 is processed to avoid crosstalk by
anti-crosstalk processor 224, and then supplied as
chrominance signal C to the input of the chroma modulator 226
of chrominance/motion signal combining circuit 116, to be
frequency down-converted, e.g. to a 629 KHz color-under
signal for VHS format recording by chrominance signal
modulator 226 in known manner, e.g. by heterodyning the 3.58
MHz NTSC chroma sub-carrier against a 4.21 MHz four-phase
carrier and passing the lower resultant sidebands only to
provide the color-under chrominance component signal
amplitude modulated on a 629 KHz carrier. The down-converted
chrominance component signal from modulator 226 will thus
- 32 -
2~~6144
have been first processed (to reduce adjacent line crosstalk
with the motion signal M in the composite chrominance-plus-
motion signal C+M) by anti-crosstalk element 224. Anti-
crosstalk element 224 may be, for example, a vertical high
pass filter (VHPF), which may be implemented as a three-tap
line comb low pass filter. Optionally, a vertical filtering
of the composite video signal V may precede the horizontal
bandpass filtering by horizontal BPF 206 at the chroma
separation stage. It will be noted that in Figure 3, VHPF
202 and THPF 204 are both responsive to the composite video
signal V. Because they are implemented as comb filters, they
can share delay lines.
Figure 3 illustrates a portion of encoder 10 which is
primarily applicable for processing an NTSC video signal.
One skilled in the art would understand how to construct
encoder 10 in accordance with the present embodiment for
processing a PAL video signal, a SECAM video signal or a
video signal according to any other standard. Figure 4 is a
more detailed block diagram illustrating the luminance
component separation and spatial and temporal filtering
sections and the motion representative signal generation
section of Figure 3 in more efficiently constructed form,
sharing delay lines, whenever possible.
In Figure 4 , elements which are the same as those in
Figure 3 have the same reference number designation and are
not described in further detail below. In Figure 4, an input
terminal 305 is coupled to the output terminal of A/D
converter 102 in Figure 2. Input terminal 305 is coupled to
an input of VHPF 202, to the minuend input terminal of
subtractor 208, to the minuend input terminal of subtractor
210, to a minuend input terminal of a weighted subtractor 306
whose input is weighted by 1/2, and to an input terminal of
a 1F delay device 312. Delay 312 produces a 1F-delayed
signal at its output terminal which is the signal(V) applied
at its input terminal delayed by a period of time equal to
one frame scan period (iF). An output terminal of 1F delay
- 33 -
2056'44
device 312 is coupled to a subtrahend input terminal of
weighted subtractor 306, whose input is weighted by 1/2. The
combination of 1F delay device 312 and weighted subtractor
318 forms THPF 204 configured as a frame high pass comb
filter of known design producing the temporally high pass
filtered output signal ThP.
An output terminal of VHPF 202 is coupled to the input
terminal of HHPF 212, and the derivation of spatially derived
luminance signal LS is as described previously in connection
with Fig. 3.
An output terminal of subtractor 218 is coupled to
serially coupled magnitude detector 220, horizontal spreader
318 and vertical spreader 320. The combination of horizontal
spreader 318 and vertical spreader 320 forms motion signal
spreader 222 of Figure 3 and operates as described above.
The remainder of the block diagram of Figure 4 is the
same as illustrated in a portion of Figure 3 and described
above. It will be understood that Figure 4 does not purport
to show timing accuracy, or timing matching. That is, in
Fig. 4 there are not shown any delay lines which would be
utilized for equalizing the delays along the respective
signal paths for maintaining pixel correlation. A person of
ordinary skill in the signal processing art would understand
the need for providing correction for timing mismatching, and
would also have knowledge of various ways in which such
correction could be implemented, and it is therefore not
necessary to describe such in detail here.
The horizontal HPFs 212 and 216 may be standard digital
high pass filters each having a break frequency at around 2
MHz. A 15-tap horizontal high pass filter is preferred,
yielding a response characteristic which is -6dB at 1. 75 MHz .
In regard to the spatial, e.g. diagonal pre-filtering
performed in the encoder prior to folding and also the
diagonal post-filtering after unfolding as will be described
later, the diagonal filters in the encoder and decoder are
well matched, and in the diagonal filtering process the input
- 34 -
2US6'~44
signal is vertically highpass-filtered, then the vertically
high-passed part of the signal is horizontally highpass-
filtered at around 1.7 MHz, and the resultant signal is then
subtracted from the input signal to provide a diagonally
lowpass filtered output signal which is in turn then
horizontally lowpass-filtered at around 3.3 MHz to produce
the spatially derived luminance signal (see e.g. Fig. lOf).
Figure 5 is a more detailed block diagram of the soft
switch 214 illustrated in Figure 3. Soft switch 214 is
utilized for motion-adaptive processing of the temporally and
spatially derived luminance signals LT and LS. In Figure 5,
a first signal input terminal 405 of soft switch 214 is
coupled to the output terminal of subtractor 210 of Figure 3
for receiving the temporally derived luminance signal LT.
therefrom. Input terminal 405 is coupled to a first input
terminal of a multiplier 404. An output terminal of
multiplier 404 is coupled to a first input terminal of an
adder 412. An output terminal of adder 412 is coupled to an
output terminal 435. Output terminal 435 is coupled to the
folding circuit 108 of Figure 2.
A second signal input terminal 415 of soft switch 214 is
coupled to an output terminal of subtractor 208 of Figure 3
for receiving the spatially derived luminance signal LS
therefrom. Input terminal 415 is coupled to a first input
terminal of a multiplier 408. An output terminal of
multiplier 408 is coupled to a second input terminal of adder
412. A control input terminal 425 of soft switch 214 is
coupled to the spread motion signal (M) output terminal of
signal spreader 222 of Figure 3. Input terminal 425 is
coupled to an input terminal of a look-up table 410. A first
scaling factor output terminal K of look-up table 410 is
coupled to a second input terminal of multiplier 404, and a
second scaling factor output terminal 1-K of look-up table
410 is coupled to a second input terminal of multiplier 408.
In operation, multiplier 404 scales the temporally
- 35 -
2056'44
derived luminance signal LT by a scaling factor K, and
multiplier 408 scales the spatially derived luminance signal
Lg by a scaling factor 1-K. Adder 412 sums the two scaled
signals output by the multipliers 404 and 408 to produce the
motion-adaptively spatio-temporally filtered separated
luminance signal L.
The spread motion signal M from input terminal 425 is
applied to the input of look-up table 410. Look-up table
produces two scaling factors, K and 1-K which are related to
the value of the control signal M. The first scaling factor
K is the proportion of the temporally derived luminance
signal LT which should be in the luminance output signal L.
The second scaling factor 1-K is the proportion of the
spatially derived luminance signal LS which should be in the
motion-adapted luminance output signal L. The sum of K and
1-K is one. The motion adaptive spatio-temporal luminance
processing function K(M) is selected such that when M is
equal to zero or nearly zero (corresponding to a low level of
motion in the luminance component), K is equal to one (all
temporally derived luminance) and 1-K is equal to zero (no
spatially derived luminance), and when M is at maximum or
nearly maximum (corresponding to a high level of motion in
the luminance component), K is equal to zero (no temporally
derived luminance) and 1-K is equal to one (all spatially
derived luminance). The function K(M) is continuous and may
be linear or non-linear. As the value of the motion signal
M gradually changes from zero to maximum, the proportion of
the temporally derived luminance signal LT in the luminance
output signal L gradually decreases and the proportion of the
spatially derived luminance signal LS in the luminance signal
output L gradually increases, and vice versa.
Look-up table 410 may be implemented in known manner as
a multi-bit read-only memory (ROM) having spread motion
signal M input terminal 425 coupled to its address input
terminals. A first subset of its data output terminals are
- 36 -
~0~6744
coupled to the K signal input terminal of multiplier 404, and
a second subset are coupled to the 1-K signal input terminal
of multiplier 408.
In operation, the storage locations of the ROM of LUT
410 are accessed by the motion signal M at the address input
terminals, where each separate value which the motion signal
M can assume accesses (addresses) a different storage
location of the ROM. Each storage location has stored
thereat a first data portion (which is coupled to the first
subset of data output terminals) pre-programmed with the K
value corresponding to the M value which accesses that
location, and a second data portion (which is coupled to the
second subset of data output terminals) pre-programmed with
the 1-K value corresponding to that value of the motion
signal M.
Next, the de-emphasis and folding processing in
accordance with an embodiment will be described. As
described above, in prior luminance signal folding systems,
the luminance high frequencies were folded back as aliases
into the luminance low frequency spectral bandwidth at their
original amplitude (or, if pre-emphasized in accordance with
the proposal of Howson et al., boosted to a higher
amplitude). If such folded luminance signals are recorded
and then played back on a conventional narrow bandwidth VCR
which has no provision for removing the folded high luminance
frequency aliases, highly objectionable artifacts will be
present in reproduced images, rendering such a video
recording backwardly incompatible for conventional playback
apparatus reproduction.
The present inventors have found that by appropriately
de-emphasizing the folded high frequency luminance components
in the band-limiting folding operation during the encoder
side processing, it is advantageously made possible to reduce
the artifacts in the video image displayed on playing back
the band-limited folded luminance signal on the videotape
with a conventional narrow bandwidth reproduction device to
- 37 -
~tuG~=~-
a level which is not objectionable to the viewer, thereby
providing desirable backward compatibility with existing
playback apparatus.
In. U.S. Patent No. 5,113,262 dated May 12, 1992
and Canadian patent application No. 2,054,261-6 filed
October 26, 1991, there have been earlier proposed
luminance folding/unfolding systems utilizing a de-peaking or
thresholding type of de-emphasis of the high frequency
luminance components on the record side accompanied by a re-
peaking type of re-emphasis for restoration of the de-peaked
high frequency luminance components on the playback side,
utilizing a coring operation and also utilizing a scaling
operation controlled by a scaling control signal which, in
some instances, is recorded and then recovered during
playback processing.
In the instant application there is presented, in
accordance with the present embodiment, ark advance over these
earlier proposed folding and unfolding systems, for providing
improved video image display performance when reproducing the
folded signal and recovering the unfolded signal on a
playback system operating in accordance with the present
invention, and also for providing at the same time improved
backward compatibility when reproducing the recorded folded
signal with a conventional narrow bandwidth playback device.
Figure 6a is a more detailed block diagram of the
folding circuit 108 illustrated in Figure 2. Folding circuit
108 performs several operations on the motion-adaptively
spatio-temporally processed luminance signal in order to
limit the luminance bandwidth for narrow band recording, in
3a particular performing an advantageous de-emphasizing,
preferably performed adaptively, of the amplitude of the high
frequency luminance components which axe folded into the
spectrum of the low frequency luminance components to produce
the band-limited luminance signal for recording. As will be
more fully described below, folding circuit 108 performs band
splitting of the luminance signal into low frequency
- 38 -
2056'44
components and high frequency components, appropriately de-
emphasizes the high frequency luminance components, and folds
the high frequency luminance components into the spectral
bandwidth of the low frequency luminance components to
thereby compress the full bandwidth luminance information of
the input video signal into a narrow frequency band
corresponding to the bandwidth of the narrow band video
medium.
In the folding circuit 108 shown in Figure 6a, an input
terminal 505 is coupled to the output terminal of the
adaptive luminance separator 104 of Figure 2 , that is, to the
spatio-temporally processed luminance signal L output
terminal 215 of soft switch 214 in Figure 3. Input terminal
505 receives the baseband motion-adaptively spatio-temporally
processed luminance signal L output from soft switch 214 and
couples it to an input terminal of a horizontal low pass
filter (LPF) 502 as well as to the minuend input of a first
subtractor 504. Horizontal LPF 502 may be selected to have
its -6dB point at approximately 2.5 MHz to correspond to one-
half the folding frequency. The low pass filtered luminance
signal LL output of horizontal LPF 502 will therefore contain
only low frequency luminance components below 2.5 MHz, and
this output signal LL is coupled to a f first input terminal of
an adder 506 and also to the subtrahend input terminal of
subtractor 504. Horizontal LPF 502 and subtractor 504 thus
are configured to form a band splitting filter at 2.5 MHz,
with LPF 502 outputting the low frequency luminance component
(low band luma) signal LL below 2.5 MHz and with the signal
LH at the output terminal of subtractor 504 being comprised
of the high frequency luminance component (high band luma)
above 2.5 MHz.
The output of LPF 502 passes the low band luma signal LL
to first adder 506. The output of subtractor 504 passes the
high luma signal LH to a signal input terminal of a first
multiplier 508 which performs a de-emphasis operation
- 39 -
2056'744
thereon, and also to an input terminal of a control signal
generator 510 which controls the operation of the de-emphasis
multiplier 508. The control signal generator 510 generates
a de-emphasis gain control signal (where the de-emphasis
amount D corresponds inversely to the amount of gain through
multiplier 508, i.e. D=1/Gain and Gain=1/D) which is coupled
to a gain data input terminal of de-emphasis multiplier 508.
Control signal generator 510 and de-emphasis multiplier 508
form a de-emphasis section for de-emphasizing, that is,
attenuating, the amplitude of the high frequency luminance
component signal LH. De-emphasis processing of high band
luma LH in multiplier 508 produces a de-emphasized high
frequency luminance signal L~ which is coupled to a data
input terminal of a folding modulator 512 where the de-
emphasized high band luma LHD is spectrally shifted by a form
of amplitude modulation, 4-field offset modulation, (bearing
in mind that sampling is an amplitude modulation operation)
around a folding carrier to shift the de-emphasized high band
luma L~ to the spectrum of the low band luma below 2.5 MHz,
that is, to produce a high frequency luminance component
shifted into the frequency spectrum of the low frequency
luminance component signal LL thereby providing a shifted de-
emphasized high band luminance signal LHnF~ The shifted de-
emphasized high band luminance signal L~F is then supplied
to the other input of adder 506 to be added back into the
baseband of the low frequency luminance component LL to
thereby produce the interleaved band-limited folded luminance
signal Lf having a bandwidth of e.g. 2.5 MHz, thus being
suitable for the narrow luminance component recording
bandwidth of a conventional VCR, e.g. VHS format.
Figure 6i shows an input full bandwidth signal, and Fig.
6j shows the same signal after folding of the high frequency
band by sub-Nyquist sampling to reduce it to one-half the
bandwidth of the original, the folded high frequency signal
component being indicated by the broken lines.
- 40 -
2056'44
In a simple implementation of a de-emphasis circuit for
record side band-limiting in accordance with the present
invention, multiplier 508 and control signal generator 510
may be replaced by a simple attenuator providing a fixed
amount of attenuation or amplitude reduction to high
frequency luminance signal component LH, so that the high
frequency luminance amplitude level in the folded luminance
signal Lf is maintained below a level at which objectionable
artifacts would become noticeable in the displayed narrow
bandwidth video image on reproduction by a narrow band
playback device. Figure 6k shows the folded signal of Fig. 6j
after it has undergone a fixed de-emphasis of the amplitude
of the folded high frequency components by approximately one-
half .
However, while providing against objectionable artifacts
(i.e. dot crawl) in the displayed narrow band image which
could result from high amplitude folded high frequency
components in the folded narrow band luminance signal, this
fixed form of de-emphasis can undesirably degrade the S/N
performance on the playback side because portions of the high
frequency luminance signal which are at a relatively low
amplitude, such as broad flat areas with little or no
contrast change, may also be reduced in amplitude by the
fixed de-emphasis process, degrading the S/N ratio of these
portions of the recorded luminance signal during
reproduction.
That is, because folding the high frequency luminance at
full amplitude will cause a severe disturbance in the
reproduced images on conventional playback as occurs in the
case of the system described by Faroudja, it is therefore
desirable, for preventing the folded high frequency luminance
components from manifesting as objectionable artifacts in the
displayed image when playing back an encoded recording on a
conventional narrow bandwidth VCR, to reduce the amplitude
(i.e. modulation level) of the high frequency luminance
component interleaved with the low band luminance component.
- 41 -
L~5c74~
Reducing the modulation level of the folded highs by one-half
can provide improved backward compatibility, but at the cost
of increased noise in the displayed wide bandwidth improved
image. This noise increase occurs when the attenuated highs
are boosted in the playback side decoding for restoring them
to their original level. This noise is most noticeable in
broad, low level, flat areas of the image.
In accordance with the present embodiment, the de-
emphasizing of the folded high frequency luminance component
during the record side encoding and the accompanying re-
emphasis processing for restoring the unfolded de-emphasized
high frequency luminance component to its original amplitude
during playback side decoding are preferably performed
adaptively, that is, the level of the folded high frequency
luminance component is de-emphasized adaptively during the
encoding processing, and re-emphasized adaptively during the
decoding processing. Such adaptive de-emphasis of the high
frequency luminance component during the folding process can
significantly improve image noise performance during decoded
playback of the encoded recording by a playback system in
accordance with the embodiment, while also providing enhanced
backward compatibility to the encoded recording.
The adaptive de-emphasis processing provides folding of
the high band luma LH at a full level when the high frequency
luminance component is at a very low amplitude, and folding
of the high band luma LH at a reduced level when the
luminance highs are at a high amplitude. When the high
frequency luminance is adaptively de-emphasized in this
manner during the folding process, the re-emphasis operation
during playback decoding only increases the noise level
during high frequency high amplitude transitions where it is
not noticeable on viewing the reproduced image.
Figure 61 shows the folded high frequency components of
Fig. 6j after adaptive de-emphasis in accordance with the
embodiment, from which it may be seen that the level of de
emphasis of the folded highs is varied in comparison to the
- 42 -
2056'44
fixed de-emphasis in Fig. 6k. Figure 6m corresponds to Fig.
61 but shows additionally the effect of a noise coring
operation performed in adaptively de-emphasizing the folded
high band luma.
As shown in more detail in Fig. 6b, control signal
generator 510 may be constructed of a serially connected
absolute value processor 518, horizontal lowpass filter (LPF)
520 and look-up table (LUT) 522. Absolute value processor
518 may be conveniently implemented as a full-wave rectifier.
The high band luminance signal LH supplied to absolute value
processor 518 is full-wave rectified and then applied to
horizontal LPF 520 which has a cutoff frequency of
approximately 1 MHz. The output signal EH from LPF 520
provides an accurate representation of the average energy in
the luminance high band signal LH over a time period
determined by the time constant of LPF 520. That is, the
value of signal EH represents the average "local" energy of
LH. In broad flat areas, the signal EH will be at zero,
whereas during sharp contrast, high amplitude transitions,
the signal EH will have a high amplitude. The signal EH is
applied as an address to LUT 522 whose output signal 1/D
controls the gain of the de-emphasis multiplier 508.
The gain G transfer function of LUT 522 is monotonically
decreasing in characteristic, as depicted by the thick line
in Fig. 6c where the energy level EH of the high band luma LH
applied to control signal generator 510 is shown along the
horizontal axis and the gain G through multiplier 508 is
plotted on the vertical axis. As may be seen in Fig. 6c, as
the level of the high band luma increases, the gain G through
modulator 508 correspondingly decreases monotonically, thus
providing the de-emphasis transfer function D(LH) through
multiplier 508. The de-emphasis amount D is depicted by the
thin line in Fig. 6c. Optionally, for improved noise
performan~:e, at low amplitudes of the high luma component LH
a coring function may be included in the transfer function of
- 43 -
2056'44
LUT 522 as shown by the diagonally striped area in Fig. 6c.
The operation of LUT 522 is similar to that of LUT 410 in
Fig. 5 described previously, except that in LUT 522 only one
data output signal, the de-emphasis gain control signal 1/D,
is generated from the memory location in the ROM addressed by
EH. If the measured average high luminance band energy EH is
very low or at zero, then the gain G is set near or equal to
unity, passing the high luminance signal LH through
multiplier 508 without any de-emphasis, i.e. no attenuation.
However, when EH is at a high level, the gain G is set at a
lower value, reducing the gain through multiplier 508 below
unity and thereby reducing the effective level of the
luminance high band component passed through multiplier 508
to provide maximum de-emphasis. At intermediate values of
EH, _de-emphasis gain control signal G will assume
correspondingly intermediate values between zero and one, and
an intermediate de-emphasizing of LH will thus occur. The
effect of this adaptive de-emphasis processing is to provide
little or no attenuation of the folded luminance high band
components when they are of low amplitude, but to provide
significant attenuation when the folded luma highs have a
high amplitude.
After the adaptive de-emphasis processing, the de
emphasized high band luminance output signal L~ is applied
to the signal input of amplitude modulator 512. A modulation
clock input terminal 525 of modulator 512 is coupled to a
source (not shown) of a folding carrier signal having a
frequency ff of e.g. 5 MHz for thereby providing shifting of
the high band luminance components to the low band luminance
component spectrum, e.g. below 2.5 MHz.
The de-emphasized high luminance signal LHD is then
modulated, as by a +1, -1 type modulation operation, about a
folding frequency one-half the folding carrier (sampling)
frequency ff in modulator 512. The folding frequency ff is
selected so as to maximize the distance between the folding
- 44 -
256'744
carrier and the baseband luminance signal in the temporal,
vertical and horizontal directions. The folding carrier is
preferably placed at one-half the maximum vertical frequency,
and one-half the maximum temporal frequency, that is, to
correspond to the so-called Fukinuki holes in the temporal
and vertical dimensions, and at about 5 MHz in the horizontal
directions. This maximizes the spectral distance between the
folding carrier and the vertical and temporal lower frequency
components of the luminance signal.
Modulator 512 may be a standard four quadrant
multiplier, or preferably, if the sampling frequency is
properly selected, a +1,-1 type modulator. A +1,-1 type
modulator modulates a sampled signal by a frequency equal to
one-half the sampling frequency by arithmetically negating
every other sample. For example, if the sampling frequency
is selected to be at about 10 MHz , then the folding frequency
will be about 5 MHz, with the actual frequency selected so as
to satisfy the above criteria relating to vertical and
temporal spectral distance from vertical and temporal DC.
The output signal contains a component of one-half the
sampling frequency, and upper and lower sidebands centered
around +1/2 and -1/2 the sampling frequency containing the
spectral information contained in the input signal. Thus the
+1, -1 amplitude modulation will shift (i.e. alias) the high
band luma LH to a -1/2 lower sideband in the 2.5 MHz
bandwidth of the low band luma LL.
As shown in Figure 6d, the amplitude modulator 512 of
Fig. 6a having data input and output terminals and a clock
input terminal may be implemented using a multiplexer (MUX)
524 having a first data input terminal corresponding to the
data input terminal of the modulator 512 and receiving the
signal L~. An arithmetic negator, i.e. invertor (INV), 526
is also coupled to the data output terminal of the de-
emphasis multiplier 508 for receiving the signal L~
therefrom. An output terminal of the arithmetic negator 526
- 45 -
205644
is coupled to a second data input terminal of the multiplexes
524. An output terminal of the multiplexes 524 is coupled to
an input terminal of the adder 506. A folding clock signal,
which has a frequency equal to one-half the sampling clock
frequency, is coupled to the control input terminal of the
multiplexes. This signal alternates between a logic '1'
value and a logic '0' value at the sampling frequency, and
may be generated by a flip-flop coupled to the sampling clock
signal.
In operation, when the folding clock signal is a logic
'1' signal, then the multiplexes 524 couples the non-negated
(+1) signal from the input terminal to its output terminal.
When the folding clock signal is a logic ' 0' signal, then the
multiplexes couples the negated (-1) signal from the
arithmetic negator 526 to its output terminal. In this
fashion, a (+1, -1) modulated signal is reproduced. The
lower sideband of the modulated signal contains a spectral
image of the 2.5-4.2 MHz bandwidth de-emphasized high band
luminance signal L~ but inverted in frequency. That is, the
de-emphasized high band luminance signal L,~ is folded about
the folding frequency such that the lower frequency
components of the de-emphasized luminance high band
frequencies are folded into the bandwidth below 2.5 MHz, and
the higher. frequency components of the de-emphasized high
band luminance frequencies of 4.2 MHz, for example, are
folded into the neighborhood of 800 kHz, thus producing the
folded de-emphasized high band luminance signal L~F.
The folded de-emphasized high band luminance signal L~F
is then combined with the low band luminance signal LL in
adder 506. This adder outputs the composite folded luminance
signal Lf which contains the luminance information of the
input luminance wide baseband signal L compressed within a
folded bandwidth of 2.5 Mhz, thus making it possible to
transmit the 4.2 MHz NTSC baseband luminance information via
a narrow 2.5 MHz bandwidth medium such as by a conventional
- 46 -
2055744
narrow bandwidth format VCR and videocassette.
The folded luminance signal L~ may then be supplied to
a record equalization section 514, shown in Fig. 6a, where
the signal is equalized prior to D/A conversion to pre-
y compensate for loss in the tape path and to compensate for
encoder processing losses, for example by boosting the
frequencies around the 2.5 MHz region to compensate for the
signal attenuation characteristic in the band split region of
the de-emphasis circuit band splitting filter. The folded
luminance signal Lf from the folding section 108 is then
supplied to the D/A converter 110 as shown in Fig. 2 to be
converted to an analog luminance signal L~, then applied to
luminance record circuit 20 of Fig. 1 where it frequency
modulates the recording carrier, and is ultimately recorded
onto videotape by record head 40 as a frequency modulated
narrow bandwidth luminance component.
It will be appreciated that although in the above
example the luminance high band signal LH was adaptively de-
emphasized prior to folding modulation and addition with the
low band luma signal LL in the folding circuit 108, an
equivalent result would be effected by reversing the order of
the multiplier 508 and modulator 512 in the folding circuit
108 shown in Fig. 6a, for first folding the luma high band
signal LH in the folding modulator 512 and then adaptively
de-emphasizing the folded high band output signal from the
folding modulator 512 in the de-emphasis multiplier 508, as
shown in the block diagram of Fig. 6g, where like elements
corresponding to the circuits of Figs. 6a and 6b are
designated with like reference numerals.
Furthermore, it will be appreciated that because the
folding operation according to the present embodiment is
performed only upon the high band luminance component signal,
it is not necessary to employ a low pass filter after the
adder 506 in the folding circuit 108 as would normally be
required if folding were performed on the baseband luminance
- 47 -
2056744
signal. However, the adaptive de-emphasis processing
according to the present embodiment may likewise be applied
effectively to a folding system in which folding is performed
on the baseband luminance signal L. An alternative folding
circuit of this type is shown in Fig. 6h, where the baseband
luminance signal L is applied to one input of an adder 550
and also to an adaptive de-emphasis circuit 560 where the
high frequency luminance component is adaptively de-
emphasized with a monotonically decreasing transfer function.
The de-emphasized baseband luminance signal is then applied
to folding modulator 570, where it is shifted as by +1, -1
multiplexing in accordance with a folding clock as described
above, and the de-emphasized shifted baseband luminance
signal is then applied to the other input of adder 550 to be
combined with the input baseband luminance signal. The
interleaved luminance signal from adder 550 is then passed
through a horizontal lowpass filter 580 having a cutoff of
2.5 MHz, then D/A converted and recorded as previously
described.
In the above-described folding circuit embodiments
employing a band-splitting filter in the de-emphasis
processing as shown in Figs. 6a and 6g, the bandwidth of the
compatibly-recorded luminance signal L, extends only to around
2.5 MHz, that is, the upper limit of the low band luma out of
the band splitting filter, while the luma frequencies above
2.5 MHz are carried in the folded signal. In the folding
circuit of Fig. 6h just described, the necessary use of
lowpass filter 580 after folding also limits the recorded
luminance signal L, bandwidth to 2.5 MHz. This limiting of
the recorded luminance bandwidth is of no significant
consequence when the recorded signal is reproduced by a
playback apparatus implementing a decoder in accordance with
the embodiment for unfolding the compatibly encoded recorded
folded luminance signal and reconstructing a wide bandwidth
luminance signal therefrom, since the folded luminance
frequencies extending beyond 2.5 MHz can be recovered in
- 48 -
2~~6744
playback decoding in accordance with the present embodiment in
order to display an image having full horizontal resolution.
However, when playing back the compatibly encoded recording
on a conventional playback apparatus lacking such a decoding
facility, the displayed horizontal resolution will be limited
by the limited bandwidth of the reproduced luminance signal,
since the higher luma frequencies carried in the folded
signal will not be recovered.
A further embodiment of a folding circuit in accordance
with the invention shown in Fig. 6n can improve the
horizontal resolution during compatible playback by
implementing a different band-split filtering of the
luminance signal during the encoding process. In comparison
to the folding circuit shown in Fig. 6g described previously,
in the folding circuit of Fig. 6n the HLPF 502 and HHPF 504
forming the band-splitting filter in Figs. 6a and 6g are
replaced respectively by a horizontal lowpass filter (HLPF)
1502 which may have its characteristics selected to provide
a -6dB response at around 3 MHz, and a vertical highpass
filter (VHPF) 1504 providing a -6dB response at 2 MHz, with
both HLPF 1502 and VHPF 1504 receiving the input luminance
signal L. HLPF 1502 and VHPF 1504 together perform a band-
splitting function, however, their respective output signals
LL. and LH. are not contiguous half- or split-bands but rather
are overlapping in frequency.
That is, HLPF 1502 and VHPF 1504 together form an
inverse filter (e. g. their respective response
characteristics are the inverse of one another and
symmetrical) which may be implemented as shown in more detail
in Fig. 60. The luminance signal L is input to an odd tap
adder 2510 and also to an even tap adder 2520. The output
of odd tap adder 2510 is coupled to one input of an adder
2530 and also to a subtrahend input of a subtractor 2540.
The output of even tap adder 2520 is coupled to the other
input of adder 2530 and also to a minuend input of subtractor
2540. Adder 2530 adds the outputs of odd tap adder 2520 and
- 49 -
2056'44
even tap adder 2530 together to provide a horizontally
lowpass-filtered output signal LL., and subtractor 2540
differences the outputs of odd tap adder 2520 and even tap
adder 2530 to provide a vertically highpass-filtered output
signal LH,.
The low frequency luminance signal LL. out of HLPF 1502
will thus have a bandwidth which at its upper end is 6dB down
at around 3 MHz, while the high frequency luminance signal LH.
out of VHPF 1504 will contain only those luminance
frequencies above approximately 2 MHz. The high luma signal
LH. from VHPF 1504 is band-shifted by folding modulator 1512
operating in a manner previously described, and the
resultant shifted high luma signal LHF. is adaptively de-
emphasized by a de-emphasis section 1560 in a manner
previously described to provide the band-shifted de-
emphasized high frequency luma signal L~.D. to adder 1506 for
combining with the low frequency luma signal LL. to output the
folded luma signal L~, which is then supplied to the luma
record circuit 20.
It will be appreciated that because the low band
luminance signal LL. from HLPF 1502 has a bandwidth extending
up to approximately 3 MHz, so also the folded luminance
signal Le will have a bandwidth extending to approximately 3
MHz at the high end, i.e. its frequency characteristic will
be 6dB down at 3 MHz, thereby providing the advantage of the
folded luminance signal L,, having approximately 0.5 MHz
greater bandwidth over the folded luminance signal Lf output
by the folding circuit shown in Fig. 6a. Thus, the recorded
luminance signal Lr. will contain the low band luminance
components up to 3 MHz, thereby providing a luminance signal
having its bandwidth limited to 3 MHz, as well as the band-
shifted high luminance frequencies above 2 MHz folded within
the limited bandwidth occupied by the low frequency luminance
components. Accordingly, the folded limited bandwidth
luminance signal Lf when recorded and then reproduced by a
- 50 -
J v
conventional narrow bandwidth playback apparatus lacking any
facility for recovering the folded high frequency luminance
component will provide greater horizontal resolution than
will the folded limited bandwidth luminance signal Lf provided
by the folding circuit of Fig. 6a, offering the advantage of
higher horizontal resolution in compatible playback.
Additional advantages of the use of HLPF 1502 and VHPF
1504 in the folding circuit of Fig. 6n are that in
implementing them in the fashion of the inverse filter shown
in Fig. 60, any ripple can be made symmetrical between the
low and high bands, thereby rendering the folding very
uniform, and the need for equalization of the folded
luminance signal L,. prior to recording can be reduced or even
eliminated since it is not necessary to compensate for loss
in the band-splitting filter cutoff regions.
It is helpful for an understanding of the present
invention to provide a further explanation regarding the
choice of the folding and pre-filtering processing employed,
in the context of applying the present invention to a narrow
bandwidth video recording format such as the conventional VHS
format. It has previously been proposed to shift a high
frequency luminance signal component by filtering and sub-
Nyquist sampling to insert it within spectral holes within
the spatio-temporal frequency domain occupied by the NTSC
chrominance sub-carrier component, but offset relative
thereto. See for example, T. Fukinuki et al., "Extended
Definition TV Fully Compatible with Existing Standards", IEEE
Trans. on Communications, Vol. COM-32, No. 8, August 1984,
pages 948-953; and T. Fukinuki et al., "NTSC FULL COMPATIBLE
EXTENDED DEFINITION TV PROTO MODEL AND MOTION ADAPTIVE
PROCESSING", reprinted from IEEE Communications Society "IEEE
Global Telecommunications Conference", No. 4.6, December 2-5,
1985, pages 113-117,
As described above, the luminance components above 2.5
MHz normally lost by low luma band recording systems such as
- 51 -
~~5674~
are employed in conventional VHS and Beta format VCRs are, in
accordance with the invention, folded into frequency slots
below 2.5 MHz by a form of sub-Nyquist sampling known as
four-field offset subsampling, which provides adequate
spatio-temporal performance. For still or low motion image
areas, high vertical resolution can be obtained through the
use of intra-frame processing, and a spatio-diagonal lowpass
performance is possible for image motion areas by employing
intra-field processing.
Figures 6e and 6f show the frequency characteristics of
the folding process employed in the present embodiment, in the
vertical-horizontal frequency spectrum and the vertical
frequency-temporal domains, respectively. The high band
luminance is folded into the so-called "Fukinuki" areas in
the upper left and lower right quadrants of the diamond in
Fig. 6f. Because conventional VCRs employ a component type
recording system, in applying the present embodiments to such
recording systems it would-also have been possible to fold
the high band luminance into the spectral 'holes' from which
the NTSC chrominance sub-carrier has been removed in the
upper left and lower right quadrants of the diamond.
However, because there may still be residual chroma sidebands
present in those areas which might interfere with the folding
and unfolding processes, it has been found to be advantageous
to fold the high band luma into the Fukinuki areas as shown,
with the result that by folding into these quadrants, any
residual chroma components when unfolded will be in
complementary phase on successive fields and will be
optically canceled in the display monitor. Further, it
should be noted that because the folded highs alternate in
phase at 15 Hz, motion detection becomes impractical after
folding, so that it is preferred to detect motion prior to
folding by temporal differencing and spatial lowpass
filtering of the separated baseband luminance prior to
folding.
Next, with reference to Figures 7a through 7f, the
- 52 -
2~~6744
processing of the chrominance and motion signals to provide
the composite chrominance-plus-motion signal C+M in the
encoder will be described in more detail in the exemplary
case of implementing the system of the present embodiment for
compatibility with the conventional VHS format. As is well
known, in VCRs according to the standard VHS format, the
input video chrominance (color difference) information which
is modulated on a chroma sub-carrier (nominally 3.58 MHz for
NTSC composite video) is separated from the luminance
information prior to recording and then frequency down-
converted by heterodyning against a 4.21 MHz carrier to
provide a sub-carrier at approximately 629 KHz (40 times the
horizontal scanning frequency) with the sidebands reversed in
order from the NTSC chroma component, to be directly recorded
onto the videotape as a so-called 'color-under' sub-carrier
signal, that is, in the frequency spectrum below the recorded
luminance component, i.e. below approximately 1.2-1.3 MHz.
As is also well known, in order to reduce beat and cross-talk
between mutually adjacently recorded tracks, during the
recording process the phase of the VHS format's 629 KHz
color-under carrier is shifted 90° each line in every track
(i.e. every field), so that its phase is shifted or rotated
+90° per line (advanced) on odd tracks and -90° per line
(retarded) on even tracks. See for example U.S. patents No.
3,723,638 to Fujita, No. 4,068,257 to Hirota et al. and No.
4,178,606 to Hirota.
Figure 7b depicts the relationship between the "raw"
spread motion signal M (e.g. out of signal spreader 222) and
the NTSC chrominance sub-carrier in the vertical and
horizontal frequency domain. As a consequence of the phase
shifting of the VHS format's color-under carrier, a two-
dimensional (horizontal and vertical frequency) spectral
analysis of the VHS format's 629 KHz color carrier shows that
on even tracks (fields) it occupies the first and third
quadrants and on odd tracks (fields) it occupies the second
- 53 -
~~~E144
and fourth quadrants, as depicted in Figs. 7c and 7d. Thus,
it may be seen that on even tracks the second and fourth
quadrants of the VHS format 629 MHz color-under carrier are
normally "vacant" or unused, that is, not utilized for signal
carriage, while on odd tracks the first and third quadrants
are "vacant".
In accordance with the present embodiment as exemplarily
embodied for compatibility with conventional VHS format
recording and reproduction apparatus, during the encoding
processing in the chrominance/motion combining circuit 116 of
Fig. 2 the spread motion signal M is modulated on a carrier
and combined with the down-converted VHS color-under signal
so that the modulated motion signal appears in the vacant
second and fourth quadrants of the VHS format 629 KHz color-
under carrier C on even tracks, and in the vacant first and
third quadrants of the color-under carrier on odd tracks, to
produce the composite chrominance-plus-motion signal C+M.
It should be noted that the "vacancy" in the quadrants
of the quadrature modulated chrominance signal for
accommodating the motion signal encoding therein may be
created in several ways and at different stages of the
encoding process, such as by implementing a pre-combing of
the separated NTSC chrominance signal e.g. by the use of
anti-crosstalk element 224 in the form of a 2H comb filter.
Thus for example, the NTSC chrominance signal at 3.58 MHz may
be processed by passing it through a vertical high pass
filter (such as VHPF 224), or the down-converted color-under
chrominance signal at 6~9 KHz may be passed through a
diagonal filter, or the baseband chrominance (U+V/I+Q) signal
may be vertically lowpass filtered. Each of these processes
will provide essentially an equivalent effect, namely
creating the "vacant" quadrants in the modulated chrominance
signal into which the motion signal may then be compatibly
encoded.
Figure 7a shows the chrominance/motion combining circuit
116 of Fig. 2a according to a VHS format-compatible
- 54 -
2056'44
embodiment in more detail. The spread motion signal M from
output terminal 225 is input to a modulator 610 and modulated
on a 250 KHz four-phase carrier to generate a modulated
motion signal component having a horizontal frequency of 250
KHz and with its phase shifting forward or backward 90° per
line in alternate fields in complementary fashion (phase
complement) to that of the 629 VHS KHz color-under carrier C,
so that in those fields (tracks) where the color-under
carrier C occupies the first and third quadrants, the motion
signal M occupies the second and fourth quadrants, while in
opposite fields (tracks) the color-under carrier C and motion
signal M reverse quadrants. The down-converted color-under
carrier component C from chroma modulator 226 in Fig. 3 and
the modulated motion signal component M from modulator 610
are input to respective signal inputs of an adder 620 to be
combined into the resultant chrominance-plus-motion signal
C+M. It will be understood by those skilled in the art that
the chrominance signal C may be appropriately processed for
color burst emphasis as by burst emphasis or gating circuitry
235 prior to application to adder 620. The chrominance-plus-
motion signal C+M output by adder 620 thus contains the
chrominance information C as well as the spread motion signal
information M modulated on a four-phase carrier but occupying
complementary quadrants of the carrier, advancing and
retarding 90° in phase and alternating between even and odd
quadrants in alternate fields. The chrominance-plus-motion
signal C+M from adder 620 is then filtered by a horizontal
lowpass filter (HLPF) 630 having a cutoff frequency of around
1.2 - 1.3 MHz and supplied to D/A converter 118 to be
converted to an analog signal C+Mr which is supplied to
chrominance record circuit 30 to be recorded directly onto
the videocassette tape the same as a color-under component by
the recording head 40 in conventional manner. The choice of
the 250 KHz carrier frequency for the motion signal is made
in order to reduce the visibility of interference during
- 55 -
20~614~
playback of an encoded recording on a conventional VHS format
playback device, however, it is also possible to modulate the
motion information on a 629 KHz carrier like the chroma, so
long as the quadrants occupied by the respective signals are
complementary as described above.
As is well known, in playback processing according to
the VHS format, the reproduced 629 KHz color-under carrier is
up-converted to 3.58 MHz by reversing the heterodyning
process employed on the record side. As shown in Fig. 7f,
conventional VCRs typically employ a vertical high-pass comb
filtering (line comb filtering) in a bandpass amplification
processing of the recovered 3.58 MHz chrominance sub-carrier
after the up-conversion, to cancel out the luminance and
adjacent track crosstalk signals from the chroma. Further,
many conventional television sets include a chroma bandpass
amplification circuit using such a chroma comb filter. Thus,
on reproduction, the encoded motion signal component M, being
180° out of phase with the chrominance component C, will be
canceled by the line combing and only the in-phase chroma
signal will be output from the chroma comb filter in such a
conventional VCR or TV when playing back an encoded
recording, as shown graphically in Fig. 7e, thereby ensuring
that even though the encoded motion signal will be reproduced
from the tape during playback, it will be filtered out of the
chroma signal in a conventional VCR during playback
processing (or filtered out in the television set used to
display the playback) and not generate any noise or
interference in the reproduced image.
Furthermore, as described above, because in accordance
with the embodiments the folded high frequency luminance
components which are inserted into the bandwidth of the low
band luminance are advantageously adaptively de-emphasized
during the encoding process on the record side, the amplitude
of these high luma components in the reproduced signal on
playback of the encoded recording by a conventional low band
VCR will be sufficiently below a level which would cause
- 56 -
"t.~ .J ~ f '"1' ~"P
objectionable artifacts in the displayed reproduced video
image.
The apparatus and processing described above may be used
to record a full-bandwidth video signal in encoded bandwidth
s reduced form on a standard video cassette, which encoded
narrow band recording may then be compatibly played back on
a standard narrow bandwidth VCR to reproduce a narrow
bandwidth video image without' objectionable artifacts.
Apparatus and processing as described hereafter may be used
for extracting the luminance high frequencies folded into the
luminance low frequencies and regenerating the full bandwidth
video signal upon playback of such a previously recorded
video cassette.
Figure 8 is a block diagram of a VCR playback system in
accordance with the present embodiments. !In Figure 8, a
playback head 50 is incorporated in a standard tape transport
(not shown) of a conventional narrow bandwidth (e.g. VHS
format) VCR. Playback head 50 is coupled to respective input
terminals of a luminance signal playback circuit 60 and a
chrominance signal playback circuit 80. An output terminal
of luminance signal playback circuit 60 is coupled to a first
input terminal of a decoder 70, and an output terminal of
chrominance signal playback circuit 80 is coupled to a second
input terminal of decoder 70. An output terminal of decoder
70 is coupled to a video output terminal 15. Output terminal
15 is coupled to utilization circuitry (not shown) which may
be, for example, a television receiver for reproducing the
images previously recorded on the video cassette, or a Y-C
output jack.
In operation, playback head 50 supplies its reproduction
signal to both the luminance playback circuit 60 and the
chrominance playback circuit 80, in a known manner. The
previously recorded folded luminance signal occupies a band
of frequencies at about 1.4 - 7.0 MHz, and the previously
recorded chrominance-plus-motion signal occupies a 1 MHz band
of frequencies at around 500 KHz above and below 629 kHz .
- 57 -
2oss~44
The luminance playback circuitry processes the folded
luminance signal in the usual manner (i.e. frequency
demodulation) to produce the narrow bandwidth playback folded
luminance signal LPb. The chrominance playback circuitry
processes the chrominance-plus-motion signal to produce the
playback chrominance-plus-motion signal C+Mpb. This signal is
then processed by the decoder 70 which separates the motion
signal from the chrominance signal, and uses the recovered
motion signal to aid in processing the luminance component
for reconstructing the full bandwidth luminance signal. The
reconstructed full bandwidth luminance signal and the
separated chrominance signal may then be combined to form a
composite video signal at output terminal 15.
Figure 9 is a more detailed block diagram of the decoder
70 illustrated in Figure 8. In Figure 9, an input terminal
805 is coupled to the output terminal of luminance playback
circuit 60 of Figure 8. Input terminal 805 is coupled to an
input terminal of an analog-to-digital converter (A/D) 804.
An output terminal of A/D 804 is coupled to an input terminal
of a horizontal lowpass filter HLPF 805 having its passband
selected at around 2.5 MHz or 3 MHz, depending on the
bandwidth of the folded luminance signal. An advantage of
filtering the playback luminance signal digitally is that the
group delay characteristics of the digital HLPF 805 may be
made flat, which is difficult to achieve in an analog
implementation. The output terminal of HLPF 805 is coupled
to an input terminal of a time base corrector (TBC) 806. An
output terminal of TBC 806 is coupled to a data input
terminal of an unfolding circuit 808. An output terminal of
unfolding circuit 808 is coupled to a luminance signal input
terminal of a spatio-temporal post-filter 820. An output
terminal of post-filter 820 is coupled to an input terminal
of an adaptive re-emphasis circuit 822. An output terminal
of adaptive re-emphasis circuit 822 is coupled to a luminance
signal input terminal of composite video signal generator
810. Composite video signal generator 810 may typically
- 58 -
2056'44
include a D/A converter for converting the applied digital
luminance and chrominance component signals to analog
signals. An output terminal of composite video signal
generator 810 is coupled to an output terminal 815. Output
terminal 815 is coupled to utilization circuitry (not shown)
which, for example, may be a television receiver for
reproducing the images which were previously recorded on the
cassette or a Y-C output jack.
Alternatively, the recovered luminance and chrominance
signals L' and C', which are in digital form, may be output
directly in digital form for utilization in further
processing, or may be D/A converted and output directly as
analog Y and C signals for further utilization.
An input terminal 825 is coupled to the output terminal
of the chrominance playback circuit 80 of Figure 8. Input
terminal 825 is coupled to an input terminal of an analog-to
digital converter (A/D) 814. An output terminal of A/D 814
is coupled to an input terminal of a time base corrector
(TBC) 816. An output terminal of TBC 816 is coupled to an
input terminal of a chrominance/motion signal separator 818.
A chrominance signal output terminal of the
chrominance/motion signal separator 818 is coupled to a
chrominance input terminal of the composite video signal
generator 810. A motion signal output terminal of the
chrominance/motion signal separator 818 is coupled to a
control input terminal of the spatio-temporal pre-filter 820.
In operation, the upper elements in Figure 9 operate to
extract the full-bandwidth luminance signal from the reduced
bandwidth luminance signal previously recorded on the
cassette. A/D 804 produces a sampled multi-bit digital
signal representing the playback folded luminance signal.
The TBC 806 operates to correct any timing inaccuracies which
are introduced by fitter in the tape mechanism or any other
source of timing inaccuracy, and produces the recovered
folded luminance signal Lf* (where "'" indicates a playback
signal representing the same signal as previously recorded on
- 59 -
2~~~7~~
the cassette).
The lower elements in Figure 9 operate the extract the
chrominance-plus-motion (C+M) signal previously recorded on
the cassette. The A/D 814 produces a sampled multi-bit
digital signal representing the chrominance-plus-motion
signal and the TBC 816 operates to correct any timing
inaccuracies in this signal, and produces the recovered
chrominance-plus-motion signal C+M'.
When recorded, the chrominance signal and the luminance
signals were in phase synchronism. However, they are passed
through two separate independent paths in the record
circuitry (illustrated in Figure 1) and are frequency
division multiplexed on the cassette. This separate
processing may introduce phase inaccuracies between the two
signals which are not compensated for in the two separate
TBCs 806 and 816. U.S. Patent No. 5,083,197 dated
January 21, 1992 describes in detail apparatus for restoring
the proper phase relationship between the chrominance and
luminance signals.
Chrominance/motion separator 818 processes the recovered
chrominance-plus-motion signal C+M' to produce a recovered
motion signal M', which is supplied to the control input
terminal of the spatio-temporal post-filter 820, and
chrominance signal C', which is supplied to the chrominance
signal input terminal of the composite video signal generator
810.
The unfolding circuit 808 unfolds (i. e. re-shifts) the
luminance high band frequencies which were previously folded
into the luminance low band frequency spectrum and combines
the low and high band luminance signals to output the full
bandwidth unfolded luminance signal Luf. This full bandwidth
unfolded luminance signal L"f is supplied to the spatio-
temporal post-filter 820 where the unfolded full bandwidth
luminance signal L~f is motion-adaptively spatio-temporally
filtered to provide the de-emphasized luminance signal LD'
- 60 -
2056'44
having the high frequency luminance components still de-
emphasized due to the record side de-emphasis processing.
This unfolded de-emphasized luminance signal LD' is supplied
to the adaptive re-emphasis circuit 822 where the de-
emphasized high frequency components are adaptively re-
emphasized to restore them to their original amplitude to
provide the recovered full bandwidth luminance signal L' with
proper amplitude relationship. The recovered full bandwidth
luminance signal L' is supplied to the luminance signal input
terminal of the composite video signal generator 810.
Composite video signal generator 810 operates in a known
manner to combine the luminance signal L' and chrominance
signal C' to form a standard (digital or analog) composite
video signal. This signal may be used by any equipment which
utilizes such a signal, for example, a television receiver or
display monitor.
Figure l0a is a block diagram of a portion of the
luminance recovery section illustrated in the upper half of
Fig. 9, showing in more detail the unfolding circuit 808 and
the spatio-temporal post-filter 820. After time base
correction by TBC 806, the folded luminance signal Lf' is
applied to one input of unfolding circuit 808, which may be
implemented by a modulator 902 to which is also supplied an
unfolding carrier having a frequency f". A three-dimensional
spectral plot of folded luminance signal Lf is depicted in
Fig. lOd, with the low band luminance component appearing in
the foreground (i.e. temporal zero frequency) and the folded
luminance high band component appearing in the background
(i.e. 15 Hz temporal frequency) . The folded luminance signal
Lf is unfolded by direct or "straight" sub-Nyquist sampling
(as contrasted to the ~offset~ technique employed during
folding) around the unfolding frequency (selected to be e.g.
5 MHz in accordance with the criteria described above in the
description of the folding modulator 512 of Figure 6a) by the
modulator 902, to provide the unfolded luminance signal L"f~
- 61 -
2056'44
Unfolding modulator 902 may be constructed in a known manner
using a four quadrant multiplier, and is preferably a +1, 0
type modulator operating to insert zero values replacing odd
or even samples depending on the unfolding phase, driven by
a clock signal at one-half the sampling frequency, which in
this example is a sampling frequency of 10 MHz.
The unfolded (i.e. remodulated) luminance signal L"f is
then applied to the input terminal of spatio-temporal post-
filter 820 for removal of byproducts of the unfolding process
prior to re-emphasis of the unfolded high frequency luminance
component. The post-filter 820 includes a temporal lowpass
filter (TMF) 904 configured as a frame comb lowpass filter
(which may be identical in structure and operation to THPF
204, HHPF 216 and subtractor 210 in Figs. 3 and 4) which
provides frame averaging and removes components all the way
down to spatial DC from the unfolded luminance signal L"f for
providing temporally derived unfolded luminance signal LT'.
Temporal filter 904 is arranged in parallel with a spatial
filter (SPF) 906 (which may be identical in structure and
operation to VHPF 202, HHPF 212, subtractor 208 and HLPF 209
in Figs. 3 and 4) acting as a diagonal lowpass filter for
providing spatially derived unfolded luminance signal LS'.
A soft switch 914 having its data inputs connected to
the LT' and LS' outputs of TMF 904 and SPF 906, respectively,
varies its data output proportionally between the temporally
filtered and spatially filtered unfolded luminance signals
applied at its data inputs from TMF 904 and SPF 906, under
control of motion-adaptive scaling factor signals K' and 1-K'
which are applied to control input terminals of soft switch
914 from a look-up table (LUT) 910. LUT 910 generates
scaling factor signals K' and 1-K' in accordance with the
recovered motion signal M' applied to its input terminal from
chrominance/motion separation circuit 818, for performing
motion-adaptive post-filtering of the unfolded luminance
signal L"f prior to the re-emphasis stage processing. The
- 62 -
~~~~~74~f
output terminal of soft switch 914 is coupled to an output
terminal of post-filter 820 at which is provided the spatio-
temporally post-filtered unfolded de-emphasized luminance
signal LD' which is in turn coupled to the luminance signal
input terminal of adaptive re-emphasis circuit 822.
As noted, SPF 906 may correspond in structure and
operation to the luminance spatial filter section of the
adaptive luma separation circuit 104 formed by VHPF 202, HHPF
212, subtractor 208 and HLPF 209 shown in Fig. 3, providing
a diagonal lowpass filtering response for spatially
processing the unfolded signal to remove unfolding artifacts,
i.e. remodulation byproducts and residual unfolding carrier
which may be present during image motion and which manifest
strongly in the diagonal. This correspondence permits
utilization of the same filter in encoding and decoding.
The unfolded luminance signal L"f from modulator 902 of
unfolding circuit 808 is applied to the input of post-filter
820 and is commonly supplied to the inputs of both TMF 904
and the SPF 906. The temporally derived unfolded luminance
signal LT' output from TMF 904 is supplied to a first data
input terminal of soft switch 910. The spatially derived
unfolded luminance signal LS' output from SPF 906 is supplied
to a second data input terminal of soft switch 914.
The unfolding carrier frequency in the decoder
corresponds to the folding carrier frequency in the encoder.
As described above with respect to the folding carrier, the
carrier frequency is selected to maximize the distance
between the baseband luminance signal and the luminance image
signal in the temporal, vertical and horizontal directions.
But the spectral characteristics of the recorded luminance
signal effect the spectral shape of the unfolded luminance
signal and the image signal. Thus, the spectral
characteristics of the recorded luminance signal must be used
to adaptively filter the reproduced video signal to leave
only the full bandwidth unfolded luminance signal.
- 63 -
f r.
~~~~b/
Canadian patent application No. 2,036,177 filed
February 12, 1991 describes in detail techniques for spatial
filtering of an unfolded luminance signal to remove unwanted
diagonal frequency components present during image motion.
When the level of motion in the video image is low, the
unfolded luminance signal lies at temporal low frequencies,
close to temporal DC, and the luminance component lies close
in the temporal direction to the unfolding carrier, which was
selected to be far away from temporal DC. In the presence of
motion in the video image, the luminance component occupies
a wider temporal bandwidth. In this case modulation
byproducts may overlap temporally with the unfolded luminance
signal and must be removed spatially. This is graphically
depicted in Figs. l0e and lOf. In Fig. l0e is shown three-
dimensionally the spatio-temporal (F~~~, F~,~, F~,~) spectrum of
the unfolded luminance signal L"f output from unfolding
circuit 808. During still image areas (no motion), the
baseband luminance (low band and unfolded high band luma) is
2 0 in the tempora 1 foreground ( i . a . at zero tempora 1 frequency ) ,
while in the temporal background (i.e. at 15 Hz temporal
frequency) where the unfolding carrier is shown are the
folded highs and remodulated lows.
In the case of still motion areas of the video image,
the frame averaging (frame lowpass filtering) performed by
TMF 904 will cancel out all of the components in the temporal
background by the unfolding carrier (e. g. at 15 Hz) by frame
combing. In this manner, a 3 dB reduction of low spatial
frequency noise components can be obtained. However, during
image motion, it becomes necessary to employ spatial
filtering of the unfolded luminance signal as shown in Fig.
lOf, in order to notch out remodulation byproducts which
manifest in the diagonal, thereby maintaining the baseband
luminance including the unfolded highs in the temporal
foreground.
The selection between what amounts of the temporally-
- 64 -
2o5s~44
and spatially-filtered unfolded luminance signals LT' and LS.
which will be output is performed by soft switch 914 under
control of the recovered motion signal M'. Recovery of the
motion signal in the decoder 70 will be described in more
detail later.
Soft switch 914 controls the proportion of the
temporally derived and spatially derived unfolded full
bandwidth luminance signals LTi and LS~ to be included in the
spatio-temporally post-filtered unfolded de-emphasized
luminance signal LD' in response to the recovered motion
control signal M'. When the level of image motion is zero or
nearly zero, the output of the soft switch 914 consists
completely of temporally derived unfolded luminance signal LTy
from TMF 904, and does not contain any of spatially derived
unfolded luminance signal LS'. As the magnitude of motion in
the image gradually increases, the proportion of the
temporally derived luminance signal LT' input from the TMF 904
in the output of the soft switch 914 correspondingly
decreases and the proportion of the spatially derived
luminance signal LS' input from SPF 906 correspondingly
increases. In the presence of relatively high levels of
motion, the output from the soft switch 914 will consist
entirely of the spatially derived signal LS~ from SPF 906.
Referring now to Fig. lOg, there is shown a more
detailed block diagram of the soft switch 914 of the post
filter 820 in Fig. 10a. It will be seen that soft switch 914
is constructed in identical fashion to soft switch 214 shown
in Fig. 5. In Fig. 10g, input terminals 1005 and 1015 for
temporally derived unfolded luminance signal LT' and spatially
derived unfolded luminance signal LS' respectively correspond
to input terminals 405 and 415 in Fig. 5. Multipliers 1004
and 1008 correspond to multipliers 404 and 408, respectively,
in Fig. 5. Likewise, LUT 910 in Fig. 10f having motion
signal M~ input terminal 1025 corresponds to LUT 410 in Fig.
5 having motion signal M input terminal 425. Similarly,
- 65 -
2oss~~4
adder 1012 for adding the outputs of modulators 1004 and 1008
to provide thereby the motion adaptively spatio-temporally
filtered unfolded de-emphasized luminance signal LD' at output
terminal 1012 in Fig. 10g corresponds to adder 412 in Fig. 5.
Generation of the scaling factor signals K' and 1-K' in LUT
910 in accordance with the applied recovered motion signal M*
is performed in the same manner as was previously described
above with regard to the operation of soft switch 214 in Fig.
5 for generating scaling factors K and 1-K in accordance with
motion signal M, and will not be described in further detail
here.
Advantageously, the ROM employed for LUT 910 may be the
same as in LUT 410. Furthermore, because the encoder 10 and
decoder 70 will typically not be operated simultaneously, for
advantages of convenience and economy the pre-filter 820 may
share many common elements such as filter blocks and the soft
switch with the encoder side luminance separation circuit
104.
Next, the re-emphasis processing of the post-filtered
unfolded luminance signal will be described with reference to
Figs. l0a through 10c. After motion-adaptive spatio-temporal
filtering by post-filter 820, the unfolded baseband luminance
signal LD' contains a reduced amplitude high frequency
luminance component LH' (the de-emphasized high horizontal
detail signal) as a result of the adaptive de-emphasis
processing during the encoding on the record side, as shown
in Fig. lOb. In order to restore the unfolded high luminance
component to its original amplitude, the adaptive re-emphasis
circuit 822 shown in more detail in Fig. lOb is employed for
performing the inverse operation to the de-emphasis
processing of the high luma by the adaptive de-emphasis
circuit in the encoder.
Adaptive re-emphasis circuit 822 is constructed in
similar manner to the adaptive de-emphasis circuit 108 shown
in Figs. 6a and 6b, and for convenience the two circuits may
- 66 -
2056'44
advantageously share many common elements. The post-filtered
unfolded luminance signal LD' from the output terminal 435 of
soft switch 914 is applied to a 2.5 MHz band-splitting filter
configured by a 2.5 MHz horizontal lowpass filter (HLPF) 1102
and a subtractor 1104, corresponding to HLPF 502 and
subtractor 504 in Fig. 6a. The low band luminance signal LL'
from HLPF 1102 is supplied to the subtrahend input terminal
of subtractor 1104 and also to one input of an adder 1106
corresponding to adder 506 in Fig. 6a. At the subtractor
1104, the low band luma LL' is subtracted from the de-
emphasized baseband luma LD' which is applied to the minuend
input terminal of subtractor 1104, to output the recovered
de-emphasized high band luminance component LHD' corresponding
to the de-emphasized high band luminance component L~ output
at the multiplier 508 in the adaptive de-emphasis circuit 108
during encoding. The de-emphasized high band luminance
component L~ output by subtractor 1104 is applied to the
data input of the re-emphasis multiplier 1108 corresponding
to de-emphasis multiplier 508 of de-emphasis circuit 108 in
Fig. 6a, and also to the input of a control signal generator
1110. Control signal generator 1110 includes serially
arranged absolute value circuit 1118, horizontal low pass
filter (HLPF) 1120 and look-up table (LUT) 1122, in similar
fashion to absolute value circuit 518, HLPF 520 and LUT 522
in control signal generator 510 of de-emphasis circuit 108 in
the encoder 10. However, as will be explained below, the
characteristics of LUT 1122 used in the re-emphasis
processing are reversed from those of LUT 522 used in the de-
emphasis processing.
The action of control signal generator 1110 corresponds
to that of control signal generator 510 in Fig. 6a, except
that their respective transfer functions are substantially
the inverse of one another, as may seen from comparing the
characteristic graphs in Figs. 6c and lOc. That is, whereas
the de-emphasis gain G transfer function of LUT 522 in the
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~o~s~44
de-emphasis circuit 108 is preferably monotonically
decreasing for providing luma de-emphasis, the re-emphasis
gain G' transfer function of LUT 1122 is preferably
monotonically increasing for providing unity gain through re-
emphasis multiplier 1108 at low signal levels of L~' and high
gain through re-emphasis multiplier 1108 at high signal
levels of L~'. Thus, re-emphasis circuit 822 has the
adaptive characteristic that in broad flat image areas it
provides little or no gain to the de-emphasized high band
luma signal L~', with the resultant effect that a high
frequency luminance component which was originally at a low
level in the wideband video signal input to the encoder (that
is, in high band luma signal LH in de-emphasis circuit 108)
and did not therefore undergo de-emphasis during the encoding
process is not subjected to re-emphasis during the decoding
processing but is passed through multiplier 1108 at unity
gain and output at its original amplitude. On the other
hand, for those portions of the de-emphasized high band luma
signal L~' which correspond to high frequency, high amplitude
transitions in the original input video signal (that is, in
high band luma signal LH) and were therefore de-emphasized
during encoding, the gain through multiplier 1108 is
increased to restore these high frequency luminance
components to their original level. The adaptive re-emphasis
is accomplished by measuring the average energy level (i.e.
the average "local" energy) in the de-emphasized high band
luma signal L~' by operation of absolute value circuit 1110
and LPF 1120 to derive the average energy signal EH' which is
applied as an address to LUT 1122. LUT 1122 then outputs a
re-emphasis gain control signal G' to the gain control data
input of multiplier 1108 to control the gain through
multiplier 1108, and thereby the amount of re-emphasis R
performed on the de-emphasized high band luma signal L~'.
The relation between EH', the gain G' through multiplier 1108
and the re-emphasis amount R is shown in Fig. lOc, where the
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2~56'~44
gain G" is depicted by a heavy line and the re-emphasis
amount R is shown by a thin line.
The re-emphasized unfolded high frequency luminance
component signal LH" output from re-emphasis multiplier 1108
is applied to adder 1106 to be added together with the
unfolded low frequency luminance component LL , and adder 1106
outputs the reconstructed baseband luminance signal L' with
proper amplitude relationship restored thereto and
corresponding to the full bandwidth luminance signal L in the
encoder. The reconstructed baseband luminance signal L" is
supplied to the luminance input of composite video signal
generator 810.
In practice, on the recording side, some amplitude
boosting of very low amplitude high frequency luminance
signal components may be done, allowing for some compression
during playback, thus improving the S/N ratio in broad flat
areas of the image without degrading backward compatibility
of the encoded recorded signal. Correspondingly, in the re-
emphasis circuit 822, the control signal generator 1110 may
provide a coring function, as by incorporating such a coring
function in the transfer characteristic of LUT 1122 as
depicted by the shaded area in Fig. lOc.
It will be understood that the re-emphasis processing
employed in the playback side decoding will correspond as
closely as possible to the de-emphasis processing employed in
the record side encoding. Thus, in the case where the de-
emphasis processing employed the alternative embodiment low
pass filter 1502 and high pass filter 1504 in the folding
circuit of Fig. 6n which are formed by the inverse filter of
Fig. 6o for deriving the low and high band luma signals LL
and LH, then the re-emphasis circuit 822 will employ a
corresponding lowpass and highpass filter arrangement
(preferably the same inverse filter arrangement) in place of
the band-split filter of HLPF 1102 and subtractor 1104, in
order to derive the low and high luma signals LL" and LH", and
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2Q56'~~4
similarly, because the folded signal Lf' on playback will
contain the low band luma frequencies up to 3 MHz, it will be
understood that in such case the lowpass filter HLPF 805 may
be selected to correspondingly pass frequencies below 3 MHz
rather than 2.5 MHz.
As described above, during recording it is possible to
derive a motion-representative signal during the
chrominance/luminance signal separation processing of the
input composite video signal itself. So-called "false
motion" may be introduced into this motion-representative
signal by the chrominance signal (i.e. chrominance
information which aliases as motion), but this false motion
can be largely eliminated by vertically and horizontally
lowpass filtering the temporally highpass-filtered signal (or
temporally highpass-filtering the spatially lowpass-filtered
composite signal). Because the NTSC chrominance component
sidebands do not extend down below 2 MHz, horizontal lowpass-
filtering ensures that chrominance components which might
give rise to false motion are removed from the motion-
representative signal during the motion derivation process.
As described above, the luminance high frequencies are
folded into the low frequency luminance signal spectrum by
modulating them on a folding carrier which is placed in a
Fukinuki hole, similar to the manner in which the NTSC
chrominance subcarrier is placed in the composite NTSC video
signal. However, there are no restrictions on the lower
sidebands of the folded luminance high frequencies. In fact,
diagonal detail in the high band luminance signal, when
folded into the luminance low band frequencies, can extend
all the way down to spatial DC. Because the folding carrier
is alternated on a frame-to-frame basis (to maximize the
temporal distance from DC) these diagonal details may
incorrectly get detected, i.e. mis-recognized as motion,
giving rise to false motion detection, and no degree of
spatial filtering can prevent this false motion detection.
Thus, to properly remove the folding byproducts from the
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2~ss~4~
unfolded luminance signal on playback, it is necessary to
supply a separate channel for passing the motion-
representative signal, derived during the encoding side
processing and utilized for motion-adaptively filtering the
separated luminance component signal, to the decoder side for
utilization in motion adaptively pre-filtering the unfolded
luminance signal as previously described.
One skilled in the art of video signal processor design
will recognize that providing a separate channel for
supplying the motion representative signal to the playback
circuitry allows the motion adaptive luminance reconstruction
process in the playback circuitry to mimic the motion-
adaptive processing in the chrominance/luminance signal
separator in the record circuitry. For example, if the
chrominance/luminance signal separator in the record
circuitry chose temporal processing in some region of the
image to derive the luminance signal, it would be incorrect
to choose spatial processing on the playback side to
reconstruct the full bandwidth luminance signal in the same
region of the image.
Further, the chrominance/luminance signal separation
process performed on the composite NTSC signal in the
encoder, no matter how well done, can introduce some
artifacts into the image, i.e. chrominance aliasing as
luminance and vice versa. The full bandwidth luminance
signal reconstruction process performed in the decoder can
also introduce artifacts into the image. If the second
process is independent of the first process, then the
artifacts introduced by the upstream process have artifacts
introduced upon them by the downstream process, intensifying
them. Artifact intensification can be greatly reduced if the
downstream processing can be made to faithfully follow, that
is, to parallel, the upstream processing. Providing a
separate channel in the transmission or recording medium for
the motion representative signal, for compatibly encoding the
motion signal and allowing it to be recovered on the decoding
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2G56'~44
side, allows for both the above improvements. As described
previously, it has been found advantageous to compatibly
encode the motion signal into the chrominance component
signal and then to recover it by a separation process in the
decoding processing. In a VHS format-compatible
implementation, this is done by encoding the motion signal
into vacant quadrants of the VHS format color-under carrier
on the record side. On the playback side, the encoded motion
signal must be separated from the reproduced color-under
carrier component, in order that the recovered motion signal
may be utilized for motion-adaptive filtering of the unfolded
luminance signal.
The separation of the composite chrominance-plus-motion
signal C+M* in chrominance/motion separation circuit 818 of
Fig. 9 as implemented for application to use with the
conventional VHS format will now be described in more detail
with reference to Figs. lla through lle. It is preferable,
from the viewpoint of digitally implementing the separation
process, to perform this processing directly on the C+M*
signal when it is still at the color-under frequency prior to
up-conversion, due to the reduced storage requirements for
the filter stages involved at the lower frequency. As shown
in Fig. lla, the chrominance-plus-motion signal C+M* from TBC
816 is applied to the input of a quadrant selective filter
QSF 1202 and also to the minuend input of a subtractor 1204.
QSF 1202 selects between odd and even quadrants of the input
C+M' signal and may be implemented by a diagonal filter with
2H of delay and having four unique coefficients and a
structure as shown in Fig. llb. The spatial frequency
response of QSF 1202 is shown in Figs. llc and lid, where the
negative (lower) peaks represent zero amplitude or stop bands
and the positive (upper) peaks represent pass bands. Figure
lle shows the horizontal frequency response or selectivity of
QSF 1202 , centered on 629 KHz . The bandwidth of the pass
region is approximately 1 MHz, providing 500 KHz bandwidth
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2~56'~44
for each sideband for maintaining good chroma response. Due
to the fixed nature of QSF 1202, it will be appreciated that
for those reproduced tracks (i.e. fields) from the encoded
recorded videotape in which the chrominance component of the
signal C+M' is in a quadrant of QSF 1202 with the positive
peaks (pass bands) , the filter will pass only the chrominance
component, while in the next track (field) only the motion
signal component will be passed. Accordingly, QSF passes the
motion component or the chroma component, depending on what
track, odd or even, is being decoded.
For each track (field/channel) reproduced by the
playback head 50, QSF 1202 will pass either the C* or the M'
component of the reproduced composite C+M* signal to its
output, depending on which component is located in the
filter's pass region during that track. The output of QSF
1202 is coupled to the subtrahend input of subtractor 1204 in
order to obtain the opposite signal component, M' or C*, by
differencing the filter output against the composite C+M*
signal applied at the minuend input of subtractor 1204. The
output of QSF 1202 (i.e. M* or C*) and the output of
subtractor 1204 (i.e. C' or M') are coupled to respective
inputs of a multiplexes (MUX) 1206 having a pair of inputs
and a pair of outputs which are switched at the f field rate of
Hz under control of a signal (e. g. field pulse or channel
25 1-channel 2 switching signal) which may be generated in known
manner by a conventional head switching circuit (not shown)
of the VCR associated with the playback head 50, or by
conventional channel switching circuitry associated with
playback preamplifier circuits. One output of MUX 1206
30 therefore will continuously provide only the separated
chrominance component C' for each track/channel reproduced,
while the other output of MUX 1206 will continuously provide
only the separated motion signal component M'.
The separated motion signal M' is coupled to the input
of an absolute value circuit (ABS) 1208 which may be
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~0~6i ~.4
conveniently implemented by a full wave rectifier. The
output of ABS 1208 is coupled to the input of a horizontal
lowpass filter (HLPF) 1210 which may be implemented with 15
taps at about 500 KHz . It will be appreciated that the order
of ABS 1208 and HLPF 1210 may be reversed if desired without
affecting the resultant signal. The recovered spread motion
signal M' output from HLPF 1210 is supplied to the motion
signal input of LUT 910 of the spatio-temporal post-filter
820 for controlling the motion-adaptive filtering of the
unfolded luminance signal L"~ as previously described.
The separated chrominance signal component C' output by
MUX 1206 may be D/A converted to an analog signal and then
processed by a conventional VHS chroma recovery circuit in
known manner to obtain the 3.58 MHz NTSC chroma component.
This may in fact be preferable from the viewpoint of chroma
phase control processing during picture search and still
modes, which is more complex if performed digitally.
However, the separated chrominance signal component C' output
by MUX 1206 may be also be digitally processed by a digital
implementation of a conventional VHS chroma recovery circuit
employing a modulator 1220 (e. g. a multiplier) supplied with
a 4.21 MHz four-phase carrier, whereby the 629 KHz chroma
component carrier is frequency up-converted to 3.58 MHz by
heterodyning, then passed through a 3.58 MHz bandpass filter
(BPF) 1222 to pass the 3.58 MHz chroma. The chrominance
component may be filtered further for removing residual up-
conversion carrier and modulation byproducts, and processed
for burst de-emphasis if desired. The recovered digital 3.58
MHz chrominance sub-carrier component signal may then be D/A
converted to an analog NTSC chroma component signal, supplied
to composite video signal generator 810 (which may include a
D/A conversion facility), or utilized in further processing.
Having now described the details of a video signal
system according to the present invention and its processes,
it will be appreciated that the invention is amenable to many
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2656'744
modifications and alternative implementations, for example,
application to a broadcast or other transmission or storage
medium rather than a recording medium, or implementation
according to different video signal conventions and formats,
or varying the respective transfer functions of the adaptive
de-emphasis and adaptive re-emphasis processing, and such
modifications are considered to fall within the scope of the
invention which is intended to be limited only by the
appended claims.
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