Note: Descriptions are shown in the official language in which they were submitted.
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RESONANT CAVITIES FOR NMR
The present invention relates to resonant cavities
for nuclear magnetic resonance (NMR) and more particularly to
resonant arrays for receiver and transmitter probes for use at
high frequencies.
Resonator arrays [Hayes, C., Edelstein, W., Schenk,
I., Muller, O. and Eash, M., J. Mag, Res. 63, 622-628, (1985)]
are becoming increasingly popular for receiver and transmitter
coil probes in nuclear magnetic resonance imaging and
spectroscopy. More so since ever higher magnetic fields and
therefore frequencies are being employed. The difficulty in
tuning standard multi-turn saddle coils and short solenoids
makes alternative slow wave structures and resonator arrays
more attractive. We have considered several structures which
employ the resonator array principle, for example the petal
resonator [Mansfield, P., J. Phys. D., 21, 1643-4 (1988)]. In
considering these devices it is necessary to develop the
general theory of lumped parameter circuits as applied to these
systems. We have considered the theory from a general matrix
approach (Fisher, E.M. 1955 Electronic Engineering 27, 198-204,
Sander K.F. and Reed G.A.L. 1978 Transmission and Propagation
of Electromagnetic Waves, C.U.P. Cambridge). A perturbation
approach for the analysis of non-symmetric 'bird cage'
resonators has recently been published [Tropp, J., J. Mag. Res.
82, 52-62 (1989)].
The present invention provides a resonant array for
NMR for use at high frequencies comprising two identical end
structures and a plurality of continuous electrically
conductive rods electrically connected to the identical end
structures the rods thereby supporting the end structures a
predetermined distance apart characterised in that the rods and
end structures comprise a plurality of ~ or T electrical
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circuit sections and are so shaped and dimensioned as to
substantially contribute to the electrical characteristics of
the array as a whole and to enable it to support a standing
wave.
The present invention also provides a modified
resonant array in which a first and second rod are combined in
a flat rectangular sheet and in which a third and fourth rod
are combined in an electrically conductive wire surrounding the
sheet.
In accordance with the present invention, there is
provided a resonant array for NMR use at high frequencies, said
array comprising: two similar end structures shaped and
dimensioned to define high frequency electrically resonant
cavities; a plurality of electrically conductive rods connected
between said end structures and supporting said end structures
at a predetermined distance apart to form an array including
said end structures and said electrically conductive rods; the
end structures being shaped and dimensioned and the connecting
rods being positioned in a way that is effective to cause the
array to define a plurality of electrical circuit sections
selected from the ~ and T classes of electrical circuits and is
effective to support a standing wave.
Embodiments of the present invention will now be
described by way of example with reference to the accompanying
drawings in which:-
Figure 1 shows sections of a lumped parameter
transmission line as (a) a ~-section and (b) a T-section;
Figure 2 shows a graph showing the allowed frequency
characteristics for a simple low pass filter section;
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Figure 3 shows a tuneable NMR cavity comprising a
cluster of slotted loop resonators, of effective cavity length
1, according to the present invention;
Figure 4a shows one end plate of the tuneable NMR
cavity of Figure 2;
Figure 4b shows an alternative end plate design with
greater axial access;
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Figure 4c shows a drive circuit showing quadrature and
balanced arrangements;
Figure 4d shows a flux guide sleeve or end ring thimble for
use with the alternative end plate design of figure 4b.
Figure 5 shows a plot of inductance against diameter for an
individual slotted loop resonator;
Figure 6a shows an equivalent circuit of the resonant
cavity structure, points A and B being joined;
Figure 6b shows a plot of frequency characteristics for one
section of the cavity resonator of Figure 6a;
Figure 7 shows a linear regression plot of f2 against 1/1
giving asymptotic values of C,,~o ;
Figure 8 shows a plot of frequency, f, against length for
the quadrature and balanced driving modes of the cavity
resonator;
Figure 9 shows a section of an end plate for a lower
frequency resonator design leaving larger central access;
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Figure l0a shows a schematic arrangement in plan view of
slotted loop resonators with 2 layers to increase loop
inductance;
Figure lOb shows a section in side elevation;
Figure 11 shows a schematic arrangement for a 2 turn cavity
resonator;
Figure 12 shows a schematic of a section of an end plate
arrangement for a 2 turn cavity resonator;
Figure 13 shows an alternative arrangement in which the
loops are rotated out of the plate plane by 90°;
Figure 14 shows a schematic diagram of one end plate of a
high pass resonant cavity coil;
Figure 15a shows a modification of the cavity resonator of
Figure 14;
Figure 15b shows the equivalent circuit for the resonator
of Figure 15a;
Figure 16 shows a split or half resonator arrangement with
reflective conductive screen according to the present invention;
WO 91/02261 ~' I~ ~ ~ ~ PCT/GB90/01259
Figure 17a shows diagrammatically an end view of a wire
arrangement for a saddle coil, additional wires W, W' being
added for symmetry but carrying no current, i.e. at standing
wave nodes;
Figure 17b shows an equivalent split or half saddle
arrangement with reflective screen;
Figure 18a shows a sketch of a split or half saddle
arrangement with screen;
Figure 18b shows a strip coil with screen;
Figure 18c shows an equivalent circuit of a split saddle or
strip coil with drive.
THEORY
In the following we shall represent one section of our
resonant structure by a lumped parameter circuit in the form of
either an n-section or a T-section as indicated in Figure 1. We
assume that there is no interaction between sections, (see
hereinafter).
The transfer matrix A for the section, considered as part
of a longer transmission line, satisfies the propagation
equation
WO 91/02261 PCT/GB90/01259
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En En+1
- A (1)
In _In+1
~n = A~'n+1
in which En, En+1 etc. of equation 1 are the output and input
voltages and currents respectively, and ~n etc. represent the
column matrices.
For N identical sections it follows from equation 2 that
'f'o = ANN . ( 3 )
If wave propagation is sustained along the transmission
line then we can also write
~n = ~~n+1
where the matrix ~ represents the common losses and phase
changes per section. Combining equations 2 and 3 gives the
characteristic equation
A'I~ = uW ( 5 )
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The eigenvalues of equation 5 represent the sustainable
frequency and phase characteristics of the circuit. They may be
obtained by diagonalising A. The diagonal values are obtained
from the determinant
det (A - u) - ~ (6)
Since we are dealing with a passive network, det A = 1. The
eigenvalues of equation 5 are in general complex and since ulu2
- 1, may be written as
ul - ~Ao + [ (Ao/2)2 - 1]~ = e'y (7a)
~2 - ~F'o - [(Ao/2)2 - 1]~ _ e-y (7b)
where
Ao = Tr A (g)
in which Tr stands for the trace or diagonal sum. We shall see
later that'y is the transmission line propagation constant. This
is in general complex and is given by
y = tanh-1[(AO/2)2 - 1]~/(AO/2)
_ cz + iJ3.
When oc is zero corresponding to a loss-less line we see
from equations (7 - 9) that through the invariance of the trace
to the basis functions 'hn
WO 91/02261 ~ ~ ~ ~ ~ ~ PCT/GB90/01259
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Tr A = 2 cos ~3 , . ( 10 )
Let the colinearity transformation matrix which
diagonalises A be S. Then we may write
S_lAS = I~ ( 11 )
which may be inverted to give
A = SOS-1 (12)
Circuit Details
Before proceeding it is necessary to consider the transfer
matrix for the particular circuit section. For the n-section,
Figure la
1 + (Z2/Z1) Z2 (13)
A = (2/Z1) + (Z2/Z12) 1 + (Z2/Z1)
while for the T-section, Figure lb we have
1 + (Z1/Z2) 2Z1 + (Zi /Z2) (14)
A =
1/Z2 1+ (Z1/Z2)
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Provided All - A22, that is to say we have a symmetric
section, it is straightforward to show that the characteristic
impedence of the section Z~ is given by
2
ZO - A12/A21. (15)
Using the invariance of Tr A together with equations 7a, 7b
and 8, it follows that the transfer matrices, equations 13 and
14 may be written generally as
A = cosh ~ ZOsinhy (16)
(sinhy )/Z~ coshy
which may also be generated from the matrix S as follows
A = S eY S 1
a y (17)
where the matrices S, S-1 are given by
S = 1 -Z~/2 and S-1 - 1/2 Z~/2 (18)
2/Z~ 1/2 ~ -1/Z~ 1
Using S we see that
S_lANS = ey N _ eN~Y
e-y e-Ny (19)
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and inverting
AN=S eNy S-1 - cosh Ny NOsinh N'y
e- ( 1/ZO ) sinh N'~ cosh N~ ( 20 )
For a loss-less transmissionw line comprising N sections
corresponding to a wavelength P (P integer), equation 20 shows
that the phase relationship along the line satisfies the
relationship
N(3 = 2nMP ( 21 )
where j3 is the phase shift per section and M (integer) is the
resonant mode.
Using this formulation a single turn (P - 1) cyclic
transmission line has been designed which forms the basis of the
petal resonator [Mansfield, P., J. Phys. D., 21, 1643-4 (1988
)]. Other related structures are what we have called the chain
mail coil and the chain or necklace resonator useful for
studying restricted parts of the anatomy, for example, the neck.
In these designs, the current distribution in the
successive transmission line elements around a cylindrical
surface, follows a cosinusoidal or sinusoidal variation as a
function of the cylindrical azimuthal angle 8. For straight
wires lieing on the surface of a cylinder parallel to the
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cylindrical axis, a cosinusoidal current distribution will
produce a uniform magnetic field transverse to the cylindrical
axis. This will be the case if the straight wires form the
impedance elements Z1 of Figure la and the voltage around the
line follows a cosinusoidal variation about the drive point. In
this case the current flowing through the nth wire, I~ is given
as
Iwn - En~Zl. (22)
This current follows En whereas In, equations 1, 3 and 16,
varies as sin Nj3 for a loss-less line.
In NMR imaging applications, a coil structure which
produces a uniform transverse magnetic field is useful as a
transmitter coil and as a receiver coil. We also note that
multi-turn resonator structures are possible with P > 1.
Input Impedance
It is useful in designing cavity resonators to have an
expression for the input impedance of the device. This may be
obtained by initially considering the expression for an N
element transmission line equation 20 together with equations 1
and 3. Let the input and output voltage and current be V1, I1
and V2, I2 respectively. Then we obtain
WO 91/02261 2 0 6 4 7 6 3 . PCT/GB90/01259
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V1 - V2 cosh N'y + I2Z~ sinh Ny (23a)
and
I1 - (V2/Z~) sinh Ny + I2 cosh N'~. (23b)
The input impedance is therefore Zl - Vl/Il. Let the line
be terminated by Z2 - V2/I2. Substituting in above we obtain an
expression for Z1 for a discrete line which is similar to the
well known result for a continuously distributed line, i.e.
Z1 = Z [Z cosh N~ + Z sink Ny]
2 sin ~'L~ cosT-~ R~ . ( 24 )
For an open circuit line of wavelength P , Z2 - ~. In this
case
Z1 = Z~/tanh Ntf (25a)
which for a small argument Ny becomes
Zl '" Zp/~ ~ ( 25b)
It is worth pointing out that the impedance of an open
circuit P7~line is not changed if its output is connected to its
input. This means that all the resonator designs herein may be
either physically cyclic that is actually joined, head to tail,
or cut at a high impedance point. The cyclic boundary conditions
are identical in both cases. We also point out that the
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expression Z1 is developed for a n- or T- section as illustrated
in Figure 1. The actual circuits constructed are symmetrical
about the earth point, as discussed with reference to the high
frequency probe hereinafter.
Q of Cavity Resonator
The quality factor of the cavity may be found from equation
25 by substituting 'y = a + ij3 and noting that for a P'7~ line,
since Nj3 = 2nMP on resonance, a small shift s;', gives
tank N - Ncx + 2nsw/c.a ( 26 )
Using this approximation in equation 25a we note that it
produces a Lorentzian variation of Z1 versus 8 w with a line
width at half height given when
2Ncx = 4nsw/w = 2n/Q. ( 2~ )
At resonance the input res stance is
R = Z~/Na,. ( 28 )
Combining equations 27 and 28 we get for the Q factor
WO 91/02261 ~ Q ~ /~ "~ ~ ~ PCT/GB90/01259
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Q = nR/ZO
(29)
Simple resonator desiqns
Low Pass
I f Z 1 - 2 / j ~~~C and Z 2 - ( jc,rL + r ) the trans f er matrix f or
this n-section is given by
1 - W2LC/2 + jwrC/2 jc.>L + r ( 30 )
A = jcaC/2 1 - l.~ 2LC/2 + jcarC/2
From equations 10 and 30 we see that for r = 0
Tr A = 2 - (c.~/w0)2 = 2 cos J3 (31)
where
X02 - 1/LC. (32)
Plotting this transcendental equation gives for this low
pass filter section the allowed frequency response shown shaded
in Figure 2.
Combining equations 21 and 31 we see that the condition for
a sustained standing wave in an open circuit line or in a
resonant cyclic structure comprising line of fixed length looped
back on itself so that input and output terminals are joined, is
given by
(ta/wo)2 = 4 sin2(nMP/N) (33)
WO 91 /02261 ' ~ "~ 6 ~ PCT/G 890/01259
for 1 , MP ~ N/2. Solutions of equation 33 show that the modal
frequencies increase from the lowest mode M = 1 to the cut-off
mode M = N/2P.
We also find as expected that
cx = r{C/L)~/2 = r/2Z0 and j3 = ~/w0. (34)
In this case we obtain the relationship
R = 2Z0 /Nr. (35)
From equations 27 and 35 we also obtain the Q value
Q = 2nZ0/Nr (36)
i.e. the Q is determined by the characteristics of a single
section resistance.
High Pass Line
In this arrangement Z 1 - 2 ( j L + r ) and Z 2 - 1 / jwC . With
these parameters we find for r = 0.
(WO%2) - 4 sin2(nMP/N). ~ (37)
WO 91 /0226 ~ ~ ~ ~ r~
PCT/GB90/01259
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As in the case for the low pass section ZO = (L/C)~ at high
frequencies. Also when loss is included
= 4[r/2Zo]sin2(nMP/N) and j3 = -~p/W. (38a)
For small J3 this leads to a Q factor of
Q=NZO/2nr.
(38b)
In the "birdcage" resonator design (Hayes et al, 1985),
regarded as a series of n-sections Z2 is inductive while Z1
comprises an inductor and a capacitor in series. This has a
constructional disadvantage in so far as curcuits with many
elements require many bulky, high voltage capacitors for
individual tuning.
Hiqh Frequency Probe
Miniaturising RF probes for high frequency operation can be
problematical because dimensions are often limited by the use of
lumped element components.
In this work, a new tuneable RF cavity design is
introduced, which has application at RF frequencies around 500
MHz or above, but could be scaled and adapted for lower
frequencies.
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The RF cavity design was inspired by the microwave
magnetron cavity resonator. It is similar to the birdcage
resonator (Hayes et al, 1985), but has the advantage that it may
be accurately constructed from machined solid copper and rods.
The theoretical basis of our approach is as presented
hereinbefore.
The resonator consists of two end plates 10, each having a
symmetric cluster of slotted loop resonators 20 joined by a
number of rod inductors 30, Figure 3. The plan view of one end
plate is shown in Figure 4a. In a particular embodiment each
slotted loop resonator 20 is 10 mm in diameter with an
inductance of 11 nH and a gap 40 corresponding to a capacitance
of 12 pF giving a resonant frequency of 438 MHz . Leadless chip
capacitors may also be used to increase gap capacitance. An
alternative end ring arrangement 10' is shown in Figure 4b. This
provides a larger axial access.
In the arrangement of figure 4b the end ring comprises an
annulus 10' which has loop resonators 20' with slots or gaps
40'. The annulus has an inner surface 41 and an outer surface
42 the circular apertures 20' being formed therebetween. Rods
30' are joined to one end surface 43 at positions intermediate
the resonators 20'. A second series of slots 45 is formed in
the second end surface 44 towards the first end surface to
accommodate a flux guide or sleeve as shown in figure 4(d).
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PCT/G B90/01259
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To prevent inductive coupling of the slotted loop
resonators, a flux guide sleeve or end ring thimble 50 may be
fitted over each end ring 10. Such a sleeve arrangement is
shown in Figure 4d. This comprises two short coaxial conducting
cylinders 11, 12, the inner cylinder 12 being held centrally
within the outer cylinder by 'a series of conductive metallic
spacers or fins 13. The disposition of the fins 13 is arranged
to engage in the slots 14 between the slotted loop resonators of
Figure 4b. The slots 14 must be insulated so that neither the
fins 13 nor the flux guide rings 11, 12 touch the resonator 10,
Figure 4b. A suitable insulating material could be an
insulating tape or a lacquer. With this or a similar
arrangement, magnetic flux from one slotted loop resonator 20~is
prevented from coupling with other slotted loop resonators.
Provided the annular space between the end rings 11, 12 of the
thimble 50 is large enough, a flux return path is provided for
each slotted loop resonator 20, thereby maintaining its self-
inductance close to the unscreened value.
The characteristics of an individual slotted loop resonator
were assessed empirically by measuring the inductance of a
single loop, formed by drilling a hole in a copper block.
Inductance versus the hole diameter is plotted in Figure 5. When
assembled, the complete coil resonates over a range of
frequencies around 500 MHz. Tuning in this design may be done
WO 91/02261 ~ ~ ~ ~3 r~ ~ ~ PCT/GB90/01259
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manually by sliding one end plate along the rod inductors. The
equivalent circuit for the resonator coil is shown in Figure 6a
with A and B joined.
The allowed frequency response for one section of this
circuit (rl - r2 - 0) is shown shaded in Figure 6b. the stop
bandwidth frequency is set by the parallel resonant elements,
see below.
From the circuit parameters of Figure 6a and the above
analysis with rl - r2 - 0 it is found that the resonant angular
frequency of the cavity,Ct~, (when correctly driven) is given by
2 = ca 22 1 + L2 / 4 L1 Sin2 ( nM/N ) ( 3 9 )
For the principal mode, M = 1 and for N - 6 sections this
reduces to
2 = c~2 1 + L2/L1 (40a)
where 122 - 1/L2C = 4n2f2~ and where the rod inductance L1 - kl
in which k is a constant and 1 is the length of the rods . With
these substitutions, Eq. (40a) becomes
f2 = f2 {1 + (L2/kl)~ ~ (40b)
WO 91 /02261
PCT/GB90/01259
in which f is the:.:cavity frequency. There will in general be
mutual inductance between the rods, but this is small and is
therefore ignored in this application.
From equations 14 and 15 we find that for W/c,.~2 > 1, ZO =''
(L1/C)~. If we assume that resistive losses in the cavity arise
essentially in the rods, a reasonable assumption since the end
plates are machined from solid copper, r2 - 0 in which case for
ca/W 2 > 1
oc = [rl/2ZOl(~2/w)2~ (41a)
If however rl - 0 and the losses arise in the slotted loop
resonators, we find that
cx = [r2/2ZOl(L1/L2)~1 + (L1/L2)~(~2/W)2~ (41b)
In both cases j3 = W1/G~
Results
Measurements of resonant frequency for the principal mode
(M = 1) were taken for different cavity lengths and it was found
that the resonant behaviour of the coil was in accordance with
theory when driven correctly, thus supporting our initial
assumption that mutual inductive effects between sections may be
WO 91/02261 ~ ;~ PCT/GB90/01259
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ignored in this case. The coil is driven from one end in
balanced mode, as outlined below. With N slotted loop resonators
and N rod inductors, N/2 resonant modes are observed.
Circuit Drive
The cavity may be most easily driven from one end plate,
Figure 4a, across AB or A'B' through the split capacitor
arrangement, Figure 4c. The centre point is earthed and the
coaxial drive connected to A or alternatively C. (Alternative
drive and connection points are indicated with primes). To
ensure proper balance, a single capacitor of ~ Cd should be
placed across the corresponding points at the other end of the
cavity. The square of the cavity frequency, f, is plotted versus
1/1 in Figure 7 according to the linear regression, equation
40b. the intercept at the origin gives an experimental value for
the base frequency f2 = 424 MHz.
The data are also plotted as f versus 1 in Figure 8. Using
the measured parameters enables the theoretical curve (solid
line) to be drawn.
It is also noted that when the input drive is connected to
point A, Figure 4a, it forms an anti-node or high voltage point.
Apart from an RF carrier phase shift, the rod currents follow
the end plate voltage, see equation 22. The drive point
therefore corresponds to a rod current anti-node. Point B has
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the opposite RF phase. The additional capacitance Cd/2 is
distributed around a.l'1 slotted loop resonators to produce an
additional slotted loop capacitance of Cd/2N. This will
therefore affect the base frequency f2 in equation 40b and
Figure 7. This behaviour is confirmed experimentally. This may
therefore be used as a fine cavity tuning adjustment.
Alternatively, the cavity length 1 may be reduced to restore the
desired operating frequency.
At 500 MHz, the Q of the coil was ~ 160 and the cavity
length was 4.0 cm. From equation 40b we deduce that L1 - 7.48 nH
producing a characteristic impedance ZO - 22.3 S2. From equation
29 the input resistance R = 1135 S2. The cavity was matched to 50
S2 with a 6.8 pF variable matching capacitor Cm of Figure 4c with
Cd - 3 pF. Allowing for Cd, the theoretical slotted loop
resonance frequency is 429 MHz, in agreement with the measured
value from Figure 7.
We stress that the input impedance Z1 is correct for end
plate drive arrangements between points A and C but will be 4
times greater in balanced drive across A and B. For the slotted
loop cavity resonator the drive circuit, Figure 4c, affects f2
in the expected manner by virtue of the additional distributed
capacitance, Cd/2N, introduced in each slotted loop resonator.
To this extent the drive capacitance will change Z1 in a
predictable fashion by virtue of its effect on Z0, equation 15.
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Alternative Cavity Desiqns
In an alternative low frequency cavity design, a section of
the end plate of which is shown in Figure 9, the slotted loop
capacitance is increased by adding a segmented guard ring 21.
Alternatively, actual small capacitors 22 may be inserted
between the loop slots.
In a further arrangement, the loop inductance may be
increased by stacking machined end plates 10, 10' slightly
displaced as in the schematic of Figure 10. In this arrangement,
successive layers of inductance must be coupled effectively in
series as indicated. In order that successive loop displacements
do not block the hole, the second 10' and subsequent layers of
loops must be suitably elongated.
All the arrangements discussed so far are "single turn"
cavities in which a standing wave around the transmission line
structure obeys the phase relationship NJ3 = 2nMP where J3 is the
phase shift per section, N the number of sections and M and P
are integer. For a ~ line the operating frequency, P=1. The
principal mode M - 1 means that there is just 2 n radians of
phase shift. However, if P = 2 at the same frequency, then this
would imply a two turn structure as sketched in Figure 11. Such
an arrangement would produce double the RF field per unit
current at the coil centre and would thus offer a means of
effectively increasing the resonator impedence through mutual
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coupling of the inductive elements. A sketch of part of an end
plate arrangement for a two turn cavity resonator is shown in
Figure 12.
All the inductive loops so far are either flat and in the
plane of the end plate or within an end ring arrangement.
However in an alternative arrangement of loops they may be
rotated out of the plate plane by 90°, Figure 13. In this
arrangement, loop magnetic flux forms a toroidal shape, which is
effectively contained in a torus for a large number of elements.
The loops must be sufficiently spaced to give effectively no
mutual inductance. Alternatively, flux guard plates may be
introduced to separate the loops magnetically.
In a further modification of the cavity resonator the
equivalent n circuit per section Figure la comprises a capacitor
in series with an inductor for Z2 and inductor for Z1. This
gives a high pass transmission line section, in this case the
cavity end-plates could be made as a segmented ring joined by
rod inductors. One end-plate is sketched in Figure 14. Each
segment 60 is machined from solid block and the spaces ?0 form a
ring of series capacitors. The segments may be suitably spaced
with a dielectric material.
The above arrangement may be further modified as in Figure
15a. Here the block inductance may be increased by forming a
slotted loop. The equivalent circuit for this arrangement is
WO 91/02261 PCT/GB90/01259
shown in Figure 15b. When C is small, by having a wide slot
(i.e. C = 0), the equivalent circuit reduces to that for Figure
14.
RF Screening
In some situations it may be desirable to surround the
cavity resonator with an RF screening can thereby making the
resonator characteristics independent of surrounding metal
structures. The effect of the can will be to reduce the rod
inductance and also introduce a stray capacitance Cs which
shunts each rod. The shunt capacitance may be readily
incorporated in the theory. The net effect is to increase the
operating frequency f for a given cavity length. The magnitude
of the effect depends on the proximity of the screen. For a
screen/ cavity diameter ratio of 1.25 the frequency change is
around 15$. This may be compensated by either increasing the
resonator length or by increasing Cd which lowers f2 as outlined
in the Circuit Drive description.
Slit Resonator Designs
The resonator coil designs described so far are all cage-
like arrangements which completely surround the specimen around
the cylindrical axis. However, there are a number of situations
where it is more convenient to have a split coil system
WO 91/02261 PCT/GB90/01259
'~~ ~'
26
providing easy access for'the specimen. Such an arrangement is
desirable in the case of very small specimens and also in
clinical imaging for easy access of limbs, torso, head etc.
Such a new split resonator coil arrangement is sketched in
Figure 16. Here, as an example, we take a half cavity array 100
only which is simply placed close to, but not necessarily
touching, a large earthed conducting metal sheet 102. Two */4
standing waves are generated about the drive point provided all
four corners of the half cage are earthed. Because of the
particular symmetry of the resonator wires, a magnetic field
parallel to the conductor surface will be doubled and rendered
uniform due to the induced image currents in the sheet.
Magnetically and electrically, the arrangement will behave as
though it were a single cylindrical resonator as described
earlier, since the boundary conditions for an open circuit /2
line are the same as for an open circuit or cyclic line.
Practically, however, it is possible to lift the coil in the
manner of a cake or cheese dish cover allowing straightforward
specimen access. A half-cage coil design without the conducting
plate has been described by Ballon, D., Graham, M.C., Devitt,
B.L., Koutcher, J.A. Proc., Soc., Mag. Res. in Med. 8th Annual
meeting, Amsterdam 2,951 (1989), but is less valuable because of
poor RF homogeneity and a lower signal response.
WO 91/02261 ~ ~" PCT/GB90/01259
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27
The arrangement sketched, Figure 16 is semicircular in
cross-section. But in general it is possible to generate uniform
transverse magnetic fields with a semi-elliptical structure in
which the elliptical axes are 2a and 2b. Such an arrangement
could be extremely convenient as a head coil, or as a leg or
knee coil. The corners A,B may be joined by an inner return
wire 104 providing current path continuity around the end plate.
Corners P and Q should be similarly joined by a wire 106.
If the number of wires in a circular cage is reduced to
six, the coil structure becomes effectively a saddle coil
arrangement, Figure 17a, where wires w,wl carry no current, i.e.
are at positions corresponding to wave nodes. If a reflective
screen is introduced, as in Figure 17b, wires 1 and 4 form an
earth while wires 2 and 3 may be connected since their currents
are in phase. The arrangement is sketched in Figure 18a. To
force wires 1 and 4 in the nodal plane, the circuit must be
driven as indicated. If the device is not actually touching
the screen plate, current return paths may be provided between
points A and B and points P and Q in the form of inner return
wires 104', 106' as sketched. In a further modification the
pair of wires are replaced by a single strip of conductor 110
Figure 18b and rods 2 and 3 are combined together in a flat
conductive sheet 112. In either case, this arrangement offers a
demountable coil system useful for flat samples in either
microscopy or whole body imaging.
WO 91/02261 PCT/GB90/01259
28
The presence of the conducting plate serves to symmetrize
the arrangement, thereby increasing the field and at the same
time making it more uniform. The equivalent circuit and drive
arrangement are shown in Figure 18c.
The alternative end plate design of Figure 4b may be halved
across its diameter to produce a split or half resonator design
replacing the two half end plates shown in Figure 16. To
obviate mutual inductance between loops, the end ring thimble
Figure 4d may also be halved to fit the half resonator end
rings.
In the embodiment of Figures 16 and 18 the conductive sheet
102 is preferably not a continuous sheet but may comprise a
plurality of strips as indicated by dotted lines 102' in Figure
16. This is in order to satisfy the boundary conditions for RF
currents but to block other induced currents at lower
frequencies which would otherwise be caused by the switched
gradients used in NMR.
The strips may be formed by commencing with a continuous
sheet and slitting it at appropriate distances. Alternative
arrangements comprise cutting the sheet into suitably shaped
flat loops which follow the induced RF current contours in an
otherwise continuous conductive sheet.
WO 91/02261 PCT/GB90/01259
~~~'~'
29
In a further modification, the passive conductive sheet is
replaced by an actively driven flat wire array provided with
current to simulate the induced. screening currents in a flat
passive conductive sheet.
Using the matrix approach we have designed a cavity
resonator type NMR coil operating at 500 MHz. The analysis
developed ignores mutual inductive effects between transmission
line sections. Experimental results obtained with the resonator
coil confirm the theoretical expectations for the fundamental
mode. The general frequency characteristics of the higher
resonant modes are also reasonably well described by the theory,
although there are differences between the observed relative
frequencies and those predicted by the theory. These could well
be ascribable to the ignored mutual inductive effects. However,
in NMR applications we are generally only interested in the
fundamental mode since the higher order modes in these resonant
structures produce spatially inhomogeneous RF fields. A
symmetrical balanced drive arrangement is found to give best
performance of the circuit when applied from one end of the
cavity resonator.
The idea of a split resonator design is introduced in which
half a resonator array is placed close to but not touching a
flat conducting plate. Since the split coil cross-section may be
WO 91/02261 PCT/GB90/01259
semi-circular or semi-elliptical and is not fixed to the plate,
the whole assembly is demountable thereby allowing easy access
for limbs, head or whole body imaging.
A variant of the split coil design is also described which
corresponds to a half saddle arrangement. This may be further
modified to produce a strip coil in proximity to an isolated
conducting plate.
