Note: Descriptions are shown in the official language in which they were submitted.
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SYSTEM AND METHOD FOR RADAR-VISION
FOR VEHICLES IN TRAFFIC
BACKGROUND OF THE INVENTION
1. Field of the Invention
The present invention is directed generally to the field of radar, and in
particular to
a radar system with improved interference characteristics. More particularly,
the radar
system belongs to the category known as "low probability of interception - iow
probability of exploitation" (LPI-LPE) radar. More particularly still, the
preferred
field of application of the present system is in the field of vehicular
traffic on the
ground and on roads, where fog and the like conditions obstruct normal vision.
But
it is, of course, also applicable to other situations; for example, for boat
traffic in
harbours, rivers and canals. The system is based on phased-array, monostatic
limited
scan volumetric radar. The intended radar range is between zero and a few
hundred
meters.
2. Prior Art of the Invention
Radar systems using phased-array antennae are well known, for example as in
systems
CA 02135215 2005-03-11
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known as "side looking airborne radar" (SLAB), where carefully shaped radar
beams
permit imaging of the ground underneath an air~~raft in flight. In such and
similar
systems, interference from other radars is not a problem. And while such
systems
could, in principle, be used to permit "vision less" driving in normal
vehicular traffic
on roads on the ground, in practice this would only be possible where few
vehicles
~e thus equipped eadi having v~ll sepma~ad radar fi~ As soon as indeterminate
or
significant numbers of vehicles are involved the problem of interference
becomes
intractable.
The known closest prior art to the present invention is disclosed in European
Patent
Application No. 92100969.2 published as No. 0 501 135 A2 on September 2, 1992
entitled "Broad-band mobile radio link for high-volume transmission in an
environment
with multiple reflectors".
In the above referenced European and United States patent application, which
are
particularly suitable for mobile radio/data links, the problems of
interference are
mitigated by providing two orthogonalities between different vehicle signals.
The first orthogonality being that of code-divis ion multiple access (CDMA),
and the
second being a small frequency separation. These techniques improve radio/data
,,~, 2i.3~~15
3
communications links under high-density usage conditions, but would not be
sufficient
to permit safe and reliable "radar-vision" to drivers in road traffic, or the
like
applications.
A further patent of interest is United States Patent No. 5,031,193 granted
July 9, 1991
to Frederick G. ATKINSON et al, and entitled °Method and apparatus for
diversity
reception of time-dispersed signals". The patent teaches as follows:
A method and apparatus for diversity reception in a communication
system wherein at least a dual branch receiver is provided with a stored
replica of expected reference information that is correlated with the
received time-dispersed signals to obtain an estimate of the transmission
channel's impulse response as seen by each branch, and determine,
among other things, phase error between the branch local oscillators
and the time-dispersed signals. Matched filters are constructed which
then coherently align the time-dispersed signals from each branch with
that branch's local oscillator, also constituting the first part of the
equalization. The diversity processing stage may perform bit by bit
selection on the re-aligned signals, maximal ratio combining of the re-
aligned signals, or equal gain combining of the re-aligned signals,
following each by a sequence estimation which uses similarly selected
or combined channel distortion compensation parameters to complete
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the equalization process on the new signal. In digital modulated carrier
systems, providing expected reference information eliminates the need
for carrier recovery feedback for each branch while performing part of
the equalization process.
Thus, this United States patent stores a replica of expected reference
information and
correlates it with actually received information to provide an estimate of the
equalization necessary in the diversity receive channel to permit better
reception of
unexpected information.
An earlier United States Patent No. 4,291,410, granted September 22, 1981 to
Edgar
L. CAPLES et al, is entitled "Multipath diversity spread spectrum receiver" .
The
patent discloses a multipath diversity receiver utilizing decision directed
coherent
integration with post detection correlation techniques.
United States Patent Nos. 5,031,193 and 4,291,410 are incorporated herein by
reference, where permitted.
In a recent article in the "New Scientist" (15 October 1994, No. 1947) titled
"CARS
THAT DRIVE THEMSELVES" it is stated (page 38)"
"Controlling the car's speed is more difficult. The major challenge is
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building a sensor that can monitor the precise distance of the vehicle
ahead and its closing speed, over a range of 100 metres to less than 1
metre. Such a sensor must be able to detect everything in the lane
ahead while ignoring vehicles in other lanes. It must work accurately
in all weathers and be reasonably cheap to build. 'We thought the
aerospace industry might have all the answers, but even military radars
cannot, do all the things we need,' ... For the moment researchers
make do with hand-built radars, but they work only when the vehicle
ahead is a few metres away."
SUMMARY OF THE INVENTION
The present invention utilizes monostatic phased-array antennas to advantage.
In
addition to well known advantages (cf. S. Drabowitch et F. Gauthier "Antennes-
Reseauz Phasees: des principes aux Realisations", Revue Technique Thomson -
CSF,
Mars 1980), other advantages are:
- variable dwell time;
- flexible beam shaping for transmission, reception or both;
- use of leakage canceller correlation loop;
- use of limited scan with attendant significant cost
reduction (cf. J.M. Howell: Limited Scan Antennas,
6
IEEE AP-5, Inf. Symp. 1972); and
antenna can be made conformaI.
The phased-array antenna with the limited scan, narrow beam shape, provides an
additional signal orthogonality to curtail interference.
In a broad aspect of the present system, radar in a mobile vehicle is used in
combination with roads equipped with radar reflectors (e.g. Luneberg lenses),
which
mark road-limits, to provide radar-vision in conditions where natural vision
is
destructed without interference from other vehicles' radars.
In another aspect, interference from other radar equipped vehicles is
curtailed by the
implementation of three signal orthogonalities provided by:
(a) code-division multiple access (CDMA);
(b) small frequency separation (o fJ; and
(c) an angle-of arrival (A-o-A) defining very narrow beam phased-array
antenna.
In yet another aspect, the (vehicular) radar receivers utilizes autocorreladon
to detect
and identify the reflected waveform echo.
~~3~2~.~
Preferably, and where possible, the radar antenna is positioned at or near the
windshield of the vehicle. Further, a passive (or also active) rear antenna is
used to
enable computation of an estimate of distance and relative speed of rear
vehicles.
BRIEF DESCRIPTION OF THE DRAWINGS
The preferred embodiment of the present invention will now be described in
conjunction with the annexed drawings, in which:
Figure 1 is an example of a pseudorandom, direct sequence-code division
multiple
access (DS-CDMA) waveform generated in the radar system of the present
invention;
Figure 2 is an example of the intermediate frequency (IF) output of a segment
of the
received radar signal echo, which is input to a reception correlator in the
radar system;
Figure 3a depicts the output of the correlator, or matched filter, when the IF
signal
applied to it is an echo of the signal radiated by the particular radar
system;
Figure 3b is a photograph of an actual test result of the output shown in
Figure 3a;
Figure 4 is a block schematic of the radar system of the present invention;
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Figure 5 is a schematic depicting use of an equalizer to compensate the
leakage of the
transmitted radar signal of the radar receiver;
Figure 6 is a schematic depicting a preferred arrangement for leakage
compensation;
Figure 7 is a front elevation illustrating use of a conformal phased array
antenna at the
(front) windshield of the vehicle in the present invention;
Figure 8 is a front elevation illustrating use of a conformal phased array
antenna at the
rear of the vehicle in the present invention;
Figure 9 depicts a plan view of a road equipped with demarcating Luneberg
lenses to
reflect radar signals; and
Figure 10 is high-level flow chart illustrating operation of the radar system
of the
present invention.
DESCRIPTION OF THE PREFERRED EMBODIMENT
The preferred radar system of the present invention utilizes three
orthogonalities, as
mentioned earlier, in order to curtail interference from other vehicles' radar
systems.
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The first (time-waveform) orthogonality is provided by the computational
generation
of direct sequence (DS-CDMA) pseudo-random code sequences. An example of an
arbitrary code waveform of duration T is shown in Figure 1 of the drawings.
These
sequentially generated pseudorandom code waveforms should have: a duty cycle
close
to unity (greater than 0.95); constant power; and a nearly flat power density
spectrum
(PDS). The flat PDS assures, in reception, dme resolution between echoes close
to
the Heisenberg limit.
The time-waveform orthogonality means that out-of phase autocorrelation of an
arbitrary code waveform is negligible compared to the in-phase
autocorrelation; and
that cross-correlating with any phase, of any two members of the CDMA universe
is
also negligible compared to in-phase autocorrelation. In the frequency domain,
for
this family of CDMA code waveforms, the in-phase autocorrelation is obtainable
as
Parseval's integral. Thus, a shift in frequency equal to 1/T should yield zero
for in-
phase autocorrelation (Parseval's integral); for example, given a T - 100
microseconds, a shift in frequency (e f) by a mere 10 kHz would (still) yield
zero for
in-phase autocorreladon (thereby curtailing interference). In practice,
assuming T =
100 microseconds and a f = 500 kHz (at, say, a radar carrier frequency of 70
GHz),
the level of an interfering signal as given by
sin ~r . n f. T
~ . a f. T
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.....
would be less than -44 dB. This is adequate frequency domain orthogonality. Of
course, interference levels decrease as a f and T increase.
As the echo of the radar signal modulated by the particular pseudorandom
waveform
just transmitted is being received, it is (as usual) mixed down to an IF
frequency, and
5 input to a correletor. The IF signal has the form of a bipolar phase-shift
keyed
(BPSK) signal, as shown in Figure 2, and has 180 phase transitions where the
code
waveform has transitions. Thus if the waveform received is an echo of the one
transmitted, the correletor output would be an autocorrelation of the waveform
and,
in theory, be a signal as shown in Figure 3a, while an actual test result is
shown in
10 Figure 3b. All other waveforms received would yield a cross-correlation
signal,
whose maximum envelope amplitude would be less than that of the
autocorrelation
envelope, shown in Figure 3a as unity for reference. The correlator's output
is
sampled and identified as a legitimate echo only if the envelope maximum
exceeds a
preselected minimum value above the prevailing background noise level (which
is not
shown in Figure 3).
Figure 4 shows a block schematic of the radar system. It comprises a phased
array
antenna 10, a system computer 11, and a screen display 12. A clock 13
generates a
clock frequency at 200 mHz (giving a chip-time of 5 nanoseconds), which clocks
a
waveform generator 14 (which is a bank of programmable shift registers) to
produce
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a pseudorandom sequence, an example of which is shown in Figure 1. The
generation
of pseudorandom sequences is well-known, and many millions of orthogonal
waveforms can thus be produced.
The pseudorandom waveform generated is applied to a modulator 15 and modulates
a carrier frequency generated by a local oscillator {LOl) 16. The output of
the
modulator 15 is band-pass filtered in filter (F 1 ) 17 before being applied to
the phased
array 10. The returning (echo) waves, are superimposed on the non-compensated
leakage from the transmitted waves, and are applied to band-pass filter (F2)
and
amplifier 18, the output of which is applied to mixer 19 and is heterodyned
downward
to the IF frequency by means of local oscillator (L02) 20. The IF signal is
band-pass
filtered in filter (F3) 21 and applied to low noise amplifier 22 to compensate
for the
anticipated loss in correlator 23, the correlated output of which is applied
to
(sampling) detection and identification circuit 24, which is controlled by the
computer
11. The latter computes the distances travelled by expected echoes and
controls the
display i2 to show the echo reflecting objects in real-time. The computer 11
also
controls the incremental sweep of the narrow radar beam radiated by the phased
array
10 within the desired horizontal angle, (more than one array may be used to
cover
3600. Assuming a position of the phased array 10 antenna approximately 2
meters
behind the front bumper of the vehicle, the distances measured w.r.t. the
bumper
range from - 2 meters to a few hundred meters.
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The phased array 10 in Figure 4 comprises, as is known in the art, the
necessary
dividers, diplexers, and phase-shifters, which are controlled by local
microprocessors,
in order to permit the efficient forming of the requisite very narrow
beamwidth. But
in addition, in order to reduce the leakage via the diplexers from transmitter
to
receiver below -30 dB, equalizers are used as shown in Figure 5 within the
phased
array 10.
In Figure 6 is shown a preferred arrangement for compensating the leakage
coupling
from the transmit side to the receive side of a circulator 26 (diplexer). The
transmit
signal is applied to a directional coupler 27 before reaching the circulator
26 and a
small amount of the transmit signal is tapped by the directional coupler 27
and applied
to a complex multiplier 28, to the other input of which is applied the
integrated (by
1_ block)
s
output of the correlator 23. The output of the complex multiplier 28 is
applied to the
reference input of the correlator 23, as well as to the negative input of a
summer 29,
the positive input of which receives the radar echo as supplied by the
circulator 26
(plus the leaked coupling from the transmit side). The output of the summer 29
is
input to the correlator 23, which signal. The output of the summer 29 is also
used in
beam forming. Thus the transmit signal is tapped, and the level of the tapped
signal
13 adjusted by means of the complex multiplier 28 under control of the
computer 11
(once or twice per hour), and subtracted from the input to the summer 29 to
cancel
(in a least square error sense) the leakage through the circulator 26, which
least square
2~~~2~.~
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error cancellation is indicated when the correlator output falls below the set
correlation
detection threshold. The leakage coupling through the circulator 26, of
course,
adversely affects the signal-to-noise ratio (SNR), which is (at best) equal to
the
isolation of the circulator 26, when the echo experiences zero geometric
attenuation
(i.e. at zero distance). As is explained by the flow chart shown in Figure 10,
the
system adaptively adjusts the leakage cancellation by means of the complex
multiplier
28 until the output of the correlator 23 falls below threshold, at which point
the
multiplier 28 is frozen at its present setting. Thus the system is self
calibrating.
Figure 7 shows the preferred phased array antenna, which is shaped as a
rectangular
border of dimensions equal to or greater than fhe windshield of the vehicle.
The
antenna pattern will depend on the frequency allocated. At 4 mm wavelength (75
GHz
band), for example, the antenna will have a horizontal beamangle of no more
than 10
mR (milli-radians), and a vertical beamangle of about 25 mR. This would permit
sufficient resolution to distinguish a cyclist one meter removed beside a bus
at a
distance of over 50 meters ahead of the antenna.
Figure 8 shows a rear array which may be used passively to make an estimate
(for
example, using least square error methods) of distance and speed of an
approaching,
radar equipped, vehicle. The length of this lateral array would preferably be
greater
than 1.5 meters.
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Figure 9 depicts a road equipped with Luneberg lenses 3U, which are "tennis-
ball" like
objects mounted on guard-rail or road signs, marking its boundaries, which
would
appear as road boundaries on the display screen 12 positioned in front of the
driver
of the vehicle. The screen 12 may be built in the windshield of the vehicle as
an LCD
display screen.
