Note: Descriptions are shown in the official language in which they were submitted.
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System and Device for, and Method of, Processing
Baseband Signals to Combat ISI and Non-linearities in
a Communication System
Cross-Reference to Related Applications
This application is related to the following U.S. patents
all of which are owned by the same assignee as the assignee of
this application:
U.S. Pat. No. 5,818,879, "Device, System and Method for
Spectrally Shaping Transmitted Data Signals", to Vedat Eyuboglu
and Pierre Humblet, filed on even date herewith;
U.S. Pat. No. 5,875,229, "System and Device for, and
Method of, Detecting, Characterizing, and Mitigating Deterministic
Distortion in a Communications Network", to Vedat Eyuboglu and
Pierre Humblet, filed on even date herewith.
Background
1. Field of the Invention
The invention relates generally to communication systems
and, more particularly, to high-speed modem communications
over a telephone network, possibly having an analog local loop.
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2. Discussion of Related Art
There is an increasing demand for data communications
and, in particular, for communication systems with
increasingly higher transmission rates. With the advent of the
Internet and multimedia, this demand is not expected to wane
any time soon.
To date, communication systems have evolved with
attempts to meet these demands. For example, one of the more
popular communication paradigms uses modems connected to
local loops of a conventional telephone network. These
conventional systems are largely desirable because of their
leverage of existing telephone network infrastructure. In
short, a user needs to make only a relatively small investment
for a modem and has to pay relatively modest line charges. To
meet user demand for higher transmission rates, modem-
communication standards have evolved with each generation
including capabilities to support higher transmission rates.
Unfortunately, the rate of growth of the transmission
rates of modems has slowed as transmission rates approach
the information theoretic limits of the telephone channel.
Consequently, users who want higher transmission rates are
forced to use alternative communication networks, such as
ISDN, in their homes or offices, rather than the conventional
public switched telephone network (PSTN) with analog local
loops. Though these alternative arrangements provide higher
transmission rates, the equipment and line charges are high.
There is, therefore, a need in the art for a device and
system for, and method of, communicating at higher
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information rates on conventional telephone networks having
conventional analog local loops.
Summary
The invention includes a method of, and system and
device for, forming an actual alphabet of symbols to be used in
a communication system. In this fashion, the communication
system may use the actual alphabet to detect symbols rather
than use a proscribed alphabet. To do this, the invention
IO causes the communication system to transmit a predefined
sequence of symbols. These symbols are then received and
processed to form an actual alphabet of estimate symbols of
the transmitted symbol. The actual alphabet may deviate from
the proscribed alphabet. Nonetheless, the actual alphabet is
the one used in the communication system, and the forming of
an actual alphabet may be exploited to decode the transmitted
symbols to prevent known types of erroneous symbol
interpretations.
Among other things, the invention allows information to
be transmitted on a conventional local loop at rates higher
than previously considered as the theoretical limits. An
exemplary embodiment includes an analog adapter that
responds to baseband fine interface signals transmitted on to a
channel and produces an estimate of the most-probably-
transmitted line interface signal therefrom. The decoder
includes an alphabet of symbols, stored in a storage medium.
Each symbol is an estimate of a corresponding line interface
signal from the set of fine interface signals transmittable on
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to the channel. One embodiment uses a subset of the
potentially transmittable symbols to form a signal
constellation of the baseband line interface signals actually
transmitted on the channel. The analog adapter further
includes a decoder that cooperates with the alphabet and
responds to the received line interface signal to output the
estimate of the most-probably-transmitted symbols line interface.
In this fashion, the line interface can produce !ow
fidelity versions of signals specified by a predefined
companding algorithm and . the analog adapter will reduce the
likelihood of erroneous predictions possible if the alphabet
held high fidelity versions of the signals.
An exemplary embodiment includes mechanisms for
equalizing sampled versions of the received line interface
1 S signal in which the mechanisms are adaptable to an error
signal indicative of the accuracy of the above estimates.
In this fashion, the analog adapter may accurately
equalize signals to combat ISi and other forms of noise, while
also operating in an environment with low fidelity line
interface signals.
According to another aspect of the invention a method of compensating for
accumulated noise during transmission over a public switched telephone network
(PSTN), the PSTN comprising a backbone network, a conventional line interface
coupled to the backbone network, an analog adapter connected by a local loop
to
the line interface, the method is provided. The method comprises the steps of:
(a)
receiving, at the line interface, a sequence of octets representing an
approximation
of a binary information contained in octets originally transmitted over the
backbone
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4A
network and performing inverse quantization for outputting discrete baseband-
modulated
signals s(t) according to the equation: s(t) _ ~ a(Yn) ~g(t - nT~
n
where a(Y ) is a transformation of an octetY obtained with a non-uniform
n n
quantization rule at the time "n", "T" is the sampling interval, and "g(t)" is
the interpolation
function, (b) transmitting the discrete baseband-modulated signals over the
local loop,
and (c) at the analog adapter, sampling the discrete baseband-modulated
signals using a
one-dimensional signal constellation to create a sampled version of the
discrete
baseband-modulated signals distorted by the accumulated noise, and equalizing
the
sampled version to generate an equalized version of digital signals
substantially identical
to the octets originally transmitted.
According to another aspect of the invention a communication system for
compensating accumulated noise during transmission over a public switched
telephone
network (PSTN), the PSTN having a backbone network, the system is provided.
The
system comprises: a conventional line interface coupled to the backbone
network for
receiving a sequence of octets representing an approximation of a binary
information
contained in octets originally transmitted over the backbone network,
performing inverse
quantization for outputting discrete baseband-modulated signals s(t) according
to the
equation: s(t) _ ~ a(Y ) ~g(t - nT)
n
n
where a(Y ) is a transformation of an octetY obtained with a non-uniform
quantization
n n
rule at the time "n", "T" is the sampling interval, and "g(t)" is the
interpolation function and
transmitting the baseband-modulated signals over a local loop, and an analog
adapter
connected to the line interface through the local loop and including a
sampling section and a
detection-interpretation section, the sampling section comprising: means for
sampling the
baseband-modulated signals using a one-dimensional signal constellation to
create a
sampled version of the baseband-modulated signals distorted by the accumulated
noise,
and the detection-interpretation section comprising: means for equalizing the
sampled
version to generate an equalized version of digital signals substantially
identical to the
octets originally transmitted.
Brief Description of the Drawing
In the drawing,
Figure 1 shows a conventional telephone system having local loops;
Figure 2 is an architectural diagram of an exemplary embodiment of
the invention;
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Figures 3A-B are architectural diagrams showing a
conventional line interface, in part, a local loop, and a decoder
of an exemplary embodiment of the invention; and
Figures 4A-4B are architectural diagram of a decoder of
5 an exemplary embodiment of the invention.
Detailed Description
This invention allows binary information to be
transmitted on conventional telephone networks that include
conventional digital backbones, line interfaces, and analog
local loops at transmission rates higher than presently
achievable with existing modem standards such as V.34. This
is achieved by viewing the conventional network from a new
perspective, in which certain sources of "noise" which limit
the achievable bit rate are avoided with new processing
techniques in an analog adapter at a user site. The invention
improves the system's information capacity and concomitantly
achieves higher transmission rates without requiring costly
infrastructure, such as ISDN lines or the like, at the user site.
To better understand the invention, certain aspects of a
conventional telephone network are described. This is done to
explain the various sources and forms of "noise" that limit the
information capacity of a conventional arrangement and that
are addressed with the invention. Afterwards, the
architecture and operation of the invention are described,
followed by a description of the invention's mechanisms for
combating particular forms of "noise," in particular, nonlinear
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distortion by the line interface and intersymbol interference
(IS/).
A conventional telephone network 100 is shown in figure
1.
What are typically interpreted as analog signals enter and exit
the network 100 at "local loops" 140 and 150. Each signal on
loop 140 and 150 is received by a corresponding line interface
120 and 130, or local switch, and each line interface
communicates with another via a backbone digital network
110.
Under conventional operation, a signal 175 is sent to a
first site 170, which emits an analog signal, for example,
representative of a voice signal or a binary information, on the
local loop 140. The line interface 120 samples and quantizes
the analog signal and outputs an octet 125, representative of
the analog signal 140.
More specifically, the analog signal 140 is quantized
according to a known set of rules, or a companding algorithm,
such as p.-law or A-law, which specifies the quantization's
amplitude levels. The ~.-law and A-law quantization rules
involve unequally-spaced quantization steps, i.e., non-linear
quantization, that were chosen to map to the inherent
characteristics of speech. The quantized signal is then
encoded into octets 125.
The backbone 110 receives the octet 125 and, though not
shown, also receives octets from other sources, such as other
fine interfaces. Using known techniques, the backbone 110
merges octets from the various sources and transmits and
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routes the data to various line interfaces, e.g., 130. Modern
backbones transmit data at a rate of 64,000 bits per second
(8,000 octets per second). Eventually an octet 125', which is
similar but not necessarily identical to the original octet 125,
is transmitted to the line interface 130 corresponding to the
signal's destination site 160.
The line interface 130 essentially inverse quantizes and
further processes the received octet 125' to create on loop
150 an analog signal, which is an "estimate" of originally-
transmitted signal 140. Loop signal 150 is called an
"estimate" because information may have been lost in the
quantization and inverse quantization processes of the line
interfaces. Signal 150 is then transmitted to the site 160,
where it may be used to recreate a voice signal or a binary
sequence.
Information may analogously flow in the opposite
direction. Destination 160 provides analog signal on loop 150
to line interface 130. Line interface 130 samples and
quantizes the signal on loop 150 to provide a sequence of
octets 135 to backbone 110. Backbone 110 routes these octets
and provides a similar sequence of octets 135' to line
interface 120. Line interface 120 provides analog signal on
loop 140 to be received by the site 170.
When used for conventional data communications, as
opposed to voice communications, the sites 160 and 170 may
each include a modem for modulating and demodulating the
analog signals on the local loops 140 and 150. A conventional
modem at the site 170, for example, will receive a sequence of
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bits 175 from some form of an information source (e.g., a
server) and modulate the bits and transmit the modulated
signal, according to a communication standard, such as V.34.
The modulation technique may use the information contained in
code 175 to alter the amplitude and phase of the signal to be
sent on loop 140. The modulated signal is routed to the line
interface 120 where it is sampled and quantized, as outlined
above. Eventually a representative signal is received by the
other modem at site 160, where it may be demodulated and
transmitted to computer 180.
Systems following the above conventional arrangement
have achieved transmission rates of approximately 30 Kb/s,
the conventionally-accepted view of the telephone channel's
capacity. This accepted limit of capacity is dependent on the
"noise" in the system, and in particular, the quantization noise
of the line interfaces.
The invention attains higher transmission rates yet
operates in arrangements having conventional analog local
loops, unlike the ISDN and similar approaches, outlined above.
In short, the invention is able to attain these advantages by
considering the conventional network from a new perspective.
Under this new paradigm, the invention reconsiders, and where
appropriate combats with new processing techniques, the
various forms of "noise" that limit the information capacity.
More specifically, the invention treats a signal s (t ) on
the local loop 150 as a discrete baseband signal and the
inverse quantization process in line interface 130 as a
baseband modulation that yields the baseband line interface
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signal s (t ). The modulation technique akin to PAM in that a
signal's amplitude is modulated but different than PAM in that
the amplitudes of adjacent signal points differ non-linearly.
The signal s (t ) is in the form
s (t ) = E~ a (v~)9 (t - nT) ( 1 )
In equation (1 ), v~ represents the octets 125' received from
the digital backbone network 110; a (v~) represents the
transformation of that signal 125' according to the relevant
quantization rules, e.g., ~,-law; T equals the sampling interval
of the system, e.g., 125 ~.s; and g (t ) is an interpolation
function, which is bandlimited to approximately 4000 Hz.
The new perspective yields powerful results the most
important of which is that, unlike conventional systems,
embodiments of the invention are not limited in their capacity
to carry information by the quantization noise inherent in the
line interfaces. An exemplary embodiment attains 56 Kb/s.
To better understand the new paradigm, refer to system
200 shown in figure 2. In system 200, backbone 110, line
interface 130, and computer 180 remain unchanged from the
conventional components, outlined above. A first site 270,
such as an Internet server site, communicates with a digital
adapter 220, or digital modem, by sending signals over a high
speed link 240. The digital adapter 220 sends a sequence of
octets 225 to backbone 110. Analogously to that described
above, backbone 110 sends a similar sequence of octets 225' to
conventional line interface 130. Line interface 130 then
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inverse quantizes octets 225' and transmits the baseband-
modulated, line interface signals, outlined above, on loop 250.
Analog adapter 260 receives the baseband signal and may, in
turn, possibly equalize and sample the baseband signal, detect
S the binary information in the demodulated signal, and sends the
results to computer 180. A reverse path from analog adapter
260 to digital adapter 220 may be constructed using
conventional modem techniques, for example, using V.34
technology.
10 In an exemplary embodiment, useful for description,
signal 240 represents a sequence of bits. These bits are
encoded in digital adapter 220 into a sequence of octets 225
which travel to the line interface 130 with minimal alteration.
At the line interface 130, the received octets 225' are used to
construct an analog baseband modulated signal on loop 250
according to equation (1 ) and as specified by the relevant ~.-
law or A-law rules. This latter step, from the perspective of
the invention, is now considered as baseband modulation, or a
variant of PAM, in which the signal constellation corresponds
to a subset of the quantization levels of the ~.-law or A-law
rules (more below about subset}. The analog baseband signal is
received by the analog adapter 260, which then samples the
received baseband signal at the symbol rate and, possibly after
equalization, detects the binary information in the sampled
signal with the results being sent to computer 180, for
example. Among other things, the exemplary arrangement 200,
unlike the conventional arrangement 100, takes advantage of
the fact that there is no analog local loop on one side of the
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connection and concomitantly avoids quantization noise as a
limiting factor to the system's transmission capacity.
Under the new paradigm of figure 2, the system 200 is
theoretically capable of transmitting data at rates of 64,000
b/s and more precisely at the rate of the backbone 110, i.e.,
8,000 octets per second. (Consequently, if the backbone
operated at a faster rate, the transmission rate of the
invention could scale correspondingly) To achieve the 64,000
b/s rate, however, all of the quantization levels must be used
in modulating the baseband signal; that is, each of the
quantization levels would correspond to a signal point of a 255
point, one-dimensional constellation. (~.-law and A-law have
255 quantization levels)
An exemplary embodiment trades some of the
theoretically possibly bandwidth for noise resistance. In
particular, though quantization noise is alleviated, noise
resistance may help combat other noise in the telephone
channel.
More specifically, the spacing between some of the
adjacent quantizer levels in ~-law and A-law is relatively
small. Consequently, the "minimum distance," or dmin, is small
of a signal constellation that includes these small p.-law and
A-law levels as signal points. (dm;n is a known parameter for
characterizing the performance of a signal constellation in an
uncoded system, and in short, dmin refers to the shortest
"distance" between different levels in a signal constellation.
The distance may be measured according to different known
metrics, such as Euclidean distance or Hamming distance.)
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An exemplary embodiment of the invention, exploits the
inherent non-linear characteristics of the ~.-law or A-law
rules to achieve an acceptable dr,~,in yet retain substantially
improved transmission rates. The above exploitation may be
best illustrated by comparing a uniformly spaced (PAM) signal
constellation, with a non-uniform p.-law or A-law signal
constellation. To double dm;n of a uniformly spaced (PAM)
signal constellation or the non-uniform p-law or A-law signal
constellation, the amplitude difference between the closest
IO signal points needs to be doubled. To do this for a uniformly
spaced (PAM) signal constellation, requires that every other
level would need to be eliminated. To do this for non-uniform
~,-law or A-law signal constellation, a significantly smaller
percentage of levels needs to be eliminated. This is so,
because the spacing between adjacent levels grows non-
linearly and rapidly. Much less than half of the levels need to
be eliminated before the smallest spacing between remaining
levels is doubled.
To this end, the above embodiment attains an
advantageous trade-off of transmission rate for noise
resistance by using a signal constellation that excludes some
of the quantization levels of the line interface 130. That is,
the alphabet used by the system 200 will exclude some octets
225 that would be otherwise inverse quantized to levels that
would result in small spacings relative to other symbols in the
alphabet.
Consequently, a preferred embodiment uses a subset of
the ~.-law or A-law quantization levels as valid levels in the
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signal constellation. Using a subset allows the system to
attain transmission rates approaching 56 Kb/s, yet attain
desirable levels of noise resistance.
The above system 200 and corresponding paradigm
departs from the conventional arrangement 100 to attain
significant advantages, but it also creates design problems and
issues with no parallel in the conventional arrangement.
Among other things, the new arrangement creates problems of
1. ensuring that the signal 225 is appropriately
modified, or spectrally shaped, to improve overall
performance;
2. ensuring that the analog adapter 260 has precise
enough timing to properly sample the baseband
modulated signals received on loop 250;
3. combating certain distortion introduced by the
digital backbone network such as "robbed bit
signaling," which otherwise would effectively act
as a form of noise limiting the system's capacity;
4. handling intersymbol interference (1S1) generated
by the line interface 130 and the loop 250 so that
the transmitted binary information sent by the
source 270 may be recovered; and
5. combating various forms of system-introduced
noise, such as memory-less nonlinear distortion
from the line interface 130, so that the binary
information transmitted sent by the source 270
may be recovered.
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The Digital Ada~er ail Spectral Shad it g
The digital adapter 220 receives data from the site 270,
for example, in the form of a bit stream from a Local Area
S Network (LAN) or the Internet. The digital adapter 220 encodes
the incoming bit stream 240 into a sequence of octets 225,
which are transmitted to the backbone 110.
The line interface 130 converts the sequence of received
octets 225' into a sequence of quantization levels. In certain
situations, it is desirable to shape the frequency spectrum of
this sequence to combat the effects of certain forms of
distortion. For example, it may be desirable to avoid placing
any energy at DC to avoid certain distortion that may be
created by such energy. Although such distortion may be
1 S relatively tolerable for voice communications, it may present
a significant impairment to data communications.
The system uses a novel mechanism to spectrally shape
the sequence of quantization levels to be transmitted. The
spectral shaping assures that the data attain the desired
characteristics, while minimizing the shaping's impact on
achievable transmission rates. This aspect is described in the
U.S. Pat. 5,818,879 entitled Device, System and Method for Spectrally
Shaping Transmitted Data Signals.
II. The Analog Adapter and Timing Recovery to Properly
Sample Si_ana_!s
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Referring briefly to figure 3B, the analog adapter 260
includes a section 440 for sampling the baseband signal
received from the local loop 250 possibly after equalization
5 and a section 450 for detecting, or estimating, the binary
information in the demodulated signal 445.
The system includes novel mechanisms for providing the
timing signals used for sampling the signals 250. ..
!!L Combating Robbed-Bit Si n~ling
I S Robbed bit signaling is a technique used in the telephone
network to accomplish various signaling functions. Robbed bit
signaling can modify the octets as they ,are being transmitted
across the digital network. in this regard, robbed bit signaling
is a form of distortion or noise that can limit the capacity of
the system.
The description above alluded to this aspect when stating
that the sequence of octets 225 entering the backbone 110 is
not necessarily the same as the sequence of octets 225'
exiting the backbone 110. The two sequences may differ
depending upon the presence and type of robbed bit signaling.
The system includes novel mechanisms for handling
robbed bit signaling. This aspect is described in the U.S. Pat.
No. 5,875,229 entitled System and Device for, and Method of, Detecting,
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Characterizing, and Mitigating Deterministic Distortion in a
Communications System.
l V.. Controlling Intgr-S~rmbol Interference (ISII
The backbone 110 and line interface 130 were designed
and constructed for voice communications. One consequence of
the design is that an interpolation filter (420, see fig. 3A)
typically found in the line interface 130 does not satisfy
Nyquist's criterion when signaling at 8000 baud, causing ISI on
the signal received by the analog adapter 260.
To handle ISI, the invention uses a novel arrangement of
an~ equalizer and a level decoder. Because the inventive
arrangement for controlling ISI is also used to combat system
introduced noise, to avoid a redundant description, the
arrangement is discussed in the next section only.
V. Combating System-Introduc~_ Noisy
To better understand the invention's novel mechanisms
for combating ISI and other noise, refer to figures 3A-B. In
figure 3A, only the parts of the line interface 130 and loop 250
that are material to understanding the invention are shown. In
figure 3B, the analog adapter 260 is shown as a high-level
architectural diagram:
The fine interface 130 includes a digital-to-analog
converter (D/A converter) 410 and a low pass filter (LPF) 420,
or interpolation filter. The D/A converter 410 is responsible
for converting the received sequence of octets 225' into a
sequence of quantization levels as outlined above. That is, the
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D/A converter 410 will receive an octet 225', v~, and construct
a signal 415, a (v~), having an .amplitude level corresponding to
the octet 225' and the relevant ~-law or A-law rules (more
below). The resulting sequence of levels 47 5 is then sent to
LPF 420, which shapes the sequence and sends tile resulting
line interface signals 416 on to the channel 430. For
de~i~ti~Ee p+.~Roees, -the -c#ar~neE 430-- ray be--r~o~efed as
having an impulse response g (t ). Thus, the signal exiting the.
channel at this point is modeled as s (f ), described above in
I0 equation (1). Signal s (t ) is subject to the addition 435 of a
noise component n (t ), yielding the analog signal received on
loop 250 by the analog adapter 260. .
The analog adapter 260, shown in figure 3B, includes a
section 440 that is responsible for sampling the signal 250.
The various components 441-443, responsible for timing
recovery, will not be described here. Suffice it to say that demodulated
signal 445 is a sampled version of loop signal 250.
The analog adapter 260 also includes an equalization and
detection section 450 that is responsible for compensating for
the linear distortion and then "interpreting" the resulting
sampled equalized sequence 465, r~. In this regard,
"interpretation" means analyzing the sequence 465 to detect
which sequence 225' of octets were sent. Since this sequence
is nearly identical to the originally-transmitted sequence of
octets 225, the original transmitted binary information can be
recovered. (The handling of robbed bit signaling function
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compensates for any discrepancies between 225 and 225'.)
Among other things, this detection must account for the noise
n (t ) on the channel, the effects of the channel g (t ), the
presence of lSl, and the effects of non-linearities in the D/A
converter 410.
Beginning with the latter, the invention assumes that
real-world systems will not precisely follow the ~.-law or A-
Law quantization levels proscribed in ITU Recommendation
6.711. Instead, the invention assumes that the line interface
130 will be low fidelity with regard to the accuracy of the
transmitted levels a (v~) vis-a-vis the proscribed levels.
In other words, the line interface 130 will produce, in
response to a given octet 225', a level not having the precise
amplitude specified by the p-law or A-taw rules as
I S corresponding to the given octet. Instead, the invention
assumes that the line interface 130 will produce a level having
an amplitude level that varies from the specified amplitude
and, moreover, that the amount of variation between the real
amplitude and the specified amplitude will depend on the
specified amplitude level. For convenience, using y(i) to
designate a level taken from the A-law or p-law
specifications, representing the i'th level in the signal
constellation where i is an integer between 0 and M-1, the
actual level 415 produced by D/A converter 415 may be
mathematically described as follows:
X(i) = Y(i) + ~(Y(i)) (2)
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In equation (2), the error component o(y(i)) describes how
much the actual level x(i) varies from the level y(i) specified
in A-law or p.-law. Moreover, although the error component is
described as a function of the particular specified level, it
should be appreciated that the error component is not known a
-p.ria~i a-nd that the ~nderlyi~ng relat;~+~s-h~i-p, ~e#i~i-rig the
function ~(y(i)), may change slowly over time and is likely to
change from one connection on the telephone network to the
next.
Without more, the error component is a source of non-
linear noise that could affect the system's information-
carrying capacity. Unless corrective steps are taken, the low
fidelity line interface 130 could cause incorrect predictions,
or interpretations, of the line interface signal 416 that was
transmitted on to the channel 430.
- The problem of possible erroneous prediction is
illustrated with figures 4A-6. Figure 4A sF~ows two points
y(1 ) and y(2) corresponding to two of the "signal points" of the
one-dimensional constellation of an exemplary embodiment.
Each signal point corresponds to an amplitude level and may be
considered as an information-carrying "symbol." The set of
symbols may be considered as an "alphabet." Graphing all
symbols of an alphabet constructs a "constellation,"
representative of the code.
Assuming that the alphabet mirrors the ~.-law or A-law
specified amplitude levels, y(1 ) corresponds to one of the 256
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specified amplitude and y(2) corresponds to another, possibly
adjacent specified amplitude level. In a noise-free,
distortion-free system, the signals 250 received possibly
equalized and sampled by the analog adapter 260 will only have
5 the amplitude levels y(1 ) . , y(2) and so on.
As alluded to above, however, the signal 250 includes
additive noise n(t), non-linear distortion due to errors in the
D/A converter, ISI, and the like. Thus, the eventually received
and sampled sequence r~~ should not be expected to fall right on
10 a signal point of the constellation.
In a conventional system, a decoder will analyze the
received sequence r~, for example, using distance metrics, to
detect the sequence of levels that most probably was sent that
would have yielded the received. signal. For example, a
15 conventional decoder, responsible for detecting the
transmitted signals of the simple, memory-less signal
constellation of figure 4A, will predict that the transmitted
signal was y(1 ), because the distance d[rn, y(1 )J between rn and
y(1 ) is smaller than the distance d[rn, y(2)J between r~ and
20 y(2). The smaller distance d[rn, y(1 )] is interpreted as meaning
that it was more likely that y(1 ) was transmitted. When
coding is employed, e.g., trellis codes, analysis of metrics
would involve comparing distances between sequences of
levels rather than individual levels.
Figure 4B illustrates how the conventional arrangement
may yield incorrect predictions when the characteristics. of a
low fidelity Fine interface 130 are considered. Figure 4B '
includes alf of the items of figure 4A and alsq illustrates two
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new items x(1 ) and x(2). x(1 ) and x(2) represent the actual
levels of the transmitted signal point-416, i.e., the signals
described by equation (2). Thus, x(1) and. x(2) represent the
true levels the line interface 130 actually uses. In other
words, when interface 130 means y(1 ) it actually sends x(1 )
and when it means y(2) it sends x(2).
_ _ _ . _. _. A.s _ shn~arn,__if ~_. dec~ader_ o.p_erate~ .on the_
ideal_aecarld
alphabet, specified with y(1 ) and y(2) as a partial alphabet, it
will predict that y(1 ) was transmitted, because the distance
d[r~, y(1 )] is smaller than the distance d[rn, y(2)]. This
determination would be incorrect, however. As shown,
appreciating the actual distortion and its effect on the real-
world alphabet, specified with x(1 ) and x(2) as a partial
alphabet, it is more likely that the transmitter sent symbol
x(2) because the distance d[rn, , x(2)] between the received
sequence rn and the actual signal point x(2) is smaller than the
distance d[rn, x(11].
To address this problem of error, an exemplary
embodiment of the invention estimates the actual alphabet,
i.e., the actual signals transmitted, and then uses those
estimates in an improved level decoder to predict the most-
probably-transmitted signal. In this fashion, the exemplary
decoder will avoid the erroneous interpretations illustrated in
figure 4B. As will be explained below, the exemplary decoder
computes metrics with respect to the estimate of the actual
signal constellation used by the line interface, rather than the
pre-specified A-law or p.-law quantization levels specified in
Recommendation 6.711.
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Another aspect of the invention is to estimate the actual
alphabet in the presence of other linear distortion. An
exemplary embodiment of the invention compensates for such
distortion with an equalizer, e.g., a linear equalizer or a
decision-feedback equalizer. On severely distorted channels, a
linear equalizer may be combined with a maximum-likelihood
sequence decoder (MLSD) or a suboptimum version thereof, in
which the linear equalizer removes part of the ISI and the
Viterbi equalizer removes the rest.
First consider the somewhat idealized case in which
there is no linear distortion in the system, or the distortion is
already equalized by some other means. In such a system, the
received sequence r~ will be of the form:
r~, = x~ + N~, ( 3 )
where x~ is the actually-transmitted symbol, also represented
by the index i". The decoder 470 keeps in storage an alphabet
471 dX~ _ {dx(0), dx{1 ),.....,dx(M)} of symbols which represent
the decoder's estimate of the actual alphabet. The decoder
includes logic to compute metrics based on the received
symbol r~ and symbols dx~ = dx(i~) from the estimated alphabet.
An exemplary embodiment includes logic to generate an
estimate of the actual alphabet, as follows. During a training
period the digital adapter 220 sends a known sequence of
octets 225, corresponding to signal point indices i~. The
decoder 470 eventually receives a corresponding baseband
signal on loop 250 and maintains an ordinary average of the
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received level. When performed over many symbols, the
averages wilt accurately represent the values of the actual
levels.
Such averages can also be determined iteratively using
the Least-Mean Square (LMS) algorithm. In this case, the
decoder would compute the error e~, between the received
signal r~ and the estimate dx(i") of the actual signal x(i~) taken
from the present alphabet, and will then update this estimate
according to:
dx(i~) -> dx(i~) + a en, (4)
where a is a small update coefficient. The value of a depends
on the signal constellation, but is typically a small fraction of
the average magnitude of the actual signal points.
When the transmitted index sequence i~ is unknown in the
receiver, as it would normally be during data transmission, the
decoder can compute the error signal as the difference
between the received signal r~ and the decoders estimated
level dx(i~) from the present alphabet 471. That value of that
level is then updated according to the averaging scheme or the
LMS update formula given above in equation (4).
When the above scheme is used with an adaptive linear
equalizer, a new problem arises. When the equalizer
coefficients 461 and the decoder alphabet 471 are updated
jointly, both the estimates in the alphabet and the equalizer
coefficients may converge towards zero. An exemplary
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24
embodiment of the invention avoids this undesirable operation
by constraining the sum of the magnitudes of the equalizer
coefficient to be equal to some non-zero value.
As shown in figure 3B, the equalizer 460 outputs the
sequence r~ 465 which is equalized using a transversal fitter
arrangement with tap coefficients w. The difference between
the equalized signal, i.e., one that is processed to control ISI
and the like, and the estimate dx(i") of the actual level that
was most probably sent to yield that equalized signal may be
expressed with the following equation:
a = r - dx(i~) (5)
As mentioned above, one known solution to minimize the
1 S energy of the error described in equation (5) would force dx(i~)
and tap coefficients w to zero. Though this solution minimizes
- the - average e-rro-r, ft also ~e-~roves III tn-fir-r-na-tio-r~ end tftust,
therefore, be guarded against.
To protect against the uninformative solution, an
exemplary embodiment imposes the constraint that the tap
coefficients w have a norm equal to 1. That is,
(6)
liwll ~1
(The mathematical concept of a norm is known).
To update the decoder's estimate of the alphabet 471, the
decoder could use the same iterative LMS algorithm described
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earlier with equation (4). In this fashion, the decoder's 470
alphabet 471 is updated iterativefy. The initial alphabet could
correspond to the nominal values proscribed by the ITU
Recommendation and the initial update rnay take part from a
5 training sequence included in an initialization procedure. For
example, all symbols in the alphabet may be sent many times
to make sure that the system is sufficiently exercised to
create an initial real-world alphabet 471 in decoder 470.
Analogously to the above, the adaptive equalization of
10 signals is best accomplished when performed using the
symbols from the estimated alphabet as a reference to form
the error signal e~. That is, the quality of the equalization is
judged by comparing the equalized signal to an estimate of the
signals actually transmitted rather than by comparing the
15 equalized signal to some pre-defined ideal value.
Conventionally, equalization would compare the output of the
equalizer with a corresponding expected value of an output,
such as a prescribed value of the alphabet. As stated above,
the invention dynamically builds an alphabet 471 corresponding
20 to the symbols actually used by the line interface 130, rather
than using the pre-specified amplitude levels of the ~.-law
rules, for example.
Thus, an exemplary embodiment updates the i'th equalizer
coefficient according to the following iterative procedure
25 using known techniques.
wi(n + 1 ) = wi(n) + ~d xi,n e~. (7)
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Here, xi,n represents the equalizer input signal at the i'th
coefficient, and e~ is the error signal described earlier. E
will be empirically determined, and its value is typically much
smaller than the average energy of the equalizer input signal.
To prevent the equalizer coefficients from converging towards
zero, the coefficients need to be scaled up. This can be
accomplished by multiplying all coefficients once in a while by
a scale factor f3, a number slightly greater than 1.0, or by
multiplying wi(n) by a similar scale factor f3.
The above-described use of creating an error signal by
comparing the equalized signal with a corresponding symbol
Y
from a dynamically-built estimate of the alphabet may also be
used as an error signal when the linear equalizer is followed
by a Viterbi decoder (for example, in partial-response
systems), or when a decision-feedback equalizer is used. The
only difference is that in these cases, the reference signal
used in computing the error signal may depend on more than
one symbol.
The equalization and detection algorithms can be
implemented on conventional DSP hardware or on PC
processors using well-known programming techniques.
A prior embodiment of the above invention was described
in French application No. 95-12672 on Oct. 23, 1995. to Pierre
Humblet.
Although the invention was described in the context of a
particular preferred embodiment, namely one that uses the D/A
converter in line interface card, it has much broader
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applicability. in fact, the invention can be used in any digital
communication system, where the generation of the
symbols in the transmitter introduces a non-linearity which
causes the transmitted symbols to deviate from their pre-
y specified values. The invention also applies to situations
where the transmission medium further introduces distortion,
thus requiring an equalizer in the receiver.
The present invention may be embodied in other specific forms
without departing from the spirit or essential characteristics.
The described embodiments are to be considered in all respects
only as illustrative and not restrictive.
