Note: Descriptions are shown in the official language in which they were submitted.
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CIRCUIT FOR LINEARIZING ELECTRONIC DEVICES
[0001] This application claims priority from U.S. Provisional Application
Serial No. 60/245,707, entitled "CIRCUIT FOR LINEARIZING ELECTRONIC
DEVICES," filed November 3, 2000, the content of which is incorporated herein
by
reference in its entirety.
Field of the Invention:
[0002] The present invention relates to electronic circuits. More
specifically,
the present invention relates to systems and methods for linearizing
electronic,
devices, circuits and systems.
Description of the Related Art:
[0003] The front ends of radio frequency (RF) communication systems
typically include amplifiers and mixers. The receiver front-end amplifier
serves to
boost a received RF signal and the mixers serve to downconvert the signal from
high
RF frequencies to lower frequencies more suitable for processing and output.
Unfortunately, when the RF amplifiers and mixers handle desired received
signals in
a presence of strong interferers, distortion often results from the strong
interferers that
can degrade the quality of the desired signal. The distortion products that
fall in the
band of the received signal are most dangerous. They are typically generated
due to
third degree nonlinearities and called third-order intermodulation (IM3)
distortion.
With two strong interferers at f~ = 1000 MHz and f2 = 1001 MHz, for example,
the
intermodulation distortion products will be generated at (2f1 - fz) and (2fz -
f~).
Hence, for the exemplary signals centered at 1000 MHz and 1001 MHz the
distortion
signals will be centered at 999 MHz and 1002 MHz. If one of these signals
falls in
band of the desired signal, it may degrade receiver sensitivity.
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[0004] Conventionally, it is difficult to achieve low levels of the distortion
inasmuch as the third degree nonlinearity of the circuit is an inherent
property of
amplifying active devices. Typically, a lower distortion is achieved at the
expense of
an increased DC current consumption. The latter is undesirable especially in a
wireless communication system powered by a battery where the higher current
consumption results in a shorter battery life and, thus, a shorter system
operation (for
example, the talk time of the cellular phones).
[0005] Hence, a need remains in the art for a system or method for improving
the linearity of amplifiers and mixers used in RF receivers and other systems
without
a significant rise in the DC current consumption.
SUMMARY OF THE INVENTION
[0006] The need in the art is addressed by the system and method of the
present invention. In an illustrative application, the invention is realized
as a radio
frequency amplifier. The inventive amplifier includes a first transistor
having first,
second and third terminals. In the illustrative embodiment, the first
transistor is
bipolar with the first terminal being an input terminal and the second
terminal being
an output terminal and the third terminal being a common terminal. A
linearization
circuit is included having first and second terminals. The first terminal is
connected to
the common terminal of the transistor and the second terminal is connected to
the
input terminal of the transistor. The linearization circuit doesn't interfere
with the
transistor amplification of an input RF signal. But, in the presence of two
input signals
or an input modulated carrier, the linearization circuit forces the control
voltage
between the input terminal and the common terminal of the transistor to be
zero at the
difference frequency of the two input signals or at a modulation frequency of
the input
modulated carrier.
[0007] In a specific embodiment, the linearization circuit is implemented as a
non-inverting unity gain buffer with an input terminal connected to the common
terminal of the transistor and an output terminal connected to the input
terminal of the
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transistor. The buffer is designed so that its gain is unity and its output
impedance is
low at the difference frequency of the two input signals or at the modulation
frequency of the input modulated carrier. It forces the transistor input
voltage to
follow the common terminal voltage at these low frequencies. The buffer gain
is
sufficiently low and its output impedance is high in the amplifier operating
frequency
band to allow the amplification of input signals by the transistor in this
band.
[0008] In another specific embodiment, the linearization circuit includes a
non-inverting unity gain buffer with an input and output terminals and a radio
frequency choke coil. The input terminal of the buffer is connected to the
common
terminal of the transistor and the output terminal is connected to the first
terminal of
the choke coil. The second terminal of the choke coil is connected to the
input
terminal of the transistor. In accordance with the inventive teachings, the
choke coil
has high impedance in the operating frequency band of the amplifier and, thus,
isolates the buffer output from the amplifier input in this band. The choke
coil has a
low impedance at the difference frequency of the two input signals or at the
modulation frequency of the input modulated Garner. Thus, the choke doesn't
prevent
the buffer from forcing the transistor input voltage to follow the common
terminal
voltage at the mentioned frequencies.
[0009] In alternative embodiments, circuitry is shown for providing a DC
offset at the input of the transistor. As another alternative, the
linearization circuit
consists of series inductor and capacitor connected between the common and
input
terminals of the transistor. This series LC circuit acts as an open circuit in
the
operating frequency band of the amplifier and as an AC short circuit at the
difference
frequency of the two input signals or at the modulation frequency of the input
modulated Garner.
[0010] In yet another embodiment, the linearization circuit consists of the
first
and the second series inductor and capacitor circuits. The first series LC
circuit is
connected between the common terminal of the transistor and ground and the
second
series LC circuit is connected between the input terminal of the transistor
and ground.
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[0011] The inventive method provides increased linearity and minimal third-
order distortion in amplifiers, mixers, and other circuits used in high
frequency
circuits and systems.
BRIEF DESCRIPTION OF THE DRAWINGS
[0012] Fig. la is a graph of output power verses input power for a typical
amplifier showing the third order intercept point between the linearly
extrapolated
output power associated with fundamental signals and the linearly extrapolated
output
power associated with third-order intermodulation distortion resulting
therefrom.
[0013] Fig. 1b is a typical RF amplifier.
[0014] Fig. lc is a graph of output voltage verses frequency for the typical
RF
amplifier of Fig. la showing the fundamental tones at f~ and f2 and 2°d
and 3rd order
distortion resulting therefrom.
[0015] Fig. 2 is a schematic diagram of an arrangement for mitigating third-
order intermodulation products constructed in accordance with conventional
teachings.
[0016] Fig. 3 is a simplified schematic diagram of a linearized circuit
including a linearization circuit for mitigating third-order intermodulation
distortion
in accordance with the teachings of the present invention.
[0017] Fig. 4 shows a linearized circuit with a linearization circuit
implemented as a non-inverting unity gain buffer
[0018] Fig. 5 is similar to Fig. 4 with the exception that a choke coil is
inserted between the output of the unity gain buffer and the base of Q1.
[0019] Fig. 6 is similar to Fig. 5 with the exception that an arrangement for
providing a DC offset between the terminals of the linearization circuit.
[0020] Fig. 7 is similar to Fig. 6 with the exception that the base of Q2 is
connected to the output of the unity-gain buffer.
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[0021] Fig. 8 is similar to Fig. 7 with the exception that Q3 emitter follower
replaces the unity gain buffer in accordance with the teachings of the present
invention.
(0022] Fig. 9 depicts a first alternative embodiment of the linearized circuit
of
the present invention.
[0023] Fig. 10 depicts a second alternative embodiment of the linearized
circuit of the present invention.
DESCRIPTION OF THE INVENTION
[0024] Illustrative embodiments and exemplary applications will now be
described with reference to the accompanying drawings to disclose the
advantageous
teachings of the present invention.
[0025] The level of the third-order intermodulation (IM3) distortion product
generated by a nonlinear circuit at small input power levels is usually
estimated from
a third-order intercept point (TOIP) measured with a two-tone input signal.
This is
illustrated in Figs. la and 1b.
[0026] Fig. la is a graph of output power verses input power for a typical RF
amplifier showing the third order intercept point between the extrapolated
output
power associated with fundamental tones and the extrapolated output power
associated with intermodulation distortion resulting therefrom. When the two
fundamental tones (fl and f2) representing two interferers are applied to a
transistor,
its nonlinear characteristics generate spurious responses in the output
voltage. This is
depicted in Fig. 1b.
[0027] Fig. 1b is a graph of output voltage verses frequency for the typical
RF
amplifier of Fig. la showing the fundamental tones at f~ and f2 and 2°d
and 3rd order
distortion resulting therefrom. Similar distortion products will be generated
in the
input voltage of the amplifier due to its nonlinear input impedance. On the
simultaneous application of the fundamental tones to a typical RF amplifier,
the
second order mixing term at f2 - fl appearing in the input voltage modulates
the
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amplifier bias and adds to the IM3 distortion products at 2f~ - f2 and 2f2 -
f~ appearing
in the output voltage.
[0028] To prevent the difference-frequency mixing term from being generated
in the input voltage, the input bias circuit is typically designed to provide
a very low
impedance at the modulation frequency (f2 - f,). An example of such a bias
circuit is
shown in Fig. 2.
[0029] Fig. 2 is a schematic diagram of an arrangement for mitigating
intermodulation products constructed in accordance with conventional
teachings. In
Fig. 2, a transistor Q1 has its emitter connected to ground through an emitter
degeneration circuit 12. The emitter degeneration circuit 12 is not always
required and
is shown for generality. A bias circuit 14 generates the input DC bias voltage
applied
to the base of the transistor Q1.
[0030] The impedance looking into the output of the bias circuit 14 is
designed to be very small at low frequencies and, specifically, at the
difference
frequency (f2 - fl). Thus, the base voltage distortion products at this
frequency are
significantly attenuated. The output impedance of the bias circuit 14 is
designed to be
much higher than the transistor input impedance in the operating frequency
band to
prevent an input RF signal from being shorted by the output of the bias
circuit.
[0031] The approach taken in Fig. 2 is a typical method of designing low-
distortion RF amplifiers so that the amplifiers are not modulated by the input
interferers. Unfortunately, with this approach, although modulation by (f2 -
f,) is
inhibited at the input of the circuit (the base terminal of the transistor
Ql), the emitter
is free to be modulated if the emitter-degeneration circuit has a nonzero
impedance at
the modulation frequency.
[0032] The collector current of Q1 varies with Vbe in a nonlinear fashion. The
component of the collector current that depends only on the squared base-
emitter
voltage Vbe2 is called the second-degree nonlinearity. It mixes together the
fundamental tones in the input voltage and produces second-order distortion
products.
In particular, there will be a distortion product at the difference frequency
of f2 - f,
which will be particularly problematic. If the emitter degeneration circuit
has a nearly
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zero ohm impedance at (f2 - fl), the distortion current at (f2 - fl) will not
create a
voltage drop across it and the base-emitter junction of Q1 will not be
modulated and a
low IM3 distortion level is achieved. If the emitter degeneration circuit is
not zero
ohms at (f2 - fl), the difference-frequency product of the collector current
creates a f2 -
f~ spurious response in the emitter voltage Ve. Even though the base voltage
Vb = 0 at
f2 - f,, the nonzero Ve at f2 - fl modulates Vbe adding to the third-order
distortion.
[0033] The present invention provides a system and method for holding the
base-emitter voltage Vbe of a bipolar junction transistor constant at the
difference
frequency (f2 - fl) and reducing the IM3 distortion.
[0034] Fig. 3 is a simplified schematic diagram of a linearized RF circuit 10'
having an arrangement for mitigating intermodulation distortion in accordance
with
the teachings of the present invention. The circuit 10' of Fig. 3 is similar
to the
circuit 10 of Fig. 2 with the exception that the bias circuit 14 is replaced
by the
linearization circuit 14' with two terminals one of which is connected to the
emitter of
Q1 and the other one is connected to the base of Q1. The purpose of the
linearization
circuit 14' is to force the base-emitter voltage Vbe to be zero at the
difference
frequency (f2 - f,). The linearization circuit is designed so that it doesn't
interfere with
Q1 amplification of an input RF signal.
[0035] Fig. 4 shows the linearized circuit 10" of the present invention with
the linearization circuit 14" implemented as a non-inverting unity gain buffer
15".
The input of the buffer is connected to the emitter of Q1 and the output is
connected
to the base of Q1. The buffer is designed so that its gain is unity and its
output
impedance is low at the difference frequency (f2 - f~). It forces the base
voltage of Q1
to follow the emitter voltage of Q1 resulting in a zero Vbe at this frequency.
The
buffer gain is sufficiently low and its output impedance is high at the
frequencies of
the input signals f~ and f2 to allow their amplification by Q1.
[0036] The requirement to the buffer output impedance to be low at low
frequencies and high at high frequencies can be met if the output impedance is
designed to be inductive. There are several approaches to achieve this design
goal.
The simplest one is to add an inductive choke coil in series with the buffer
output
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designed to have a low impedance in a wide range of frequencies as shown in
Fig. 5.
The other possible approach is to implement the unity gain buffer as an
'active
inductor'.
[0037] Fig. 5 is similar to Fig. 4 with the exception that the choke coil L is
inserted between the output of the unity gain buffer 15"' and the base of Q1.
Since
the inductor L appears as an open circuit at RF frequencies isolating the
output of the
unity gain buffer from the base of Q1, the buffer output impedance is no
longer
required to be high at these frequencies. At the difference frequency (f2 -
fl), the
inductor L has a very low impedance that doesn't prevent the buffer from
forcing the
base voltage of Q 1 to follow its emitter voltage. The buffer is still
required to have a
low impedance at the difference frequency to short out the distortion products
in the
base of Q 1.
[0038] To set the desired bias point of the transistor, the linearization
circuit
14"' should have a DC offset between its terminals. More specifically, the
terminal
that is connected to the base of Q1 should be at a higher potential than the
terminal
connected to the emitter of Ql. Figs. 6, 7 and 8 show how the invention can be
used
in combination with a desired Vbe bias circuit.
[0039] Fig. 6 shows the linearized circuit 20 of the present invention with an
arrangement for providing a DC offset between the terminals of the
linearization
circuit 24. The unity gain buffer 25 forces the DC base voltage of Q2 set by
the
reference current Iref to appear at the base of Q1. Consequently, Vbe of Q1 is
equal to
Vbe of Q2 at DC or close to DC. If Q1 and Q2 have equal emitter area, both of
them
will draw the same DC current i.e. Iref~ Typically though, the emitter area of
Q2 (often
referred to as the reference device) is n times smaller than the emitter area
of Q1. In
this case, Q1 draws the DC current of nIret. The current Iref is supplied by a
current
source 26 such as a P-MOS transistor with a fixed gate source voltage.
[0040] In operation, Q2 acts as a level-shifting diode. On the application of
the two tones to Ql, there will be a (fz-fl) mixing product in the emitter
voltage of Q1.
This product will be shifted in its DC level by Q2 and then forced by the
unity gain
buffer 25 at its output. The inductor L will act as a short at (fz-f~)
transferring the
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buffer output directly to the base of Q1. So, the base voltage of Ql is forced
to follow
the emitter voltage of Ql at (fz-f~) resulting in a constant Vbe. At RF
frequencies, the
inductor L appears as an open circuit and the emitter voltage of Q1 is
isolated from
the base of Q1 which allows Q1 to act as a transconductor i.e. to convert the
input RF
voltage across the base-emitter junction into the output RF current.
[0041] One of the drawbacks of the DC bias approach in Fig. 6 is that not all
of the reference current Iref flows through the collector of Q2 but only that
portion of it
left after supplying the Q2 base current Ib.Qz. Therefore, the collector
current drawn by
Q1 is n(Iref - Ib.Qz). Since Ib.Qz=Iref/(~ where (3 is the Q2 forward DC
current gain, the
DC collector current of Q1 is nI~e~{1-1/(3) i.e. a function of the Q2 beta.
The latter
significantly (50°l0 or more) varies over process and temperature
causing a varying
DC current of Q1.
[0042] The circuit in Fig. 7 is similar to the circuit of Fig. 6 with the
exception
that the base of Q2 is connected to the output of the unity-gain buffer 25'.
Provided
that the input impedance of the buffer is very high, all of Iref flows through
the
collector of Q2. The DC collector current of Q1 is then exactly nIrer~ The
other
advantage of the linearization circuit in Fig. 7 is that the unity gain buffer
25' and the
transistor Q2 form a negative feedback loop that reduces the output impedance
of the
buffer by (1+AoL). AoL is the open-loop gain computed as the voltage gain from
the
base of Q2 to its collector.
[0043] The circuit of Fig. 8 is similar to the circuit of Fig. 7 with the
exception
that the unity gain buffer 25" is implemented as Q3 emitter follower. In Fig.
8, Q2
and Q3 are connected in a current mirror configuration with Q2 being the
current
mirror reference device and Q3 being a 'beta helper'. Q2, Q3 and the Iref
current
source act as a unity-gain buffer 15"' in Fig. 5 with an input DC offset. The
emitter
terminal of Q2 is the input and the emitter of Q3 is the low-impedance output
of the
buffer. Due to Q3 providing a closed DC-feedback loop around Q2, there will be
a
difference in voltages between the base and the emitter of Q2 which acts as a
DC
offset that is required for Q2 to conduct non-zero current Iref. The same DC
voltage is
applied to the base-emitter junction of Q1 setting its DC collector current at
nI~ef.
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[0044] Figs. 9 and 10 depict alternative embodiments of the linearized circuit
of the present invention. In the circuit of Fig. 9, the linearization circuit
34 is
implemented as a capacitor C connected in series with the RF choke inductor L
between the base and the emitter of the transistor Ql. The capacitor C should
be large
enough to create a low impedance path between the base and the emitter of Q1
at low
frequencies. The bias for Q1 is generated by other circuitry that is not shown
in Fig. 9
for generality. The linearization principle of the circuit in Fig. 9 is based
on the
capacitor C and the inductor L acting as an AC short between the base and
emitter of
Q1 at low-frequencies and, specifically, the difference frequency (f2-fl). At
RF
frequencies, the high impedance of the inductor L prevents the base and
emitter of Q1
to be AC shorted by the capacitor C and Q1 acts as a transconductor.
[0045] In Fig. 10, the linearization circuit 34' consists of a first series LC
network L1/C1 connected between the emitter of Q1 and ground, and a second
series
LC network L2/C2 connected between the base of Q1 and ground. The capacitors
C1
and C2 are DC blocking capacitors whose capacitance should be large enough to
ensure a low impedance to ground at the difference frequency (f2 - fl). Since
the
inductors L1 and L2 are nearly DC shorts at f2 - f,, the emitter and base of
Q1 are
individually kept constant at f2-fl by the capacitors C1 and C2 respectively.
Consequently, Vbe is also constant. The output of Q1 is its collector. The
difference in
the approach in Fig. 10 is that there is no connection between the emitter and
base of
Q1 in the linearization circuit 34'.
[0046] Thus, the present invention has been described herein with reference to
a particular embodiment for a particular application. Those having ordinary
skill in
the art and access to the present teachings will recognize additional
modifications,
applications and embodiments within the scope thereof. For example, transistor
Ql
may be a cascode transistor and the emitter-degeneration circuit may be its
transconductor, the invention could operate with an RF mixer or other circuit,
and/or
transistor Q1 could be an input transistor of an amplifier or an active mxer.
Furthermore, the invention is not limited to a bipolar junction transistor
implementation. Heterojunction bipolar transistors (HBT's) or field-effect
transistors
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such as MOSFET's, PN junction FET's, MESFET's and PHEMT's may be used
without departing from the scope of the present teachings.
[0047] It is therefore intended by the appended claims to cover
any and all such applications, modifications and embodiments within the scope
of the
present invention.
