Note: Descriptions are shown in the official language in which they were submitted.
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NON-ZERO COMPLEX WEIGHTED SPACE-TIME CODE FOR MULTIPLE
ANTENNA TRANSMISSION
FIELD OF THE INVENTION:
[0001] This invention relates to a method and apparatus for achieving
transmit diversity in telecommunication systems and, more particularly, to a
method and apparatus for non-zero complex weighting and space-time coding
signals for transmission on multiple antennas.
BACKGROUND OF THE INVENTION:
[0002] As wireless communication systems evolve, wireless system design
has become increasingly demanding in relation to equipment and performance
requirements. Future wireless systems, which will be third and fourth
generation systems compared to the first generation analog and second
generation digital systems currently in use, will be required to provide high
quality high transmission rate data services in addition to high quality voice
services. Concurrent with the system service performance requirements there
will be equipment design constraints, which will strongly impact the design of
mobile fierminals. The third and fourth generation wireless mobile terminals
will
be required to be smaller, lighter, more power-efficient units that are also
capable of providing the sophisticated voice and data services required of
these
future wireless systems.
[0003] Time-varying multi-path fading is an effect in wireless systems
whereby a transmitted signal propagates along multiple paths to a receiver
causing fading of the received signal due to the constructive and destructive
summing of the signals at the receiver. Several methods are known for
overcoming the effects of multi-path fading, such as time interleaving with
error
correction coding, implementing frequency diversity by utilizing spread
spectrum
techniques, or transmitter power control techniques. Each of these techniques,
however, has drawbacks in regard to use for third and fourth generation
wireless systems. Time interleaving may introduce unnecessary delay, spread
spectrum techniques may require large bandwidth allocation to overcome a
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large coherence bandwidth, and power control techniques may require higher
transmitter power than is desirable for sophisticated receiver-to-transmitter
feedback techniques that increase mobile terminal complexity. All of these
drawbacks have negative impact on achieving the desired characteristics for
third and fourth generation mobile terminals.
[0004] Antenna diversity is another technique for overcoming the effects of
multi-path fading in wireless systems. In diversity reception, two or more
physically separated antennas are used to receive a transmitted signal, which
is
then processed by combining and switching to generate a received signal. A
drawback of diversity reception is that the physical separation required
between
antennas may make diversity reception impractical for use on the forward link
in
the new wireless systems where small mobile terminal size is desired. A
second technique for implementing antenna diversity is transmit diversity. In
transmit diversity a signal is transmitted from two or more antennas and then
processed at the receiver by using e.g. maximum likelihood sequence estimator
(MLSE), minimum mean square error (MMSE) receivers, Maximum-a Posteriori
receivers, or their approximations. Transmit diversity has more practical
application to the forward link in wireless systems in that it is easier to
implement multiple antennas in the base station than in the mobile terminal.
[0005] Transmit diversity for the case of two antennas is well studied.
Alamouti has proposed a method of transmit diversity for two antennas that
offers second order diversity for complex valued signals. S. Alamouti, "A
Simple Transmit Diversity Technique for IlVireless Communications," IEEE
Journal on Selected Areas of Communications, pp. 1451-1458, October 1998.
The Alamouti method involves simultaneously transmitting two signals from two
antennas during a symbol period. During one symbol period, the signal
transmitted from a first antenna is denoted by SO and the signal transmitted
from the second antenna is denoted by S1. During the next symbol period, the
signal -S1* is transmitted from the first antenna and the signal SO* is
transmitted
from the second antenna, where * is the complex conjugate operator. A similar
diversity transmission system may also be realized in code domain. As an
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example, two copies of the same symbol can be transmitted in parallel using
two orthogonal Walsh codes. Similar techniques can be also used to construct
a space-frequency coding method.
[0006] Extension of the Alamouti method to more than two antennas is not
straightforward. Tarokh et al. have proposed a method using rate ='/z, and 3/
SpaceTime Block codes for transmitting on three and four antennas using
complex signal constellations. V. Tarokh, H. Jafarkhani, and A. Calderbank,
"Space-Time Block Codes from Orthogonal Designs," IEEE Transactions on
Information Theory, pp. 1456-1467, July 1999. This method has a
disadvantage in a loss in transmission rate and the fact that the multi-level
nature of the ST coded symbols increases the peak-to-average ratio
requirement of the transmitted signal and imposes stringent requirements on
the linear power amplifier design. Additional techniques that mitigate these
problems are proposed in O. Tirkkonen and A. Hottinen, "Complex space-time
block codes for four Tx antennas," Proc. Globecom 2000, November 2000, San
Francisco, USA. Other methods proposed include a rate = 1, orthogonal
transmit diversity (OTD) + space-time transmit diversity scheme (STTD) four
antenna method. L. Jalloul, K. Rohani, K. Kuchi, and J. Chen, "Performance
Analysis of CDMA Transmifi Diversity Methods," Proceedings of IEEE Vehicular
Technology Conference, Fall 1999, and M. Harrison, K. Kuehi, "Open and
Closed Loop Transmit Diversify at High Data Rates on 2 and 4 Elements,"
Motorola Contribution to 3GPP-C30-19990817-017. This method requires an
outer code and offers second order diversify due to the STTD block (Alamouti
block) and a second order interleaving gain from use of the OTD block. The
performance of this method depends on the strength of the outer code. Since
this method requires an outer code, it is not applicable to encoded systems.
For the case of rate = 1l3 convolutional code, the performance of the OTD +
STTD method and the Tarokh rate = 3/ method ST block code methods are
about the same. Another rate 1 method is proposed in O. Tirkkonen, A. Boariu,
and A. Hottinen, "Minimal non-orthogonality rate 1 space-time block code for
3+
Tx antennas," in Proc. ISSSTA 2000, September 2000. The method proposed
in this publication attains high performance but requires a complex receiver.
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(0007] It would be advantageous, therefore, to have a method and apparatus
that provided the advantage of transmit diversity on greater than two antennas
while at the same time not greatly increasing the complexity of system design.
SUMMARY OF THE INVENTION
(0008] The present invention presents a method and apparatus for non-zero
complex weighting and space-time coding signals for transmission on multiple
antennas. The method and apparatus provides expansion of an N x N' space-
time block code, where N is the number of transmit paths and N' is the number
of output symbols per transmit path, to a M x M' space-time block code, where
M>N, generated by using repetition and non-zero complex weighting of the
symbols within the N x N' space time block code, to allow transmission of the
space time block code on a number M of diversity transmit paths. The diversity
transmit paths may comprise separate antennas or beams. The temporal
length of the larger code M', may equal the temporal length of the original
code,
N'. In the method and apparatus, a transform is performed on an input symbol
stream, to generate a transform result comprising a space-time block code.
The N output streams of the space-time block code, each consisting of N'
output
symbols, are then repeated and at least one of the repeated streams non-zero
complex weighted over time to generate M streams of N' output symbols for
transmission on M diversity transmit paths. The non-zero complex weighting
may include phase shifting.
[0010] In an embodiment, N is at feast 2 and M is at least 3. At least two of
the N streams of N' output symbols, corresponding to the original N streams of
N' output symbols, are then each transmitted on a first at least one antenna
and
at least one of the M-N non-zero complex weighted streams of N' symbols are
transmitted on one of a second at least one antenna. The first at least one
antenna and second at least one antenna may comprise of any one of the M
antennas.
[0011] In another embodiment, the method and apparatus may be
implemented in a transmitter having common or dedicated pilot channels that
enable efficient channel estimation of the coefficients that are required to
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decode the space-time code. In this embodiment the common and dedicated
pilot channels may be implemented alone or both together in the transmitter.
In
one alternative of this embodiment, training symbols are transmitted on N
transmit diversity paths, making it possible to estimate the N independent
diversity transmit paths. For this, a dedicated pilot channel code sequence
may
be multiplexed into each of the N streams of N' output symbols of the original
space-time block code, to generate N streams of N' output symbols and pilot
channel sequence. Repetition and non-zero complex weighting may then be
applied to generate M phase shifted streams of N' symbols and pilot channel
sequence. At least two of the N original streams of N' output symbols and
pilot
channel sequence are then transmitted on one of the first at least one antenna
and at least one of the M-N complex weighted streams of N' output symbols and
pilot channel sequence are transmitted on one of the second at least one
antenna. Another way of enabling estimation of N channels is to transmit
common pilot channels so that N common pilot channel are transmitted on each
of the first at least one antenna, and M-N complex weighted copies of some of
the N common pilot channels are transmitted on each of the second at least one
antenna. The complex weighting factors used for the common channels on
each of the second at least one antenna are the same as the ones used~to
construct the M-N additional complex weighted streams of N' output symbols
from the original N streams of N' output symbols. In these embodiments, the
receiver may or may not know the method used to expand the N x N' space-
time block code to an M x N' space-time block code, and the temporal weighting
sequences employed.
(0012] In other embodiments, where N is at least 2 and M may be at least 3,
the pilot channels may be arranged to enable estimation of at feast N+1
diversity transmit paths. At least one of the N streams of N' output symbols,
corresponding to the original N streams of N' output symbols, are then each
transmitted on a first at least one antenna and at feast one of the M-N
complex
weighted streams of N' symbols are each transmitted on one of a second at
least one antenna. Different common pilot channels are transmitted on each of
the first at least one antenna and on at least one of the second at least one
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antenna. In these embodiments, the receiver needs at least partial
knowledge of the method used to expand the N x N' space-time block code to
an M x N' space-time block code, and the temporal weighting sequences
employed.
Complex weighting in the various embodiments may be applied by
applying a periodic or random complex weighting pattern to each of the
symbol streams that are complex weighted. The relationship between the
complex weights of the symbol streams transmitted on the various antennas
may also be predefined.
Accordingly, in one aspect of the present invention there is provided a
method for transmitting a signal from a plurality of antennas, said method
comprising the steps of:
receiving a symbol stream at a transmitter;
performing a transform on the input symbol stream to generate a
transform result, the transform result comprising an N x N' orthogonal space-
time block code, and generating N first signals;
non-zero complex weighting, over time, at least one of the N first
signals of the transform result to generate at least one second signal, each
of
said at least one second signals being phase shifted relative to the one of
the
N first signals from which it was generated, and wherein the non-zero
complex weighting comprises phase shifting at least one of the N first signals
by at least a first predetermined hopping sequence, wherein hopping weights
for the predetermined hopping sequence are derived from a PSK constellation
having 8 states, and wherein the predetermined hopping sequence in degrees
is (0, 135, 270, 45, 180, 315, 90, 225); and,
transmitting, substantially simultaneously, each of the N first signals of
said transform result on a first antenna set comprising at least one antenna
and, each of said at least one second signals on a second antenna set
comprising at least one antenna, said N first signals and said at least one
second signal together comprising M signals wherein M is greater than N.
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According to another aspect of the present invention there is provided
an apparatus for transmitting a signal, the transmitter comprising:
an input symbol stream;
a processor for performing a transform on said input symbol stream to
generate a transform result, the transform result comprising an N x N'
orthogonal space-time block code, and generating N first signals;
at least one weighter for, non-zero complex weighting, over time, at
least one of the N first signals of said transform result to generate at least
one
second signal, each of said at least one second weighted signals phase
shifted relative to the one of the N first signals from which it was
generated,
and wherein the non-zero complex weighting comprises phase shifting at
least one of the N first signals by at least a first predetermined hopping
sequence, wherein hopping weights for the predetermined hopping sequence
are derived from a PSK constellation having 8 states, and wherein the
predetermined hopping sequence in degrees is (0, 135, 270, 45, 180, 315, 90,
225); and,
a transmitter for transmitting, substantially simultaneously, each of the
N first signals of said transform result on a first antenna set comprising at
least one antenna, and each of the N second signals on a second antenna set
comprising at least one antenna, said N first signals and said at least one
second signal together comprising M signals wherein M is greater than N.
BRIEF DESCRIPTION OF THE FIGURES:
FIG 1 a shows a block diagram of a transmitter according to an
embodiment of the invention;
FIG. 1 b shows a block diagram of portions of a common pilot channel
STTD transmitter according to an embodiment of the invention;
FIG. 2 shows a block diagram of portions of a common pilot channel
STTD transmitter according to another embodiment of the invention;
FIG. 3 shows a block diagram of portions of a dedicated pilot channel
STTD transmitter according to a further embodiment of the invention;
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6b
FIG. 4 shows a block diagram of portions of an embodiment of a
receiver for use with the transmitter of FIG.1;
FIG. 5 shows a block diagram of portions of an embodiment of a
receiver for use with the transmitter of FIG.2 or the transmitter of FIG. 3;
FIG. 6 shows rake finger embodiment of STTD demodulator 508 of
FIG. 5;
FIG.7 shows a block diagram of portions of an STS transmitter
according to an embodiment of the invention;
FIG. 8 shows a block diagram of portions of an OTD transmitter
according to an embodiment of the invention;
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[0023] FIG 9 shows a block diagram of portions of an embodiment of a
receiver for use with the transmitter of FIG. 7;
[0024] FIG. 10 shows a block diagram of portions of an embodiment of a
receiver for use with the transmitter of FIG. 8;
[0025] FIG. 11 shows a block diagram of portions of a long ST block code
transmitter according to an embodiment of the invention;
[0026] FIG. 12 shows a block diagram of portions of a comr~ion/dedicated
pilot channel STTD transmitter according to another embodiment of the
invention;
j0027] FIG. 13 shows a block diagram of portions of a receiver for use with
the transmitter of FIG. 12; and
[0028] FIG. 14 shows a block diagram of portions of a receiver for use in
power control of the transmitter of FIG. 12.
[0029] FIG. 15 shows a constellation defining a phase shifting pattern that
may be used in various embodiments of the invention.
DETAILED DESCRIPTION OF THE INVENTION:
[0030] Referring now to FIG. 1 a, therein is shown a block diagram of a
transmitter 150 according to an embodiment of the invention. Transmitter 150
includes input 152 for receiving an input symbol stream, block code processor
154 for performing a transform on the input symbol stream to generate a
transform result representable by an orthogonal space time block code and
outputting 2 streams of symbols of the transform result, non-zero complex
weighter 156 for non-zero complex weighting a first one of the two streams of
symbols, a non-zero complex weighter 158 for non-zero complex weighting the
second of the two streams of symbols, an RF transmitter 160 for transmitting
the first stream of symbols on Ant. 1, RF transmitter 162 for transmitting the
non-zero complex weighted stream of symbols on Ant. 2, RF transmitter 164 for
transmitting the second stream of symbols on Ant. 3, and RF transmitter 166
for
transmitting the phase shifted second stream of symbols on Ant, 4. The
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antennas Ant. 1 - Ant. 4 may be polarized relative to one another to provide
enhanced diversity reception. For example, Ant. 1 or Ant. 2 may be vertically
polarized relative to a horizontal polarization of Ant. 3 or Ant. 4,
respectively.
The embodiment of transmitter 150 of FIG. 1 a may be implemented in various
forms suitable for different technologies and systems to expand a 2 X N' block
code for transmission over 4 transmit diversity paths. In transmitter 150,
each
of the 4 transmit diversity paths includes a separate antenna, Ant. 1 - Ant.
4.
This may include code division multiple access (CDMA) systems, time division
multiple access (TDMA) systems, or any other type of digital communications
system into which transmit diversity may be introduced. In an alternative of
the
embodiment of FIG. 1 a, the non-zero complex weighting may be all performed
on selected ones of the transmission paths to create relative phase shifts
between the transmissions on Ant. 1 and Ant. 2 or on Ant. 3 and Ant. 4. For
example, non-zero complex weighting could also be applied before the inputs to
RF transmitters 160 and 164, creating a non-zero complex weighted version of
each of the symbol streams, but maintaining a relative phase shift between the
transmitted signals. An alternative of the embodiment of transmitter 150 may
be implemented using less than 4 antennas, to implement the 4 diversity paths.
As an example, the signals input to RF transmitters 164 or 166 may be
connected together and transmitted on a single antenna. Also other
alternatives are possible in which less than 4 diversity paths are used, for
example, only one of the 2 data streams may be non-zero complex weighted
and transmit on two diversity paths. In an alternative embodiment of FIG. 1 a,
the non-zero complex weighing operation may be performed after RF
transmitter blocks 160, 162, 164, 166, i.e., non-zero complex weighing could
be
implemented as a continuous phase sweep after the modulation, and baseband
filtering of Space-Time coded symbols.
[0031 The non-zero complex weighting for these transmissions on Ant. 2
and Ant. 4 may be performed according to various alternatives. For example, a
phase pattern Wi(t)=exp(j*pi*phase_in degrees/150) used on Ant. 2 may be
applied and the phase pattern - W~(t), which is 150 degrees out of phase with
W~(t) may be used on Ant. 4. Examples of this would be a phase pattern of
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shifts in degrees of {0, 90, 180, 270} on Ant. 2 and {180, 270, 0, 90} on Ant.
4
for 4PSK constellation. Other example patterns {0, 45, 90, 135, 180, 225, 270,
315} for 8PSK and {0, 22.5, 45, 67.5,.....337.5} for 16PSK. FIG. 15 shows a
constellation defining another phase shifting pattern that may be used in
various
embodiments of the invention. This sequence of shifts in degrees of {0, 135,
270, 45, 180, 315, 90, 225} may be transmitted on antenna 2 while using the
pattern of shifts in degrees of X180, 315, 90, 225, 0, 135, 270, 45} on
antenna 4.
The phase shifting may be periodic or random. Periodic phase shifting refers
to
a predefined phase pattern for e.g. complex weight W 1 (t) repeated
periodically.
The complex weights can be defined so that the sequence of complex weights
defines a maximal length path, to make successive samples of the effective
channel as independent as possible. This can make interleaving redundant and
thus enable low delay transmission. Pseudo-random phase shifting used may
be a sequence of random phase selections from a MPSK constellation.
Alternatively, another non-zero complex weighting scheme where the phase
difference between successive phase states is as small as possible is
advantageous when estimating channel coefficients or metrics related to power
control from a non-zero complex weighted channel. In this case, the phase
states may still cover 360 degrees during the duration of one encoding block.
Channel interleaving may be used in the embodiments as in conventional
systems. It is also possible to implement the non-zero complex weighting
sequence and the interleaver jointly, so that the symbols at the output of the
interleaver are as independent as possible. Furthermore, by changing the
relative phase between antennas 1 a,nd 2, and 3 and 4, respectively, the
method can be implemented so that there is a phase shift or sweep in all
antenna elements, but relative phase shifts between antennas 1 and 2, and 3
and 4 are maintained. As an example, with phase sweep, one may have a 50
Hz phase sweep on antenna 1 and -50 Hz phase sweep on antenna 2, in order
to implement a 100 Hz effective sweep. Similarly for antennas 3 and 4.
[0032] The phase rotation may be changed every T seconds. The choice of
T depends on total time duration of the data symbols and the method used for
estimating the channel coefficients. The phase may be kept constant for the
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total time duration occupied by the data symbols within at least one space-
time
encoding block and the corresponding dedicated or common pilot
sequence/training sequence can be used to enable proper channel estimation.
The pilot sequence could be a walsh code, as used in CDMA systems, or
sequence of training symbols with good correlation properties used for channel
estimation in TDMA. The pilot symbols may apply the same non-zero complex
weighting coefficients as the data within the space-time block. Alternatively,
the
pilots may be transmitted without phase hopping. In this case the effective
channel for the data can be derived jointly from the a priori known hopping
pattern and the channel estimate obtained from a non-hopping channel. In
cases where non-zero complex weighting is applied to common pilots, the same
or different phase pattern may be applied to both data and common pilots.
Channel estimation using non-hopping pilot or training sequences (either
transmitted on common or dedicated channels) provides better channel
estimates as the channel is more stationary.
(0033] Referring now to FIG. 1 b, therein is a block diagram of portions of a
common pilot channel space time transmit diversity (STTD) transmitter 100
according to an embodiment of the invention. Transmitter 100 may operate as
a 4-antenna transmit diversity extension to release 99 of the wideband CDMA
(WCDMA) third generation system standard. Transmitter 100 comprises input
126, block code processor 124, traffic channel symbol stream processing
branch inputs 102a - 102d, antenna gain blocks 104a, - 104d, phase shifters
106a and 106b, phase shifter inputs 112a and 112b, Code multipliers 108a -
108d, pilot sequence processing branch inputs 114a - 114d, antenna gain
blocks 116a -116d, code multipliers 118a - 1.18d, RF transmitter 128,
including
RF transmitters 1282 -128d, and antennas Ant.1 - Ant. 4.
[0034] In FIG. 1 b, data to be transmitted including a channel coded and
interleaved input symbol stream X(t) comprising the symbols S1S2 is received
at input 126. Block code processor 124 performs a transform on every two
received symbols S1S2 to generate a transform result comprising a 2x2
orthogonal space-time block code. In the embodiment, block code processor
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124 may perform an Alamouti transform to generate the block code in the form
represented by the following matrix:
51~: S~ ( 1 )
S2 S1
The matrix is then divided into 4 streams of 2 symbols with each of he streams
being input to one of the traffic channel symbol stream processing branch
inputs
102a - 102d. As shown in FIG. 1, the stream S 1 S2 is input to 102a, S 1 S2 is
input to 102b, -S2*S1* in input to 102c, and to -S2*S1* is input to 102d. The
non-zero complex weighting is performed by antenna gain blocks 104x -104d
and phase shifters 106a and 106b. Antenna gain for each of the processing
branches is adjusted in antenna gain blocks 104a -104d. After antenna gain is
adjusted, phase shifters 106a and106b apply a phase shift to the stream S1S2
output from antenna gain block 104b and stream -S2*S1 * output from antenna
gain block 104d. The phase shifter control blocks 112a and 112b may control
phase shifters 106a and 106b by causing shifting using a continuous or
discrete
phase hopping pattern. A CDMA scrambling code is then input to code
multipliers 108a - 108d to generate the stream S1S2 to RF transmitter 128a for
transmission on Ant. 1, S1S2 (expQ~k1)) to RF transmitter 128b for
transmission Ant. 2, -S1 *S2* to RF transmitter 128c for transmission on Ant.
3
and -S2*S1* (exp(j~k2)) to RF transmitter 128d for transmission on Ant. 4. The
RF transmitters may perform of baseband pulse shaping, modulation, and
carrier up conversion. In some implementations one may choose to apply
phase hopping or sweep after baseband pulse shaping and modulation steps.
[0035] Common pilot channel sequences X1 - X4 are input to pilot sequence
processing branch inputs 114a -114d. The pilot sequences are then
separately processed through antenna gain blocks 116a -116d, and code
multipliers 118a - 118d. The coded outputs from code multipliers 118a - 118d
are then input to RF transmitters 128a - 128d, respectively, of RF transmitter
130.
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[0036] The pilot sequence X1 is then transmit on Ant. 1, the pilot sequence
X2 is transmit on Ant. 2, the pilot sequence X3 is transmit on Ant. 3, and the
pilot sequence X4 is transmit on Ant. 4.
[0037] Referring now to FIG. 4, therein is a block diagram of portions of a
receiver for use with transmitter 100 of FIG. 1 b. FIG. 4 shows the signal
processing for one rake finger receiver section of a receiver. The received
pilot
sequences X1 - X4 transmit from transmitter 100 are received and input to
channel estimation processing branch 402a - 402d, respectively. Channel
estimator 404 then performs a channel estimation function, for example a low
pass filter moving average function, for each of channel 1 - channel 4. The
estimates of channel 1 - channel 4 are then output from outputs 406a - 406d to
summer 410a, phase shifter 408a, summer 410b and phase shifter 408b.
Phase shifter 408a receives input from phase shifter control block 414a and
shifts the estimate for channel 2 by the same phase shift used on the traffic
channel symbols S1 S2 transmit from Ant. 2 in transmitter 100. Phase shifter
408b receives input from phase shifter control block 414b shifts the estimate
for
channel 4 by the same phase shift used on the traffic channel symbols -S2* S1
transmit from Ant. 4 in transmitter 100. The phase shifted version of the
estimate for channel 2 is combined with the estimate for channel 1 by summer
410a, and the phase shifted version of the estimate for channel is combined
with the estimate for channel 3 in summer 410b. The combined estimate for
channels 1 and 2 (412a) and the combined estimate for channel 3 and 4 (412b)
are then input to STTD demodulator 418, which processes the received traffic
signals from input 416 using the channel estimates. The demodulated signal is
then processed in rake combiner, deinterleaver and channel decoder 420 to
generate the received symbols S1S2.
[0038] In an alternative common pilot channel embodiment for 4-antenna
diversity, common pilot channels are phase shifted in the same manner as the
traffic channels before transmission. Referring now to FIG. 2, therein is a
block
diagram of portions of a common pilot channel STTD transmitter 200 according
to another embodiment of the invention. Transmitter 200 comprises input 226,
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block code processor 224, traffic channel symbol stream processing branch
inputs 202a - 202d, antenna gain blocks 204a, - 204d, phase shifters 206a and
206b, phase shifter inputs 212a and 212b, Code multipliers 208a - 208d, Code
multiplier input 210, pilot sequence processing branch inputs 214a - 214d,
antenna gain blocks 216a - 216d, phase shifters 218a and 218b, phase shifter
control blocks 224a and 224b, code multipliers 220a - 220d, code multiplier
input 222, RF transmitter 228, including RF transmitters 228a - 228d, and
antennas Ant 1 - Ant. 4.
(0039] The traffic channel processing and transmission in transmitter 200 is
performed in the same manner as used for the traffic channel processing in
transmitter 100 of FIG. 1. Transmitter 200, however, uses common pilot
channels, which are phase shifted. Common pilot channel sequence P1 is input
to pilot sequence processing branch inputs 214a and 214b and common pilot
channel sequence P2 is input to pilot sequence processing branch inputs 214c
and 214d. The pilot sequences are then separately processed through antenna
gain blocks 216a - 216d. The pilot sequence P1 output from antenna gain block
216a is input to code multiplier 220a. The pilot sequence P2 output form
antenna gain block 216c is input to code multiplier 220c. Pilot sequence P1
output from antenna gain block 216b is input to phase shifter 218a. Pilot
sequence P2 output from antenna gain block 216d is input to phase shifter
218b. Phase shifter 218a and 218b apply a phase shift under the control of
phase shifter control block 224a and 224b, respectively. The phase shift may
be the same continuous or discrete phase hopping pattern used for the traffic
channels. The phase shifted pilot sequence P1 output from phase shifter 218a
is then input to code multiplier 220b and the phase shifted pilot sequence P2
output from phase shifter 218b is then input to code multiplier 220d. The
coded
pilot sequence P1 output from code multiplier 220a is then input to RF
transmitter 228a for transmission on Ant. 1. The coded phase shifted pilot
sequence P1 output from code multiplier 220b is input to RF transmitter 228b
for transmission on Ant. 2, the coded pilot sequence P2 output from code
multiplier 220c is input to RF transmitter 228c for transmission on Ant. 3,
and
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the coded phase shifted pilot sequence P2 output from code multiplier 220d is
input to RF transmitter 228d for transmission on Ant. 4.
[0040] The phase shifting performed by phase shifters 218a and 218b may
according to various alternatives, for example, as described for the phase
shifting performed in the embodiment of FIG. 1.
[0041] Referring now to FIG. 5, therein is a block diagram of portions of an
embodiment of a receiver 500 for use with the transmitter of FIG. 2. Receiver
500 comprises channel 1 and channel 2 estimate processing branch input 502a
and channel 3 and channel 4 estimate processing branch input 502b, channel
estimator 504, STTD demodulator 508, traffic signal input 510 and rake
combiner, deinterleaver and channel decoder 512.
[0042] The received pilot sequence P1 (ch1+ch2Q~) received on channels 1
and 2 from Ant. 1 and Ant. 2, respectively, of transmitter 200 is input to
input
502a. The received pilot sequence P2 (chi+ch4~) received on channels 3 and
4 from Ant. 3 and Ant. 4, respectively, of transmitter 200 is input to input
502b.
Channel estimator 504 performs channel estimation using, for example, a low
pass filter moving average function, and outputs combined estimate for
channels 1 and 2 (chest 1,2), and a combined estimate for channels 3 and 4
(chest 3,4). The channel estimates are then input to STTD demodulator 508,
which processes the received traffic signals from input 510 using the channel
estimates. The demodulated signal is then processed in rake, combiner,
deinterleaver and channel decoder 512 to generate the received symbols S1 S2.
FIG. 6 shows an embodiment of a rake finger of STTD demodulator 508 of FIG.
that utilizes chest1,2 and chest3,4 for demodulating the received traffic
signals.
[0043] In another embodiment for 4-antenna diversity, dedicated pilot
channels may be implemented in a WCDMA version of transmitter 150 of FIG.
1. Referring now to FIG. 3, therein is a block diagram of portions of a
dedicated
pilot channel STTD transmitter 300 according to a further embodiment of the
invention Transmitter 300 comprises input 318, block code processor 316,
channel symbol stream processing branch inputs 302a - 302d, antenna gain
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blocks 304a, - 304d, phase shifters 306a and 306b, phase shifter inputs 312a
and 312b, code multipliers 308a - 308d, code multiplier input 310, and
antennas
Ant. 1 - Ant 4.
[0044) Transmitter 300 of FIG. 3 is an implementation that uses dedicated
pilot channels that are transmitted by embedding pilot sequences in the
traffic
channel symbol stream. Input 318 and block code processor 316 function in the
same manner as input 126 and block code processor 124 of FIG. 1. In
transmitter 300, as symbols S1S2 are input to symbol stream processing
branch inputs 302a and 302b, pilot channel sequence U1 is input into inputs
302a and 302b multiplexed between the symbol sets of S 1 S2. Also, -S2*S 1 *
is
input to symbol stream processing branch inputs 302c and 302d, and pilot
channel sequence U2 is input onto inputs 302c and 302d, and multiplexed
between the symbol sets of -S2*S1 *. Another possibility is to define 4
different
dedicated pilot sequences, one for each transmit antenna
(0045] The multiplexed symbol streams at inputs 302a - 302d are then input
to antenna gain blocks 304a - 304d, respectively. Channel gain is applied in
antenna gain blocks 304a - 304d. The stream comprising S1 S2 and pilot
sequence U1 is output from antenna gain block 304a to code multiplier 308a.
The stream comprising S1S2 and pilot sequence U1 is output from antenna
gain block 304b to phase shifter 306a, where it is phase shifted according to
input from phase shifter control block 312a and then input to code multiplier
308b. The stream comprising -S2*S1 * and pilot sequence U2 is output from
antenna gain block 304c to code multiplier 308c, and the same stream, -S2*S1*
and pilot sequence, is output from antenna gain block 304d to phase shifter
306b, where it is phase shifted according to input from phase shifter control
block 312b and then input to code multiplier 308d. Code multipliers 308a -
308d multiply the appropriate stream by a scrambling code. The code
multiplied stream S1 S2 and pilot sequence U1 is then input to RF transmitter
314a for transmission on Ant. 1. The code multiplied phase shifted stream
S1 S2 and pilot sequence U1 is input to RF transmitter 314b for transmission
on
Ant. 2. The code multiplied stream -S2*S1* and pilot sequence U2 is input to
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RF transmitter 314c for transmission on Ant. 3, and the code multiplied phase
shifted stream -S2*S1* and pilot sequence U2 is input to RF transmitter 314d
for transmission on Ant. 4. RF transmitter 314a - 314d perform modulation and
carrier up conversions before transmitting the streams on Ant. 1 - Ant. 4. The
RF transmitters may perform of baseband pulse shaping, modulation, and
carrier up conversion. In some implementations one may choose to apply the
non-zero weighting after baseband pulse shaping and modulation.
[0046] The receiver of FIG. 5 may be modified for use with transmitter 300 of
FIG. 3. In this case, receiver 500 would function similarly but inputs 502a
and
502b would input U 1 (Ch 1 +Ch2fZ~) and U2(Ch3+Ch4~), respectively, to channel
estimator 504c.
[0047] In another embodiment for 4 antenna diversity, dedicated pilot
channels and common pilot channels may be implemented in a combined
embodiment. Referring now to FIG. 12, therein is a block diagram of portions
of
a dedicated/common pilot channel STTD transmitter 1200 according to another
embodiment of the invention.
[0048] Transmitter 1200 functions essentially in the same manner as
transmitter 300 of FIG. 3 with the exception being that common pilot channels
are added on Ant. 1 and Ant. 3. Common pilot channel sequences P1 and P2
are input to pilot sequence processing branch inputs 1218a and 1218b,
respectively. The pilot sequences are then separately processed through
antenna gain blocks 1220a and 1220b, and code multipliers 1222a and 1222b.
The coded outputs from code multipliers1222a and 1222b are then input to RF
transmitters 1214a and 1214c, respectively, of RF transmitter 1214. The RF
transmitters may perform baseband pulse shaping, modulation, and carrier up
conversions. In some implementations one may choose to apply the non-zero
weighting after baseband pulse shaping and modulation.
[0049] Transmitter 1200 of FIG. 12 provides non-hopped common pilot
channels on Ant. 1 and Ant. 3 and dedicated pilot channels on Ant. 1, Ant. 2,
Ant. 3, and Ant. 4. The pilot sequences may be multiplexed within one slot,
for
example in an embodiment where there are 15 slots in a transmission frame.
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Antenna gains may be set different for the common and dedicated control
channels, The antenna gains can also be time varying.
[0050] Referring now to FIG. 13, therein is a block diagram of portions of a
receiver 1300 for use with the transmitter of FIG. 12. Receiver 1300 comprises
channel 1 and channel 2 processing branch having inputs 1302a and 1302b,
and channel 3 and channel 4 processing branch having inputs 1302c and
1302d. Phase shifter input 1304, channel estimator 1306, STTD demodulator
1310, traffic signal input 312, and deinterleaver and decoder 1314.
[0051] The received pilot sequences P1, U1, P2, and U2 and input to inputs
1302a, 1302b, 1302c, and 1302d, respectively, of receiver 1300. Channel
estimator 1306 performs channel estimation using, for example, a low pass
filter
having average function, and outputs a combined estimate for channels 1 and 2
(chest 1,2) 1308a, and a combined estimate for channels 3 and 4 (chest 3,4)
1308b. The channel estimates are then input to STTD demodulator 1310,
which processes the received traffic signals from input 1312 using the channel
estimates. The demodulated signal is then processed in rake, combiner,
deinterleaver, and channel decoder 1314 to generate the received symbols S1,
S2.
[0052] A prior knowledge of the phase hopping may be used for power
control purposes. Referring now to FIG. 14, therein are shown portions of a
receiver for estimating power control, according to an embodiment of the
invention. Receiver 1400 includes channel estimator 1402, channel estimating
branch,inputs 1404a-1404d, phase shifter inputs 1408a and 1408b, phase
shifter 1406a and 1406b, channel estimate output 1410a and 1410b, squaring
blocks 1412a and 1412b, and power control processor 1414.
[0053] Channel estimator 1402 computes channel coefficients from the
common or dedicated channels from, for example transmitter 1200, for all four
antennas during a given slot "t". This may be a channel prediction for slot
t+1,
alternatively the channel estimate for slot t may be used in slowly fading
channels . These channel coefficients are denoted by chanest#1 (t),
chanest#2(t), chanest#3(t), and chanest#4(t) at inputs 1404a-1404d,
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respectively. For multiple rake fingers, e.g. chariest#1 (t) is a vector
channel
estimate corresponding to all rake fingers from Ant. 1.
[0054] Using the prior knowledge of phase hopping in phase shifter inputs
1408a and 1408b and knowledge of channel estimate for the current slot "t",
channel coefFicients for slot "t+1" are estimated:
chariest#12(t + 1) = chariest#1(t) + chariest#2(t)e~'2~'+'>
chariest#34(t + 1) = chariest#3(t) + chariest#4(t)e~3arr+~> (2)
Where ~12,~34are known a priority.
Received signal power estimate for slot (t+1 ) can be done based on chariest
#12(t+1 ) and chariest #12(t+1 );
rieceived - poweri(t + 1 ) = II chariest#12(t + 1)~IZ + "chariest#34(t + 1)IIZ
[0055] A power control command is generated by processor 1414 using the
received power estimate.
[0056] The method and apparatus of the invention may also be implemented
with diversity in the Walsh code domain. Referring now to FIG.7, therein is a
block diagram of portions of an space time spreading (STS) transmitter 700
according to an embodiment of the invention;
[0057] Transmitter 700 is a STS embodiment of transmitter 150 of FIG. 1a in
which the space time block processor perForms the transform in the Walsh
Code domain. The STS block code matrix used may be represented as:
S1W - S2*Wz where W = [W W ] W = [W -W ] (3)
S2W1 +Sl*WZ i i i z i i
[0058] As is done for the embodiment of FIG. 1a, each row of the matrix and
its phase shifted version are each transmitted on separate antennas Ant. 1 -
Ant. 4. The symbols S1 and S2 in each row are each transmitted
simultaneously over two symbol periods, rather than sequentially. Data
symbols are input to transmitter 700 at input 718 of channel coder 720.
Channel coder 720 codes, punctures, interleaves, and formats the input data
symbols. and outputs every other coder output symbol S1 as even data and
every other coder output symbol S2 as odd data. The even data is then
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processed through symbol repetition blocks 702a,b,e,f, Walsh function blocks
704b and 704d, Walsh multipliers 706a,b,e,f, summers 708a - 708d and
complex adders 710a and 710b. The odd data is processed through symbol
repetition blocks 702c,d,g,h, Walsh function blocks 704b and 704d, Walsh
multipliers 706c,d,g,h, summers 708a - 708d, and complex adders 71 Oa and
710b. The result at the output of complex adder 710a is the matrix row
S1W - S2*Wz and the result at the output of complex adder 710b is the matrix
row S2W +Sl*W2.
[0059] S1W -S2*W2 is then input to complex multiplier 712a to generate
fSIW, -S2*WZ}e'~1 and S2W, +Sl*Wz is input to complex multiplier 712b to
generate ~S2W +Sl*W2}e'~2 . S1W -S2*Wz is then input to RF transmitter 714a
for transmission on Ant. 1, ~S1W -S2*Wz~e'~' is input to RF transmitter 714b
for
transmission on Ant. 2, S2W, +Sl*WZ is input to RF transmitter 714c for
transmission on Ant. 3., and {S2W, +S1*WZ}e'~2 is input to RF transmitter 714d
for transmission on Ant. 4.
[0060] Referring now to FIG. 9, therein is shown a block diagram of portions
of an embodiment of a receiver 900 for use with transmitter 700 of FIG. 7.
Transmitter 700 comprises input 912, Walsh function blocks 902b and 902d,
Walsh multipliers 902a and 902c, channel multipliers 904a - 904d, complex
adders 906a and 906b, multiplexer (Mux) 908, and output 910. A received input
signal is received at input 912, is processed by the STS demodulator. The
pilot
channel transmission and channels estimation procedures can be same as
explained in STTD case. The channel estimates 904c and 904b can be same
as 412a, 412b from Fig. 4 for non-hopping common pilot channel case. For the
case of hopping common pilots or dedicated pilot transmission the channel
estimates can be obtained from channel estimation block 504 from Fig. 5.
These channel estimates are input to the STS demodulator in Fig. 9 as h1 and
h2. h1 corresponds to combined channel estimate from Ant. 1, Ant. 2 and h2
corresponds to channel estimate from Ant. 3, Ant 4. After STS demodulation
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using 902a,b,c,d and 904a,b,c,d, and 906a,b the output from 908 is the STS
demodulated signal to be sent to rake combiner, deinterleaver, and channel
decoder block 512 from Figs.
[0061] The proposed invention can also be implemented in an orthogonal
transmit diversity (OTD) embodiment of the invention. Referring now to FIG. 8,
therein is a block diagram of portions of an OTD transmitter 800 according to
an
embodiment of the invention. Transmitter 800 comprises input 822, channel
coder 820, symbol repetition blocks 802a - 802d, Walsh function blocks 804a
and
804b, Walsh multipliers 806a - 806d, complex adders 808a - 808b, complex
multipliers 810a and 810b, RF transmitters 812a - 812d. Transmitter is an
orthogonal transmit diversity (OTD) embodiment of transmitter 150 of FIG. 1 a
in
which the space time block processor performs the transform in the Walsh code
domain. The OTD block code matrix used may be represented as:
S1W_ where W, _ [W1 W, ] Wz = [W, - W, ] (4)
~~~z
[0062] As is done for the embodiment of FIG. 1 a, each row of the matrix and
its phase shifted version are each transmitted on separate antennas Ant. 1 -
Ant. 4. Data symbols are input to transmitter 800 at input 822 of channel
coder
820. Channel coder 820 codes, punctures, interleaves, and formats the input
data symbols and outputs every other coder output symbol S1 as even data and
every other coder output symbol S2 as odd data. The even data is then
processed through symbol repetition blocks 802a and 802b, Walsh function
block 804a, Walsh multipliers 806a and 806b, and complex adder 808a. The
odd data is processed through symbol repetition blocks 802c and 802d, Walsh
function block 804b, Walsh multipliers 806c and 806d, and complex adder
808b. The result at the output of complex adder 808a is S1W1 and the result at
the output of complex adder 808b is S2Wz . SIWi is then input to complex
multiplier 818a to generate fslW1}e'~I and SZWz is input to complex multiplier
818b to generate {S2Wz)e'~2. S1W is then input to RF transmitter 812a for
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transmission on Ant. 1, ~S1W, ~e'~' is input to RF transmitter 812b for
transmission on Ant. 2, SZWZ is input to RF transmitter 812c for transmission
on
Ant. 3, and fS2YY2~e'~2 is input to RF transmitter 812d for transmission on
Ant. 4.
[0063j Referring now to FIG. 10, therein is shown a block diagram of
portions of an embodiment of a receiver 1000 for use with transmitter 800 of
FIG. 8. Transmitter 800 comprises input 1010, Walsh function blocks 1002a
and 1002b, Walsh multipliers 1010a and 1010b, multipliers 1004a and 1004b,
multiplexers 1006 and output 1008. A received input signal is received at
input
912 is demodulated using a OTD demodulator 1000 using the knowledge of
channel coefficients h1* and h2*. The channel coefficients h1 and h2 for this
OTD block are derived in the same as explained in Fig4 and Figs. The OTD
demodulator 1000 is implemented using 1010, 1010a,b and 1012a,b and
1004a,b and 1006. The OTD demodulated output 1008 is sent to rake
combiner, deinterleaver, and channel decoder block 512 from Figs.
[0064] The embodiment of FIG. 7 may also be implemented in a TDMA
transmitter for operation in an EDGE system. Referring now to FIG. 11, therein
is a block diagram of portions of a long ST block code transmitter according
to
an embodiment of the invention. Transmitter 1100 comprises input 1118, 1120,
symbol stream processing branch inputs 1116a -1116d, time reversal blocks
1102 and 1104, complex conjugate blocks 1106a and 1106b, multiplier 1108,
phase multiplier 1110a and 1110b, phase multiplier control blocks 1112a and
1112b, and antennas Ant. 1, Ant. 2, Ant. 3, and Ant. 4. Channel coder 1120
codes, punctures, interleaves, and formats a symbol stream received at input
1118, Channel coder 1120 also splits the input symbol stream into odd and
even data streams. The even data stream is input to branch input 1116a and
RF transmitter 1122a for transmission on Ant. 1 during the first half of a
data
burst and the odd data stream is input to branch input 1116c and RF
transmitter
1112c for transmission on Ant. 2 during the first half of the data burst.
During
the second half of a burst, the even data stream is input to branch input
116b,
time received on time reversal block 1102, complex conjugated in complex
conjugate block 1106a and sent to RF transmitter 1122c for transmission on
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Ant. 3. The odd data stream is input to branch input 1116d, time reversed in
time reversal block 1104, complex conjugated in complex conjugate block
1106b, multiplied by a negative in multiplier 1108 and sent to RF transmitter
1122d for transmission on Ant. 4 during the second half of the data burst. A
training sequence SEQ1 is embedded in the middle of the burst transmit on Ant.
1 and a training sequence SEQ2 is embedded in the middle of burst transmit on
Ant. 2. Phase multipliers 1112a and 1112b, phase shifts the inputs to RF
transmitters 1122b and 1122d, using the multiplication blocks 1110a and 1110b
respectively. The output of phase multiplier 1112a is then input to RF
transmitter 1122b for transmission on Ant. 2 and the output of phase
multiplier
1112b is input to RF transmitter 11224 for transmission on Ant. 4. The RF
transmitters may perform of baseband pulse shaping, modulation, and carrier
up conversion. In some implementations one may choose to apply the phase
multiplication after baseband pulse shaping and modulation steps.
[0065] The phase rotation applied in phase multipliers 1122a and 1122b is
kept constant during the burst length, with the phase being changed on a burst
by burst basis. The phase can be chosen periodically or randomly from a
MPSK constellation as explained previously. In a preferred embodiment the
phase rotation on the Ant. 4 is kept same as phase rotation on ant2 with a 180
degree shift or multiplied by -1. The phase multiplication may be done before
or after the base band pulse shaping. In an alternative embodiment of FIG. 11
the transmission on Ant. 1 and Ant. 3 may be intercharged.
j0066] The transmitter shown FIG. 3 can also be applied to EDGE with some
modification. The space-time code described in 316 is applied blockwise
instead of symbol wise for an EDGE application. The block length can be
chosen as first half of the burst. In EDGE the length of.the first half and
second
half of the bursts are equal to 58 symbols. In this case S1 and S2 denotes a
block of symbols and ()* denotes time reversal of a block of symbols and
complex conjugation operation. S1* denotes the block of symbols S1 is time
reversed and complex conjugated. -S2* denotes that the block of symbols S2 is
time reversed, complex conjugated and multiplied by -1Ø The pilot sequences
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U1 and U2 can be chosen as two training sequences such as well-known
CAZAC sequences. The spreading codes 308a,b,c,d will not be applied in
EDGE. T,he phase multiplication blocks 306a and 306b are retained.
[0067] A receiver designed for a 2-antenna space time block code may be
used as a receiver for the embodiments of FIG. 1 or FIG. 2.
[0068] From the preceding description and embodiments, one skilled in the
art will realize that, although the method and apparatus of the present
invention
has illustrated and described with regard to particular embodiments thereof,
it
will be understood that numerous modifications and substitutions may be made
to the embodiments described, and that numerous other embodiments of the
invention may be implemented without departing from spirit and scope of the
invention as defined in the following claims.
