Note: Descriptions are shown in the official language in which they were submitted.
CA 02495356 1998-08-31
Transmitter Diversity Technique for Wireless Communications
Background of the Invention
This invention relates to wireless communication and, more particularly, to
techniques for effective wireless communication in the presence of fading and
other
1 o degradations.
The most effective technique for mitigating rnultipath fading in a wireless
radio channel is to cancel the effect of fading a~ the transmitter by
controlling the
transmitter's power. That is, if the channel conditions are known at the
transmitter
(on one side of the link), then the transmitter c2n pre-distort the signal to
overcome
15 the effect of the channel at the receiver (on the other side). However,
there are two
fundamental problems with this approach. The first problem is the
transmitter's
dynamic range. For the transmitter to overcome an x dB fade, it must increase
its
power by x dB which, in most cases, is not practical because of radiation
power
limitations, and the size and cost of amplifiers. The second problem is that
the
2o transmitter does not have any knowledge of the channel as seen by. the
receiver
(except for time division duplex systems, when the transmitter receives power
from
a known other transmitter over the. same chann~:l). Therefore, if one wants to
control a transmitter based on channel characteristics, channel information
has to be
sent from the receiver to the transmitter, which results in throughput
degradation
25 and added complexity to both the transmitter aad the receiver.
Other effective techniques are time and frequency diversity. Using time
interleaving together with coding can provide diversity improvement. The same
holds for frequency hopping and spread spectnan. However, time interleaving
results in unnecessarily large delays when the channel is slowly varying.
CA 02495356 1998-08-31
Equivalently, frequency diversity techniques are ineffective when the
coherence
bandwidth of the channel is large (small delay spread).
It is well known that in most scattering environments antenna diversity is the
most practical and effective technique for reducing the effect of multipath
fading.
The classical approach to antenna diversity is to use multiple antennas at the
receiver and perform combining (or selection) to improve the quality of the
received
signal.
The major problem with using the receiver diversity approach in current
wireless communication systems, such as IS~ 136 and GSM, is the cost, size and
~ o power consumption constraints of the receivers. For obvious reasons, small
size,
weight and cost are paramount. The addition of multiple antennas and RF chains
(or selection and switching circuits) in receivers is presently not be
feasible. As a
result, diversity techniques have often bean E.pplied only to improve the up-
link
(receiver to base) transmission quality with multiple antennas (and receivers)
at the
15 base station. Since a base station often serves thousands of receivers, it
is more
economical to add equipment to base station; rather than the receivers
Recently, some interesting approaches for transmitter diversity have been
suggested. A delay diversity scheme was proposed by A. Wittneben in "Base
Station Modulation Diversity for Digital SIMULCAST," Proceeding of the 1991
2o IEEE Vehicular Technology Conference (V'l.'C 41 st), PP. 848-853, May 1991,
and
in "A New Bandwidth Efficient Transmit Antenna Modulation Di~rersity Scheme
For Linear Digital Modulation," in Proceeding of the 1993 IEEE International
Conference on Communications (IICC '93), PP.1630-1634, May 1993. The
proposal is for a base station to transmit a sequence of symbols through one
25 antenna, and the same sequence of symbols --but delayed - through another
antenna.
U.S. patent 5,479,448, issued to Nambirajan Seshadri on December 26,
1995, discloses a similar arrangement where a sequence of codes is transmitted
through two antennas. The sequence of codes is routed through a cycling switch
that directs each code to the various antenna;, in succession. Since copies of
the
CA 02495356 1998-08-31
3
same symbol are transmitted through multiple antennas at different times, both
space and time diversity are achieved. A maxirlum likelihood sequence
estimator
(MLSE) or a minimum mean squared error (Mr~ISE) equalizer is then used to
resolve multipath distortion and provide diversity gain. See also N. Seshadri,
J.H.
Winters, "Two Signaling Schemes for Improving the Error Performance of FDD
Transmission Systems Using Transmitter Antetuia Diversity,' Proceeding of the
1993 IEEE Vehicular Technology Conference (VTC 43rd), pp. 508-511, May 1993;
and J. H. Winters, "The Diversity Gain of Transmit Diversity in Wireless
Systems
with Rayleigh Fading," Proceeding of the 1994 ICClSUPERCOMM, New Orleans,
1o Vol. 2, PP. 1121-1125, May 1994.
In still another interesting approach
symbols are encoded according to the antennas through which they are
simultaneously transmitted, and are decoded using a maximum likelihood
decoder.
s
More specifically, the process at the transmitter handles the information in
blocks of
Ml bits, where M1 is a multiple ofM2, i.e., M1=k*M2. It converts each
successive
group of M2 bits into information symbols (generating thereby k information
symbols), encodes each sequence of k information symbols into n channel codes
(developing thereby a group of n channel codes for each sequence of k
information
symbols), and applies each code of a group of codes to a different ~utenna.
Summary
The problems of prior art systems are cwercome, and an advance in the art is
realized with a simple block coding arrangement where symbols are transmitted
over a plurality of transmit channels and the coding comprises only simple
arithmetic operations, such as negation and conjugation. The diversity created
by
the transmitter utilizes space diversity and eitl~~er time diversity or
frequency
diversity. Space diversity is effected by redundantly transmitting over a
plurality of
CA 02495356 1998-08-31
antennas; time diversity is effected by redundantly transmitting at different
times;
and frequency diversity is effected by redun~3antly transmitting at different
frequencies. Illustratively, using two transn-.at antennas and a single
receive
antenna, one of the disclosed embodiments provides the same diversity gain as
the
maximal-ratio receiver combining (MRRC) scheme with one transmit antenna and
two receive ~ntennas_ The novel approach d oes not require any bandwidth
expansion or feedback from the receiver to the transmitter, and has the same
decoding complexity as the MRRC. The diversity improvement is equal to
applying
maximal-ratio receiver combining (MRRC) at the receiver with the same number
of
~0 antennas. The principles of this invention are applicable to arrangements
with more
than two antennas, and an illustrative embodiment is disclosed using the same
space
block code with two transmit and two receive antennas. This scheme provides
the
same diversity gain as four-branch MRRC.
15 Brief Description of the Drawings
FIG. 1 is a block diagram of a first embodiment in accordance with the
principles of this invention;
FIG. 2 presents a block diagram of a second embodiment, where channel
estimates are not employed;
20 FIG. 3 shows a block diagram of a third embodiment, where channel
estimates are derived from recovered signals; and
FIG. 4 illustrates an embodiment wh~;re two transmitter antennas and two
receiver antennas are employed
25 Detail Description
In accordance with the principles of this invention, effective communication
is achieved with encoding of symbols that comprises merely negations and
conjugations of symbols (which really is merely negation of the imaginary
part) in
CA 02495356 1998-08-31
S
combination with a transmitter created diversity. Space diversity and either
frequency diversity or time diversity are employed.
FIG. 1 presents a block diagram of an Furangement where the two
controllable aspects of the transmitter that are 'used are space and time.
That is, the
FIG. 1 arrangement includes multiple transmitter antennas (providing space
diversity) and employs multiple time intervals. Specifically, transmitter 10
illustratively comprises antennas 11 and 12, and it handles incoming data in
blocks
n symbols, where n is the number of transmitter antennas, and in the
illustrative
embodiment of FIG. 1, it equals 2, and each block takes n symbol intervals to
1o transmit. Also illustratively, the FIG. 1 arranl;ement includes a receiver
20 that
comprises a single antenna 21.
At any given time, a signal sent by a t~~ansmitter antenna experiences
interference effects of the traversed channel, which consists of the transmit
chain,
the air-link, and the receive chain. The channel may be modeled by a complex
is muItiplicative distortion factor composed of a. magnitude response and a
phase
response. In the exposition that follows therefore, the channel transfer
function
from transmit antenna 11 to receive antenna ::1 is denoted by ho and from
transmit
antenna 12 to receive antenna 21 is denoted t~y h, , where:
ho = aoe~a°
20 h, = a~e''~' .
(1)
Noise from interference and other sources is added at the two received signals
and,
therefore, the resulting baseband signal received at any time and outputted by
reception and amplification section 25 is
25 r(t)=aae~°sl +a~e'~'sl +n(t),
(2)
where s, and s~ are the signals being sent b;y transmit antenna 11 and 12,
respectively.
CA 02495356 1998-08-31
6
As indicated above, in the two-antenna embodiment of FIG. 1 each block
comprises two symbols and it takes two symbol intervals to transmit those two
symbols. More specifically, when symbols s; 2nd s~ need to be transmitted, at
a first
time interval the transmitter applies signal s; to antenna 11 and signal s~ to
antenna
12, and at the next time interval the transmitter applies signal - s, * to
antenna 11
and signal so *., to antenna 12. This is clearly a very simple encoding
process where
only negations and conjugations are employed. As demonstrated below, it is as
effective as it is simple. Corresponding to the above-described transmissions,
in the
first time interval the received signal is
to r(t) = hos, + h,s~ + n(t),
(3)
and in the next time interval the received sign,il is
r(t+T) _-laos~ *+h,s; *+n(t+T).
(4)
Table 1 illustrates the transmission pattern over the two antennas of the FIG.
1
arrangement for a sequence of signals { so , s, , sz , s3 , s, , ss ,... } .
Table 1
Time: t t+T t+27' t+3T t+4T t+ST
Antenna * * * .....
11 SO sl s1 S3 S4 SS
Antenna s~ so s3 sz ss s4 .....
12 * * *
The received signal is applied to channel estimator 22, which provides
signals representing the channel characteristi~a or, rather, the best
estimates thereof.
2o Those signals are applied to combiner 23 and to maximum likelihood detector
24.
The estimates developed by channel estimator 22 can be obtained by sending a
known training signal that channel estimator 22 recovers, and based on the
recovered signal the channel estimates are computed. This is a well known
approach.
CA 02495356 1998-08-31
7
Combiner 23 receives the signal in the first time interval, buffers it,
receives
the signal in the next time interval, and combines the two received signals to
develop signals
s; =ho*r(t)+h,r*(t+T)
s~ =h, *r(t)-hor*(t+T)_
(5)
Substituting equation (1) into (5) yields
s; =(ao +a; )s;+ho*n(t)+h,n*(t+T)
s~ =(ao +a; )s~ -hon*(t+T)+h, *n(t),
io (6
)
where ao = hoho * and a; = h, h, * , demonstrating that the signals of
equation (6)
are, indeed, estimates of the transmitted sign,ils (within a multiplicative
factor).
Accordingly, the signals of equation (6) are : ent to maximum likelihood
detector 24.
~5 In attempting to recover s,, two kind of signals are considered: the
signals
actually received at time t and t+T, and the signals that should have been
received if
s; were the signal that was sent. As demonstrated below, no assumption is made
regarding the value of s~ . That is, a decision is made that s, = s,~ for that
value of x
for which
2o dZ[r(t),(hosX+hts~)]+dz[r(t+T~,(-h,s~*+hasr*)]
is less than
dz[r(t),(hosk +h~sj)]+d2[r(t+T),1;-h,st *+hosk*)],
where d s (x, y) is the squared Euclidean distance between signals x and y,
i.e.,
Zs dI(X,y)=~r-y~?.
CA 02495356 1998-08-31
8
Recognizing that ho = ho +nnise that i~~ independent of the transmitted
symbol, and that h, = h, +noise that is independent of the transmitted symbol,
equation (7) can be rewritten to yield
2 2 x ~ x x 2 ~
(a~ +a~ ~S~I -SjSs -S~ *Sx < (a~ -1-CC~ ~Skl -S~Sk -Sr * Sk
(g) r
where ao = haho * and a; = h,h, * ; or equiv alently,
~au+ai _l~s~~x+dx~snsX~<~ao+ai -I~SkIx+dx~s;~sk~.
(9)
In Phase Shift Keying modulation, all symbols carry the same energy, which
~o means that Isxlx = Isk Ix and, therefore, the decision rule of equation (9)
may be
simplified to
choose signal s, = sx iff d x ~s" sx ) <_ ~~x (s,,sk ) .
(10)
Thus, maximum likelihood detector 24 develops the signals sk for all values of
k,
15 with the aid of ho and h, from estimator 22, develops the distances d x (s;
, sk ) ,
identifies x for which equation (10) holds an3 concludes that s, = s,~. A
similar
process is applied for recovering s j .
In the above-described embodiment each block of symbols~is recovered as a
block with the aid of channel estimates ho arid h, . However, other approaches
to
2o recovering the transmitted signals can also be employed. Indeed, an
embodiment
for recovering the transmitted symbols exist;> where the channel transfer
functions
need not be estimated at all, provided an initial pair of transmitted signals
is known
to the receiver (for example, when the initial -pair of transmitted signals is
prearranged). Such an embodiment is shown in FIG. 2, where maximum likelihood
25 detector 27 is responsive solely to combiner 26. (Elements in FIG. 3 that
are
CA 02495356 1998-08-31
9
referenced by numbers that are the same as reference numbers in FIG. 1 are
like
elements.) Combiner 26 of receiver 30 develops the signals
ro =r(t)=hoso +h,s, +rto
r, =r(t+T~=h,so *-hos, *+n,
ri =r(t+2T~=hosZ +h,s~ +ni
r3 =r(t+3T7=hisZ *-~tpS3 *+ns,
(1
1)
then develops intermediate signals A and B
A = ror3 * -r2r,
B = riro * +r,r3 * ,
(12)
and finally develops signals
s2 = As, * +Bso
s3 = -Aso * +Bs, ,
(13)
where N3 and Nt are noise terms. It may be noted that signal ri is actually
rz = host + h,s3 = host + h,s3 + nz, and similf~rly for signal r3 . Since the
makeup of
signals A and B makes them also equal to -
2o A=(ao +a~ Xs~s, -s,so)+N,
B=(ao +a; )(s~so *+s;s,*)+N~,
(14)
where N 1 and N2 are noise terms, it follows that signals s2 and s3 are equal
to
sZ = (ao + a; )(~so ~2 + ~s, ~2 )sz + N3
s, _ (ao + a~ )~so h + Is, IZ )s~ + N4 .
(15)
CA 02495356 1998-08-31
When the power of all signals is constant (and normalized to 1) equation (15)
reduces to
s~ _ (ao + a; )sz + N3
s; _ (ao + a; )s3 + N, .
5 (16)
Hence, signals s2 and s3 are, indeed, estimatfa of the signals s2 and s3
(within a
multiplicative factor). Equation (15) dern~nstrates the recursive aspect
of equation
(13), where signal estimates s~ and s3 are ev,iluated with the aid of
recovered
signals sa and s, that are fed back from the output of the maximum likelihood
1 o detector.
Signals s2 and s3 are.applied to maximum likelihood detector 24 where
recovery is effected with the metric expressed by equation (10) above. As
shown in
FIG. 2, once signals sz and sz are recovered, they are used together with
received
signals r~ , r3 , r4 , and rs to recover signals s4 and s s , and the process
repeats.
is FIG. 3 depicts an embodiment that does not require the constellation of the
transmitted signals to comprise symbols of edual power. (Elements in FIG. 3
that
are referenced by numbers that are the same ;is reference numbers in FIG. 1
are like
elements.) In FIG. 3, channel estimator 43 of receiver 40 is responsive to the
output
signals of maximum likelihood detector 42. Having access to the recovered
signals
2o so and s, , channel estimator 43 forms the estimates
_ roso *-r,si _ so *no +s~n,
ISOIZ +IslI2 ~ ~0I2 +~S~I2
ros, *-r,so s, *no +son,
he = Isolz +'n'Iz =~ + IsoIZ +Isch
(1'~
and applies those estimates to combiner 23 and to detectoi 42. Detector 24
recovers
25 signals s~ and s3 by employing the approach used by detector 24 of FIG. 1,
except
CA 02495356 1998-08-31
11
that it does not employ the simplification of equation (9). The recovered
signals of
detector 42 are fed back to channel estimator ~.3, which updates the channel
estimates in preparation for the next cycle.
The FIGS. 1-3 embodiments illustrate the principles of this invention for
arrangements having two transmit antennas anal one receive antenna. However,
those principles are broad enough to encompass a plurality of transmit
antennas and
a plurality of receive antennas. To illustrate,1~IG. 4 presents an embodiment
where
two transmit antennas and two receive antennas are used; to wit, transmit
antennas
31 and 32, and receive antennas S1 and S2. The signal received by antenna S1
is
t0 applied to channel estimator S3 and to combir~er SS, and the signal
received by
antenna 52 is applied to channel estimator 54 and to combiner 55. Estimates of
the
channel transfer functions ho and h, are applied by channel estimator 53 to
combiner SS and to maximum likelihood detector 56. Similarly, estimates of the
channel transfer functions hz and h3 are appl ied by channel estimator 54 to
combiner 55 and to maximum likelihood detector S6. Table 2 defines the
channels
between the transmit antennas and the receivE: antennas, and table 3 defines
the
notion for the received signals at the two receive antennas.
Table :?
Antenna Antenna
'~ 1 52
Antenna ho hi
31
Antenna h, hj
32
Table 3
Antenna Antenna
'i 1 52
Time t
ro r2
Time t+T
r, r3
Based on the above, it can be shown that the received signals are
ro = hoso + h,s~ + no
r, =-hos, *+h,sp *+n,
CA 02495356 1998-08-31
12
rZ = h2so + has, + n2
r3 = -h2s, * +h3so * +n3
(18)
where no,n~,n" and n3 are complex random variables representing receiver
thermal noise, interferences, etc.
In the FIG. 4 arrangement, combiner S'_. develops the following two signals
that are sent to the maximum likelihood detector:
so=ho*ro+h, r,*+h2*rz+h~r3*
si = h, * ro - hors * +h3 * rz - h2r3 * .
to (19)
Substituting the appropriate equations results i:n
so = (ao +a; +a2 +a3 )so +ho * np + h,n, *+h2 * n2 +h3n3
s~ _ (ao + a; + aZ + a; )s, + h, * no - Jan, * +h3 * n~ - h~ ns * ,
(~0)
which demonstrates that the signalsso and s~ are indeed estimates of the
signals sa
and si. Accordingly, signals so and s~ are sent to maximum likelihood decoder
56,
which uses the decision rule of equation (10) to recover the signals so and
s,.
As disclosed above, the principles of tlus invention rely on the transmitter
to
force a diversity in the signals received by a receiver, and that diversity
can be
2o effected in a number of ways. The illustrated embodiments rely on space
diversity -
effected through a multiplicity of transmitter antennas, and time diversity -
effected
through use of two time intervals for transmitting the encoded symbols. It
should be
realized that two different transmission frequencies could be used instead of
two
time intervals. Such an embodiment would double the transmission speed, but it
would also increase the hardware in the receiver, because two different
frequencies
need to be received and processed simultaneo,.~sly.
CA 02495356 1998-08-31
13
The above illustrated embodiments are., obviously, merely illustrative
implementations of the principles of the invent ion, and various modifications
and
enhancements can be introduced by artisans without departing from the spirit
and
scope of this invention, which is embodied in the following claims. For
example,
all of the disclosed embodiments are illustrated for a space-time diversity
choice,
but as explained above, one could choose the space-frequency pair. Such a
choice
would have a direct effect on the construction of the receivers.
