Note: Descriptions are shown in the official language in which they were submitted.
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INGRESS SUPPRESSION FOR COMMUNICATION SYSTEMS
This invention relates to a method for use in a receiver of a
communication system for suppressing one or several narrow band interferences
present in a communication channel
BACKGROUND OF THE INVENTION
Narrow band interferences also referred to as Ingress are often
present in the return path of cable systems. Ingress noise enters the cable
system through defective shields, and couples with the signal. It consists of
narrow band signals with power as high as +10 dB over the carrier power and
bandwidth typically less than 20 KHz. Sources of ingress include ham radio
transmissions and therefore can be intermittent. Ingress may slowly drifts in
frequency. Channel impairments also include channel amplitude and phase
= distortions, burst / impulse noises, and thermal noises.
Cable Modems (CMs) use the return path to transmit short signals,
referred to as bursts, to the Cable Modem Termination System (CMTS) located at
the head end. The Media Access Control (MAC) layer of the CMTS uses Time
Division Multiple Access (TDMA) or Synchronous Code Division Multiple Access
(S-CDMA) to control the access to the return path by the CMs, and avoid burst
collisions. Transmissions occur inside time slots. The CMs must start
transmitting
their bursts early enough so they reach the CMTS at the time slots that were
allocated to them. The CMTS synchronizes to the incoming bursts prior to
extracting the data. Synchronization is facilitated by means of a preamble.
The
preamble represents a sequence of symbols that is known to the CMTS.
To facilitate inter-operability between CMTS and CMs, a Standard
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known as Data Over Cable Service Interface Specification (DOCSIS) and which
is employed by many cable operators, was developed to define the
communication protocols between CMTS and CMs. In DOCSIS, two modes of
operations are used in the return path: ranging and traffic modes. Ranging is
used when a CM joins the network and also for periodic maintenance. The
process of ranging is similar for TDMA and S-CDMA modes, except that timing
synchronization requires higher accuracy in S-CDMA. If TDMA is selected, a CM
which has ranged successfully enters the traffic mode to transmit TDMA bursts.
Similarly, if S-CDMA is selectad, a CM which has ranged successfully enters
the
traffic mode to transmit S-CDMA bursts.
Different preambles are used in ranging and traffic modes. In
ranging mode the preamble is inserted at the beginning of the burst, and
includes
a sync sequence and a training sequence. The sync sequence is used by the
CMTS to detect the start of the burst, recover timing, measure the signal
level,
and correct for phase and frequency offsets. The training sequence is used for
equalizing the channel. During the ranging mode, the CMTS processes the
received burst transmitted by the CM to both measure the CM synchronization
parameters (i.e., power level, carrier frequency and timing offsets), and
compute
the coefficients of the CM pre-equalizer. Following processing of the burst,
the
CMTS transmits the information back to the CM using the downstream channels,
so the CM can adjust its synchronization parameters and configure its pre-
equalizer before transmitting a new burst in the return path (i.e., upstream
channels). Pre-equalization at the CM allows to cancel the effect of the
channel
at the CMTS. The CMTS informs the CM to operate in traffic mode when the
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received burst is determined to be of sufficient quality for communicating in
TDMA or S-CDMA, whichever modes has been selected. CMs periodically return
to ranging for adjustments of the pre-equalizer coefficients and/or
synchronization
parameters.
In TDMA traffic mode, the preamble is inserted at the beginning of
the burst, and consists of a sync sequence that is used to estimate residual
timing offset, phase offset and signal level. In S-CDMA traffic mode, the
preamble
Is interleaved with the data, and is used to estimate phase offset and signal
level.
Since ingress does not depend on which CM transmits, it is natural
to suppress the narrow band interferences (i.e., ingress) at the CMTS, so in
the
receiver. There are several methods to suppress the narrow band interferences.
One method uses a prediction filter to extract the interferences so they can
be
subtracted from the communication signal. A prediction filter is an adaptive
FIR
notch filter whose coefficients can be adjusted using a gradient algorithm
such as
for example the Least Mean Square (LMS) algorithm. The prediction filter
coefficients are usually adjusted when no CM transmits so only ingress noise
and
broadband noises are present in the channel, or in other words, there is no
communication signal. The coefficients are kept unchanged when the CMs
transmit. There is provision in the DOCSIS Standard to reserve time slots when
no CM transmits to allow adjustments of the prediction filter coefficients.
The suppression of interferences using a prediction filter when the
communication signal is present causes distortions to the communication
signal.
A Decision Feedback Equalizer (DFE) can be used to compensate for the
distortions introduced by the prediction filter.
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An alternative technique to suppress ingress noise is to use an
Infinite Impulse Response (IIR) notch filter instead of an FIR notch filter.
IIR notch
filters can filter interferences with significantly less coefficients than
their FIR
counterparts.
IIR notch filters can cause severe distortions to the communication
signal, while suppressing the interferences. As in the case of an FIR filter,
a DFE
can be used to compensate for the signal distortions introduced by the notch
filter, as described in the paper by G.Redaelli, et al. :"Advanced Receiver to
Dip
Ingress Noise in HFC Return Channel", ISPACS 2000 conference proceedings.
The cascade of an IIR/FIR notch filter and a DFE is an effective
solution to suppress narrow band interferences but has two main problems.
Problem 1: Synchronization to the burst signal is required for the DFE to
properly
operate, however high-power interferences mask the burst signal and must be
notched in order to synchronize. Notching the interferences causes severe
distortions to the burst, which adds to the distortions caused by the channel.
This
plurality of distortions renders synchronization very complex to achieve.
Problem
2: The DFE simultaneously compensates for both channel distortions and
distortions caused by the notch filter. This is expensive to implement as
different
DFE coefficients are needed for each CM, since channel distortions depend on
which CM transmits. Also, the DFE coefficients must be re-computed for each
CM every time the interferences change, which adds significant overhead to the
system.
An effective approach to solve problem 1 is to "build" a notch filter
that minimizes distortions to the communication signal in order to
synchronize.
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Distortions introduced by an IIR notch filter are mostly phase distortions. A
method was proposed to build a zero phase shift IIR notch filter (i.e., a
filter which
only causes amplitude distortions to the signal) in the paper by J. J.
Kormylo, et
al., "Two-Pass Recursive Digital Filter with Zero Phase Shift", IEEE
transactions
5 on acoustics, speech, and signal processing, October 1974. As described in
Kormylo's paper, a zero phase shift 1IR notch filter can be built from a
conventional IIR notch filter by performing a two-pass filtering of the
received
burst signal through the conventional IIR notch filter, the first pass
occurring in
the forward direction as the burst signal is received and the second pass in
the
time reverse direction after receiving the full burst and storing the notch
filter
output in a memory so it can be read backward from the memory. The output of
the second-pass filtering is also stored in memory, since it has to be read
backward again for processing the burst. This method is quite expensive and
time consuming since it requires two times buffering of the complete burst
before
the burst can be processed to recover the data.
An effective approach to solve problem 2 is to isolate / decouple
channel equalization from the suppression of interferences. A solution was
proposed in the case of a prediction filter in cascade with a DFE in US patent
7843847 (Quigley) issued November 2010 to Broadcom. The coefficients of the
prediction filter are adjusted when no CM transmits so only when narrow band
interferences are present in the channel. A DFE comprises a feedforward
equalizer and a feedback equalizer. The feedback equalizer is used to
compensate for the distortions introduced by the prediction filter. This is
achieved
by setting the feedback equalizer coefficients equal to those of the
prediction filter
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coefficients. The feedforward equalizer is solely used to compensate for the
channel distortions. The coefficients of the feedforward equalizer are
adjusted
during the ranging mode, while feedback equalizer and prediction filter
coefficients are kept unchanged. This mechanism allows to decouple channel
equalization from ingress noise removal in the case of a FIR notch filter in
cascade with a DFE. The coefficients of the feedforward equalizer are sent to
the
CM for configuration of its pre-equalizer.
The following references are relevant to this field:
US Patent 7843847 (Quigley) issued November 2010 to Broadcom
Corporation discloses a number of features for enhancing the performance of a
communication system, in which data is transmitted between a base station and
a plurality of subscriber statirns located different distances from the base
station.
The power transmission level, slot timing, and equalization of the subscriber
stations are set by a ranging process. Data is transmitted by the subscriber
stations in fragmented form. Various measures are taken to make transmission
from the subscriber stations robust. The uplink data transmission is
controlled to
permit multiple accesses from the subscriber stations
US Patent 7826569 (Popper) issued May 2010 to Juniper
Networks, Inc. discloses a device for reducing ingress noise in a digital
signal,
comprising a noise predictor for predicting an amount of ingress noise in the
digital signal based on past samples of the ingress noise, and a subtractor
for
subtracting the predicted amount of ingress noise from the digital signal.
Channel
distortion is compensated for by a noise-independent equalizer, such as a ZF
equalizer, placed upstream of the noise predictor. The device may be
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incorporated, for example, in a cable modem termination system (CMTS) of an
hybrid fiber/coax (HFC) network.
US Patent 771 671 2 (Booth) issued May 2010 to General Instrument
Corporation discloses an adaptive data stream filter which removes narrowband
interference from the CATV return path prior to these paths being combined in
the network. This provides a method for removing narrowband interference from
the CATV return path which detects potential narrowband interference in real-
time and adapts a filter to remove this potential narrowband interference. The
method uses previously created filters that are combined based on detected
interference in an adaptive manner to continually adapt to new interference
sources. Another embodiment calculates new filter coefficients for the data
stream filter based on detected interference. In another embodiment, two
filters
are operated in a ping-pong manner for each band of interference identified as
above threshold. This enables updating of one filter while another filter is
performing the data stream filter operation
US Patent 6360369 (Mahoney) issued March 2002 to Broadcom
Corporation discloses a system for the removing of interference (ingress) in
cable
modems without reducing the data rate or changing the frequency of the signal.
A
variable band stop filter bank and a non-linear equalization system are used.
The
band stop filter bank removes ingress while the equalization system,
comprising
an inner and an outer equalizer, removes the distortion created by the band
stop
filtering. A spectrum monitor detects both the presence of ingress and its
frequency, and then feeds the data to a digital signal processor which
calculates
the distortion removal settings for the equalizers.
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US Patent 6285718 (Reuven) issued September 2001 to Orckit
Communication Ltd discloses an apparatus for transmission of high speed data
over communication channels including a modulator which modulates an
outgoing stream of digital data to generate an outgoing signal, and a
demodulator
which demodulates an incoming signal to generate an incoming stream of digital
data, wherein the modulator comprises a band suppressor for suppressing
portions of the outgoing signal which have specified frequencies.
G. Redaelli, et al.,"Advanced receiver to dip ingress noise in HFC
return channel", 1SPACS 2000 conference proceedings, 2000
J. J. Kormylo, et al., 'Two-pass recursive digital filter with zero
phase shift", IEEE transactions on acoustics, speech, and signal processing,
October 1974
M. D. Macleod, "Fast nearly ML estimation of the parameters of real
or complex single tones or resolved multiple tones", IEEE Transactions on
Signal
Processing, January 1998
A.V. Oppenheim, and, R.W. Schafer, "Discrete-time signal
processing", Prentice-Hall, 1st ed, 1989, chap. 10
SUMMARY OF THE INVENTION
According to the present invention there is provided a method for
use in a receiver of a communication system for suppressing one or more narrow
band interferences present in a communication channel, the method comprising:
receiving at the receiver a communication signal from the
communication channel;
filtering in the communication signal each narrow band interference
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to be suppressed in turn with a respective one of a plurality of Infinite
Impulse
Response (IIR) notch filters configured to suppress a respective one of the
narrow band interferences;
providing a Decision Feedback Equalizer (DFE) to provide
compensation for distortions introduced by the plurality of IIR notch filters,
wherein equalization of the communication channel is isolated or decoupled
from
the suppression of the narrow band interferences;; and
providing for each of the plurality of IIR notch filters prior to said
DFE a respective one of a plurality of IIR all-pass filters devised to
compensate
for distortions introduced by a respective notch filter, to minimize the
distortions
introduced by each of the plurality of IIR notch filters to the communication
signal
prior to said DFE.
Preferably the communication channel is shared by a plurality of
transmitters by using Time Division Multiple Access (TDMA) scheme.
Preferably there is provided an algorithm to detect the narrow band
interferences in the communication channel, and configure the plurality of IIR
notch filters to suppress the detected narrow band interferences.
Preferably the algorithm comprises Fast Fourier Transforms,
frequency interpolations, filtering, down-sampling, and up-sampling
operations.
Preferably the method includes isolating or decoupling equalization
of the communication channel from suppression of the respective one of the
narrow band interferences.
Preferably the communication system is a Cable Television (CATV)
network having a plurality of Cable Modems for communicating data in a return
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to
path and the method operates on the return path to suppress narrow band
interferences that enter the return path through defective shields.
According to a second aspect of the invention there is provided a
method for use in a receiver of a communication system for suppressing one or
more narrow band interferences present in a communication channel, the method
comprising:
receiving at the receiver a communication signal from the
communication channel;
filtering in the communication signal each narrow band interference
to be suppressed in turn with a respective one of a plurality of Infinite
Impulse
Response (IIR) notch filters configured to suppress a respective one of the
narrow band interferences;
providing a Decision Feedback Equalizer (DFE) to provide
compensation for distortions introduced by the plurality of IIR notch filters,
wherein a training sequence is used for equalizing the communication channel;
and
providing for each of the plurality of IIR notch filters prior to said
DFE a respective one of a plurality of 111; all-pass filters devised to
compensate
for distortions introduced by a respective notch filter, to minimize the
distortions
introduced by each of the plurality of IIR notch filters to the communication
signal
prior to said DFE.
Preferably the communication signal is distorted using a pre-
equalizer of a transmitter to cancel effects of the communication channel at
the
receiver, and the adaptive equalizer in the receiver is used to compute
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coefficients of the pre-equalizor of the transmitter.
Preferably the communication system is a Cable Television (CATV)
network having a plurality of Cable Modems for communicating data in a return
path and the method operates on the return path to suppress narrow band
interferences that enter the return path through defective shields.
Preferably the communication system is a Cable Television (CATV)
network having a plurality of Cable Modems for communicating data in a return
path and the method operates on the return path to suppress narrow band
interferences that enter the return path through defective shields.
Preferably the communication channel is shared by a plurality of
transmitters by using Time Division Multiple Access (TDMA) scheme and wherein
a Decision Feedback Equalizer (DFE) is used to compensate distortions
introduced by the plurality of IIR notch filters not compensated by the
plurality of
IIR all-pass filters. .
Preferably the communication channel is shared by a plurality of
transmitters by using Synchronous Code Division Multiple Access (S-CDMA)
scheme.
Preferably there is provided an algorithm to detect the narrow band
interferences in the communication channel, and configure the IIR notch
filters to
suppress the detected narrow band interferences.
Preferably the algorithm comprises Fast Fourier Transforms,
frequency interpolations, filtering, down-sampling, and up-sampling
operations.
BRIEF DESCRIPTION OF THE DRAWINGS
Fig. 1 is a block diagram of a preferred embodiment of the invention
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which provides a method to suppress the narrow band interferences in the
receiver during the TDMA / S-CDMA ranging modes and the TDMA traffic mode
including components arranged to detect the interferences present in the
channel
during idle times, synchronize to the incoming bursts during the TDMA / S-CDMA
ranging modes, compute the pre-equalizer coefficients during the TDMA / S-
CDMA ranging modes, recover the data during the TDMA / S-CDMA ranging
modes, synchronize to the incoming bursts during the TDMA traffic mode, and
recover the data during the TDMA traffic mode.
Fig. 2 is a block diagram of a preferred embodiment of the
Programmable Interference Suppressor (PIS) block shown in Fig. 1. The PIS
block consists of a cascade of N Interference Suppressor Filter (ISF) blocks,
respectively labeled asisFi ISF2 ISFN.
Fig 3 is a block diagram of a preferred embodiment of the
Interference Suppressor Filter (1SF) blocks shown in Fig. 2.
Fig. 4 is a block diagram of a preferred embodiment of the IIR All-
pass filter block shown in Fig. 3.
Fig. 5 is a block diagram of a preferred embodiment of the invention
which provides a method to suppress the narrow band interferences during the S-
CDMA traffic mode including components arranged to extract the preambles after
de-spreading the incoming S-CDMA burst, and then synchronize, and recover the
data.
DETAILED DESCRIPTION
Referring to the block diagram in Fig. 1, the input indicated at Input
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signal 10, represents the received digital Quadrature Amplitude Modulated
(QAM) baseband signal at the sampling rate, where the sampling rate is
typically
4 times the symbol rate. This signal is normally obtained by sampling the RF
received signal and performing filtering and down-conversions operations which
are not shown in Fig. 1. Input signal 10 is fed to the Interference Detection
(ID)
block 11 and Programmable Interference Suppressor (PIS) block 12 by means of
Mux indicated at 13. The ID block 11 is active when no CM transmits and the
channel is idle. There is provision in the DOCSIS Standard to prevent CMs from
transmitting during arbitrary period of times. These periods of times are
referred
to in the standard as idle times (see DOCSIS 3ب MAC and Upper Layer
Protocols interface Specifications).
At the start of an idle time, the ID block 11 processes the input
signal to estimate center frequencies, bandwidths and powers of the
interferences present in the channel. No CM transmits during an idle time, so
the
input signal only consists of the interferences. Detection and
characterization of
the interferences are performed using an algorithm described as follows. It is
an
iterative algorithm wherein one of the interferences is detected at a time,
then an
IIR notch filter in PIS 12 is configured to filter the received signal and
suppress
the detected interference, then a power measurement of the filtered signal is
performed to determine whether interferences are still present and if it is
the
case, the search for one of the remaining interferences is started again on
the
filtered signal. Detection and characterization of an interference are
performed as
follows. It is a multi-stage process. In the first stage, several FFTs of size
M1 with
M1 integer are computed, and powers and center frequencies of possible
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interferences are computed at each bin of each measurement by using a
frequency interpolation technique. There are several known frequency
interpolation techniques. A suitable technique is as described in the paper by
M.
D. Macleod, "Fast nearly ML estimation of the parameters of real or complex
single tones or resolved multiple tones", IEEE Transactions on Signal
Processing, January 1998. The power estimates are processed to select a slice
of the frequency band, which is determined to contain at least one of the
interferences. This portion of the frequency band is extracted by filtering
and
down-sampling operations. The down-sampling operation increases the
frequency resolution and allows to compute better power and center frequency
estimates in the next stage.
Following down-sampling of the communication signal by an integer
L1, the second stage consists of computing as in the first stage several FFTs
of
size M2 with M2 integer (where M2 may be equal to M1) of the down-sampled
signal, computing power and center frequencies of possible interferences at
each
bin of each measurement, and processing the power estimates to select a
narrower slice of frequencies within the slice selected in the first stage.
This
narrower slice is extracted y filtering and down-sampling of the signal by an
integer L2 (where L2 may be equal to L1). Additional stages similar to stage 1
and 2 may be used to further "zoom-in" on the signal spectrum, and this until
one
of the interferences is detected and its bandwidth is estimated. A final
measurement of the center frequency and power of the detected interference is
computed by up-sampling by U with U integer the down-sampled signal which
contains the detected interference, and taking several FFTs of size M, and
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computing by frequency interpolation accurate center frequency and power
estimates for this interference.
As illustrated in Fig. 1 by the dashed arrow joining ID 11 to PIS 12,
the detection and characterization of the interferences by ID 11 involved
5 configuring as many IIR notch filters in PIS 12 as there are
interferences to
suppress.= Configuration of the notch filters consists of selecting pre-
computed
values for the coefficients; selection which is based on the bandwidth and
power
estimates computed by ID 11. A same number of all-pass filters are then
configured by selecting the pre-computed values associated with the values
that
10 were selected for the IIR notch filters.
Several other activities occur during an idle time in addition to
detecting and characterizing the interferences. Following configuration of the
PIS
notch and all-pass filters, the output of memory unit indicated at 14 is
passed to
the PIS block 12 by means of mux 13. Memory unit 14 outputs a reconstructed
15 baseband signal at the sampling rate of the training sequence of the
preamble.
U1 output is filtered by PIS 12 and matched filter 15, which is the filter
matched
to the signaling pulse, before being down-sampled to the symbol rate using the
Down-Sampler (DS) block 16. This down-sampled signal is esSentially the
preamble training sequence after being distorted by the PIS block 12. This
"distorted" training sequence is stored in a memory unit indicated at 17, from
which it is passed to a Decision Feedback Equalizer (DFE) block 18 by means of
a mux 19 to train the DFE on this newly configured PIS block 12. The main
purpose of the DFE 18 is to compensate for the distortions introduced by the
PIS
block 12 to the communication signal. The DFE 18 is trained using for example
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the LMS algorithm. The desired signal to train the DFE is the actual training
sequence of the preamble. Training the DFE may require to repeatedly feed the
distorted training sequence to the DFE. This is achieved by cycling several
times
through memory unit 17. The DFE coefficients obtained are stored in memory for
use when the communication signal is present. The input signal 10 is connected
back to the PIS block 12 by means of mux 13 upon complete processing of the
signal stored in memory 14.
Still referring to Fig. 1, an idle time is followed by an activity period
during which CMs are allowed to transmit. The PIS block 12 is not reconfigured
during an activity period, and the coefficients of its notch and all-pass
filters
remain the same. The duration of an activity periods depends on how long the
PIS block 12 can successfully suppress the interferences without changing its
configuration. This depends on the frequency drift of the interferences, and
the
rate at which new interferences appear. The intermittent
disappearance/reappearance of the interferences that are currently filtered by
the
PIS is not an issue, since these interferences are strongly notched by the PIS
block 12.
Again referring to Fig. 1, two types of bursts are received during an
activity period: ranging bursts and traffic bursts. Ranging bursts are fed to
the PIS
block 12, then the matched filter 15, then a Burst Detection (BD) block 20
which
detects the start of the burst by processing the preamble, and then the Burst
SYNChronization (BSYNC) block 21 which recovers timing, measures / adjusts
the signal level, provides coarse estimates of frequency and phase offsets,
digitally re-samples the signal, down-samples the resampled signal to the
symbol
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rate, and de-rotates the down-sampled signal using the coarse frequency and
phase estimates. The de-rotated symbols corresponding to the training sequence
are passed to the Adaptive Equalizer 22, which is used to compute the
coefficients of the pre-equalizer in the transmitter. The desired signal to
train the
adaptive equalizer is the distorted training sequence provided by memory unit
17.
Using this sequence as opposed to the actual training sequence, allows the
system to isolate / decouple the equalization of the channel from the
suppression
of the interferences. The LMS algorithm may be used to train the adaptive
equalizer 22. A Phase-Locked Loop (PLL) indicated at 23, which is internal to
the
adaptive equalizer, compensates for residual frequency offsets. The computed
values for the coefficients of the adaptive equalizer are transmitted to the
CM so
the CM may configure its pre-equalizer.
The payload of the ranging burst may be extracted using the output
of the adaptive equalizer 22, in which case the output is passed to the slicer
26 to
= recover the QAM data, by means of mux 25. Also, the coefficients of the
adaptive
equalizer are kept unchanged during reception of the payload, and the PLL 23
corrects for residual frequency offsets.
The output of the adaptive equalizer 22 may not be sufficiently
reliable for properly detecting the payload, especially in the early stages of
ranging. The DFE block 18 may then be used in place of the adaptive equalizer
22 to extract the payload. In this case, the output of the BSYNC block 21 is
fed to
the DFE 18 by means of Mux 19, and its coefficients are reset to the values
obtained by training the DFE 18 in the idle time that preceded this activity
period.
The coefficients of the DFE 18 get adjusted using the LMS algorithm during
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reception of the preamble training sequence so the DFE also compensates for
the distortions introduced by the channel. The output of the DFE is passed to
the
slicer 26 by means of mux 25. A PLL indicated at 24, which is internal to the
DFE, is used to correct for residual frequency offsets.
Still referring to Fig. 1, TDMA traffic bursts are handled as follows.
Traffic bursts are passed to the PIS 12, then the matched filter 15, then the
BD
20 and BSYNC blocks 21. The adaptive Equalizer 22 is not used in traffic mode.
The DFE 18 may or may not be used. If the DFE 18 is used, then its
coefficients
are reset to the values obtained by training the DFE in the idle time that
preceded this activity period. The DFE coefficients may be adjusted during
reception of the payload by using the LMS algorithm in decision-directed mode.
If
the DFE 18 is not used, then its feedforward linear filter may be used in
decision-
directed mode in combination with the LMS algorithm to compensate for timing
drift during reception of the burst.
Referring to Fig. 2, the PIS block 12 in Fig. 1 consists of a cascade
of Ar Interference Suppressor Filter (ISF) blocks 12A to 12N, labeled as ,
ISF.,
ISF.1 , ¨ , ISFN . An ISF block removes one of the interferences. A cascade of
N
ISF blocks allow to remove N interferences. A mux indicated at 12M is
configured so the PIS block 12 suppresses either , 1 , 2, ¨, or N
interferences.
Referring to Fig. 3, a preferred embodiment of the ISF blocks
shown in Fig. 2, for example block 12A, consists of a cascade of an IIR Notch
filter 27 and an IIR all-pass filter 28. The all-pass filter 28 is "matched"
to the
notch filter 27 in the sense that it compensates for the phase distortions
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introduced by the notch filter. The interference to be notched, say
interference
number , is shifted to DC by multiplying the input signal by the complex
operand, 49-127'ftr' where n is an estimate of the center frequency of
interference
number , which was provided by the ID block 11 shown in Fig. I. The all-pass
filter output, An], is shifted in frequency by multiplying it with el2fffic'.
The notch filter output, An] , is given by
y [71 = x[n] E(cAtt_g_ dolit-13),
where ct and di, are real coefficients. Values for ci and 411 are pre-
computed and stored in memory for different ranges of bwi and Pi .
The all-pass filter output, An] , is given by
P-i
4- I kp_, y[n ¨ Ekzzin-1], (1)
where kz, 1 are real coefficients.
Each notch filter has a corresponding all-pass filter, and thus
coefficients ki , kP of the
all-pass filters are pre-computed for each notch
filter. The calculation of these coefficients involves the numerical
evaluation of an
equation that uniquely relates the magnitude and phase of a complex cepstrum
signal that is causal. Such equation, which can be found in Section 10.3 of
the
first edition of the textbook by A.V. Oppenheim, and, R.W. Schafer, "Discrete-
Time Signal Processing", Prentice-Hall, 1989, is reproduced as:
r -
1ogJ C(eica)1 = -n D4C(eje)cot1-)de, (2)
2
where Cleft()) = 210g2ICCeft()1 4-ihC(efti) is the Fourier transform
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of the complex spectrum, , of a
causal sequence, c[n]. Using equation (2),
the magnitude of the complex cepstrum signal is evaluated at M evenly spaced
frequency points, 0,27r/M, 47r/M, ¨ 1)/M ,
where M > P , after taking
4C(eig) equal to the phase response of the notch filter times a constant .
5 Magnitude and phase vectors of e(e1'4) are generated, where the phase
vector is
obtained by evaluating the phase response of the notch filter times a constant
at
the same frequency points as the ones used to construct the magnitude vector.
ex p eica)
Taking the inverse Discrete Fourier Transform of using the
magnitude and phase vectors computed previously, produces values for did at
10 n = 0, ¨,M ¨ 1 . Coefficients kJ , are set equal to ki
= 411 fort = 1, .
For the all-pass filter 28 to have a phase response that well
approximates the negative of the notch filter phase response requires P to be
large. This renders the implementation of the the all-pass filter 28 non
practical
unless some sort of approximation can be made. For example, P=100 requires
15 100multiplications and 200 additions per filter output.
Good approximate values of coefficients kt, 1 = , are
obtained with piecewise linear approximations. If the curve, ki = , is
linearly approximated by Q segments of equal length where P and Q are chosen
such that Q is an integer, then the coefficients, k, l =1,--P are approximated
20 by
= ai(1.)n 4- at. (2), t = n = 1., (3)
where ai(1) and ai(2) are the coefficients describing segment i , and ai(l)
and
CA 02746702 2015-04-29
21
cte(2) are obtained using a least squares method. Also, coefficients, b(1) and
b2), associated with the Q segments approximating the curve, ,
= P are given by
bi(1) =
bi(2) ______________________________ + at2.444(2) (4)
Any combinations of P and Q can be used, the smaller the Q , the
better, since the less the number of multipliers and adders are required to
implement the IIR all-pass filter.
It is appreciated that the present invention is disclosed using the
simplest IIR all-pass filter, which is obtained when Q =1 . However, the
present
invention may also be used for Q > 1 . The reason for choosing Q =1 is that in
practice a single linear approximation does not cause significant performance
degradation and the all-pass filter remains stable.
Fig. 4 is a block diagram of the all-pass filter 28 for the case Q = .
If Q =1 , the subscript of a and b in (3) and (4) can be dropped for clarity
purposes and (3) and (4) become
= a(1)n a(2). n = 1,...,P
1.(1) = ¨a(1)
b(2) = a.(1)? a(2). (5)
Referring to Fig. 4, input samples YEA , Yrn ¨ 1.,¨ , ¨ 11 are
stored in memory 29 and output samples An ¨ , ¨ , , P 13 are
CA 02746702 2015-04-29
22
stored in memory 30, since from (1) they are needed to compute zfrd.
Computation of Arti involves computing the quantities, Yrtap, Art -1,1)2P-2. ,
- P + 1.1=ki But from (5), yingp -
*k()+ Anib(2) ,
yin = -2y(n - 11(1) + yin -1,1)5(2), -
>En - P + = -PAIL P +
1la(1) + yEn P + 1)). Quantities An3b(2) and
-Yin]a(1) are computed upon reception of a new sample, y[n].
Referring to Fig. 4, quantity
-a(1) x ()[n] + An - lj + + - P + 1.3)
is stored in accumulator a ,indicated
at 31 and
quantity, /(2) x (An3+ An - 1.3 + + An P + 13) , is stored in
accumulator b indicated at
32. This is made possible by updating
accumulators a and b upon reception of a new sample An], where the
update consists of adding the quantities--YErda(1) and Anib(2) to accumulators
E3
a and b and
subtracting the quantities, -An-Pio:1), and, An - Plb(2)
Z.E1
from accumulators a and b .
Similarly, computation of Ard involves computing the quantities
. zIn - zEn 21ii2, - P112 p But from
(5),
= An - 1ia(1) + - 112(2) , - = 2itit - 21a(1) + An- 2k42) -
An- Pgp = PAn na(1) + - P3a(2) Quantities Ank(2) and zinia(1) are
computed upon computation of filter output zini .
CA 02746702 2015-04-29
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Referring to Fig. 4, quantity cz(1) (z[n] + ZE1L ¨ 1.1 z[n PD , is
EL:
stored in accumulator a , and quantity, (K2) x (z[n] + z[n + z[n PD
, is
Ez
stored in accumulator b . This is made possible by updating accumulators a
and b upon reception of a new sample Yin] , where the update consists of
ET2 E72
adding the quantities An¨ lia(1) and An 1la(2) to accumulators d and b ,
and subtracting the quantities, zin¨P-11a(1), and, ZEN ¨ P-111(2) from
DI Et:
accumulators a and b .
Referring again to Fig. 4, the output, An], is obtained by adding the
E'L
content of accumulator, a , to accumulator Er' indicated at 33 and summing
accumulator , accumulator b and Att n . Also quantity
P x a(1) x (¨y[n ¨ 1:] ¨ P 11) must be subtracted from accumulatorEc .
Referring to Fig. 5, the input signal 10 consists of an S-CDMA traffic
burst, which is typically made of signal bursts transmitted simultaneously
from
several transmitters using S-CDMA codes. Signal 10 is passed to the PIS block
12, then the matched filter 15 and the down-sampler 16. A S-CDMA despreader
block 34 despreads the output of the down-sampler using the S-CDMA codes to
extract the signal bursts. This is illustrated in Fig. 5 by showing several
arrows
leaving the despreader block 34, where each arrow represents a signal burst.
CA 02746702 2015-04-29
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The signal bursts (at the symbol rate) are processed to extract their preamble
using a preamble extraction block 35. The preambles are then used to recover
phase and measure/adjust the level of the signal bursts using a phase recovery
block 36 and a signal level adjustment block 37, so the data in each signal
burst
can be properly recovered using a slicer 38.
