Note: Descriptions are shown in the official language in which they were submitted.
CA 02792702 2012-10-17
CA Application
Blakes Ref.: 78680/00001
1 STABILIZED HIGH-VOLTAGE POWER SUPPLY
2
3 TECHNICAL FIELD
4 The present invention relates to a high-voltage power supply comprising a
plurality of DC power
modules connected in series, each module providing a DC link voltage that may
be selectively
6 switched. This general setup is often called a "pulse-step modulator"
(PSM).
7
8 PRIOR ART
9 Pulse-step modulators are widely used as voltage supplies for high-power
vacuum tubes in
various applications, in particular, as modulation amplifiers in AM
transmitters or radar systems.
11 An early example of a pulse-step modulator is disclosed, e.g., in EP 0
066 904.
12
13 A typical prior-art pulse-step modulator is illustrated in Fig. 1. The
modulator comprises a
14 plurality of identical DC power modules 10 connected to the secondaries
of a multi-secondary
transformer 14. Each module comprises a rectifier circuit 11 and a smoothing
capacitance 12
16 for providing a DC link voltage, and a switching circuit 13 for
selectively providing the DC link
17 voltage to the output of the module. A free-wheeling diode in each
switching circuit ensures that
18 an unidirectional current may flow through each module even when the
output switch of the
19 module is open.
21 The total output voltage Vtot of such a PSM-type power supply can be
modulated by switching
22 the DC link voltages to the outputs of the modules 10 in an on/off
fashion. This 30 provides a
23 rather coarse, step-wise modulation of the total output voltage, the
steps being determined by
24 the individual output voltages of the modules ("coarse-step modulation",
CSM). For more
accurate control, pulse-width modulation (PWM) at the outputs of the modules
is often
26 additionally employed. Intricate switching schemes have been suggested
to ensure that the
27 power load is distributed equally over the modules, and to increase the
effective PWM
28 frequency without increasing the actual switching frequency of each
module. Since PWM occurs
29 at much higher frequencies than the mains frequency, PWM is also usually
employed to remove
voltage ripple at multiples of the mains frequency.
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1 Whereas PWM allows for a very accurate control of the total output
voltage, any PWM
2 switching scheme requires a PWM output filter to eliminate voltage ripple
at the PWM
3 frequency.
4
One potential application of a PSM-type power supply is its use as the cathode
power supply
6 (main power supply, MPS) of a gyrotron. A gyrotron is a particular type
of vacuum tube, which
7 emits millimeter-wave electromagnetic radiation. Typical beam output
powers range from some
8 tens of kilowatts well into the megawatt range. Gyrotrons are used, inter
alia, in nuclear fusion
9 research to heat plasmas.
11 A gyrotron typically comprises an electron gun, an acceleration chamber,
a resonance cavity
12 immersed in a strong magnetic field, and an electron collector. An
electron beam is accelerated
13 to relativistic energies and subjected to the magnetic field. The
electrons gyrate around the
14 magnetic field lines and emit electromagnetic radiation. By interaction
of the relativistic
electrons with the radiation field, amplification of the electromagnetic
radiation occurs.
16 Gyrotrons are as such well known and are available commercially from a
variety of
17 manufacturers.
18
19 A gyrotron, together with a typical configuration of its power supplies,
is illustrated in highly
schematic form in Fig. 2. A typical gyrotron comprises a filament F heating a
cathode K for
21 emitting electrons. The electrons are accelerated in the acceleration
chamber past an anode A
22 and past a body electrode B to reach the resonance cavity. The electrons
finally hit a collector
23 C, which is typically kept at a lower potential than the body electrode
to decelerate the
24 electrons ("depressed collector"). Several power supplies are employed
to operate the
gyrotron. A filament power supply FPS powers the filament F. A cathode power
supply MPS
26 provides a negative voltage between the collector C and the cathode K,
this voltage being in
27 the range of several tens of kilovolts. A body power supply BPS provides
a positive voltage,
28 which is also typically in the range of a few tens of kilovolts, between
the collector C and the
29 body electrode B. An anode power supply APS allows to selectively switch
a voltage between
the cathode K and the anode A to modulate the beam current. Other
configurations of
31 electrodes and power supplies have been suggested, which however need
not be discussed
32 here.
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1 An extremely important parameter for the operation of the gyrotron is the
so-called beam
2 voltage between the cathode K and the body electrode B. The electrons are
accelerated
3 between the cathode and the body electrode by this voltage, i.e., the
kinetic energy of the
4 electrons and therefore their relativistic mass increase is determined by
this voltage. Through
the relativistic mass increase, this voltage influences the cyclotron
frequency of the electrons.
6 Since the cavity of a gyrotron typically has a high quality factor 0,
already small variations in
7 the cyclotron frequency can dramatically influence the output power of
the gyrotron. The beam
8 voltage should therefore be as accurate as possible and should have as
little ripple as possible.
9
In the configuration of Fig. 2, the quality of the beam voltage is determined
both by the quality
11 of the cathode power supply MPS and of the body power supply BPS.
Whereas the current
12 load on the body power supply BPS is relatively low (typically in the
range of a few tens of
13 milliamperes), the current load on the cathode power supply MPS is
typically in the range of
14 several tens of amperes. In addition, the cathode power supply MPS must
be capable of rapidly
responding to large, rapid current changes while maintaining the beam voltage
at a
16 predetermined value as accurately as possible. This presents a
considerable challenge to the
17 successful design of a cathode power supply.
18
19 In particular, if PWM is employed for regulating the MPS output voltage,
the PWM output filter
must be designed to ensure both a rapid response to a change in current load
and a low voltage
21 ripple due to PWM. These two requirements are difficult if not
impossible to satisfy
22 simultaneously. Furthermore, parasitic capacitances of the transformer
may propagate the
23 PWM frequency and its harmonics back into the mains grid, which may lead
to EMC problems.
24
If, on the other hand, only coarse-step modulation is used without employing
PWM,
26 considerable ripple from the mains may result, and voltage regulation
can only occur stepwise.
27
28 In J. Alex et al., "A new klystron modulator for XFEL based on PSM
technology", Proceedings of
29 PAC07, Albuquerque, New Mexico, USA (2007), a particular type of PSM has
been suggested
to drive a klystron tube. A plurality of DC power modules are connected in
series. Each module
31 comprises a rectifier with a smoothing capacitance, followed by a boost
converter (step-up
32 converter), acting as a constant power converter to charge a large
output capacitance (20 mF).
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1 The voltage across the output capacitance can be selectively connected to
the output of the
2 module by an output switching circuit. The power supply acts to provide
short high-voltage
3 pulses to a load (specifically, a klystron connected to the power supply
via a pulse transformer).
4 During each pulse, power is drawn from the output capacitances of the
modules. Consequently,
the voltage at the output of each module drops significantly during each
pulse. Subsequently the
6 output capacitance of each module is recharged by the boost converter.
The boost converter is
7 controlled in a manner to ensure that constant power is drawn from the
mains supply to keep
8 power variations ("flicker") on the mains grid due to the pulses as low
as possible.
9
A further development of this type of power supply is disclosed in J. Alex et
al., "A new
11 prototype modulator for the European XFEL Project in pulse step
modulator technology",
12 Proceedings of PAC09, Vancouver, BC, Canada, May 4-8, 2009. Again, each
module employs
13 a boost converter to ensure constant power consumption on the mains
supply. Pulse-width
14 modulation is employed to compensate for the voltage droop on the
storage capacitance during
the pulse in order to improve the flatness of the pulse.
16
17 A high-voltage power supply of the same general type, wherein each
module comprises a
18 boost converter acting as a constant-power converter, is also disclosed
in EP 2 099 127 Al.
19
These power supplies cannot overcome the deficiencies noted above in
connection with the
21 more traditional kinds of PSM power supplies if both a rapid response to
current changes and
22 low voltage ripple are needed, since stabilization of the total output
voltage still either requires
23 PWM or, in the alternative, can be only carried out stepwise.
24
SUMMARY OF THE INVENTION
26 It is therefore an object of the present invention to provide a high-
voltage power supply that is
27 capable of providing a stable total output voltage having low voltage
ripple while 5 being able to
28 handle large, rapid current changes.
29
This object is achieved by a high-voltage power supply as specified in claim
I. The invention
31 further provides a method of operation of such a power supply, as laid
down in claim 11, a
32 method of use as laid down in claim 13, and a gyrotron employing such a
power supply as laid
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1 down in claim 14. Further embodiments of the invention are laid down in
the dependent claims.
2
3 According to the present invention, a high-voltage power supply is
provided, comprising:
4 a plurality of DC power modules having their outputs connected in a
series configuration,
each power module comprising a DC voltage source, a DC-DC converter receiving
an
6 input voltage from said DC voltage source and providing a DC link
voltage, and an output
7 switching circuit for selectively connecting said DC link voltage to the
output of said power
8 module,
9 characterized in that the power supply comprises, for each DC power
module, a
feedback control circuit operable to provide driving signals to the DC-DC
converter of said
11 power module in a manner that regulates said DC link voltage to a
predetermined reference
12 voltage.
13
14 Therefore, in contrast to the prior art, the DC-DC converter is not
operated to ensure that
constant power is drawn from the DC voltage source of the module, but it is
operated as a
16 voltage regulator, ensuring that the DC link voltage takes a
predetermined reference value. The
17 reference voltages of all involved modules are preferably identical,
such that the total output
18 voltage of the power supply is the reference voltage, multiplied by the
number of modules
19 involved.
21 Each module will therefore normally comprise a DC link voltage sensor
for measuring the DC
22 link voltage, and the feedback control circuit will normally receive the
measured DC link
23 voltage from the sensor, optionally process the measured voltage (e.g.,
by subjecting it to a
24 low-pass filter), and compare it to the (optionally also pre-processed)
reference voltage. A
difference signal will be fed to a suitable controller, e.g., a PI or PID
controller. The controller
26 output will be fed to an actuator, which calculates the driving signals
for the DC-DC converter.
27
28 Advantageously, when deriving the driving signals, the actuator may
directly take into account
29 the major disturbance variables of the control loop, in particular, the
actual output current of the
power modules and/or the actual input voltage of the power modules. In this
manner, a very
31 rapid response to changes in these disturbance variables may be
achieved. In particular, the
32 power supply may comprise at least one current sensor for measuring an
output current of said
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1 DC power modules, and the feedback control circuit may then be adapted to
derive driving
2 signals for the DC-DC converter of each power module taking into account
the measured
3 output current. A single current sensor for the complete power supply may
be sufficient;
4 however, it is preferred that each module comprises its own current
sensor. Furthermore, each
power module may comprise an input voltage sensor for determining an input
voltage of its
6 DC-DC converter, and the voltage control circuit may then be adapted to
derive driving signals
7 for the DC-DC converter of each power module taking into account the
measured input voltage.
8 The DC-DC converter will normally be a switched converter operable at a
variable duty cycle.
9 The feedback control circuit may then be adapted to calculate the duty
cycle taking into
account the measured disturbance variables, i.e., the measured output current
and/or the
11 measured input voltage, and to operate the DC-DC converter at that duty
cycle. A possible
12 formula for the duty cycle in the case of a boost converter will be
given in section "Detailed
13 Description of Preferred Embodiments" below; however, other
relationships taking into account
14 these disturbance variables are also conceivable.
16 Preferably, the DC-DC converter of each power module is a boost
converter. However, it is
17 conceivable to employ any other type of switched DC-DC converter, such
as, e.g., a buck
18 converter, a buck-boost converter, or a SEPIC. For practical reasons, it
is preferred that the
19 converter is capable of delivering an output voltage that is higher than
the input voltage, which
would render buck converters less preferred. Since it is furthermore preferred
to minimize any
21 losses, a boost converter with its simple topology is the
22 preferred converter type.
23
24 A boost converter comprises at least one output capacitance and at least
one converter
switching element cooperating with at least one choke inductance and at least
one diode (or
26 any other unidirectional switching element) to charge the output
capacitance to the DC link
27 voltage. The feedback control circuit is then operable to control the DC
link voltage by switching
28 the converter switching element at a variable duty cycle.
29
An interleaved topology of the boost converter may be chosen to reduce AC
currents in the
31 input and output capacitances, and to further reduce voltage ripple. The
boost converter of each
32 power module may therefore comprise at least two interleaved boost
converter circuits adapted
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1 to charge a common output capacitance. The control circuit is preferably
operable to operate
2 the interleaved boost converter circuits in a synchronous but phase-
shifted manner.
3 In a concrete setup, such an interleaved boost converter topology may
comprise at least one
4 common output capacitance, at least one first converter switching element
(possibly two or
more such elements connected in series) cooperating with at least one first
choke inductance
6 (possibly two or more such inductances in a series configuration with the
first switch elements)
7 and diode (possibly two or more diodes in a series configuration with the
first choke
8 inductances) to charge said output capacitance to the DC link voltage,
and at least one second
9 converter switching element (possibly two or more such elements connected
in series)
cooperating with at least one second choke inductance (possibly two or more
such inductances
11 in a series configuration with the second switch elements) and diode
(possibly two or more
12 diodes in a series configuration with the second choke inductances) to
charge the same output
13 capacitance. The control circuit is then operable to control the DC link
voltage by driving the
14 first and second converter switching elements at a variable duty cycle,
preferably in a
synchronous but phase-shifted manner.
16
17 In order to ensure a rapid response to current changes and to reduce
switching losses, the
18 control circuit preferably operates the boost converter of each power
module in discontinuous
19 mode during voltage regulation. This can be ensured by appropriately
choosing the choke
inductance to be lower than the so-called critical choke inductance in the
intended operating
21 regime of the module. An example for the calculation of the critical
choke inductance is
22 provided below in section "Detailed Description of Preferred
Embodiments".
23
24 The control circuits of the individual modules may be commonly
controlled by a main control
system for the modules. In particular, the overall step-response behavior of
the power supply to
26 disturbances such as current changes may be improved by driving the DC-
DC converters of
27 different DC power modules in a synchronous but phase-shifted manner. In
this manner, the
28 DC-DC converters of some modules in the appropriate phase within the
converter cycle time
29 can react to disturbances earlier than those lagging behind in phase.
31 Preferably, the total output voltage of the power supply during each
time period in which the
32 power supply is supposed to deliver a certain predetermined, non-zero
voltage value ("voltage
=
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1 pulse") is regulated by only controlling the DC-DC converters of the
individual power modules,
2 without applying coarse-step modulation, and without applying pulse-width
modulation. In other
3 words, the modules are preferably operated in a manner in which the
output switching circuits
4 of all power modules remain in the same (active or passive) state over
prolonged times, in
particular, as long as the desired total output voltage remains unchanged,
unless a fault
6 condition occurs. Such time periods will normally cover at least 10,
often at least 50 switching
7 cycles of the DC-DC converter. They can last, e.g., from less than 1 ms
to several seconds or
8 even longer.
9
In order to enable a rapid discharge of the output capacitance in cases such
as a missing load,
11 each DC power module may comprise a discharge resistor and a discharge
switch for
12 selectively discharging the output capacitance.
13
14 In preferred embodiments, the power source may comprise at least one
multi-secondary
transformer having a plurality of sets of secondary windings, and each DC
power module may
16 then comprise a rectifier circuit connected to one set of secondary
windings. The DC voltage
17 source of each module may thus be considered to comprise this set of
secondary windings and
18 the corresponding rectifier circuit. Alternatively, a separate
transformer may be used for each
19 module. Depending on the field of application, it is even conceivable to
use other kinds of DC
voltage sources, even batteries.
21
22 To improve power ratio, the high-voltage power supply may comprise a
first and a second
23 multi-secondary transformer, the first transformer and the second
transformer being configured
24 to provide secondary voltages that are phase-shifted between the
transformers. In the case of a
three-phase mains grid, this may be accomplished, e.g., by using a star-star
configuration for
26 the first transformer and a delta-star configuration for the second
transformer, resulting in
27 twelve-pulse rectification. Other suitable schemes for achieving a phase
shift are well known in
28 the art.
29
The present invention also provides a method of operating a high-voltage power
supply as
31 described above. The method comprises, for each power module:
32 setting the reference voltage;
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1 measuring and processing the actual DC link voltage;
2 comparing the DC link voltage and the reference voltage to derive a
difference signal;
3 from the difference signal, deriving an actuating signal;
4 measuring an actual output current and/or the actual input voltage;
from the actuating signal, deriving driving signals for the DC-DC converter,
taking into
6 account the measured output current andlor input voltage; and
7 driving each DC-DC converter by said driving signals to actively control
said output
8 voltage.
9
In particular, as outlined above, the driving signals may implement a duty
cycle determined,
11 inter alia, by the actuating signal, the output current and the input
voltage.
12
13 The high-voltage power supply described above may advantageously be used
as a power
14 supply for a gyrotron, in particular, as its cathode power supply.
Accordingly, the present
invention also relates to a gyrotron. Generally, a gyrotron has at least a
cathode, a body
16 electrode, and a collector electrode, and advantageously the power
supply is connected
17 between the cathode and the collector electrode.
18
19 While a particular application for a gyrotron has been described, there
are also other fields of
applications where a highly stable high-voltage power supply capable of
handling rapid current
21 changes is needed, and the power supply of the present invention may be
employed in such
22 other fields as well.
23
24 BRIEF DESCRIPTION OF THE DRAWINGS
Preferred embodiments of the invention are described in the following with
reference to the
26 drawings, which are for the purpose of illustrating the present
preferred embodiments of the
27 invention and not for the purpose of limiting the same. In the drawings,
28
29 Fig. 1 shows a schematic block diagram of a PSM power supply
according to the prior
art;
31
32 Fig. 2 shows a highly schematic sketch of a typical gyrotron together
with its
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1 associated power supplies;
2
3 Fig. 3 shows, in a highly schematic fashion and not to scale, a typical
sequence of
4 voltages, currents and rf power in a gyrotron;
6 Fig. 4 shows a schematic block diagram of a power supply according to the
present
7 invention;
8
9 Fig. 5 shows a simplified circuit diagram of a single module of the
power supply of Fig.
4; and
11
12 Fig. 6 shows a diagram illustrating the control loop for controlling
the output voltage of
13 a module as shown in Fig. 5.
14
DESCRIPTION OF PREFERRED EMBODIMENTS
16 Figure 3 illustrates, in a highly schematic fashion and not to scale, a
typical sequence of the
17 various voltages and currents involved in the operation of a gyrotron.
At the beginning of the
18 sequence, the cathode power supply MPS and the body power supply BPS are
switched to their
19 nominal output voltages to provide a predetermined beam voltage VBK.
Both voltages are
actively controlled to ensure stability of these voltages. The anode-cathode
voltage vAK provided
21 by the anode power supply APS is initially kept at a value which avoids
any significant beam
22 current between the cathode K and the collector C. Only when the
gyrotron is to generate
23 electromagnetic radiation, the anode-cathode voltage vAK is switched to
a positive value, leading
24 to a rapid rise of the beam current IK and to the emission of
electromagnetic radiation with
power P .
26
27 From this diagram it is apparent that the cathode power supply MPS must
be capable of rapidly
28 reacting to large variations in beam current IK while providing a stable
output 5 voltage with low
29 voltage ripple.
31 Figure 4 shows a simplified block diagram of a high-voltage power supply
according to the
32 present invention, which is adapted to satisfy these requirements. The
general setup is similar
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1 to the setup of Fig. 1. However, instead of one single multi-secondary
transformer, in the
2 present example two multi-secondary transformers 44, 45 in different
configurations (YY vs.
3 DY) are used to improve overall power factor. Both transformers 44, 45
are fed by a
4 three-phase mains voltage Vm. The three primary windings of the first
transformer 44 are
connected in a star ("Y") configuration, while the three primary windings of
the second
6 transformer 45 are connected in a delta ("D") configuration. Each
transformer comprises a
7 plurality of sets of three secondaries (typically 20-30 such sets), each
set being connected in a
8 star configuration. Overall, the first transformer 44 thus has a star-
star ("YY") configuration,
9 while the second transformer has a delta-star ("DY") configuration. By
this overall configuration
of the two transformers 14, 15, the secondary voltages of the two transformers
are
11 appropriately phase-shifted with respect to each other to achieve twelve-
pulse rectification,
12 which leads to a high power factor of typically more than 0.95.
13
14 The power supply comprises a plurality of identical DC power modules 40
connected to the
secondaries of the transformers 44, 45. Each DC power module comprises a
rectifier circuit 41,
16 a boost converter 42, and an output switching circuit 43. The outputs of
the modules 40 are
17 connected in a series configuration.
18
19 Fig. 5 shows a simplified circuit diagram for a single module 40, which
will now be described in
more detail.
21
22 Input terminals X10, XI 1, X12 are connected to a set of three
secondaries of one of the
23 transformers 44, 45. The rectifier circuit 41 in the form of a full-
bridge rectifier, consisting of six
24 diodes D1, rectifies the three-phase secondary voltage.
26 The boost converter 42 comprises two interleaved boost converter
circuits having a common
27 input capacitance and charging a common output capacitance, each of the
interleaved
28 converter circuits being in turn designed with two switching elements in
series to reduce the
29 required voltage rating of each switching element. In more detail, the
common input capacitance
of the boost converter is formed by a network of capacitors C10, C11,
connected in series to
31 enable the use of capacitors having a reduced voltage rating, and
symmetrizing resistors R10,
32 R11 in parallel to the two capacitors. The voltage across this input
capacitance is called the
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1 input voltage V. The common output capacitance of the boost converter is
formed by two
2 series-connected capacitors C20, C21.Each of the upper and lower
terminals of the output
3 capacitance is connected to the input capacitance via two groups of
elements consisting of a
4 series-connected choke inductance and diode each, the two groups being
connected in parallel.
The diodes are forward-biased to allow energy to flow from the input
capacitance through the
6 choke inductances into the output capacitance, but not in the reverse
direction from the output
7 capacitance back into the input capacitance. Two pairs of series-
connected actively controlled
8 converter switches V10, V11 and V20, V21, respectively, here in the form
of IGBTs, connect the
9 connection node between an upper choke inductance and upper diode with
the connection node
between a lower choke inductance and lower diode. By closing these switches,
current can flow
11 through these choke inductances while the output capacitance is
bypassed. As a result, the
12 upper and lower choke inductance together are subjected to the input
voltage, and inductive
13 energy builds up in the choke inductances due to this voltage. When the
switches are opened
14 again, the stored energy in the choke inductances is transferred to the
output capacitance
through the diodes. The resulting output voltage (DC link voltage) Vout across
the output
16 capacitance can be much larger than the input voltage. This principle of
operation of a boost
17 converter is as such well known in the art.
18
19 A discharge resistor R40 and a switch K20 enable a rapid discharge of
the output capacitance if
needed. Output switches V30, V31 in the form of IGBTs selectively provide the
DC link voltage
21 Vo to the output terminals X20, X21 of the module. Two reverse-biased,
series-connected
22 freewheeling diodes D30, D31 enable a unidirectional current to flow
between the output
23 terminals even when the output switches V30, V31 are open. The common
node between the
24 two diodes D30, D31 is connected to the common node between the
converter switches VIO,
V11 and V20, V21, respectively, of each pair of these switches as well as to
the common node
26 between the capacitors C20, C21 to provide improved symmetrization.
Additional
27 symmetrization resistors (not shown in Fig. 5) may optionally be
provided in parallel to the
28 capacitors C20, C21.
29
A module controller 51 is fed from input terminals X10, X11 via a small
transformer T10.
31 Control signals and diagnostic signals are exchanged between the module
controller 51 and an
32 external main control system 53 via a fiber-optic link 52 ensuring
galvanic isolation. The module
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1 controller 51 controls the booster switches V10, V11, V20, V21, the
discharge switch K20, and
2 the output switches V30, V31 via leads that have been omitted in Fig. 5
for the sake of clarity.
3 In the present example, all switches are implemented as semiconductor
switches, in particular,
4 as IGBTs, which will usually be equipped with additional reverse-biased
freewheeling diodes in
parallel to the collector-emitter path (not shown). However, other types of
actively controlled
6 semiconductor switches may be employed, depending on the actual load
requirements, such
7 as power MOSFETs etc.
8
9 The power supply is operated as follows: Depending on the desired total
output voltage Vtot and
on the status of the modules, the main control system selects whether all or
only a part of the
11 modules shall be involved in providing the desired total output voltage
Vtot. The main control
12 system accordingly provides control signals to the module controllers 51
of the individual
13 modules via the fiber-optic link 52. Each module controller drives the
converter switches
14 V10-V21 of its associated module to charge the output capacitance C20,
C21 to a reference
DC link voltage Vref determined by the main control system. This reference
voltage is set to the
16 same value in all modules. It corresponds to the desired total output
voltage Vtot, divided by the
17 number of involved modules. The output switches of the involved modules
are then closed to
18 provide the DC link voltages of the modules to their outputs, so as to
provide the sum of the DC
19 link voltages at the output of the power supply. During normal
operation, and in particular
during individual current pulses delivered by the power supply, the output
switches remain
21 closed and are not operated, in contrast to prior-art devices, where PWM
is implemented at the
22 output. switches to provide voltage regulation. Instead, regulation of
the total output voltage is
23 carried out by regulating the DC link voltages Vow supplied by the boost
converters in a
24 feedback control circuit implemented in module controller 51.
26 The feedback control circuit 60 is illustrated schematically in Fig. 6.
The actual DC link voltage
27 Võt (in terms of control theory, the controlled variable of the control
loop) is determined by a
28 suitable voltage sensor 65. The measured DC link voltage is subjected to
a low-pass filter 64 to
29 filter out the switching frequency of the boost converter and its
harmonics, and the filtered DC
link voltage is compared to the reference voltage Vref in a comparator 61. The
difference Vdiff of
31 these two voltages is fed to a PI controller 62, whose controller output
signal Vctri (in terms of
32 control theory, the actuating variable of the control loop) is fed to a
calculating unit 63 (the
22294238.2 13
CA 02792702 2012-10-17
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1 actuator of the control loop). An input voltage sensor 64 measures the
input voltage Vin, and
2 an output current sensor measures the output current lout. These signals
(corresponding to the
3 main disturbance variables of the control loop) are also fed to the
calculating unit 63. From the
4 controller signal V et rt, from the input voltage V1, and from the output
current km, the
calculating unit 63 calculates a duty cycle (i.e., the ratio of the time
during which the switches of
6 the converter are closed and the cycle time of the converter) and drives
the converter switches
7 V 1 0- V 2 1 according to this duty cycle.
8
9 The controlled variable Võt is a quadratic function of the duty cycle. In
order to achieve a linear
control path, the duty cycle should therefore be a square root function of the
actuating variable
11 Vcid. In this manner, a linear dependence between controller signal
Vctrl (actuating variable) and
12 DC link voltage Vnut (controlled variable) results. As will be detailed
further below, the boost
13 converter is preferably operated in discontinuous mode (i.e., the
current in the booster choke
14 inductances substantially decreases to zero before the next booster
cycle starts). The duty
cycle in discontinuous mode may be calculated according to the following
formula:
16
2 = iota = Livoster (Vcrri ¨
D
Tbooster = V2
in
17 (Equation 1).
18
19 Here, Tbooster is the booster cycle time (the inverse of the boost
converter operating frequency,
which is kept constant during operation), and Lbooster is the total choke
inductance of the boost
21 converter (in the specific arrangement illustrated here, Lbooster = L10
=L11=L20=L21). Both
22 pairs of converter switches in the interleaved converter circuits are
operated at the same duty
23 cycle, but phase-shifted by 1800 relative to each other.
24
An important property of the calculating unit is that the most important
disturbance variables,
26 lnut and V,n, directly act on the actuator of the control loop. For
example, in the case of a sudden
27 rise of the output load, the output current will also rise rapidly,
while the DC link voltage will
28 drop slowly due to the presence of the large output capacitance. Since
the increased output
29 current directly acts on the actuator, the duty cycle will be increased
almost instantaneously,
and the voltage drop will stop within a single booster cycle time Tbooster.
The PI controller can
31 now correct the (relatively small) voltage drop that has already
occurred.
22294238.2 14
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1 In order to ensure a rapid response to load changes, the boost converter
should be operated in
2 discontinuous mode during normal operation. This measure also improves
stability of the
3 regulation of the DC link voltage, and minimizes switching losses on the
converter switches,
4 since these are always switched on at zero current. This poses certain
restrictions on the
booster choke inductances. In particular, the total choke inductance should
not exceed a
6 certain critical value, which is well known in the art and depends on the
desired operating point
7 as follows:
8
i14,1 = (Vour ¨ Vin) = Tbooster
Latt = _______________ 2
9 2 = /out Voia (Equation2).
11 On the other hand, the choke inductance should not be too small in order
to keep the input
12 current to low figures. This calls for a choice of choke inductance
below but close to the critical
13 choke inductance Lcrit=
14
The PI controller 62 can be disabled selectively in certain situations where
feedback control
16 would be inappropriate. One such situation is the case of a missing
output load (no-load case),
17 if at the same time the DC link voltage V01 is higher than the reference
voltage Vref . While the
18 controller is disabled, the output capacitance may be discharged down to
the reference voltage
19 by closing the discharge switch K20. Once the reference voltage Vref is
reached again, the
controller is enabled again.
21
22 In order to enable controlled ramp-up of the output voltage in the no-
load case, e.g., after a
23 positive change of the reference voltage Vref, particular measures are
required, since Equation
24 (1) implies that the duty cycle will be zero as long as the output
current is zero. In order to
overcome this problem, it is possible to set the current in Equation (1) to
some predetermined
26 minimum value (e.g., 1-2% of the maximum output current) if the actual
measured output
27 current is smaller than this minimum value.
28
29 Particular measures are required for power-up. A possible power-up
sequence may be
implemented as follows: The input capacitance CIO, 011 and the output
capacitance C20, C21
31 are initially charged to the nominal voltage of Vin via step-start
switches and a charging resistor
22294238.2 15
CA 02792702 2012-10-17
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1 (not shown in Fig. 5), as they are well known in the art. The boost
converter switches remain
2 disabled until all capacitances are charged to approximately the nominal
voltage of Võ, . Only
3 then the boost converter starts to operate. The boost converter is
initially operated at constant
4 duty cycle, until the reference voltage is reached across the output
capacitance. Only then
closed-loop control starts.
6
7 Operating conditions of the modules are continuously supervised, and any
module is switched
8 off and possibly replaced by another (so far idle) module if a fault
condition is detected. In
9 particular, a fault condition is assumed if the booster input voltage or
the booster output voltage
is outside a predetermined range, or if the output current exceeds a
predetermined maximum
11 value. In addition, temperature, desaturation etc. may also be
supervised.
12
13 The output of each module may be provided with a small output snubber
(not shown in Fig. 5) in
14 order to limit the current surge in the module in the case of a short
circuit. This snubber should,
however, be kept as small as possible in order not to compromise the step
response under load
16 changes.
17
18 Actual values of capacitances, inductances, resistors etc. will largely
depend on the concrete
19 application and on the desired operating point.
21 The above description is only for illustrative purposes, and a number of
modifications can be
22 made without departing from the scope of the present invention. In
particular, the boost
23 converter design can be different from the design as described above. In
the simplest case, a
24 single converter switch may be used in conjunction with a single choke
inductance, a single
diode and a single output capacitor, as it is well known in the art and
illustrated schematically in
26 the box symbolizing module 40 in Fig. 6. Different controller types than
PI controllers may be
27 employed, such as PID controllers. All diodes (acting as passive
switches) may be replaced by
28 active switching elements such as transistors if desired. The rectifier
circuit may be designed
29 differently, e.g., as an actively controlled thyristor rectifier
circuit. Instead of single-quadrant
output switching circuits, as in the above-described embodiment, which allow
only for unipolar
31 voltage and unidirectional current, also two-quadrant output switching
circuits allowing for
32 bipolar voltages at unidirectional current or for unipolar voltage at
bidirectional currents or even
22294238.2 16
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1 four-quadrant output switching circuits allowing for arbitrary sign of
both output voltage and
2 output current may be employed. Two-quadrant switching may be useful,
e.g., for inverse
3 voltage operation to reduce currents after a short circuit has occurred,
or for driving capacitive
4 loads such as a control electrode of a vacuum tube. Suitable output
switching circuits for
two-quadrant or four-quadrant operation are disclosed, e.g., in EP 2 099 127
Al, in particular in
6 its Figures 5-7, and the disclosure of that document is incorporated
herein by reference in its
7 entirety for teaching suitable output switching circuits for two- and
four-quadrant operation.
8 Suitable output switching circuits and modes of operation of two- and
four-quadrant output
9 switching circuits are also disclosed in WO 95/10881 Al and EP 1 553 686
Al.
11 In other embodiments, depending on the intended field of use, the boost
converter may be
12 replaced by any other form of switched DC-DC converter. This might be a
buck converter, a
13 buck-boost converter, a SEPIC etc. Such switched-mode DC-DC converters
are well known in
14 the art. The operating principles as outlined above remain the same with
such DC-DC
converters. In particular, also with other types of DC-DC converters it is
possible to regulate the
16 DC link voltage of each module by controlling the DC-DC converter,
instead of employing PWM
17 and/or CSM schemes to regulate the total output voltage of the complete
power supply.
18
19 The proposed power supply may not only be employed as the main power
supply of a gyrotron,
but may be used in any application which require a stabilized high voltage
which is stable even
21 under rapid load changes. Examples include the cathode or anode power
supply of any other
22 type of vacuum tube having a control electrode which may rapidly change
the current in the
23 tube.
22294238.2 17
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1 List of references
2
3 10 : power module
4 11 : rectifier circuit
12 : smoothing capacitance
6 13 : output switching circuit
7 14 : transformer
8 2 : gyrotron
9 FPS : filament power supply
MPS : cathode power supply
11 BPS : body power supply
12 APS : anode power supply
13 F : filament
14 K : cathode
A : anode
16 B : body electrode
17 C : collector
18 VK : cathode voltage
19 VBK : beam voltage
VAK : anode-cathode voltage
21 IK : beam current
22 RI : radiated power
23 40 : power module
24 41: rectifier circuit
42 : boost converter
26 43 : output switching circuit
27 X10, X11, X12 : input terminals
28 X20, X21 : output terminals
29 Dl: rectifier diode
C10, C11 : input capacitors
31 R10, R11 : divider resistors
32 L10, L11, L20, L21 :choke inductances
22294238.2 18
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1 V10, V11, V20, V21 : converter switches
2 D10, D11, D20, D21 : converter diodes
3 020, 021 : output capacitors
4 R20 : dissipating resistor
K40 : dissipating switch
6 V30, V31 : output switches
7 D30, D31 : freewheeling diodes
8 51 : module controller
9 52 : fiber optic link
V,n : input voltage
11 Vont : DC link voltage
12 Vtot : total output voltage
13 Vref : reference voltage
14 Vdiff : voltage difference
Vctri : controller output signal
16 lout : output current
17 60 : control circuit
18 61 : comparator
19 62: PI controller
63 : calculating unit
21 64 : input voltage sensor
22 65 : output voltage sensor
23 66 : output current sensor
22294238.2 19
