Note: Descriptions are shown in the official language in which they were submitted.
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Descripdon
MULTICHANNEL AUDIO CODING
This is a divisional of Canadian National Phase Patent Application Serial
No. 2,556,575 filed February 28, 2005.
Technical Field
The invention relates generally to audio signal processing. The invention is
particularly useful in low bitrate and very low bitrate.audio signal
processing. More
particularly, aspects of the invention relate to an encoder (or encoding
process), a decoder
= (or decoding processes), and to- an encode/decode system (or
encoding/decoding process)
=
=
for audio signals in which.a plurality of audio channels is represented by a
composite
= monophonic (imono") audio chmmel and auxiliary ("sidechain") information.
Alternatively, the plurality of audio channels is represented by a plurality
of audio =
channels and sidechain information. Aspects of the invention also relate to a
multichannel to composite monophonic channel downmixer (or downmix process),
to a =
monophonic channel to multichannel upmixer (or uptnixer process), and to a
monophonic
channel to multichannel decorrelator (or .decorrelation process). Other
aspects of the
invention relate to a multichannel-to-multichannel dowmnixer (or downmix
process), to a
multichannel-to-multichannel upmixer (or upmix process), and to a decorrelator
(or
decorrelation process).
Backgroimd Art
In the AC-3 digital audio encoding and decoding system, charmels may be
selectively combined or "coupled" at high frequencies when the system becomes
starved
for bits. Details of the AC-3. system are well known in the art ¨ see, for
example: ATSC
= Standard A52/A: Digital Audio Compression Standard (AC-3), Revision A,
Advanced
= Television Systems Committee, 20 Aug. 2001. The A/52A document is
available on the
World Wide Web at http://www.atsc.org/standardshtml.
The frequency above which the AC-3 system combines channels on demand is .
=
referred to as the "coupling" frequency. Above the coupling frequency, the
coupled
channels are combined into a "coupling" or composite channel. The encoder
generates
= "coupling coordinates" (amplitude scale factors) for each subband
above the coupling =
frequency in each channel. ' The coupling coordinates indicate the ratio of
the original
=
=
= =
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energy of each coupled channel subband to the energy of the corresponding
subband in
the composite channel. Below the coupling frequency, channels are encoded
discretely.
The phase polarity of a coupled channel's subband may be reversed before the
channel is
combined with.one or more other coupled channels in order to reduce out-of-
phase signal
component cancellation. The composite channel along with sidechain information
that
includes, on a per-subband basis, the coupling coordinates and whether the
channel's
phase is inverted, are sent to the decoder. In practice, the coupling
frequencies employed
in commercial embodiments of the AC-3 system have ranged from about 10 kHz to
about
3500 Hz. U.S. Patents 5,583,962; 5,633,981, 5,727,119, 5,909,664, and
6,021,386
include teachings that relate to the combining of multiple audio channels into
a composite
channel and auxiliary or sidechain information and the recovery therefrom of
an
approximation to the original multiple channels.
.Disclosure of the Invention
= Aspects of the present invention may be viewed as improvements upon the
"coupling" techniques of the AC-3 encoding and decoding system and also upon
other =
= techniques in which multiple channels of audio are combined either to a
monophonic
composite signal or to multiple channels of audio along with related auxiliary
information
and from which multiple channels of audio are reconstructed. Aspects of the
present
invention also may be viewed as improvements upon techniques for downmixing
multiple
audio channels to a monophonic audio signal or to multiple audio channels and
for
decorrelating multiple audio channels derived from a monophonic audio channel
or from
=
multiple audio channels.
Aspects of the inv,ention may be employed in an N:1:N spatial audio coding
technique (where "N" ikthe number of audio channels) or an M:1:N spatial audio
coding
= technique (where "M" is the number of encoded audio channels and "N"
is the number of =
decoded audio channels) that improve on channel coupling, by providing, among
other
things, improved phase compensation, decorrelation mechanisms, and signal-
dependent
variable time-constants. Aspects of the present invention may also be employed
in N:x:N
and M:x:N spatial audio coding techniques wherein "X" may be 1 or greater than
1.
= Goals include the reduction of coupling cancellation artifacts in the
encode process by=
= adjusting relative interchannel phase before downmixing, and improving
the spatial
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dimensionally of the reproduced signal by restoring the phase angles and
degrees of
decorrelation in the decoder. Aspects of the invention when embodied in
practical
embodiments should allow for continuous rather than on-demand channel coupling
and lower
coupling frequencies than, for example in the AC-3 system, thereby reducing
the required
data rate.
According to one aspect of the present invention, there is provided a method
for decoding M encoded audio channels representing N audio channels, where N
is two or
more, and a set of one or more spatial parameters having a first time
resolution, the method
comprising: a) receiving said M encoded audio channels and said set of spatial
parameters
having the first time resolution, b) employing interpolation over time to
produce a set of one
or more spatial parameters having the first time resolution, c) deriving N
audio signals from
said M encoded channels, wherein each audio signal is divided into a plurality
of frequency
bands, wherein each band comprises one or more spectral components, and d)
generating a
multichannel output signal from the N audio signals and the one or more
spatial parameters
having the second time resolution, wherein M is two or more, at least one of
said N audio
signals is a correlated signal derived from a weighted combination of at least
two of said M
encoded audio channels, said set of spatial parameters having the first
resolution includes a
first parameter indicative of the amount of an uncorrelated signal to mix with
a correlated
signal and step d) includes deriving at least one uncorrelated signal from
said at least one
correlated signal, and controlling the proportion of said at least one
correlated signal to said at
least one uncorrelated signal in at least one channel of said multichannel
output signal in
response to one or ones of said spatial parameters having the second
resolution, wherein said
controlling is at least partly in accordance with said first parameter.
According to another aspect of the present invention, there is provided an
apparatus comprising means adapted to carry out the method as described above
or below.
According to another aspect of the present invention, there is provided a
computer-readable medium having stored thereon computer-executable
instructions that,
when executed by a computer, cause the computer to implement the method as
described
above or below.
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According to another aspect of the present invention, there is provided a
method for decoding M encoded audio channels representing N audio channels,
where N is
two or more, and a set of one or more spatial parameters, wherein one or more
of said spatial
parameters are differentially encoded, the method comprising: a) receiving
said M encoded
audio channels and said set of spatial parameters, b) applying a differential
decoding process
to the one or more differentially encoded spatial parameters, c) deriving N
audio signals from
said M encoded channels, wherein each audio signal is divided into a plurality
of frequency .
bands, wherein each band comprises one or more spectral components, and d)
generating a
multichannel output signal from the N audio signals and the spatial
parameters, wherein M is
two or more, at least one of said N audio signals is a correlated signal
derived from a weighted
combination of at least two of said M encoded audio channels, said set of
spatial parameters
includes a first parameter indicative of the amount of an uncorrelated signal
to mix with a
correlated signal and step d) includes deriving at least one uncorrelated
signal from said at
least one correlated signal, and controlling the proportion of said at least
one correlated signal
to said at least one uncorrelated signal in at least one channel of said
multichannel output
signal in response to one or ones of said spatial parameters, wherein said
controlling is at least
partly in accordance with said first parameter.
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Description of the Drawings
FIG. 1 is an idealized block diagram showing the principal functions or
devices of
an N:1 encoding arrangement embodying aspects of the present invention.
FIG. 2 is an idealized block diagram showing the principal functions or
devices df
a 1:N decoding arrangement embodying aspects of the present invention.
FIG. 3 shows an example of a simplified conceptual organization of bins and
subbands along a (vertical) frequency axis and blocks and a frame along a
(horizontal)
time axis. The figure is not to scale.
FIG. 4 is in the nature of a hybrid flowchart and functional block diagram
showing encoding steps or devices performing functions of an encoding
arrangement
embodying aspects of the present invention.
FIG. 5 is in the nature of a hybrid flowchart and functional block diagram
showing decoding steps or devices performing functions of a decoding
arrangement
embodying aspects of the present invention.
FIG. 6 is an idealized block diagram showing the principal fimctions or
devices of
a first N:x encoding arrangement embodying aspects of the present invention.
FIG. 7 is an idealized block diagram showing the principal functions or
devices of
an x:M decoding arrangement embodying aspects of the present invention.
FIG. 8 is an idealized block diagram showing the principal functions or
devices of
a first alternative x:M decoding arrangement embodying aspects of the present
invention.
FIG. 9 is an idealized block diagram showing the principal functions or
devices of
a second alternative x:M decoding arrangement embodying aspects of the present
invention.
= = = Best Mode for carrying Out the Invention
Basic N:1 Encoder
Referring to FIG. 1, an N:1 encoder function or device embodying aspects of
the
present invention is shown. The figure is an example of a function or
structure that
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performs as a basic encoder embodying aspects of the invention. Other
functional or
structural arrangements that practice aspects of the invention may be
employed, including
alternative and/or equivalent functional or structural arrangements described
below.
Two or more audio input channels are applied to the encoder. Although, in
principle, aspects of the invention may be practiced by analog, digital or
hybrid
analog/digital embodiments, examples disclosed herein are digital embodiments.
Thus,
the input signals may be time samples that may have been derived from analog
audio
signals. The time samples may be encoded as linear pulse-code modulation (PCM)
signals. Each linear PCM audio input channel is processed by a filterbank
finiction or
device having both an in-phase and a quadrature output, such as a 512-point
windowed
forward discrete Fourier transform (DFT) (as implemented by a Fast Fourier
Transform
(FFT)). The filterbank may be considered to be a time-domain to frequency-
domain
transform.
FIG. 1 shows a first PCM channel input (channel "1") applied to a filterbank
function or device, "Filterbank" 2, and a second PCM channel input (channel
"n")
applied, respectively, to another filterbank function or device, "Filterbank"
4. There may
be "n" input channels, where "n" is a whole positive integer equal to two or
more. Thus,
there also are "n" Filterbanks, each receiving a unique one of the "n" input
channels. For
simplicity in presentation, FIG. 1 shows only two input channels, "1" and "n".
When a Filterbank is implemented by an FFT, input time-domain signals are
segmented into consecutive blocks and are usually processed in overlapping
blocks. The
FFT's discrete frequency outputs (transform coefficients) are referred to as
bins, each
having a complex value with real and imaginary parts corresponding,
respectively, to in-
phase and quadrature components. Contiguous transform bins may be grouped into
subbands approximating critical bandwidths of the human ear, and most
sidechain
information produced by the encoder, as will be described, may be calculated
and
transmitted on a per-subband basis in order to minimize processing resources
and to
reduce the bitrate. Multiple successive time-domain blocks may be grouped into
frames,
with individual block values averaged or otherwise combined or accumulated
across each
frame, to minimize the sidechain data rate. In examples described herein, each
filterbank
is implemented by an FFT, contiguous transform bins are grouped into subbands,
blocks
are grouped into frames and sidechain data is sent on a once per-frame basis.
=
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Alternatively, sidechain data may be sent on a more than once per frame basis
(e.g., once
per block). See, for example, FIG. 3 and its description, hereinafter. As is
well known,
there is a tradeoff between the frequency at which sidechain information is
sent and the
required bitrate.
A suitable practical implementallon of aspects of the present invention may
employ fixed length frames of about 32 milliseconds when a48 kHz sampling rate
is
employed, each frame having six blocks at intervals of about 5.3 milliseconds
each
(employing, for example, blocks having a duration of about 10.6 milliseconds
with a 50%
overlap). However, neither such timings nor the employment of fixed length
frames nor
their division into a fixed number of blocks is critical to practicing aspects
of the
invention provided that information described herein as being sent on a per-
frame basis is
sent no less frequently than about every 40 milliseconds. Frames may be of
arbitrary size
and their size may vary dynamically. Variable block lengths may be employed as
in the
AC-3 system cited above. It is with that understanding that reference is made
herein to
"frames" and "blocks."
In practice, if the composite mono or multichannel signal(s), or the composite
mono or multichannel signal(s) and discrete low-frequency channels, are
encoded, as for
example by a perceptual coder, as described below, it is convenient to employ
the same '
frame and block configuration as employed in the perceptual coder. Moreover,
if the
coder employs variable block lengths such that there is, from time to time, a
switching
from one block length to another, it would be desirable if one or more of the
sidechain
information as described herein is updated when such a block switch occurs. In
order to
minimize the increase in data overhead upon the updating of sidechain
information upon
the occurrence of such a switch, the frequency resolution of the updated
sidechain
information may be reduced.
FIG. 3 shows an example of a simplified conceptual organization of bins and
subbands along a (vertical) frequency axis and blocks and a frame along a
(horizontal)
time axis. When bins are divided into subbands that approximate critical
bands, the
lowest frequency subbands have the fewest bins (e.g., one) and the number of
bins per
subband increase with increasing frequency.
Returning to FIG. 1, a frequency-domain version of each of the n time-domain
input channels, produced by the each channel's respective Filterbank
(Filterbanks 2 and 4
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in this example) are summed together ("downmixed") to a monophonic ("mono")
composite audio signal by an additive combining function or device "Additive
Combiner"
6.
The downmixing may be applied to the entire frequency bandwidth of the input
audio signals or, optionally, it may be limited to frequencies above a given
"coupling"
frequency, inasmuch as artifacts of the downmixing process may become more
audible at
middle to low frequencies. In such cases, the channels may be conveyed
discretely below
the coupling frequency. This strategy may be desirable even if processing
artifacts are
not an'issue, in that mid/low frequency,subbands constructed by grouping
transform bins
into critical-band-like subbands (size roughly proportional to frequency) tend
to have a
small number of transform bins at low frequencies (one bin at very low
frequencies) and
may be directly coded with as few or fewer bits than is required to send a
dowmnixed
mono audio signal with sidechain information. A coupling or transition
frequency as low
as 4 kHz, 2300 Hz, 1000 Hz, or even the bottom of the frequency band of the
audio
signals applied to the encoder, may be acceptable for some applications;
particularly those
in which a very low bitrate is important. Other frequencies may provide a
useful balance
between bit savings and listener acceptance. The choice of a particular
coupling
frequency is not critical to the invention. The coupling frequency may be
variable and, if
variable, it may depend, for example, directly or indirectly on input signal
characteristics.
Before downmixing, it is an aspect of the present invention to improve the
channels' phase angle alignments vis-à-vis each other, in order to reduce the
cancellation
of out-of-phase signal components when the channels are combined and to
provide an
improved mono composite channel. This may be accomplished by controllably
shifting
over time the "absolute angle" of some or all of the transform. bins in ones
of the
channels. For example, all of the transform bins representing audio above a
coupling
frequency, thus defining a frequency band of interest, may be controllably
shifted over
time, as necessary, in every channel or, when one channel is used as a
reference, in all but
the reference channel.
The "absolute angle" of a bin may be taken as the angle of the magnitude-and-
angle representation of each complex valued transform bin produced by a
filterbank.
Controllable shifting of the absolute angles of bins in a channel is performed
by an angle
rotation function or device ("Rotate Angle"). Rotate Angle 8 processes the
output of
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Filterbank 2 prior to its application to the downmix summation provided by
Additive
Combiner 6, while Rotate Angle 10 processes the output of Filterbank 4 prior
to its
application to the Additive Combiner 6. It will be appreciated that, under
some signal
conditions, no angle rotation may be required for a particular transform bin
over a time
period (the time period of a frame, in examples described herein). Below the
coupling
frequency, the channel information may be encoded discretely (not shown in
FIG. 1).
In principle, an improvement in the channels' phase angle alignments with
respect
to each other may be accomplished by shifting the phase of every transform bin
or
subband by the negative of its absolute phase angle, in each block throughout
the
frequency band of interest. Although this substantially avoids cancellation of
out-of-
phase signal components, it tends to cause artifacts that may be audible,
particularly if the
resulting mono composite signal is listened to in isolation. Thus, it is
desirable to employ
the principle of "least treatment" by shifting the absolute angles of bins in
a channel only
as much as necessary to Minimize out-of-phase cancellation in the downmix
process and
minimi7e spatial image collapse of the multichannel signals reconstituted by
the decoder.
Techniques for determining such angle shifts are described below. Such
techniques
include time and frequency smoothing and the manner in which the signal
processing
responds to the presence of a transient.
Energy normalization may also be performed on a per-bin basis in the encoder
to
reduce further any remaining out-of-phase cancellation of isolated bins, as
described
further below. Also as described further below, energy normalization may also
be
performed on a per-subband basis (in the decoder) to assure that the energy of
the mono
composite signal equals the sums of the energies of the contributing channels.
Each input channel has an audio analyzer function or device ("Audio Analyzer")
associated with it for generating the sidechain information for that channel
and for
controlling the amount or degree of angle rotation applied to the channel
before it is
applied to the downmix summation 6. The Filterbank outputs of channels 1 and n
are
applied to Audio Analyzer 12 and to Audio Analyzer 14, respectively. Audio
Analyzer
12 generates the sidechain information for channel 1 and the amount of phase
angle
rotation for channel 1. Audio Analyzer 14 generates the sidechain information
for
channel n and the amount of angle rotation for channel n. It will be
understood that such
references herein to "angle" refer to phase angle.
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The sidechain information for each charinel generated by an audio analyzer for
each channel may include:
an Amplitude Scale Factor ("Amplitude SF"),
an Angle Control Parameter,
a Decorrelation Scale Factor ("Decorrelation SF"),
a Transient Flag, and
optionally, an Interpolation Flag.
Such sidechain information may be characterized as "spatial parameters,"
indicative of
spatial properties of the channels and/or indicative of signal characteristics
that may be
relevant to spatial processing, such as transients. In each case, the
sidechain information
applies to a single subband (except for the Transient Flag and the
Interpolation Flag, each
of which apply to all subbands within a channel) and may be updated once per
frame, as
in the examples described below, or upon the occurrence of a block switch in a
related
coder. Further details of the various spatial parameters are set forth below.
The angle
rotation for a particular channel in the encoder may be taken as the polarity-
reversed
Angle Control Parameter that forms part of the sidechain information.
If a reference channel is employed, that channel may not require an Audio
. Analyzer or, alternatively? may require an Audio Analyzer that generates
only Amplitude
Scale Factor sidechain infonnation. It is not necessary to send an Amplitude
Scale Factor
if that scale factor can be deduced with sufficient accuracy by a decoder from
the
Amplitude Scale Factors of the other, non-reference, channels. It is possible
to deduce in
the decoder the approximate value of the reference channel's Amplitude Scale
Factor if
the energy normalization in the encoder assures that the scale factors across
channels
within any subband substantially sum square to 1, as described below. The
deduced
approximate reference channel Amplitude Scale Factor value may have errors as
a result
of the relatively coarse quantization of amplitude scale factors resulting in
image shifts in
the reproduced multi-channel audio. However, in a low data rate environment,
such
artifacts may be more acceptable than using the bits to send the reference
channel's
Amplitude Scale Factor. Nevertheless, in some cases it may be desirable to
employ an
audio analyzer for the reference channel that generates, at least, Amplitude
Scale Factor
sidechain information.
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FIG. 1 shows in a dashed line an optional input to each audio analyzer from
the
PCM time domain input to the audio analyzer in the channel. This input may be
used by
the Audio Analyzer to detect a transient over a time period (the period of a
block or
frame, in the examples described herein) and to generate a transient indicator
(e.g., a one-
bit "Transient Flag") in response to a transient. Alternatively, as described
below in the
comments to Step 408 of FIG. 4, a transient may be detected in the frequency
domain, in
which case the Audio Analyzer need not receive a time-domain input.
The mono composite audio signal and the sidechain information for all the
channels (or all the channels except the reference channel) may be stored,
transmitted, or
stored and transmitted to a decoding process or device ("Decoder").
Preliminary to the
storage, transmission, or storage and transmission, the various audio signals
and various
sidechain information may be multiplexed and packed into one or more
bitstreams
suitable for the storage, transmission or storage and transmission medium or
media. The
mono composite audio may be applied to a data-rate reducing encoding process
or device
such as, for example, a perceptual encoder or to a perceptual encoder and an
entropy
coder (e.g., arithmetic or Huffman coder) (sometimes referred to as a
"kissless" coder)
prior to storage, transmission, or storage and transmission. Also, as
mentioned above, the
mono composite audio and related sidechain information may be derived from
multiple
input channels only for audio frequencies above a certain frequency (a
"coupling"
frequency). In that case, the audio frequencies below the coupling frequency
in each of
the multiple input channels may be stored, transmitted or stored and
transmitted as
discrete channels or may be combined or processed in some manner other than as
described herein. Such discrete or otherwise-combined channels may also be
applied to a
data reducing encoding process or device such as, for example, a perceptual
encoder or a
perceptual encoder and an entropy encoder. The mono composite audio and the
discrete
multichannel audio may all be applied to an integrated perceptual encoding or
perceptual
and entropy encoding process or device.
The particular manner in which sidechain information is carried in the encoder
bitstream is not critical to the invention. If desired, the sidechain
information may be
carried in such as way that the bitstream is compatible with legacy decoders
(i.e., the
bitstream is backwards-compatible). Many suitable techniques for doing so are
known.
For example, many encoders generate a bitstream having unused or null bits
that are
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..ignored by the decoder. An example of such an arrangement is set forth in
United States
Patent 6,807,528 DI of Truman et al, entitled "Adding Data to a Compressed
Data
Frame," October 19, 2004.
Such bits may be replaced with the sidechain information. Another example is
, 5 that the sidechain information may be steganographically encoded in
the encoder's =
bitstream. Alternatively, the sidechain information may be stored or
transmitted
separately from the backwards-compatible bitstream by any technique that
permits the
transmission or storage of such information along with a mono/stereo bitstream
compatible with legacy decoders.
= 10 =Basic 1:N and 1:M Decoder
= Referring to FIG. 2, a decoder function or device ("Decoder") embodying
aspects
of the present invention is shown. The figure is an example of a function or
structure that
performs as a basic decoder embodying aspects of the invention. Other
functional or
structural arrangements that practice aspects of the invention may be
employed, including
15 alternative and/or equivalent functional or structural arrangements
described below. =
The Decoder receives the mono composite audio signal and the sidechain
information for all the channels or all the channels except the reference
channel. If
necessary, the composite audio signal and related sidechain information is
demultiplexed, =
unpacked and/or decoded. Decoding may employ a table lookup. The goal is to
derive
20 = from the mono composite audio channels a plurality of individual
audio channels
= approximating respective ones of the audio chatmels applied to the
Encoder of FIG. 1, =
subject to bitrate-reducing techniques of the present invention that are
described herein.
= Of course, one may choose not to recover all of the channels applied to
the
encoder or to use only the monophonic composite signal. Alternatively,
channels in
25 addition to the ones applied to the Encoder may be derived from the
output of a Decoder
according to aspects of the present invention by employing aspects of the
inventions
= described in International .Application PCT/US 02/03619, filed February
7, 2002,
published August 15,.2002, designating the United States, and its resulting
U.S. national
= application S.N. 10/467,213, filed August 5, 2003, and in International
Application.
30 PCT/US03/24570, filed August 6, 2003, published March 4, 2001 as WO
2004/019656,
designating the United States, and its resulting U.S. national application
S.N. 10/522,515,
filed Jannary 27, 2005.
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Channels recovered by a Decoder practicing aspects of the present invention
are
particularly useful in connection with the channel multiplication techniques
of the cited
applications in that the recovered channels not only have useful
interchannel amplitude relationships but also have useful interchannel phase
relationships.
Another alternative for channel multiplication is to empley a matrix decoder
to derive
= additional channels. The interchannel amplitude- and phase-preservation
aspects of the
present invention make the output. channels of a decoder embodying _aspects of
the .
present invention particularly suitable for application to an amplitude- and
phase-sensitive
matrix decoder. Many such matrix decoders employ wideband control circuits
that
operate properly only when the signals applied to them are stereo throughout
the signals'
bandwidth.. Thus, if the aspects of the present invention are embodied in an
N:1:N system
= in which. N is 2, the two channels recovered by the decoder may be
applied to a 2:M .
active matrix decoder. Such channels may have been discrete channels below a
coupling
= frequency, as mentioned above. Many suitable active matrix deceders are
well known in
the art, including, for example, matrix decoders known as "Pro Logic" and "Pro
Logic II"
= = decoders ("Pro Logic" is a trademark of Dolby Laboratories
Licensing Corporation).
= Aspects of Pro Logic decoders are disclosed in U.S. Patents 4,799,260 and
4,941,177.
=
Aspects of Pro Logic II
= decoders are disclosed in pending U.S. Patent Application S.N. 09/532,711
of Fosgate,
entitled "Method for Deriving at Least Three Audio Signal N from Two Input
Audio
Signals,' filed March 22, 2000 and published as WO 01/41504 on June 7, 2001,
and in
pending U.S. Patent Application S.N. 10/362,786 of Fosgate et al, entitled
"Method for
Apparatus for Audio Matrix Decoding," filed February 25, 2003 and published as
US
200410125960 Al' on July 1, 2004.
Some aspects of the operation of Dolby Pro Logic and Pro Logic II
=
decoders are explained, for example, in papers available on the Dolby
Laboratories'
website (www.dolby.com): "Dolby Surround Pro Logic Decoder Principles of
Operation," by Roger Dressler, and "Mixing with Dolby Pro Logic II Technology,
by Jim
Hilson. Other suitable active matrix decoders may include those described in
one or more
Of the following U.S. Patents and published International Applications (each
designating
the United States) t
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5,046,098; 5,274,740; 5,400,433; 5,625,696; 5,644,640; 5,504,819; 5,428,687;
5,172,415;
and WO 02/19768.
Referring again to. FIG. 2, the received mono composite audio channel is
applied
to a plurality of signal paths from which a respective one of each of the
recovered
multiple audio channels is derived. Each channel-deriving path includes, in
either order,
an amplitude adjusting function or device ("Adjust Amplitude") and an angle
rotation
function or device ("Rotate Angle").
= The Adjust Amplitudes apply gains or losses to the mono composite signal
so that,
= under certain signal conditions, the relative output magnitudes (or
energies) of the output
channels derived from it are similar to those of the channels at the input of
the encoder.
Alternatively, under certain signal conditions when "randomi7ed" angle
variations are
imposed, as next described, a controllable amount of "randomized" amplitude
variations
may also be irnposed on the amplitude of a recovered channel in order to
improve its
decorrelation with respect to other ones of the recovered channels.
The Rotate Angles apply phase rotations so that, under certain signal
conditions,
the relative phase angles of the output channels derived from the mono
composite signal,
are similar to those of the channels at the input of the encoder. Preferably,
under certain
signal conditions, a controllable amo-unt of "randomized" angle variations is
also imposed
on the angle of a recovered channel in order to improve its decorrelation with
respect to
other ones of the recovered channels.
As discussed further below, "randomized" angle amplitude variations may
include
not only pseudo-random and truly random variations, but also deterministically-
generated
variations that have the effect of reducing cross-correlation between
channels. This is
discussed further below in the Comments to Step 505 of FIG. 5A.
Conceptually, the Adjust Amplitude and Rotate Angle for a particular channel
scale the mono composite audio DFT coefficients to yield reconstructed
transform bin
values for the channel.
The Adjust Amplitude for each channel may be controlled at least by the
recovered sidechain Amplitude Scale Factor for the particular channel or, in
the case of
the reference channel, either from the recovered sidechain Amplitude Scale
Factor for the
reference channel or from an Amplitude Scale Factor deduced from the recovered
sidechain Amplitude Scale Factors of the other, non-reference, channels.
Alternatively,
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to enhance decorrelation of the recovered channels, the Adjust Amplitude may
also be
controlled by a Randomized Amplitude Scale Factor Parameter derived from the
recovered sidechain Decorrelation Scale Factor for a particular channel and
the recovered
sidechain Transient Flag for the particular channel.
The Rotate Angle for each channel may be controlled at least by the recovered
sidechain Angle Control Parameter (in which case, the Rotate Angle in the
decoder may
substantially undo the angle rotation provided by the Rotate Angle in the
encoder). To
enhance decorrelation of the recovered channels, a Rotate Angle may also be
controlled
by a Randomized Angle Control Parameter derived from the recovered sidechain
Decorrelation Scale Factor for a particular channel and the recovered
sidechain Transient
Flag for the particular channel. The Randomized Angle Control Parameter for a
channel,
and, if employed, the Randomized Amplitude Scale Factor for a channel, may be
derived
from the recovered Decorrelation Scale Factor for the channel and the
recovered
Transient Flag for the channel by a controllable decorrelator function or
device
("Controllable Decorrelator").
Referring to the example of FIG. 2, the recovered mono composite audio is
applied to a first channel audio recovery path 22, which derives the channel 1
audio, and
to a second channel audio recovery path 24, which derives the channel n audio.
Audio
path 22 includes an Adjust Amplitude 26, a Rotate Angle 28, and, if a PCM
output is
desired, an inverse filterbank function or device ("Inverse Filterbank") 30.
Similarly,
audio path 24 includes an Adjust Amplitude 32, a Rotate Angle 34, and, if a
PCM output
is desired, an inverse filterbank function or device ("Inverse Filterbank")
36. As with the
case of FIG. 1, only two channels are shown for simplicity in presentation, it
being
understood that there may be more than two channels.
The recovered sidechain information for the first channel, channel 1, may
include
an Amplitude Scale Factor, an Angle Control Parameter, a Decorrelation Scale
Factor, a.
Transient Flag, and, optionally, an Interpolation Flag, as stated above in
connection with
the description of a basic Encoder. The Amplitude Scale Factor is applied to
Adjust
Amplitude 26. If the optional Interpolation Flag is employed, an optional
frequency
interpolator or interpolator function ("Interpolator") 27 may be employed in
order to
interpolate the Angle Control Parameter across frequency (e.g., across the
bins in each
subband of a channel). Such interpolation may be, for example, a linear
interpolation of
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the bin angles between the centers of each subband. The state of the one-bit
Interpolation
Flag selects whether. or not interpolation across frequency is employed, as is
explained
further below. The Transient Flag and Decorrelation Scale Factor are applied
to a
Controllable Decorrelator 38 that generates a Randomized Angle Control
Parameter in
response thereto. The state of the one-bit Transient Flag selects one of two
multiple
modes of randomized angle decorrelation, as is explained further below. The
Angle
Control Parameter, which may be interpolated across frequency if the
Interpolation Flag
and the Interpolator are employed, and the Randomized Angle Control Parameter
are
summed together by an additive combiner or combining function 40 in order to
provide a
= 10 control signal for Rotate Angle 28. Alternatively, the Controllable
Decorrelator 38 may
also generate a Randomized Amplitude Scale Factor in response to the Transient
Flag and
Decorrelation Scale Factqr, in addition to generating a Randomized Angle
Control
Parameter. The Amplitude Scale Factor may be summed together with such a
Randomized Amplitude Scale Factor by an additive combiner or combining
function (not
shown) in order to provide the control signal for the Adjust Amplitude 26.
Similarly, recovered sidechain information for the second channel, channel n,
may
also include an Amplitude Scale Factor, an Angle Control Parameter, a
Decorrelation
Scale Factor, a Transient Flag, and, optionally, an Interpolate Flag, as
described above in
connection with the description of a basic encoder. The Amplitude Scale Factor
is
applied to Adjust Amplitude 32. A frequency interpolator or interpolator
function
("Interpolator") 33 may be employed in order to interpolate the Angle Control
Parameter
across frequency. As with channel 1, the state of the one-bit Interpolation
Flag selects
whether or not interpolation across frequency is employed. The Transient Flag
and
Decorrelation Scale Factor are applied to a Controllable Decorrelator 42 that
generates a
Randomized Angle Control Parameter in response thereto. As with channel 1; the
state of
the one-bit Transient Flag selects one of two multiple modes of randomized
angle
decorrelation, as is explained further below. The Angle Control Parameter and
the
Randomized Angle Control Parameter are summed together by an additive combiner
or
combining function 44 in order to provide a control signal for Rotate Angle
34.
Alternatively, as described above in connection with channel 1, the
Controllable
Decorrelator 42 may also generate a Randomized Amplitude Scale Factor in
response to
the Transient Flag and Decorrelation Scale Factor, in addition to generating a
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Randomized Angle Control Parameter. The Amplitude Scale Factor and Randomized
Amplitude Scale Factor may be summed together by an additive combiner or
combining
function (not shown) in order to provide the control signal for the Adjust
Amplitude 32.
Although a process or topology as just described is useful for understanding,
essentially the same results may be obtained with alternative processes or
topologies that
achieve the same or similar results. For example, the order of Adjust
Amplitude 26 (32)
and Rotate Angle 28 (34) may be reversed and/or there may be more than one
Rotate
= Angle ¨ one that responds to the Angle Control Parameter and another that
responds to
= the Randorni7ed Angle Control Parameter. The Rotate Angle may also be
considered to
be three rather than one or two functions or devices, as in the example of
FIG. 5 described
= below. If a Randomized Amplitude Scale Factor is employed, there may be
more than
one Adjust Amplitude ¨ one that responds to the Amplitude Scale Factor and one
that
responds to the Randomized Amplitude Scale Factor. Because of the human ear's
greater
sensitivity to amplitude relative to phase, if a Randomized Amplitude Scale
Factor is
employed, it may be desirable to scale its effect relative to the effect of
the Randomized
Angle Control Parameter so that its effect on amplitude is less than the
effect that the
Randomized Aligle Control Parameter has on phase angle. As another alternative
process.
or topology, the D.ecorrelation Scale Factor may, be used to control the ratio
of
randornind phase angle versus basic phase angle (rather than adding a
parameter
representing a randomized phase angle to a parameter representing the basic
phase angle),
and if also employed, the ratio of randomized amplitude shift versus basic
amplitude shift
(rather than adding a scale factor representing a randomized amplitude to a
scale factor
representing the basic amplitude) e., a variable crossfade in each case).
If a reference channel is employed, as discussed above in connection with the
basic encoder, the Rotate Angle, Controllable Decorrelator and Additive
Combiner for
that channel may be omitted inasmuch as the sidechain information for the
reference
channel may include only the Amplitude Scale Factor (or, alternatively, if the
sidechain
information does not contain an Amplitude Scale Factor for the reference
channel, it may
be deduced from Amplitude Scale Factors of the other channels when the energy
normalization in the encoder assures that the scale factors across channels
within a
subband sum square to 1). An Amplitude Adjust is provided for the reference
channel
and it is controlled by a received or derived Amplitude Scale Factor for the
reference
=
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channel. Whether the reference channel's Amplitude Scale Factor is derived
from the ,
sidechain or is deduced in the decoder, the recovered reference channel is an
amplitude-
scaled version of the mono composite channel. It does not require angle
rotation because
it is the reference for the other channels' rotations.
Although adjusting the relative amplitude of recovered channels may provide a
modest degree of decorrelation, if used alone amplitude adjustment is likely
to result in a
reproduced soundfield substantially lacking in spatiali7ation or imaging for
many signal
conditions (e.g., a "collapsed" soundfield). Amplitude adjustment may affect
interaural
level differences at the ear, which is only one of the psychoacoustic
directional cues
employed by the ear. Thus, according to aspects of the invention, certain
angle-adjusting
techniques may be employed, depending on signal conditions, to provide
additional
decorrelation. Reference may be made to Table 1 that provides abbreviated
comments
useful in understanding the multiple angle-adjusting decorrelation techniques
or modes of
operation that may be employed in accordance with aspects of the invention.
Other
decorrelation techniques as described below in connection with the examples of
FIGS. 8
and 9 may be employed instead of or in addition to the techniques of Table 1.
In practice, applying angle rotations and magnitude alterations may result in
circular convolution (also known as cyclic or periodic convolution). Although,
generally,
it is desirable to avoid circular convolution, undesirable audible artifacts
resulting from
circular convolution are somewhat reduced by complementary angle shifting in
an
encoder and decoder.. In addition, the effects of circular convolution may be
tolerated in
low cost implementations of aspects ofthe present invention, particularly
those in which
the downmixing to mono or multiple channels occurs only in part of the audio
frequency
band, such as, for example above 1500 Hz (in which case the audible effects of
circular
convolution are minimal). Alternatively, circular convolution may be avoided
or
minimized by any suitable technique, including, for example, an appropriate
use of zero
padding. One way to use zero padding is to transform the proposed frequency
domain
= variation (representing angle rotations and amplitude scaling) to the
time domain, window
it (with an arbitrary window), pad it with zeros, then transform back to the
frequency
domain and multiply by the frequency domain version of the audio to be
processed (the
audio need not be windowed).
Table 1
Angle-Adjusting Decorrelation Techniques
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Technique 1 Technique 2 Technique 3
Type of Signal Spectrally static Complex continuous Complex
impulsive
(typical example) source signals signals (transients)
Effect on Decorrelates low Decorrelates non- Decorrelates
Decorrelation frequency and impulsive complex impulsive high
steady-state signal signal components frequency signal
components components
Effect of transient Operates with Does not operate Operates
present in frame shortened time
constant
What is done Slowly shifts Adds to the angle of Adds to the angle
of
(frame-by-frame) Technique 1 a time- Technique 1 a
bin angle in a = invariant rapidly-changing
channel randomized angle (block by block)
on a bin-by-bin randomized angle
basis in a channel on a subband-by-
subband basis in a
channel
Controlled by or Basic phase angle is Amount of = Amount of=
Scaled by controlled by Angle randomized angle is randomized angle
is
Control Parameter scaled directly by scaled indirectly
by
Decorrelation SF; Decorrelation SF;
same scaling across same scaling across
= subband, scaling
'subband, scaling
= updated every frame updated every frame
Frequency = Subband (same or Bin (different Subband (same
Resolution of angle interpolated shift randomized shift randomized shift
shift value applied to all value applied to value applied to
all
, bins in each each bin) bins in each
subband) subband; different
randomized shift
= value applied to
= each subband in
= channel)
Time Resolution Frame (shift values Randomi7ed shift Block (randomized
updated every values remain the shift values
updated
frame) same and do not every block)
change
For signals that are substantially static spectrally, such as, for example, a
pitch
pipe note, a first technique ("Technique 1") restores the angle of the
received mono
composite signal relative to the angle of each of the other recovered channels
to an angle
similar (subject to frequency and time granularity and to quantization) to the
original
angle of the channel relative to the other channels at the input of the
encoder. Phase angle =
differences are useful, particularly, for providing decorrelation of low-
frequency signal
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components below about 1500 Hz where the ear follows individual cycles of the
audio
signal. Preferably, Technique 1 operates under all signal conditions to
provide a basic
angle shift.
For high-frequency signal components above about 1500 Hz, the ear does not
follow individual cycles of sound but instead responds to waveform envelopes
(on a
critical band basis). Hence, above about 1500 Hz decorrelation is better
provided by
differences in signal envelopes rather than phase angle differences. Applying
phase angle
shifts only in accordance with Technique 1 does not alter the envelopes of
signals
sufficiently to decorrelate high frequency signals. The second and third
techniques
("Technique 2" and "Technique 3", respectively) add a controllable amount of
randomized angle variations to the angle determined by Technique 1 under
certain signal
conditions, thereby causing a controllable amount of randomized envelope
variations,
which enhances decorrelation.
Randomized changes in phase angle are a desirable way to cause randomized
changes in the envelopes of signals. A particular envelope results from the
interaction of
a particular combination of amplitudes and phases of spectral components
within a
subband. Although changing the amplitudes of spectral components within a
subband
changes the envelope, large amplitude changes are required to obtain a
significant change
in the envelope, which is undesirable because the human ear is sensitive to
variations in
spectral amplitude. In contrast, changing the spectral component's phase
angles has a
greater effect on the envelope than changing the spectral component's
amplitudes ¨
spectral components no longer line up the same way, so the reinforcements and
subtractions that define the envelope occur at different times, thereby
changing the
envelope. Although the human ear has some envelope sensitivity, the ear is
relatively
phase cleat so the overall sound quality remains substantially similar.
Nevertheless, for
some signal conditions, some randomization of the amplitudes of spectral
components
along with randomization of the phases of spectral components may provide an
enhanced
randomization of signal envelopes provided that such amplitude randomization
does not
cause undesirable audible artifacts.
. Preferably, a controllable amount or degree of Technique 2 or Technique 3
= .
operates along with Techin. 'que 1 under-certain signal conditions. The
Transient Flag
. selects Technique 2 (no transient present in the frame or block,
depending on whether the
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Transient Flag is sent at the frame or block rate) or Technique 3 (transient
present in the
frame or block). Thus, there are multiple modes of operation, depending on
whether or
not a transient is present. Alternatively, in addition, under certain signal
conditions, a
controllable amount or degree of amplitude randomization also operates along
with the
amplitude scaling that seeks to restore the original channel amplitude.
Teolmique 2 is suitable for complex continuous signals that are rich in
harmonics,
such as massed orchestral violins. Technique 3 is suitable for complex
impulsive or
transient signals, such as applause, castanets, etc. (Technique 2 time smears
claps in
applause, making it unsuitable for such signals). As explained further below,
in. order to
minimize audible artifacts, Technique 2 and Technique 3 have different time
and
frequency resolutions for applying randomized angle variations ¨ Technique 2
is
selected when a transient is not present, whereas Technique 3 is selected when
a transient
is present.
Technique 1 slowly shifts (fraMe by frame) the bin angle in a channel. The
amount or degree of this basic shift is controlled by the Angle Control
Parameter (no shift
if the parameter is zero). As explained further below, either the same or an
interpolated
parameter is applied to all bins in each subband and the parameter is updated
every frame.
Consequently, each subband of each channel may have a phase shift with respect
to other
channels, providing a degree of decorrelation at low frequencies (below about
1500 Hz).
However, Technique 1, by itself, is unsuitable for a transient signal such as
applause. For
such signal conditions, the reproduced channels may exhibit an annoying
unstable comb-
filter effect. In the case of applause, essentially no decorrelation is
provided by adjusting
only the relative amplitude of recovered channels because all channels tend to
have the
same amplitude over the period of a frame.
Technique 2 operates when a transient is not present. Technique 2 adds to the
angle shift of Technique 1 a randomized angle shift that does not change with
time, on a
bin-by-bin basis (each bin has a different randomized shift) in a channel,
causing the
envelopes of the channels to be different from one another, thus providing
decorrelation
of complex signals among the channels. Maintaining the randomized phase angle
values
constant over time avoids block or frame artifacts that may result from block-
to-block or
frame-to-frame alteration of bin phase angles. While this technique is a very
useful
decorrelation tool when a transient is not present, it may temporally smear a
transient
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(resulting in what is often referred to as "pre-noise" ¨ the post-transient
smearing is
masked by the transient). The amount or degree of additional shift provided by
Technique 2 is scaled directly by the Decorrelation Scale Factor (there is no
additional
shift if the scale factor is zero). Ideally, the amount of randomized phase
angle added to
the base angle shift (of Technique 1) according to Te hnique 2 is controlled
by the
Decorrelation Scale Factor in a manner that minimizes audible signal warbling
artifacts.
Such minimization of signal warbling artifacts results from the manner in
which the
Decorrelation Scale Factor is derived and the application of appropriate time
smoothing,
as described below. Although a different additional randomized angle shift
value is
applied to each bin and that shift value does not change, the same scaling is
applied
across a subband and the scaling is updated every frame.
Technique 3 operates in the presence of a transient in the frame or block,
depending on the rate at which the Transient Flag is sent. It shifts all the
bins in each
subband in a channel from block to block with a unique randomized angle value,
common
to all bins in the subband, causing not only the envelopes, but also the
amplitudes and
phases, of the signals in a channel to change with respect to other channels
from block to
block. These changes in time and frequency resolution of the angle randomizing
reduce
steady-state signal similarities among the channels and provide decorrelation
of the
channels substantially without causing "pre-noise" artifacts. The change in
frequency
resolution of the angle randomizing, from very fine (all bins different in a
channel) in
Technique 2 to coarse (all bins within a subband the same, but each subband
different) in
Technique 3 is particularly useful in minimizing "pre-noise" artifacts.
Although the ear
does not respond to pure angle changes directly at high frequencies, when two
or more
channels mix acoustically on their way from loudspeakers to a lisiener, phase
differences
may cause amplitude changes (comb-filter effects) that may be audible and
objectionable,
and these are broken up by Technique 3. The impulsive characteristics of the
signal
minimize block-rate artifacts that might otherwise occur. Thus, Technique 3
adds to the
phase shift of Terhnique 1 a rapidly changing (block¨by-block) randomized
angle shift
on a subband-by-subband basis in a channel. The amount or degree of additional
shift is
scaled indirectly, as described below, by the Decorrelation Scale Factor
(there is no
additional shift if the scale factor is zero). The same scaling is applied
across a subband
and the scaling is updated every frame.
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Although the angle-adjusting techniques have been characterized as three
techniques, this is a matter of semantics and they may also be characterized
as two
techniques: (1) a combination of Technique 1 and a variable degree of
Technique 2,
which may be zero, and (2) a combination of Technique 1 and a variable degree
Technique 3, which may be zero. For convenience in presentation, the
techniques are
treated as being three techniques.
Aspects of the multiple mode decorrelation techniques and modifications of
them
may be employed in providing decorrelation of audio signals derived, as by
upmixing,
from one or more audio channels even when such audio channels are not derived
from an
encoder according to aspects of the present invention. Such arrangements, when
applied
to a mono audio channel, are sometimes referred to as "pseudo-stereo" devices
and
functions. Any suitable device or function (an "upmixer") may be employed to
derive
multiple signals from a mono audio channel or from multiple audio channels.
Once such
multiple audio channels are derived by an upmixer, one or more of them may be
decorrelated with respect to one or more of the other derived audio signals by
applying
the multiple mode decorrelation techniques described herein. In such an
application, each
derived audio channel to which the decorrelation techniques are applied may be
switched
from one mode of operation to another by detecting transients in the derived
audio
channel itself. Alternatively, the operation of the transient-present
technique (Technique
3) may be simplified to provide no shifting of the phase angles of spectral
components
when a transient is present.
Sidechain Information
As mentioned above, the sidechain information may include: an Amplitude Scale
Factor, an Angle Control Parameter, a Decorrelation Scale Factor, a Transient
Flag, and,,
optionally, an Interpolation Flag. Such sidechain information for a practical
embodiment
of aspects of the present invention may be summarized in the following Table
2.
Typically, the sidechain information may be updated once per frame.
Table 2
Sidechain Information Characteristics for a Channel
Sidechain Represents Quantization
Primary
Information Value Range (is "a measure Levels
Purpose
of')
Subband Angle 0 4+27c Smoothed time 6 bit (64 levels) Provides
Control average in each basic
angle
Parameter subband of
rotation for
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Side,chain Represents Quantization Primary
Information Value Range (is "a measure Levels Purpose
of')
difference each bin in
between angle of . channel
each bin in
subband for a
channel and that
of the
corresponding bin
in subband of a
reference channel
Subband 0 41 Spectral- 3 bit (8 levels) Scales
Decorrelation The Subband steadiness of randomized
Scale Factor Decorrelation signal angle
shifts
Scale Factor is characteristics added to
high only if over time in a basic angle
both the subband of a rotation, and,
Spectral- channel (the if employed,
Steadiness Spectral- also scales
Factor and the Steadiness =randomized
Interchannel Factor) and the Amplitude
Angle consistency in the Scale Factor
Consistency same subband of added to
Factor are low. a channel of bin basic
angles with Amplitude
respect to Scale Factor,
corresponding = and,
bins of a optionally,
reference channel scales degree
(the Interchannel of
Angle reverberation
Consistency
Factor)
Subband 0 to 31 (whole Energy or 5 bit (32 levels) Scales
Amplitude integer) amplitude in Granularity is amplitude of
Scale Factor 0 is highest subband of a 1.5 dB, so the
bins in a
amplitude channel with range is 31*1.5 = subband in a
31 is lowest respect to energy 46.5 dB
plus channel
amplitude or amplitude for final value = off
same subband
across all
channels
_
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Sidechain Represents Quantization Primary
Information Value Range (is ,"a measure Levels Purpose
of')
Transient Flag 1, 0 = Presence of a 1 bit (2 levels)
Determines
(True/False) transient in the which
(polarity is frame or in the technique for
arbitrary) block adding
randomized =
angle shifts,
or both angle
shifts and
amplitude
shifts, is
employed
Interpolation 1, 0 A spectral peak 1 bit (2 levels)
Detemiines
Flag (True/False) near a subband if the basic
(polarity is boundary or angle
arbitrary) phase angles rotation is
within a channel interpolated
have a linear across
progression frequency
In each case, the sidechain information of a channel applies to a single
subband
(except for the Transient Flag and the Interpolation Flag, each of which apply
to all
subbands in a channel) and may be updated once per frame. Although the time
resolution
(once per frame), frequency resolution (subband), value ranges and
quantization levels
indicated have been found to provide useful performance and a useful
compromise
between a low bitrate and performance, it will be appreciated that these time
and
frequency resolutions, value ranges and quantization levels are not critical
and that other
resolutions, ranges and levels may employed in practicing aspects of the
invention. For
example, the Transient Flag and/or the Interpolation Flag, if employed, may be
updated
once per block with only a minimal increase in sidechain data overhead. In the
case of
the Transient Flag, doing so has the advantage that the switching from
Technique 2 to
Technique 3 and vice-versa is more accurate. In addition, as mentioned above,
sidechain
inforrnation may be updated upon the occurrence of a block switch of a related
coder.
It will be noted that Technique 2, described above (see also Table 1),
provides a
bin frequency resolution rather than a subband frequency resolution (i.e., a
different
pseudo random phase angle shift is applied to each in rather than to each
subband) even
though the same Subband Decorrelation Scale Factor applies to all bins in a
subband. It
=
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will also be noted that Technique 3, described above (see also Table 1),
provides a block
frequency resolution (i.e., a different randomized phase angle shift is
applied to each
block rather than to each frame) even though the same Subband Decorrelation
Scale
Factor applies to all bins in a subband. Such resolutions, greater than the
resolution of the
sidechain information, are possible because the randomized phase angle shifts
may be
generated in a decoder and need not be known in the encoder (this is the case
even if the
encoder also applies a randomind phase angle shift to the encoded mono
composite
signal, an alternative that is described below). In other words, it is not
necessary to send
sidechain information having bin or block granularity even though the
decorrelation
techniques employ such granularity. The decoder may employ, for example, one
or more
lookup tables of randomized bin phase angles. The obtaining of time and/or
frequency
resolutions for decorrelation greater than the sidechain information rates is
among the
aspects of the present invention. Thus, decorrelation by way of randomized
phases is
performed either with a fine frequency resolution (bin-by-bin) that does not
change with
time (Technique 2), or with a coarse frequency resolution (band-by-band) ((or
a fine
frequency resolution (bin-by-bin) when frequency interpolation is employed, as
described
further below)) and a fine time resolution (block rate) (Technique 3).
It will also be appreciated that as increasing degrees of randomized phase
shifts
are added to the phase angle of a recovered channel, the absolute phase angle
of the
recovered channel differs more and more from the original absolute phase angle
of that
channel. An aspect of the present invention is the appreciation that the
resulting absolute
phase angle of the recovered channel need not match that of the original
channel when
signal conditions are such that the randomized phase shifts are added in
accordance with
aspects of the present invention. For example, in extreme cases when the
Decorrelation
Scale Factor causes the highest degree Of randomized phase shift, the phase
shift caused
by Technique 2 or Technique 3 overwhelms the basic phase shift caused by
Technique 1.
Nevertheless, this is of no concern in that a randomized phase shift is
audibly the same as
the different random phases in the original signal that give rise to a
Decorrelation Scale
Factor that causes the addition of some degree of randomized phase shifts.
As mentioned above, randomized amplitude shifts may by employed in addition to
randomized phase shifts. For example, the Adjust Amplitude may also be
controlled by a
Randomized Amplitude Scale Factor Parameter derived from the recovered
sidechain
=
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Decorrelation Scale Factor for a particular channel and the recovered
sidechain Transient
Flag for the particular channel. Such randomized amplitude shifts may operate
in two
modes in a manner analogous to the application of randomi7ed phase shifts. For
example,
in the absence of a transient, a randomi7ed amplitude shift that does not
change with time
may be added on a bin-by-bin basis (different from bin to bin), and, in the
presence of a
transient (in the frame or block), a randomized amplitude shift that changes
on a block-
by-block basis (different from block to block) and changes from subband to
subband (the
same shift for all bins in a subband; different from subband to subband).
Although the
amount or degree to which randomized amplitude shifts are added may be
controlled by
the Decorrelation Scale Factor, it is believed that a particular scale factor
value should
cause less amplitude shift than the corresponding randomind phase shift
resulting from
the same scale factor value in order to avoid audible artifacts.
When the Transient Flag applies to a frame, the time resolution with which the
Transient Flag selects Technique 2 or Technique 3 may be enhanced by providing
a
supplemental transient detector in the decoder in order to provide a temporal
resolution
finer than the frame rate or even the block rate. Such a supplemental
transient detector
may detect the occurrence of a transient in the mono or multichannel composite
audio
signal received by the decoder and such detection information is then sent to
each
Controllable Decorrelator (as 38, 42 of FIG. 2). Then, upon the receipt of a
Transient
Flag for its channel, the Controllable Decorrelator switches from Technique 2
to
Technique 3 upon receipt of the decoder's local transient detection
indication. Thus, a
substantial improvement in temporal resolution is possible without increasing
the
sidechain bitrate, albeit with decreased spatial accuracy (the encoder detects
transients in
each input channel prior to their downmixing, whereas, detection in the
decoder is done
after downmixing).
As an alternative to sending sidechain information on a frame-by-frame basis,
sidechain information may be updated every block, at least for highly dynamic
signals.
As mentioned above, updating the Transient Flag and/or the Interpolation Flag
every
block results in only a small increase in sidechain data overhead. In order to
accomplish
such an increase in temporal resolution for other sidechain information
without
substantially increasing the sidechain data rate, a block-floating-point
differential coding
arrangement may be used. For example, consecutive transform blocks may be
collected
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in groups of six over a frame. The full sidechain information may be sent for
each
subband-channel in the first block. In the five subsequent blocks, only
differential values
may be sent, each the difference between the current-block amplitude and
angle, and the =
equivalent values from the previous-block. This results in very low data rate
for static
signals, such as a pitch pipe note. For more dynamic signals, a greater range
of difference
values is required;but at less precision. So, for each group of five
differential values, an
exponent may be sent ftrst, using, for example, 3 bits, then differential
values are
quantized to, for example, 2-bit accuracy. This arrangement reduces the
average worst-
case sidechain data rate by about a factor of two. Further reduction may be
obtained by
omitting the sidechain data for a reference channel (since it can be derived
from the other
channels), as discussed above, and by using, for example, arithmetic coding.
Alternatively or in addition, differential coding across frequency may be
employed by
sending, for example, differences in subband angle or amplitude.
Whether sidechain information is sent on a frame-by-frame basis or more
frequently, it may be useful to interpolate sidechain values across the blocks
in a frame.
Linear interpolation over time may be employed in the manner of the linear
interpolation
across frequency, as described below.
One suitable implementation of aspects of the present invention employs
processing steps or devices that implement the respective processing steps and
are
= functionally related as next set forth. Although the encoding and decoding
steps listed
below may each be carried out by computer software instruction sequences
operating in
the order of the below listed steps, it will be understood that equivalent or
similar results
may be obtained by steps ordered in other ways, taking into account that
certain quantities
are derived from earlier ones. For example, multi-threaded computer software
instruction
= 25 sequences may be employed so that certain sequences of steps are
carried out in parallel.
Alternatively, the described steps may be implemented as devices that perform
the
described functions, the various devices having functions and functional
interrelationships
as described hereinafter.
Encoding
= 30 The
encoder or encoding function may collect a frame's worth of data before it
derives sidechain information and downmixes the frame's audio channels to a
single
monophonic (mono) audio channel (in the manner of the example of FIG. 1,
described
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above), or to multiple audio channels (in the manner of the example of FIG. 6,
described
below). By doing so, sidechain information may be sent first to a decoder,
allowing the
decoder to begin decoding immediately upon receipt of the mono or multiple
channel
audio information. Steps of an encoding process ("encoding steps") may be
described as
follows. With respect to encoding steps, reference is made to FIG. 4, which is
in the =
nature of a hybrid flowchart and functional block diagram. Through Step 419,
FIG. 4
shows encoding steps for one channel. Steps 420 and 421 apply to all of the
multiple
channels that are combined to provide a composite mono signal output or are
matrixed.
together to provide multiple channels, as described below in connection with
the example
of FIG. 6.
Step 401, Detect Transients
a. Perform transient detection of the PCM values in an input audio channel.
b. Set a one-bit Transient Flag True if a transient is present in any block of
a frame
for the channel.
Comments regarding Step 401:
The Transient Flag forms a portion of the sidechain information and is also
used
in Step 411, as described below. Transient resolution finer than block rate in
the decoder
may improve decoder performance. Although, as discussed above, a block-rate
rather = ,
than a frame-rate Transient Flag may form a portion of the sidechain
information with a
modest increase in bitrate, a similar result, albeit with decreased spatial
accuracy, maybe
accomplished without increasing the sidechain bitrate by detecting the
occurrence of
transients in the mono composite signal received in the decoder.
There is one transient flag per channel per frame, which, because it is
derived in
the time domain, necessarily applies to all subbands within that channel. The
transient
detection may be performed in the manner similar to that employed in an AC-3
encoder
for controlling the decision of when to switch between long and short length
audio
blocks, but with a higher sensitivity and with the Transient Flag True for any
frame in
- which the Transient Flag for a block is True (an AC-3 encoder detects
transients on a
block basis). In particular, see Section 8.2.2 of the above-cited A/52A
document. The
sensitivity of the transient detection described in Section 8.2.2 may be
increased by
adding a sensitivity factor F to an equation set forth therein. Section 8.2.2
of the A/52A
document is set forth below, with the sensitivity factor added (Section 8.2.2
as reproduced
=
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=
below is corrected to indicate that the low pass filter is a cascaded biquad
direct form II =
ER filter rather than "form I" as in the published A/52A document; Section
8.2.2 was
coned in the earlier A/52 document); Although it is not critical, a
sensitivity factor of
0.2 has been found to be a suitable value in dpractical embodiment of aspects
of the =-
present invention.
= Alternatively, a similar transient detection technique described in U.S.
Patent
5,394,473 may be employed. The '473 patent describes aspects of the A/52A
document ,
transient detector in greater detail. .
. .
As another alternative, transients may be detected in the frequency domain
rather
= than in the time domain (see the Comments to Step 408). In that case,
Step 401 may be
omitted and an alternative step emploYed in the frequency domain as deScribed
below.
Step 402. Window and DfT. =
= -= Multiply overlapping blocks of PCM time Samples by a time window and
convert
them to complex frequency values via a DFT as implemented b an.FFT.
= = Step 403. Convert Complex Values to Magnitude Ind Angle.
Convert each frequency-domain complex transfonn bin value (a +fb) to a
magnitude and angle representation using standard complex manipulations:
a. Magnitude = square root.(a2+ b2) =
. b. Angle = arctan (b/tt) ==
Comments regarding Step 403:
= Some of the following Steps use or may use, as an alternative, the energy
of a bin, ==
defined as the above magnitude squared (1.e, energy = (a2'+ b2).
Step 404. Calculate Subband Energy. ===
a. Calculate the subband energy per block-by adding bin energy values within
= = . each subband (a,summation across frequency).
b. Calculate the subband energy per frame by averaging or accumulating the
=energy in all the blocks in a frame (an averaging / accumulation across
time).
c. If the coupling frequency of the encoder is below about.1000 Hz, apply the
subband frame-averaged or frame-accumulated energy to a time smoother that
operates -
on all subbands below that frequency and-above the coupling frequency.
Comments regarding Step 404e:
=
= =
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Time smoothing to provide inter-frame smoothing hi low frequency subbands may
be useful. In order to avoid artifact-causing discontinuities between bin
values at subband
boundaries, it may be useful to apply a progressively-decreasing time
smoothing from the
lowest frequency subband encompassing and above the coupling frequency (where
the
smoothing may have a significant effect) up through a higher frequency subband
in which
the time smoothing effect is measurable, but inaudible, although nearly
audible. A
suitable time constant for the lowest frequency range subband (where the
subband is a =
= single bin if subbands are critical bands) may be in the range of 50 to
100 milliseconds,
= for example. Progressively-decreasing time smoothing may continue up
through. a
= 10 ,subband encompassing about 1000 Hz where the time constant may be
about 10
=
milliseconds, for example.
Although a first-order smoother is suitable, the smoother may be a two-stage
= smoother that has a variable time constant that short= its attack and
decay time in
response to a transient (such a two-stage smoother may be a digital equivalent
of the
analog two-stage smoothers describedin U.S. Patents 3,846,719 and 4,922,535).
In other words, the steady-state
= time constant may be scaled according to frequency and may also be
variable in response
to transients. Alternatively, such smoothing may be applied in Step 412.
- Step 405. Calculate Sum of Bin Magnitudes.
= a. Calculate the sum per block of the bin magnitudes (Step 403) of each
subband
(a summation' across' frequency).
= b. Calculate the sum per frame of the bin magnitudes of oath subband by
= averaging or accumulating the magnitudes of Stop 405a across.the blocks
in a frame (an
averaging / accumulation across time). These sums are used to calculate an
Interchannel
Angle Consistency Factor in Step 410 below.
c. If the coupling frequency of the encoder is below about 1000 Hz, apply the
subband frame-averaged or frame-accumulated magnitudes to a time smoother that
, operates on all subbands below that frequency and above the
coupling frequency. = =
Comments regarding Step 405c: See comments regarding step 404c eicept that
in,the case of Step 405c, the time sm.00thing may alternatively be performed
as part. of
= Step 410.
= Step 406. Calculate Make Interchannel Bin Phase Angle.
=
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Calculate the relative interchannel phase angle of each transform bin of each
block
by subtracting from the bin angle of Step 403 the corresponding bin angle of a
reference
channel (for example, the first channel). The result, as with other angle
additions or
subtractions herein, is taken modulo (n, -n) radians by adding or subtracting
2n until the
result is within the desired range of¨it to +7E.
Step 407. Calculate Interchannel Subband Phase Angle.
For each channel, calculate a frame-rate amplitude-weighted average
interchannel
phase angle for each subband as follows:
a. For each bin, construct a complex number from the magnitude of Step 403
and the relative interchannel bin phase angle of Step 406.
b. Add the constructed complex numbers of Step 407a across each subband (a
summation across frequency).
Comment regarding Step 407b: For example, if a subband has two bins and
one of the bins has a complex value of 1 + jl and the other bin has a complex
value of 2 +j2, their complex,sum is 3 +j3.
c. Average or accumulate the per block complex number sum for each
subband of Step 407b across the blocks of each frame (an averaging or
= accumulation across time).
d. If the coupling frequency'of the encoder is below about 1000 Hz, apply the
subband frame-averaged or frame-accumulated complex value to a time smoother
that operates on all subbands below that frequency and above the coupling
frequency.
Comments regarding Step 407d: See comments regarding Step 404c except
that in the case of Step 407d, the time smoothing may alternatively be
performed
as part of Steps 407e or 410.
e. Compute the magnitude of the complex result of Step 407d as per Step 403.
Comment regarding Step 407e: This magnitude is used in Step 410a below.
In the simple example given in Step 407b, the magnitude of 3 + j3 is square
root
(9 + = 4.24.
f. Compute the angle of the complex result as per Step 403.
Comments regarding Step 407f: In the simple example given in Step 407b,
the angle of 3 +j3 is arctan (3/3) = 45 degrees = n/4 radians. This subband
angle
=
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is signal-dependently time-smoothed (see Step 413) and quantized (see Step
414)
to generate the Subband Angle Control Parameter sidechain information, as
=
described below.
Step 408. Calculate Bin Spectral-Steadiness Factor
For each bin, calculate a Bin Spectral-Steadiness Factor in the range of 0 to
1 as
follows:
a. Let xm = bin magnitude of present block calculated in Step 403.
b. Let ym = corresponding bin magnitude of previous block.
C. If x. > ym, then Bin Dynamic Amplitude Factor = (ym/xm)2;
d. Else if ym > xm, then Bin Dynamic Amplitude Factor = (xm/ym)2,
e. Else if ym = xm, then Bin Spectral-Steadiness Factor = 1.
Comment regarding Step 408:
"Spectral steadiness" is a measure of the extent to which spectral components
(e.g., spectral coefficients or bin values) change over time. A Bin Spectral-
Steadiness
Factor of 1 indicates no change over a given time period.
Spectral Steadiness may also be taken as an indicator of whether a transient
is
present. A transient may cause a sudden rise and fall in spectral (bin)
amplitude over a
time period of one or more blocks, depending on its position with regard to
blocks and
their boundaries. Consequently, a change in the Bin Spectral-Steadiness Factor
from a
high value to a low value over a small number of blocks may be taken as an
indication of
the presence of a transient in the block or blocks having the lower value. A
further
confirmation of the presence of a transient, or an alternative to employing
the Bin
Spectral-Steadiness factor, is to observe the phase angles of bins within the
block (for
example, at the phase angle output of Step 403). Because a transient is likely
to occupy a
single temporal position within a block and have the dominant energy in the
block, the
existence and position of a transient may be indicated by a substantially
uniform delay in
phase from bin to bin in the block -- namely, a substantially linear ramp of
phase angles as
a function of frequency. Yet a further confirmation or alternative is to
observe the bin
amplitudes over a small number of blocks (for example, at the magnitude output
of Step
403), namely by looking directly for a sudden rise and fall of spectral level.
Alternatively, Step 408 may look at three consecutive blocks instead of one
block.
If the coupling frequency of the encoder is below about 1000 Hz, Step 408 may
look at
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more than three consecutive blocks. The number of consecutive blocks may taken
into
consideration vary with frequency such that the number gradually increases as
the
subband frequency range decreases. If the Bin Spectral-Steadiness Factor is
obtained
from more than one block, the detection of a transient, as just described, may
be
determined by separate steps that respond only to the number of blocks useful
for
detecting transients.
As a further alternative, bin energies may be used instead of bin magnitudes.
As yet a further alternative, Step 408 may employ an "event decision"
detecting
technique as described below in the comments following Step 409.
Step 409. Compute Subband Spectral-Steadiness Factor.
Compute a frame-rate Subband Spectral-Steadiness Factor on a scale of 0 to 1
by
forming an amplitude-weighted average of the Bin Spectral-Steadiness Factor
within each
subband across the blocks in a frame as follows:
a. For each bin, calculate the product of the Bin Spectral-Steadiness Factor
of Step
408 and the bin magnitude of Step 403.
b. Sum the products within each subband (a summation across frequency).
c. Average or accumulate the summation of Step 409b in all the blocks in a
frame
(-an averaging / accumulation across time).
d. If the coupling frequency of the encoder is below about 1000 Hz, apply the
subband frame-averaged or frame-accumulated summation to a time smoother that
operates on all subbands below that frequency and above the coupling
frequency.
= Conaments regarding Step 409d: See comments regarding Step 4040 except
that
in the case of Step 409d, there is no suitable subsequent step in which the
time
smoothing may alternatively be performed.
e. Divide the results of Step 409c or Step 409d, as appropriate, by the sum of
the
bin magnitudes (Step 403) within the subband.
Comment regarding Step 409e: The multiplication by the magnitude in Step
= 409a and the division* the sum of the magnitudes in Step 409e provide
amplitude
weighting. The output of Step 408 is independent of absolute amplitude and, if
not
amplitude weighted, may cause the output or Step 409 to be controlled by very
small
amplitudes, which is undesirable.
f. Scale the result to obtain the Subband Spectral-Steadiness Factor by
mapping
_ _
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the range from {0.5_1} to {0...1}. This may be done by multiplying the result
by 2,
subtracting 1, and limiting results less than 0 to a value of O.
Comment regarding Step 409f: Step 409f may be useful in assuring that a
channel of noise results in a Subband Spectral-Steadiness Factor of zero.
=
=Conunents regarding Steps 408 and 409: =
The goal of Steps 408 and 409 is to measure spectral steadiness ¨ changes in
=
= spectral composition over time in a subband of a channel. Alternatively,
aspects of an
"event decision" sensing such as described in International Publication NuMber
WO
02/097792 Al (designating the United States) may be employed to measure
spectral
= 10 steadiness instead of the approach just described in
connection with Steps 408 and 409.
U.S. Patent Application S.N. 10/478,538, filed November 20, 2003 is the United
States'
= national application of the published PCT Application WO 02/097792 Al.
According to these above-mentioned applications, the magnitudes of the =
=
complex FFT coefficient of each bin are calculated and normalized (largest
magnitude is
set to a value of one, for example). Then the magnitudes of corresponding bins
(in dB) in
consecutive blocks are subtracted (ignoring signs), the differences between
bins are
summed, and, if the sum exceeds a threshold, the block boundary is considered
to be an
auditory event boundary. Alternatively, changes in amplitude from block to
block may
also be considered along with spectral magnitude changes (by looking at the
amount of
= normalization required).
If aspects of the above-mentioned event-sensing applications are employed to
measure
spectral steadiness, normalization may not be required and the changes in
spectral
magnitude (changes in amplitude would not be measured if normalization is
omitted)
=
preferably are considered =on a subband basis. Instead of performing Step 408
as .
=
= indicated above, the decibel differences in spectral magnitude between
corresponding
bins in each subband may be_summed in accordance with the teachings of said
applications. Then, each of those sums, representing the degree of spectral
change from
block to block may be scaled so that the result is a spectral steadiness
factor having a
range from 0 to 1, wherein a value of 1 indicates the highest steadiness, a
change Of 0 dB
= from block to block for a given bin. A value of 0, indicating the lowest
steadiness, may
be assigned to decibel changes equal to or greater than a suitable amount,
such as 12 dB,
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for example. These results, a Bin Spectral-Steadiness Factor, may be used by
Step 409 in
the same manner that Step 409 uses the results of Step 408 as described above.
When
Step 409 receives a Bin Spectral-Steadiness Factor obtained by employing the
just-
described alternative event decision sensing technique, the Subband Spectral-
Steadiness
Factor of Step 409 may also be used as an indicator of a transient. For
example, if the
range of values produced by Step 409 is 0 to 1, a transient may be considered
to be
present when the Subband Spectral-Steadiness Factor is a small value, such as,
for
example, 0.1, indicating substantial spectral unsteadiness.
=
It will be appreciated that the Bin Spectral-Steadiness Factor produced
by Step =
= 10 408 and by the just-described alternative to Step 408 each inherently
Provide a variable
threshold to a certain degree in that they are based on relative changes from
block to
block. Optionally, it may be useful to supplement such inherency by
specifically
providing a shift in the threshold in response to, for example, multiple
transients in a
frame or a large transient among smaller transients (e.g., a loud transient
coming atop
mid- to low-level applause). In the ease of the latter example, an event
detector may
initially identify each clap as an event, but a loud transient (e.g., a drum
hit) may make it =
desirable to shift the threshold so that only the drum hit is identified as an
event =
Alternatively, a randomness metric may be employed (for example, as described
in U.S. Patent Re 36,714) instead of a measure of spectral-steadiness over
time.
= Step 410. Calculate Interchannel Angle Consistency Factor.
For each subband having more than one bin, calculate a frame-rate Interchannel
Angle Consistency Factor as follows:
= a. Divide the magnitude of the complex sum of Step 407e by the sum of the
= magnitudes of Step 405. The resulting "raw" Angle Consistency
Factor is a
number in the range of 0 to 1. =
= b. Calculate a correction factor: let n = the number of values across the
= subband contributing to the two quantities in the above step (in other
words, "n" is
the number of bins in the subband). If n is less than 2, let the Angle
Consistency
Factor be 1 and go to Steps 411 and 413.
= = c. Let r = Expected Random Variation = 1/n. Subtract r from the
result of the
Step 4101).
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- 35 -
d. Normalize the result of Step 410c by dividing by (1 r). The result has a
maximum value of 1. Limit the minimum value to 0 as necessary.
=
Commenta regarding Step 410:
Interchannel Angle Consistency is a measure of how similar the interchannel
phase angles are within a subband over a frame period. If all bin interchannel
angles of
the subband are the same, the Interchannel Angle Consistency Factor is 1.0;
whereas, if
the interchannel angles are randomly scattered, the value approaches zero.
The Subband Angle Consistency Factor indicates if there is a phantom image
between the channels. If the consistency is low, then it is desirable to
decorrelate the
channels. A high value indicates a fused image. Image fusion is independent of
other
signal characteristics.
It will be noted that the Subband Angle Consistency Factor, although an angle
parameter, is determined indirectly from two magnitudes. If the interchannel
angles are
all the same, adding the complex values and then taking the magnitude yields
the same
result as taking all the magnitudes and adding them, so the quotient is 1. If
the
interchannel angles are scattered, adding the complex values (such as adding
vectors
having different angles) results in at least partial cancellation, so the
magnitude of the
sum is less than the sum of the magnitudes, and the quotient is less than 1.
Following is a simple example of a subband having two bins:
Suppose that the two complex bin values are (3 + j4) and (6 + j8). (Same angle
each case: angle = arctan (imag/real), so anglel = arctan (4/3) and angle2 =
arctan (8/6) :=
arctan (4/3)). Adding complex values, sum = (9 +j12), magnitude of which is
square root (81+144) = 15.
The sum of the magnitudes is magnitude of (3 + j4)+magnitude of (6 +j8) = 5 +
10 = 15. The quotient is therefore 15/15 = 1 = consistency (before 1/n
normalization,
would also be 1 after normalization) (Normalized consistency = (1 - 0.5) / (1 -
0.5) = 1.0).
If one of the above bins has a different angle, say that the second one has
complex
value (6 ¨j 8), which has the same magnitude, 10. The complex sum is now (9 -
j4),
which has magnitude of square root (81 + 16) = 9.85, so the quotient is 9.85 /
15 = 0.66 =
consistency (before normalization). To normalize, subtract 1/n = 1/2, and
divide by (1-
1/n) (normalized consistency= (0.66 - 0.5) / (1 - 0.5) = 0.32.)
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Although the above-described technique for determining a Subband Angle
Consistency Factor has been found useful, its use is not critical. Other
suitable techniques
may be employed. For example, one could calculate a standard deviation of
angles using
standard formulae. In any case, it is desirable to employ amplitude weighting
to
minimize the effect of small signals on the calculated consistency value.
In addition, an alternative derivation of the Subband Angle Consistency Factor
may use energy (the squares of the magnitudes) instead of magnitude. This may
be
accomplished by squaring the magnitude from Step 403 before it is applied to
Steps 405
and 407.
Step 411. Derive Subband Decorrelation Scale Factor.
Derive a frame-rate DeCorrelation Scale Factor for each subband as follows:
a. Let x = frame-rate Spectral-Steadiness Factor of Step 409f.
b. Let y = frame-rate Angle Consistency Factor of Step 410e.
c. Then the frame-rate Subband Decorrelation Scale Factor = (1 ¨ x) * (1 ¨ y),
a number between 0 and 1.
Comments regarding Step 411:
The Subband Decorrelation Scale Factor is a function of the spectral-
steadiness of
signal characteristics over time in a subband of a channel (the Spectral-
Steadiness Factor)
and the consistency in the same subband of a channel of bin angles with
respect to
corresponding bins of a reference charmel (the Intercharmel Angle Consistency
Factor).
The Subband Decorrelation Scale Factor is high only if both the Spectral-
Steadiness
Factor and the Interchannel Angle Consistency Factor are low.
As explained above, the Decorrelation Scale Factor controls the degree of
envelope decorrelation provided in the decoder. Signals that exhibit spectral
steadiness
over time preferably should not be decorrelated by altering their envelopes,
regardless of
what is happening in other channels, as it mayresult in audible artifacts,
namely wavering
or warbling of the signal.
Step 412. Derive Subband Amplitude Scale Factors.
From the subband frame energy values of Step 404 and from the subband frame
= energy values of all other channels (as may be obtained by a step
corresponding to Step =
404 or an equivalent thereof), derive frame-rate Subband Amplitude Scale
Factors as
follows:
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a. For each subband, sum the energy values per frame across all input
channels.
b. Divide each subband energy value per frame, (from Step 404) by the sum of
the
energy values across all input channels (from Step 412a) to create values in
the range
of 0 to 1.
c. Convert each ratio to dB, in the range of ¨c to 0.
d. Divide by the scale factor granularity, which may be set at 1.5 dB, for
example,
change sign to yield a non-negative value, limit to a maximum value which may
be, for
example, 31 (i.e. 5-bit precision) and round to the nearest integer to create
the quantized
value. These values are the frame-rate Subband Amplitude Scale Factors and are
conveyed as part of the sidechain information.
e. If the coupling frequency of the encoder is below about 1000 Hz, apply the
subband frame-averaged or frame-accumulated magnitudes to a time smoother that
operates on all subbands below that frequency and above the coupling
frequency.
Comments regarding Step 412e: See comments regarding step 404e except that
in the case of Step 412e, there is no suitable subsequent step in which the
time smoothing
may alternatively be performed.
Comments for Step 412:
Although the granularity (resolution) and quantization precision indicated
here
have been found to be useful, they are not critical and other values may
provide
acceptable results.
Alternatively, one may use amplitude instead of energy to generate the Subband
Amplitude Scale Factors. If using amplitude, one would use dB=20*log(amplitude
ratio),
else if using energy, one converts to dB via dB=10*log(energy ratio), where
amplitude
ratio = square root (energy ratio).
Step 413. Signal-Dependently Thne Smooth Interehannel Subband Phase
Angles.
Apply signal-dependent temporal smoothing to subband frame-rate interchannel
angles derived in Step 407f:
a. Let v = Subband Spectral-Steadiness Factor of Step 409d.
b. Let w = corresponding Angle Consistency Factor of Step 410e.
c. Let x = (1 ¨v) * w. This is a value between 0 and 1, which is high if the
Spectral-Steadiness Factor is low and the Angle Consistency Factor is high.
=
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d. Let y = 1 ¨ x. y is high if Spectral-Steadiness Factor is high and Angle
Consistency Factor is low.
e. Let z = yeKP , where exp is a constant, which may be = 0.1. z is also in
the
range of 0 to 1, but skewed toward 1, corresponding to a slow time constant.
f. If the Transient Flag (Step 401) for the channel is set, set z = 0,
corresponding to a fast time constant in the presence of a transient.
g. Compute lim, a maximum allowable value of z, lim = 1 ¨ (0.1 * w). This
ranges from 0.9 if the Angle Consistency Factor is high to 1.0 if the Angle
Consistency Factor is low (0).
h. Limit z by lim as necessary: if (z > lim) then z = lim.
i. Smooth the subband angle of Step 407f using the value of z and a running
smoothed value of angle maintained for each subband. If A = angle of Step 407f
and RSA = running smoothed angle value as of the previous block, and NewRSA
is the new value of the running smoothed angle, then: NewRSA = RSA * z + A *
(1 ¨ z). The value of RSA is subsequently set equal to NewRSA before
processing the following block. New RSA is the signal-dependently time-
smoothed angle output of Step 413.
Comments regarding Step 413:
When a transient is detected, the subband angle update time constant is set to
0,
allowing a rapid subband angle change. This is desirable because it allows the
normal
angle update mechanism to use a range of relatively slow time constants,
minimizing
image wandering during itatic or quasi=istatic signals, yet fast-changing
signals are treated
with fast time constants.
Although other smoothing techniques and parameters may be usable, a first-
order
smoother implementing Step 413 has been found to be suitable. If implemented
as a first-
order smoother / lowpass filter, the variable "z" corresponds to the feed-
forward
coefficient (sometimes denoted "ff0"), while "(1-z)" corresponds to the
feedback
coefficient (sometimes denoted "fb1").
Step 414. Quantize Smoothed Interchannel Subband Phase Angles.
Quantize the time-smoothed subband interchatinel angles derived in Step 413i
to
obtain the Subband Angle Control Parameter:
a. If the value is less than 0, add 27c, so that all angle values to be
quantized are
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in the range 0 to 22t.
b. Divide by the angle granularity (resolution), which may be 27c / 64
radians,
and round to an integer. The maximum value may be set at 63, corresponding to
6-bit quantization.
Comments regarding Step 414:
The quantized value is treated as a non-negative integer, so an easy way to
quantize the angle is to map it to a non-negative floating point number ((add
2z if less
than 0, making the range 0 to (less than) a)), scale by the granularity
(resolution), and
round to an integer. Similarly, dequantizing that integer (which could
otherwise be done
with a simple table lookup), can be accomplished by scaling by the inverse of
the angle
granularity factor, converting a non-negative integer to a non-negative
floating point
angle (again, range 0 to 27t), after which it can be renormalized to the range
7C for further
use. Although such quantization of the Subband Angle Control Parameter has
been found
to be useful, such a quantization is not critical and other quantizations may
provide
acceptable results.
Step 415. Quantize Subband Decorrelation Scale Factors.
Quantize the Subband Decorrelation Scale Factors produced by Step 411 to, for
example, 8 levels (3 bits) by multiplying by 7.49 and rounding to the nearest
integer.
These quantized values are part of the sidechain information.
Comments regarding Step 415:
Although such quantization of the Subband Decorrelation Scale Factors has been
found to be useful, quantization using the example values is not critical and
other
quantizations may provide acceptable results.
Step 416. Dequantize Subband Angle Control Parameters.
Dequantize the Subband Angle Control Parameters (see Step 414), to use prior
to
downmixing. .
. = ,,
Comment regarding Step 416:
Use of quantized values in the encoder helps maintain synchrony between the
encoder and the decoder.
Step 417. Distribute Frame-Rate Dequantized Subband Angle Control
Parameters Across Blocks.
In preparation for downmhdng, distribute the once-per-frame dequantized
. =
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Subband Angle Control Parameters of Step 416 across time to the subbands of
each block
within the frame.
Comment regarding Step 417:
The same frame value may be assigned to each block in the frame.
Alternatively,
it may be useful to interpolate the Subband Angle Control Parameter values
across the
blocks in a frame. Linear interpolation over time may be employed in the
manner of the
linear interpolation across frequency, as described below.
Step 418. Interpolate block Subband Angle Control Parameters to Bins
Distribute the block Subband Angle Control Parameters of Step 417 for each
channel across frequency to bins, preferably using linear interpolation as
described below.
Com.ment regarding Step 418:
If linear interpolation across frequency is employed, Step 418 minimizes phase
angle changes from bin to bin across a subband boundary, thereby minimizing
aliasing
artifacts. Such linear interpolation may be enabled, for example, as described
below
following the description of Step 422. Subband angles are calculated
independently of
one another, each representing an average across a subband. Thus, there may be
a large
change from one subband to the next. If the net angle value for a subband is
applied to all
bins in the subband (a "rectangular" subband distribution), the entire phase
change from
one subband to a neighboring subband occurs between two bins. If there is a
strong
signal component there, there may be severe, possibly audible, aliasing.
Linear
interpolation, between the centers of each subband, for example, spreads the
phase angle
change over all the bins in the subband, minimizing the change between any
pair of bins,
so that, for example, the angle at the low end of a subband mates with the
angle at the
high end of the subband below it, while maintaining the overall average the
same as the
given calculated subband angle. In other words, instead of rectangular subband
distributions, the subband angle distribution may be trapezoidally shaped.
For example, suppose that the lowest coupled subband has one bin and a subband
angle of 20 degrees, the next subband has three bins and a subband angle of 40
degrees,
and the third subband has five bins and a subband angle of 100 degrees. With
no
interpolation, assume that the first bin (one subband) is shifted by an angle
of 20 degrees,
the next three bins (another subband) are shifted by an angle of 40 degrees
and the next
five bins (a further subband) are shifted by an angle of 100 degrees. In that
example,
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there is a 60-degree maximum change, from bin 4 to bin 5. With linear
interpolation, the
first bin still is shifted by 'an angle of 20 degrees, the next 3 bins are
shifted by about 30,
40, and 50 degrees; and the next five bins are shifted by about 67, 83, 100,
117, and 133
degrees. The average subband angle shift is the same, but the maximum bin-to-
bin
change is reduced to 17 degrees.
Optionally, changes in amplitude from subband to subband, in connection with
this and other steps described herein, such as Step 417 may also be treated in
a similar
interpolative fashion. However, it may not be necessary to do so because there
tends to
be more natural continuity in amplitude from one subband to the next.
Step 419. Apply Phase Angle Rotation to Bin Transform Values for Channel.
Apply phase angle rotation to each bin transform value as follows:
a. Let x = bin angle for this bin as calculated in Step 418.
b. Let y = -x;
c. Compute z, a unity-magnitude complex phase rotation scale factor with
angle y, z = cos (y) +j sin (y). =
d. Multiply the bin value (a + jb) by z.
Comments regarding Step 419:
The phase angle rotation applied in the encoder is the inverse of the angle
derived
from the Subband Angle Control Parameter.
= Phase angle adjustments, as described herein, in an encoder or encoding
process
prior to downmixing (Step 420) have several advantages: (1) they rninimi7e
cancellations
of the channels that are summed to a mono composite signal or matrixed to
multiple
channels, (2) they minimize reliance on energy normalization (Step 421), and
(3) they
prec,ompensate the decoder inverse phase angle rotation, thereby reducing
aliasing.
The phase correction factors can be applied in the encoder by subtracting each
subband phase correction value from the angles of each transform bin value in
that
subband. This is equivalent to multiplying each complex bin value by a complex
number
with a magnitude of 1.0 and an angle equal to the negative of the phase
correction factor.
Note that a complex number of magnitude 1, angle A is equal to cos(A)+j
sin(A). This
latter quantity is calculated once for each subband of each channel, with A = -
phase
correction for this subband, then multiplied by each bin complex signal value
to realize
the phase shifted bin value.
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The phase shift is circular, resulting in circular convolution (as mentioned
above).
While circular convolution may be benign for some continuous signals, it may
create
spurious spectral components for certain continuous complex signals (such as a
pitch
pipe) or may cause blurring of transients if different phase angles are used
for different
subbands. Consequently, a suitable technique to avoid circular convolution may
be
employed or the Transient Flag may be employed such that, for example, when
the
Transient Flag is True, the angle calculktion results may be overridden, and
all subbands
in a channel may use the same phase correction factor such as zero or a
randomized
value.
Step 420. Downmix.
Downmix to mono by adding the corresponding complex transform bins across
channels to produce a mono composite channel or downmix to multiple channels
by
matrixing the input channels, as for example, in the manner of the example of
FIG. 6, as
described below.
Comments regarding Step 420:
In the encoder, once the transform bins of all the channels have been phase
shifted, the channels are summed, bin-by-bin, to create the mono composite
audio signal.
Alternatively, the channels may be applied to a passive or active matrix that
provides
either a simple summation to one channel, as in the N:1 encoding of FIG. 1, or
to multiple
channels. The matrix coefficients.may be real or complex (real and imaginary).
Step 421. Normalize.
To avoid cancellation of isolated bins and over-emphasis of in-phase signals,
normalize the amplitude of each bin of the mono composite channel to have
substantially
the same energy as the sum of the contributing energies, as follows:
a. Let x = the sum across channels of bin energies (Le., the squares of the
bin
magnitudes computed in Step 403).
b. Let y = energy of corresponding bin of the mono composite channel,
calculated as per Step 403.
c. Let z = scale factor = square root (x/y). If x = 0 then y is 0 and z is set
to
1.
d. Limit z to a maximum value of, for example, 100. If z is initially greater
than 100 (implying strong cancellation from downmixing), add an arbitrary
value,
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for example, 0.01 * square root (x) to the real and imaginary parts of the
mono
composite bin, which will assure that it is large enough to be normalized by
the
following step.
e. Multiply the complex mono composite bin value by z.
Comments regarding Step 421:
Although it is generally desirable to use the same phase factors for both
encoding
and decoding, even the optimal choice of a subband phase correction value may
cause
one or more audible spectral components within the subband to be cancelled
during the
encode downmix process because the phase shifting of step 419 is performed on
a
subband rather than a bin basis. In this case, a different phase factor for
isolated bins in
the encoder may be used if it is detected that the sum energy of such bins is
much less
than the energy sum of the individual channel bins at that frequency. It is
generally not
necessary to apply such an isolated correction factor to the decoder, inasmuch
as isolated
bins usually have little effect on overall image quality. A similar
normalization may be
applied if multiple channels rather than a mono channel are employed.
Step 422. Assemble and Pack into Bitstream(s).
The Amplitude Scale Factors, Angle Control Parameters, Decorrelation Scale
Factors, and Transient Flags side channel information for each channel, along
with the
common mono composite audio or the matrixed multiple channels are multiplexed
as may
be desired and packed into one or more bitstreams suitable for the storage,
transmission
or storage and transmission medium or media.
Comment regarding Step 422:
The mono composite audio or the multiple channel audio may be applied to a
data-rate reducing encoding process or device such as, for example, a
percePtual encoder
or to a perceptual encoder and an entropy coder (e.g., arithmetic or Huffman
coder)
(sometimes referred to as a "lossless" coder) prior to packing. Also, as
mentioned above,
the mono composite audio (or the multiple channel audio) and related sidechain
information may be derived from multiple input channels only for audio
frequencies
above a certain frequency (a "coupling" frequency). In that case, the audio
frequencies
below the coupling frequency in each of the multiple input channels may be
stored,
transmitted or stored and transmitted as discrete channels or may be combined
or
processed in some manner other than as described herein. Discrete or otherwise-
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combined channels may also be applied to a data reducing encoding process or
device
such as, for example, a perceptual encoder or a perceptual encoder and an
entropy
encoder. The mono composite audio (or the multiple channel audio) and the
discrete
multichannel audio may all be applied to an integrated perceptual encoding or
perceptual
and entropy encoding process or device prior to packing.
Optional Interpolation Flag (Not shown in FIG. 4)
Interpolation across frequency of the basic phase angle shifts provided by the
Subband Angle Control Parameters may be enabled in the Encoder (Step 418)
and/or in
the Decoder (Step 505, below). The optional Interpolation Flag sidechain
parameter may
be employed for enabling interpolation in the Decoder. Either the
Interpolation Flag or
an enabling flag similar to the Interpolation Flag may be used in the Encoder.
Note that
because the Encoder has access to data at the bin level, it may use different
interpolation
values than the Decoder, which interpolates the Subband Angle Control
Parameters in the
sidechain information.
The use of such interpolation across frequency in the Encoder or the Decoder
may
be enabled if, for example, either of the following two conditions are true:
Condition 1. If a strong, isolated spectral peak is located at or near the
boundary of two subbands that have substantially different phase rotation
angle
assignments.
Reason: without interpolation, a large phase change at the boundary may
introduce a warble in the isolated spectral component. By using interpolation
to
spread the band-to-band phase change across the bin values within the band,
the
amount of change at the subband boundaries is reduced. Thresholds for spectral
peak strength, closeness to a boundary and difference in phase rotation from
subband to subband to satisfy this condition may be adjusted empirically.
Condition 2. If, depending on the presence of a transient, either the
intexchannel phase angles (no transient) or the absolute phase angles within a
channel (transient), comprise a good fit to a linear progression.
Reason: Using interpolation to reconstruct the data tends to provide a .
better fit to the original data. Note that the slope of the linear progression
need
not be constant across all frequencies, only within each subband, since angle
data -
will still be conveyed to the decoder on a subband basis; and that forms the
input
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to the Interpolator Step 418. The degree to which the data provides a good fit
to
satisfy this condition may also be determined empirically.
Other conditions, such as those determined empirically, may benefit from
interpolation across frequency. The existence of the two conditions just
mentioned may
be determined as follows:
Condition 1. If a strong, isolated spectral peak is located at or near the
boundary of two subbands that have substantially different phase rotation
angle
assignments:
for the Interpolation Flag to be used by the Decoder, the Subband Angle
Control Parameters (output of Step 414), and for enabling of Step 418 within
the
Encoder, the output of Step 413 before quantization may be used to determine
the
rotation angle from subband to subband.
for both the Interpolation Flag and for enabling within the Encoder, the
magnitude output of Step 403, the current DFT magnitudes, may be used to find
isolated peaks at subband boundaries.
Condition 2. If, depending on the presence of a transient, either the
interchannel phase angles (no transient) or the absolute phase angles within a
channel (transient), comprise a good fit to a linear progression.:
if the Transient Flag is not true (no transient), use the relative
interchannel
= bin phase angles from Step 406 for the fit to a linear progression
determination,
and
if the Transient Flag is true (transient), us the channel's absolute phase
angles from Step 403.
Decoding
The steps of a decoding process ("decoding steps") may be described as
follows.
With respect to decoding steps, reference is made to FIG. 5, which is in the
nature of a
hybrid flowchart and functional block diagram. For simplicity, the figure
shows the
derivation of sidechain information components for one channel, it being
understood that
sidechain information components must be obtained for each channel unless the
channel
is a reference channel for suCh components, as explained elsewhere.
Step 501. Unpack and Decode Sidechain Information.
Unpack and decode (including dequantization), as necessary, the sidechain data
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components (Amplitude Scale Factors, Angle Control Parameters, Decorrelation
Scale
Factors, and Transient Flag) for each frame of each channel (one channel shown
in FIG.
5). Table lookups may be used to decode the Amplitude Scale Factors, Angle
Control
Parameter, and Decorrelation Scale Factors.
Comment regarding Step 501: As explained above, if a reference channel is
employed, the sidechain data for the reference channel may not include the
Angle Control
Parameters, Decorrelation Scale Factors, and Transient Flag.
Step 502. Unpack and Decode Mono Composite or Multichannel Audio
Unpack and decode, as necessary, the mono composite or multichannel audio
signal information to provide DFT coefficients for each transform bin of the
mono
composite or multichannel audio signal.
Comment regarding Step 502:
Step 501 and Step 502 may be considered to be part of a single unpacking and
decoding step. Step 502 may include a passive or active mail-ix.
Step 503. Distribute Angle Parameter Values Across Blocks.
Block Subband Angle Control Parameter values are derived from the dequantized
frame Subband Angle Control Parameter values.
Comment regarding Step 503:
Step 503 may be implemented by distributing the same parameter value to every
block in the frame. =
Step 504. Distribute Subband Decorrelation Scale Factor Across Blocks.
Block Subband Decorrelation Scale Factor values are derived from the
dequantized frame Subband Decorrelation Scale Factor values.
Continent regarding Step 504;
Step 504 may be implemented by distributing the same scale factor value to
every
block in the frame.
Step 505. Linearly Interpolate Across Frequency.
Optionally, derive bin angles from the block subband angles of decoder Step
503
by linear interpolation across frequency as described above in connection with
encoder
Step 418. Linear interpolation in Step 505 may be enabled when the
Interpolation Flag is
used and is true.
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Step 506. Add Randomized Phase Angle Offset (Technique 3).
In accordance with Technique 3, described above, when the Transient Flag
indicates a transient, add to the block Subband Angle Control Parameter
provided by Step
503, which may have been linearly interpolated across frequency by Step 505, a
randomized offset value scaled by the Decorrelation Scale Factor (the scaling
may be
indirect as set forth in this Step):
a. Let y = block Subband Decorrelation Scale Factor.
b. Let z = yexP , where exp is a constant, for example = 5. z will also be in
the
range of 0 to 1, but skewed toward 0, reflecting a bias toward low levels of
randomized variation unless the Decorrelation Scale Factor value is high.
c. Let x = a randomi7ed number between +1.0 and 1.0, chosen separately for
each subband of each block.
d. Then, the value added to the block Subband Angle Control Parameter to add
a randomized angle offset value according to Technique 3 is x * pi * z.
Comments regarding Step 506:
As will be appreciated by those of ordinary skill in the art, "randomized"
angles
(or "randomized amplitudes if amplitudes are also scaled) for scaling by the
Decorrelation
Scale Factor may include not only pseudo-random and truly random variations,
but also
deterministically-generated variations that, when applied to phase angles or
to phase
angles and to amplitudes, have the effect of reducing cross-correlation
between channels.
Such "randomized" variations may be obtained in many ways. For example, a
pseudo-
random number generator with various seed values may be employed.
Alternatively,
truly random numbers may be generated using a hardware random number
generator.
Inasmuch as a randomized angle resolution of only about 1 degree may be
sufficient,
tables of randomized numbers having two or three decimal places (e.g. 0.84 or
0.844)
may be employed. Preferably, the randomized values (between ¨1.0 and +1.0 with
reference to Step 505c, above) are uniformly distributed statistically across
each channel.
Although the non-linear indirect scaling of Step 506 has been found to be
useful,
it is not critical and other suitable scalings may be employed ¨ in particular
other values
for the exponent may be employed to obtain similar results.
When the Subband Decorrelation Scale Factor value is 1, a full range of random
angles from -7t to + It are added (in which case the block Subband Angle
Control
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Parameter values produced by Step 503 are rendered irrelevant). As the Subband
Decorrelation Scale Factor value decreases toward zero, the randomized angle
offset also
decreases toward zero, causing the output of Step 506 to move toward the
Subband Angle
Control Parameter values produced by Step 503.
If desired, the encoder described above may also add a scaled randomized
offset
in accordance with Technique 3 to the angle shift applied to a channel before
downmildng. Doing so may improve alias cancellation in the decoder. It may
also be
beneficial for improving the synchronicity of the encoder and decoder.
Step 507. Add Randomized Phase Angle Offset (Technique 2).
In accordance with Technique 2, described above, when the Transient Flag does
not indicate a transient, for each bin, add to all the block Subband Angle
Control
Parameters in a frame provided by Step 503 (Step 505 operates only when the
Transient
Flag indicates a transient) a different randomized offset value scaled by the
Decorrelation
Scale Factor (the scaling may be direct as set forth herein in this step):
a. Let y = block Subband Decorrelation Scale Factor.
b. Let x = a randomized number between +1.0 and ¨1.0, chosen separately for
each bin of each frame.
c. Then, the value added to the block bin Angle Control Parameter to add a
randomized angle offset value according to Technique 3 is x * pi * y.
Comments regarding Step 507:
See comments above regarding Step 505 regarding the randomized angle offset.
Although the direct scaling of Step 507 has been found to be useful, it is not
critical and other suitable scalings may be employed.
To minimize temporal discontinuities, the unique randomized angle value for
each
bin of each channel preferably does not change with time. The randomized angle
values
of all the bins in a subband are scaled by the same Subband Decorrelation
Scale Factor
value, which is updated at the frame rate. Thus, when the Subband
Decorrelation Scale
Factor value is 1, a full range of random angles from -7r to +7r are added (in
which case
block subband angle values derived from the dequantized frame subband angle
values are
rendered irrelevant). As the Subband Decorrelation Scale Factor value
diminishes toward
zero, the randomized angle offset also diminishes toward zero. Unlike Step
504, the
scaling in this Step 507 may be a direct function of the Subband Decorrelation
Scale
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Factor value. For example, a Subband Decorrelation Scale Factor value of 0.5
proportionally reduces every random angle variation by 0.5.
The scaled randomized angle value may then be added to the bin angle from
decoder Step 506. The Decorrelation Scale Factor value is updated once per
frame. In
the presence of a Transient Flag for the. frame, this step is skipped, to
avoid transient
prenoise artifacts.
If desired, the encoder described above may also add a scaled randomized
offset
in accordance with Technique 2 to the angle shift applied before downmixing.
Doing so
may improve alias cancellation in the decoder. It may also be beneficial for
improving
the synchronicity of the encoder and decoder.
Step 508. Normalize Amplitude Scale Factors.
Normalize Amplitude Scale Factors across channels so that they sum-square to
1.
Comment regarding Step 508:
For example, if two channels have dequantized scale factors of -3.0 dB (= 2 *
granularity of 1.5 dB) (.70795), the sum of the squares is 1.002. Dividing
each by the
square root of 1.002 = 1.001 yields two values of .7072 (-3.01 dB).
Step 509. Boost Subband Scale Factor Levels (Optional).
Optionally, when the Transient Flag indicates no transient, apply a slight
additional boost to Subband Scale Factor levels, dependent on Subband
Decorrelation
Scale Factor levels: multiply each normalized Subband Amplitude Scale Factor
by a
small factor (e.g., 1 + 0.2 * Subband Decorrelation Seale Factor). When the
Transient
Flag is True, skip this step.
Comment regarding Step 509:
This step may be useful because the decoder decorrelation Step 507 may result
in
slightly reduced levels in the final inverse filterbank process.
Step 510. Distribute Subband Amplitude Values Across Bins.
Step 510 may be implemented by distributing the same subband amplitude scale
factor value to every bin in the subband.
Step 510a. Add Randomized Amplitude Offset (Optional)
Optionally, apply a randomized variation to the normalized Subband Amplitude
Scale Factor dependent on Subband Decorrelation Scale Factor levels and the
Transient
Flag. In the absence of a transient, add a Randomized Amplitude Scale Factor
that does
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not change with time on a bin-by-bin basis (different from bin to bin), and,
in the
presence of a transient (in the frame or block), add a Randomized Amplitude
Scale Factor
that changes on a block-by-block basis (different from block to block) and
changes from
subband to subband (the same shift for all bins in a subband; different from
subband to
subband). Step 510a is not shown in the drawings.
Comment regarding Step 510a:
Although the degree to which randomized amplitude shifts are added may be
controlled by the Decorrelation Scale Factor, it is believed that a particular
scale factor
value should cause less amplitude shift than the corresponding randomized
phpse shift
resulting from the same sale factor value in order to avoid audible artifacts.
Step 511. Upmix.
a. For each bin of each output channel, construct a complex upmix scale
factor from the amplitude of decoder Step 508 and the bin angle of decoder
Step 507: (amplitude * (cos (angle) +j sin (angle)).
b. For each output channel, multiply the complex bin value and the
complex upmix scale factor to produce the upmixed complex output bin value of
each bin of the channel.
Step 512. Perform Inverse DFT (Optional).
Optionally, perform an inverse DFT transform on the bins of each output
channel
to yield multichannel output PCM values. As is well known, in connection with
such an
inverse DFT transformation, the individual blocks of time samples are
windowed, and
adjacent blocks are overlapped and added together in order to reconstruct the
final
continuous time output PCM audio signal.
Conunents regarding Step 512:
A decoder according to the present invention may not provide PCM outputs. In
the case where the decoder process is employed only above a given coupling
frequency,
and discrete MDCT coefficients are sent for each channel below that frequency,
it may be
desirable to convert the DFT coefficients derived by the decoder upmixing
Steps 511a
and 511b to MDCT coefficients, so that they can be combined with the lower
frequency
discrete MDCT coefficients and requantized in order to provide, for example, a
bitsteam
compatible with an encoding system that has a large number of installed users,
such as a
standard AC-3 SP/DIF bitstream for application to an external device where an
inverse
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transform may be performed. Aninverse DFT transform may be applied to ones of
the
output channels to provide PCM outputs.
Section 8.2.2 of theA/52A Document
With Sensitivity Factor "F" Added
8.2.2. Transient detection
Transients are detected in the full-bandwidth channels in order to decide when
to
switch to short length audio blocks to improve pre-echo performance. High-pass
filtered
versions of the signals are examined for an increase in energy from one sub-
block time-
segment to the next. Sub-blocks are examined at different time scales. If a
transient is
detected in the second half of an audio block in a channel that channel
switches to a short
block. A channel that is block-switched uses the D45 exponent strategy [i.e.,
the data has
a coarser frequency resolution in order to reduce the data overhead resulting
from the
increase in temporal resolution].
The transient detector is used to determine when to switch from a long
transform
block (length 512), to the short block (length 256). It operates on 512
samples for every
audio block. This is done in two passes, with each pass processing 256
'samples. Transient
detection is broken down into four steps: 1) high-pass filtering, 2)
segmentation of the
block into submultiples, 3) peak amplitude detection within each sub-block
segment, and
4) threshold comparison. The transient detector outputs a flag blksw[n] for
each full-
bandwidth channel, which when set to "one" indicates the presence of a
transient in the
second half of the 512 length input block for the corresponding channel.
1) High-pass filtering: The high-pass filter is implemented as a cascaded
biquad direct form II IIR filter with a cutoff of 8 kHz.
2) Block Segmentation: The block of 256 high-pass filtered samples are
segmented into a hierarchical tree of levels in which level 1 represents the
256
length block, level 2 is two segments of length 128, and level 3 is four
segments
of length 64.
3) Peak Detection: The sample with the largest magnitude is identified foi.
each segment on every level of the hierarchical tree. The peaks for a single
level
are found as follows:
P[j][k] = max(x(n))
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and k = 1, ..., 21(j-1) ;
. where: x(n) = the nth sample in the 256 length block
j = 1, 2, 3 is the hierarchical level number
k = the segment number within level j
Note that P[j][0], (i.e., k---0) is defined to be the peak of the last
segment on level j of the tree calculated immediately prior to the current
tree. For example, P[3][4] in the preceding tree is P[3][0] in the current
tree.
4) Threshold Comparison: The first stage of the threshold comparator
checks to see if there is significant signal level in the current block. This
is done
by comparing the overall peak value P[1][1] of the current block to a "silence
threshold". If P[1][1] is below this threshold then a long block is forced.
The silence
threshold value is 100/32768. The next stage of the comparator checks the
relative
peak levels of adjacent segments on each level of the hierarchical tree. If
the peak
ratio of any two adjacent segments on a particular level exceeds a pre-defined
threshold for that level, then a flag is set to indicate the presence of a
transient in
the current 256-length block. The ratios are compared as follows:
mag(P[j][k]) x T[j] > (F * mag(P[j][(k-1)])) [Note the "F" sensitivity
factor]
where: T[j] is the pre-defined threshold for level j, defined as:
T[1] = .1
T[2] = .075
T[3] = .05
If this inequality is true for any two segment peaks on any level,
then a transient is indicated for the first half of the 512 length input
block.
The second pass through this process determines the presence of transients
in the second half of the 512 length input block.
Nall Encoding
Aspects of the present invention are not limited to N:1 encoding as described
in
connection with FIG. 1. More generally, aspects of the invention are
applicable to the
transformation of any number of input channels (n input channels) to any
number of
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output channels (m output channels) in the manner of FIG. 6 (i.e., N:M
encoding).
Because in many common applications the number of input channels n is greater
than the
n-umber of output channels m, the N:M encoding arrangement of FIG. 6 will be
referred.
=
to as "downmixing" for convenience in description.
Referring to the details of FIG. 6, instead of summing the outputs of Rotate
Angle
8 and Rotate Angle 10 in the Additive Combiner 6 as in the arrangement of FIG.
1, those
outputs may be applied to a downmix matrix device or function 6' ("Downmix
Matrix").
Downmix Matrix 6' may be a passive or active matrix that provides either a
simple
summation to one channel, as in the N:1 encoding of FIG. 1, or to multiple
e.hannels. The
matrix coefficients may be real or complex (real and imaginary). Other devices
and
functions in FIG. 6 may be the same as in the FIG. 1 arrangement and they bear
the same
reference numerals.
Downmix Matrix 6' may provide a hybrid frequency-dependent function such that
it provides, for example, mn_f2 channels in a frequency range fl to f2 and
mn_n channels
in a frequency range 12 to f3. For example, below a coupling frequency of, for
example,
1000 Hz the Downmix Matrix 6' may provide two channels and above the coupling
frequency the Downmix Matrix 6' may provide one channel. By employing two
channels
below the coupling frequency, better spatial fidelity may be obtained,
especially if the
two channels represent horizontal directions (to match the horizontality of
the human
ears).
Although FIG. 6 shows the generation of the same sidechain information for
each
channel as in the FIG. 1 arrangement, it may be possible to omit certain ones
of the
sidechain information when more than one channel is provided by the output of
the
Downmix Matrix 6'. In some cases, adceptable results may be obtained when only
the
amplitude scale factor sidechain information is provided by the FIG. 6
arrangement.
Further details regarding sidechain options are discussed below in connection
with the
descriptions of FIGS. 7, 8 and 9.
As just mentioned above, the multiple channels generated by the Downmix Matrix
6' need not be fewer than the number of input channels n. When the purpose of
an
encoder such as in FIG. 6 is to reduce the number of bits for transmission or
storage, it is
likely that the number of channels produced by downmix matrix 6' will be fewer
than the
number of input channels n. However, the arrangement of FIG. 6 may also be
used as an
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"upmixer." In that case, there may be applications in which the number of
channels m
produced by the Dowmnix Matrix 6' is more than the number of input channels n.
Encoders as described in connection with the examples of FIGS. 2, 5 and 6 may
also include their own local decoder or decoding function in order to
determine if the
audio information and the sidechain information, when decoded by such a
decoder, would
provide suitable results. The results of such a determination could be used to
improve the
parameters by employing, for example, a recursive process. In a block encoding
and
decoding system, recursion calculations could be performed for example, on
every block
before the next block ends in order to minimize the delay in transmitting a
block of audio
information and its associated spatial parameters.
An arrangement in which the encoder also includes its own decoder or decoding
function could also be employed advantageously when spatial parameters are not
stored
or sent only for certain blocks. If unsuitable decoding would result from not
sending
spatial-parameter sidechain information, such sidechain information would be
sent for the
particular block. In this case, the decoder may be a modification of the
decoder or
decoding function of FIGS. 2, 5 or 6 in that the decoder would have both the
ability to
recover spatial-parameter sidechain information for frequencies above the
coupling
frequency from the incoming bitstream but also to generate simulated spatial-
parameter
sidechain information from the stereo information below the coupling
frequency.
In a simplified alternative to such local-decoder-incorporating encoder
examples,
rather than having a local decoder or decoder function, the encoder could
simply check to
determine if there were any signal content below the coupling frequency
(determined in
, any suitable way, for example, a sum of the energy in frequency bins through
the
frequency range), and, if not, it would send or store spatial-parameter
sidechain
information rather than not doing so if the energy were above the threshold.
Depending
on the encoding scheme, low signal information below the coupling frequency
may also
result in more bits being available for sending sidechain information.
= % M:N Decoding
A more generalized form of the arrangement of FIG. 2 is shown in FIG. 7,
wherein an upmix matrix function or device ("Upmix Matrix") 20 receives the 1
to m
channels generated by the arrangement of FIG. 6. The Upmix Matrix 20 may be a
passive matrix. It may be, but need not be, the conjugate transposition (i.e.,
the
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=
=
complement) of the Downmix Matrix 6' of the FIG. 6 arrangement Alternatively,
the
= = Upmix Matrix 20 may bean active matrix ¨ a variable matrix or a passive
matrix in
= combination with a variable matrix. If an active matrix decoder is
employed, in its =
. relaxed or quiescent state it may be the complex conjugate of the Downmix
Matrix or it
may be independent of the Downmix Matrix. The sidechain information may be
applied
a shown in FIG. 7 so as to control the Adjust Amplitude, Rotate Angle, and
(optional)
Interpolator functions or devices. In that case, the Upmix Matrix; if an
active matrix,
operates independently of the sidechain information and responds only to the
channels
applied to it. Alternatively, some or all of the sidechain information may be
applied to
the active matrix to assist its operation. In that case, some or all of the
Adjust Amplitude,
= Rotate Angle, and Interpolator functions or devices may be omitted. The
Decoder
example of FIG. 7 may also employ the alternative of applying a degree of
randomized
amplitude variations under certain signal conditions, as described above in
connection =
with FIGS. 2 and 5.
When Upmix Matrix 20 is an active matrix, theiarrangement of FIG. 7 may be
characterized as a "hybrid matrix decoder" for operating in a "hybrid matrix
encoder/decoder system." "Hybrid" in this context refers to the fact that the
decoder may
derive some measure of control information from its input audio signal (i.e. ,
the active
matrix responds to spatial information encoded in the channels applied to it)
and a further
= 20 measure of control information from spatial-parameter sidechain
information. Other
elements of FIG. 7 are as in the arrangement of FIG. 2 and bear the same
reference
numerals.
=
Suitable active matrix decoders for use in a hybrid matrix decoder may
Include =
active matrix decoders such as those mentioned above, =
=
including, for example, matrix decoders known as "Pro Logic" and "Pro Logic
II"
decoders ("Pro Logic" is a trademark of Dolby Laboratories Licensing
Corporation).
= Alternative Decorrelation
FIGS. 8 and 9 show variations on the generalized Decoder of FIG. 7. In =-
particular, both the arrangement of FIG. 8 and the arrangement of FIG. 9 show
alternatives to the decorrelation technique of FIGS. 2 and 7. In FIG. 8,
respective
decorrelator functions or devices ("Decorrelators") 46 and 48 are in the time
domain,
each following the respective Inverse Filterbank 30 and 36 in their channel.
In FIG. 9,
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respective decorrelator functions or devices ("Decorrelators") 50 and 52 are
in the
frequency domain, each preceding the respective Inverse Filterbank 30 and 36
in their
channel. In both the FIG. 8 and FIG. 9 arrangements, each of the Decorrelators
(46, 48,
50, 52) has a unique characteristic so that their outputs are mutually
decorrelated with
respect to each other. The Decorrelation Scale Factor may be used to control,
for
example, the ratio of decorrelated to Icorrelated signal provided in each
channel.
Optionally, the Transient Flag may also be used to shift the mode of operation
of the
Decorrelator, as is explained below. In both the FIG. 8 and FIG. 9
arrangements, each
Decorrelator may be a Schroeder-type reverberator having its own unique filter
characteristic, in which the amount or degree of reverberation is controlled
by the
decorrelation scale factor (implemented, for example, by controlling the
degree to which
the Decorrelator output forms a part of a linear combination of the
Decorrelator input and
output). Alternatively, other controllable decorrelation techniques may be
employed
either alone or in combination with each other or with a Schroeder-type
reverberator.
Schroeder-type reverherators are well known and may trace their origin to two
journal
papers: "'Colorless' Artificial Reverberation" by M.R. Schroeder and B.F.
Logan, IRE
Transactions on Audio, vol. AU-9, pp. 209-214, 1961 and "Natural Sounding
Artificial =
Reverberation" by M.R. Schroeder, Jounial A.E.S., July 1962, vol. 10, no. 2,
pp. 219-223.
When the Decorrelators 46 and 48 operate in the time domain, as in the FIG. 8
arrangement, a single (i.e., wideband) Decorrelation Scale Factor is required.
This may
be obtsined by any of several ways. For example, only a single Decorrelation
Scale
Factor may be generated in the encoder of FIG. 1 or FIG. 7. Alternatively, if
the encoder
of FIG. 1 or FIG. 7 generates Decorrelation Scale Factors on a subband basis,
the
Subband Decorrelation Scale Factors may be amplitude or power summed in the
encoder
. 25 of FIG. 1 or FIG. 7 or in the
decoder of FIG. 8. =.
When the =Decorrelators 50 and 52 operate in the frequency domain, as in the
FIG.
9 arrangement, they may receive a decorrelation scale factor for each subband
or groups =
of subbands and, concomitantly, provide a commensurate degree of decorrelation
for such
subbands or groups of subbands.
The Decorrelators 46 and 48 of FIG. 8 and the Decorrelators 50 and 52 of FIG.
9
may optionally receive the Transient Flag. In the time-domain Decorrelators of
FIG. 8,
the Transient Flag may be employed-to shift the mode of operation of the
respective
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Decorrelator. For example, the Decorrelator may operate as a Schroeder-type
reverberator in the absence of the transient flag but upon its receipt and for
a short
subsequent time period, say 1 to 10 milliseconds, operate as a fixed delay.
Each channel
may have a predetermined fixed delay or the delay may be varied in response to
a
plurality of transients within a short time period. In the frequency-domain
Decorrelators
of FIG. 9, the transient flag may also be employed to shift the mode of
operation of the
respective Decorrelator. However, in this case, the receipt of a transient
flag may, for
example, trigger a short (several milliseconds) increase in amplitude in the
channel in
which the flag occurred.
In both the FIG. 8 and 9 arrangements, an Interpolator 27 (33), controlled by
the
optional Transient Flag, may provide interpolation across frequency of the
phase angles
= output of Rotate Angle 28 (33) in a manner as described above.
As mentioned above, when two or more chatmels are sent in addition to
sidechain
information, it may be acceptable to reduce the number of sidechain
parameters. For
example, it may be acceptable to send only the Amplitude Scale Factor, in
which case the
decorrelation and angle devices or functions in the decoder may be omitted (in
that case,
FIGS. 7, 8 and 9 reduce to the same arrangement).
Alternatively, only the amplitude scale factor, the Decorrelation Scale
Factor, and,
optionally, the Transient Flag may be sent. In that case, any of the FIG. 7, 8
or 9
arrangements may be employed (omitting the Rotate Angle 28 and 34 in each of
them).
As another alternative, only the amplitude scale factor and the angle control
=
parameter may be sent. In that case, any of the FIG. 7, 8 or 9 arrangements
may be
employed (omitting the Decorrelator 38 and 42 of FIG. 7 and 46, 48, 50, 52 of
FIGS. 8
and 9).
As in FIGS. 1 and 2, the arrangements of FIGS. 6-9 are intended to show any
number of input and output channels although, for simplicity in presentation,
only two
channels are shown.
It should be understood that implementation of other variations and
modifications
of the invention and its various aspects will be apparent to those skilled in
the art, and that
the invention is not limited by these specific embodiments described. It is
therefore
contemplated to cover by the present invention any and all modifications,
variations, or
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equivalents that fall within the true scope of the basic underlying principles
disclosed herein.
=
