Note: Descriptions are shown in the official language in which they were submitted.
WO 2015/004373 1
PCT/FR2014/051720
Optimized scale factor for frequency band extension in an audio frequency
signal
decoder
The present invention relates to the field of the coding/decoding and the
processing
of audio frequency signals (such as speech, music or other such signals) for
their
transmission or their storage.
More particularly, the invention relates to a method and a device for
determining an
optimized scale factor that can be used to adjust the level of an excitation
signal or, in an
equivalent manner, of a filter as part of a frequency band extension in a
decoder or a
3.0 processor enhancing an audio frequency signal.
Numerous techniques exist for compressing (with loss) an audio frequency
signal
such as speech or music.
The conventional coding methods for the conversational applications are
generally
classified as waveform coding (PCM for "Pulse Code Modulation", ADCPM for
"Adaptive
25 Differential Pulse Code Modulation", transform coding, etc.), parametric
coding (LPC for
"Linear Predictive Coding", sinusoidal coding, etc.) and parametric hybrid
coding with a
quantization of the parameters by "analysis by synthesis" of which CELP ("Code
Excited
Linear Prediction") coding is the best known example.
For the non-conversational applications, the prior art for (mono) audio signal
coding
20 consists of perceptual coding by transform or in subbands, with a
parametric coding of the
high frequencies by band replication.
A review of the conventional speech and audio coding methods can be found in
the works by
W.B. Kleijn and K.K. Paliwal (eds.), Speech Coding and Synthesis, Elsevier,
1995; M. Bosi,
R.E. Goldberg, Introduction to Digital Audio Coding and Standards, Springer
2002; J. Benesty,
25 M.M. Sondhi, Y. Huang (Eds.), Handbook of Speech Processing, Springer
2008.
The focus here is more particularly on the 3GPP standardized AMR-WB ("Adaptive
Multi-Rate Wideband") codec (coder and decoder), which operates at an
input/output
frequency of 16 kHz and in which the signal is divided into two subbands, the
low band (0-
30 6.4 kHz) which is sampled at 12.8 kHz and coded by CELP model and the
high band (6.4-
7 kHz) which is reconstructed parametrically by "band extension" (or BWE, for
"Bandwidth
Extension") with or without additional information depending on the mode of
the current
frame. It can be noted here that the limitation of the coded band of the AMR-
WB codec at
7 kHz is essentially linked to the fact that the frequency response in
transmission of the
35 wideband terminals was approximated at the time of standardization
(ETSI/3GPP then ITU-T)
according to the frequency mask defined in the standard ITU-T P.341 and more
specifically
by using a so-called "P341" filter defined in the standard ITU-T G.191 which
cuts the
frequencies above 7 kHz (this filter observes the mask defined in P.341).
However, in theory,
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it is well known that a signal sampled at 16 kHz can have a defined audio band
from 0 to
8000 Hz; the AMR-WB codec therefore introduces a limitation of the high band
by comparison
with the theoretical bandwidth of 8 kHz.
The 3GPP AMR-WB speech codec was standardized in 2001 mainly for the circuit
mode (CS) telephony applications on GSM (2G) and UMTS (3G). This same codec
was also
standardized in 2003 by the ITU-T in the form of recommendation G.722.2
"Wideband coding
speech at around 16 kbit/s using Adaptive Multi-Rate Wideband (AMR-WB)".
It comprises nine bit rates, called modes, from 6.6 to 23.85 kbit/s, and
comprises
continuous transmission mechanisms (DTX, for "Discontinuous Transmission")
with voice
activity detection (VAD) and comfort noise generation (CNG) from silence
description frames
(SID, for "Silence Insertion Descriptor"), and lost frame correction
mechanisms (FEC for
"Frame Erasure Concealment", sometimes called PLC, for "Packet Loss
Concealment).
The details of the AMR-WB coding and decoding algorithm are not repeated here;
a
detailed description of this codec can be found in the 3GPP specifications (TS
26.190, 26.191,
25 26.192, 26.193, 26.194, 26.204) and in ITU-T-G.722.2 (and the
corresponding annexes and
appendix) and in the article by B. Bessette et al. entitled 'The adaptive
multirate wideband
speech codec (AMR-WB)", IEEE Transactions on Speech and Audio Processing, vol.
10, no. 8,
2002, pp. 620-636 and the source code of the associated 3GPP and ITU-T
standards.
The principle of band extension in the AMR-WB codec is fairly rudimentary.
Indeed,
the high band (6.4-7 kHz) is generated by shaping a white noise through a time
(applied in
the form of gains per subframe) and frequency (by the application of a linear
prediction
synthesis filter or LPC, for "Linear Predictive Coding') envelope. This band
extension
technique is illustrated in figure 1.
A white noise u mn(n) , n = 0,===,79 is generated at 16 kHz for each 5 ms
subframe
by linear congruential generator (block 100). This noise u im(n) is formatted
in time by
application of gains for each subframe; this operation is broken down into two
processing
steps (blocks 102, 106 or 109):
= A first factor is computed (block 101) to set the white noise umi (n)
(block 102) at a
level similar to that of the excitation, u(n) , n = 0,= = =,63, decoded at
12.8 kHz in
the low band:
63 ____________________________________
Eu(02
U HB2(n) = U HBI (n)i 7,14- __________
EUHBI(1)2
/A
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It can be noted here that the normalization of the energies is done by
comparing
blocks of different size (64 for u(n) and 80 for um,i(n)) without compensation
of the
differences in sampling frequencies (12.8 or 16 kHz).
= The excitation in the high band is then obtained from block 106 or block
109
selectively by switch 110 in the form:
14uB(n)¨ g HBit HB2(n)
in which the gain g8 is obtained differently depending on the bit rate. If the
bit rate
of the current frame is < 23.85 kbit/s, the gain R is
estimated "blind" (that is to
say without additional information); in this case, the block 103 filters the
signal
decoded in low band by a high-pass filter having a cut-off frequency at 400 Hz
to
obtain a signal (n) , n = -,63 -
this high-pass filter eliminates the influence of
the very low frequencies which can skew the estimation made in the block 104 ¨
then
the 'tilt" (indicator of spectral slope) denoted effi, of the signal g hp (n)
is computed
by normalized self-correlation (block 104):
63
(n).ihp (n-1)
edit = __________ n I 63
1.i.40 (02
n=0
and finally, km, is computed in the form:
km, =wõg,+ (1¨ vt,õ)g),G
in which gsp =1¨etth is the gain applied in the active speech (SP) frames,
8BG ¨1.25g, is the gain applied in the inactive speech frames associated with
a
background (BC) noise and ws, is a weighting function which depends on the
voice
activity detection (VAD). It is understood that the estimation of the tilt
(e,1,) makes it
possible to adapt the level of the high band as a function of the spectral
nature of the
signal; this estimation is particularly important when the spectral slope of
the CELP
decoded signal is such that the average energy decreases when the frequency
increases (case of a voiced signal where era is close to 1, therefore gõ =1¨
enõ is
thus reduced). It should also be noted that the factor ,kõ in the AMR-WB
decoding
is bounded to take values within the range [0.1, 1.0]. Indeed, for the signals
whose
energy increases when the frequency increases (e1, close to -1, gsp close to
2), the
gain k'fi, is usually underestimated.
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At 23.85 kbit/s, a correction information item is transmitted by the AMR-WB
coder
and decoded (blocks 107, 108) in order to refine the gain estimated for each
subframe (4 bits
every 5 ms, or 0.8 kbit/s). The
artificial excitation umB(n) is then filtered (block 111) by
an LPC synthesis filter (block 111) of transfer function 1/ Am(z) and
operating at the
sampling frequency of 16 kHz. The construction of this filter depends on the
bit rate of the
current frame:
= At 6.6 kbit/s, the filter 1/ Am, (z) is obtained by weighting by a factor
y= 0.9 an
LPC filter of order 20, 1/ A" (z) , which "extrapolates" the LPC filter of
order 16,
/ A(z) , decoded in the low band (at 12.8 kHz) ¨ the details of the
extrapolation in
the realm of the ISF (Imittance Spectral Frequency) parameters are described
in the
standard G.722.2 in section 6.3.2.1; in this case,
1/Am(z)=1/A"(z/r)
= at the bit rates > 6.6 kbit/s, the filter 1 / AHB(z) is of order 16 and
corresponds
simply to:
1/Aim(z) =1/ A(z/ y)
in which 7=0.6. It should be noted that, in this case, the filter 1/ A(zir) is
used
at 16 kHz, which results in a spreading (by proportional transformation) of
the
frequency response of this filter from [0, 6.4 kHz] to [0, 8 kHz].
The result, siffi(n), is finally processed by a bandpass filter (block 112) of
FIR ("Finite
Impulse Response") type, to keep only the 6 ¨ 7 kHz band', at 23.85 kbit/s, a
low-pass filter
also of FIR type (block 113) is added to the processing to further attenuate
the frequencies
above 7 kHz. The high frequency (HF) synthesis is selected by switch 114 as
either
the output of block 112 or block 113. The high frequency synthesis is finally
added
(block 130) to the low frequency (LF) synthesis obtained with the blocks 120
to 122
and re-sampled at 16 kHz (block 123). Thus, even if the high band extends in
theory from
6.4 to 7 kHz in the AMR-WB codec, the HF synthesis is rather contained in the
6-7 kHz
band before addition with the LF synthesis.
A number of drawbacks in the band extension technique of the AMR-WB codec can
be identified, in particular:
= the estimation of gains for each subframe (block 101, 103 to 105) is not
optimal.
Partly, it is based on an equalization of the "absolute" energy per subframe
(block
101) between signals at different frequencies: artificial excitation at 16 kHz
(white
noise) and a signal at 12.8 kHz (decoded ACELP excitation). It can be noted in
particular that this approach implicitly induces an attenuation of the high-
band
excitation (by a ratio 12.8/16 = 0.8); in fact, it will also be noted no de-
emphasis is
performed on the high band in the AMR-WB codec, which implicitly induces an
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amplification relatively close to 0.6 (which corresponds to the value of the
frequency
response of 1/ (1-0.68z-1) at 6400 Hz). In fact, the factors of 1/0.8 and of
0.6 are
compensated approximately.
= Regarding speech, the 3GPP AMR-WB codec characterization tests documented
in the
5 3GPP report TR 26.976 have shown that the mode at 23.85 kbit/s has a
less good
quality than at 23.05 kbit/s, its quality being in fact similar to that of the
mode at
15.85 kbit/s. This shows in particular that the level of artificial HF signal
has to be
controlled very prudently, because the quality is degraded at 23.85 kbit/s
whereas
the 4 bits per frame are considered to best make it possible to approximate
the
3.0 energy of the original high frequencies.
= The low-pass filter at 7 kHz (block 113) introduces a shift of almost 1
ms between the
low and high bands, which can potentially degrade the quality of certain
signals by
slightly desynchronizing the two bands at 23.85 kbit/s - this
desynchronization can
also pose problems when switching bit rate from 23.85 kbit/s to other modes.
3.5
An example of band extension via a temporal approach is described in the 3GPP
standard TS
26.290 describing the AMR-WB+ codec (standardized in 2005). This example is
illustrated in
the bbck diagrams of figures 2a (general block diagram) and 2b (gain
prediction by response
level correction) which correspond respectively to figures 16 and 10 of the
3GPP specification
20 TS 26.290.
In the AMR-WB+ codec, the (mono) input signal sampled at the frequency Fs (in
Hz) is
divided into two separate frequency bands, in which two LPC filters are
computed and coded
separately:
= one LPC filter, denoted A(z), in the low band (0-Fs/4) - its quantized
version is
25 denoted ;1(z)
= another LPC filter, denoted Ari,(z), in the spectrally aliased high band
(Fs/4-Fs/2) -
,.
its quantized version is denoted Af.5.(z)
The band extension is done in the AMR-WB+ codec as detailed in sections 5.4
(HF coding)
and 6.2 (HF decoding) of the 3GPP specification TS 26.290. The principle
thereof is
30 summarized here: the extension consists in using the excitation decoded
at low frequencies
(LFC excit.) and in formatting this excitation by a temporal gain per subframe
(block 205) and
an LPC synthesis filtering (block 207); the processing operations to enhance
(post-
processing) the excitation (block 206) and smooth the energy of the
reconstructed HF signal
(block 208) are moreover implemented as illustrated in figure 2a.
35 It is important to note that this extension in AMR-WB+ necessitates the
transmission of
additional information: linearization, at block 209, of the coefficients of
the filter
A, (z) in 204 and a temporal formatting
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gain per subframe (block 201). One particular feature of the band extension
algorithm in
AMR-WB+ is that the gain per subframe is quantified by a predictive approach;
in other
words, the gains are not coded directly, but rather gain corrections which are
relative to an
estimation of the gain denoted g . This
estimation, g õwitch , actually corresponds to a level
equalization factor between the filters A(z) and AHF (z) at the frequency of
separation
between low band and high band (Fs/4). The computation of the factor g ch
(block 203) is
detailed in figure 10 of the 3GPP specification TS 26.290 reproduced here in
figure 2b. This
figure will not be detailed more here. It will simply be noted that the blocks
210 to 213 are
A(z)
used to compute the energy of the impulse response of , while
recalling
(1 ¨0.9 z-1)Aõ(z)
..
i0 that the filter A(z) models a spectrally aliased high band (because of
the spectral
properties of the fitter bank separating the low and high bands). Since the
filters are
interpolated by subframes at block 202, the gain g match is computed only once
per
frame, and it is interpolated by subframes.
The band extension gain coding technique in AMR-WB+, and more particularly the
compensation of levels of the LPC filters at their junction is an appropriate
method in the
context of a band extension by LPC models in low and high band, and it can be
noted that
such a level compensation between LPC fitters is not present in the band
extension of the
AMR-WB codec. However, it is in practice possible to verify that the direct
equalization of the
level between the two LPC filters at the separation frequency is not an
optimal method and
can provoke an overestimation of energy in high band and audible artifacts in
certain cases; it
will be recalled that an LPC filter represents a spectral envelope, and the
principle of
equalization of the level between two LPC filters for a given frequency
amounts to adjusting
the relative level of two LPC envelopes. Now, such an equalization performed
at a precise
frequency does not ensure a complete continuity and overall consistency of the
energy (in
frequency) in the vicinity of the equalization point when the frequency
envelope of the signal
fluctuates significantly in this vicinity. A mathematical way of positing the
problem consists in
noting that the continuity between two curves can be ensured by forcing them
to meet at one
and the same point, but there is nothing to guarantee that the local
properties (successive
derivatives) coincide so as to ensure a more global consistency. The risk in
ensuring a spot
continuity between low and high band LPC envelopes is of setting the LPC
envelope in high
band at a relative level that is too strong or too weak, the case of a level
that is too strong
being more damaging because it results in more annoying artifacts.
Moreover, the gain compensation in AMR-WB+ is primarily a prediction of the
gain known to
the coder and to the decoder and which serves to reduce the bit rate necessary
for the
transmission of gain information scaling the high-band excitation signal. Now,
in the context
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of an interoperable enhancement of the AMR-WB coding/decoding, it is not
possible to modify
the existing coding of the gains by subframes (0.8 kbit/s) of the band
extension in the AMR-
WB 23.85 kbit/s mode. Furthermore, for the bit rates strictly less than 23.85
kbit/s, the
compensation of levels of LPC filters in low and high bands can be applied in
the band
extension of a decoding compatible with AMR-WB, but experience shows that this
sole
technique derived from the AMR-WB+ coding, applied without optimization, can
cause
problems of overestimation of energy of the high band (> 6 kHz).
There is therefore a need to improve the compensation of gains between linear
prediction
filters of different frequency bands for the frequency band extension in a
codec of AMR-WB
type or an interoperable version of this codec without in any way
overestimating the energy
in a frequency band and without requiring additional information from the
coder.
The present invention improves the situation.
To this end, the invention targets a method for determining an optimized scale
factor
3.5 to be applied to an excitation signal or to a fitter in an audio
frequency signal frequency band
extension method, the band extension method comprising a step of decoding or
of extraction,
in a first frequency band, of an excitation signal and of parameters of the
first frequency
band comprising coefficients of a linear prediction filter, a step of
generation of an extended
excitation signal on at least one second frequency band and a step of
filtering, by a linear
prediction filter, for the second frequency band. The determination method is
such that it
comprises the following steps:
- determination of a linear prediction fitter called additional filter, of
lower order than the
linear prediction filter of the first frequency band, the coefficients of the
additional filter
being obtained from the parameters decoded or extracted from the first
frequency band;
and
- computation of the optimized scale factor as a function at least of the
coefficients of the
additional fitter.
Thus, the use of an additional filter of lower order than the fitter of the
first frequency
band to be equalized makes it possible to avoid the overestimations of energy
in the high
frequencies which could result from local fluctuations of the envelope and
which can disrupt
the equalization of the prediction filters.
The equalization of gains between the linear prediction filters of the first
and second
frequency bands is thus enhanced.
In an advantageous application of the duly obtained optimized scale factor,
the band
extension method comprises a step of application of the optimized scale factor
to the
extended excitation signal.
In an appropriate embodiment, the application of the optimized scale factor is
combined with the step of filtering in the second frequency band.
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Thus, the steps of filtering and of application of the optimized scale factor
are
combined in a single filtering step to reduce the processing complexity.
In a particular embodiment, the coefficients of the additional filter are
obtained by
truncation of the transfer function of the linear prediction filter of the
first frequency band to
obtain a lower order.
This lower order additional filter is therefore obtained in a simple manner.
Furthermore, so as to obtain a stable filter, the coefficients of the
additional filter are
modified as a function of a stability criterion of the additional filter.
In a particular embodiment, the computation of the optimized scale factor
comprises
the following steps:
- computation of the frequency responses of the linear prediction filters of
the first and
second frequency bands for a common frequency;
- computation of the frequency response of the additional filter for this
common
frequency;
- computation of the optimized scale factor as a function of the duly computed
frequency
responses.
Thus, the optimized scale factor is computed in such a way as to avoid the
annoying
artifacts which could occur should the higher order filter frequency response
of the first band
in proximity to the common frequency show a signal peak or trough.
In a particular embodiment, the method further comprises the following steps,
implemented for a predetermined decoding bit rate:
- first scaling of the extended excitation signal by a gain computed per
subframe as a
function of an energy ratio between the decoded excitation signal and the
extended
excitation signal;
- second scaling of the excitation signal obtained from the first scaling by a
decoded
correction gain;
- adjustment of the energy of the excitation for the current subfranne by an
adjustment
factor computed as a function of the energy of the signal obtained after the
second
scaling and as a function of the signal obtained after application of the
optimized scale
factor.
Thus, additional information can be used to enhance the quality of the
extended
signal for a predetermined operating mode.
The invention also targets a device for determining an optimized scale factor
to be
applied to an excitation signal or to a filter in an audio frequency signal
frequency band
extension device, the band extension device comprising a module for decoding
or extracting,
in a first frequency band, an excitation signal and parameters of the first
frequency band
comprising coefficients of a linear prediction filter, a module for generating
an extended
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9
excitation signal on at least one second frequency band and a module for
filtering, by
a linear prediction filter, for the second frequency band. The determination
device is
such that it comprises:
- a module for determining a linear prediction filter called additional
fitter, of
lower order than the linear prediction filter of the first frequency band, the
coefficients of the additional filter being obtained from the parameters
decoded
or extracted from the first frequency band; and
- a module for computing the optimized scale factor as a function at least
of the
coefficients of the additional filter.
The invention targets a decoder comprising a device as described.
It targets a computer program comprising code instructions for implementing
the steps of the method for determining an optimized scale factor as
described, when
these instructions are executed by a processor.
Finally, the invention relates to a storage medium, that can be read by a
processor, incorporated or not in the device for determining an optimized
scale
factor, possibly removable, storing a computer program implementing a method
for
determining an optimized scale factor as described previously.
According to one aspect of the present invention, there is provided a method
for determining an optimized scale factor to be applied to an excitation
signal or to a
filter in an audio frequency signal frequency band extension method, the band
extension method comprising a step of decoding or of extraction, in a first
frequency
band, of an excitation signal and of parameters of the first frequency band
comprising coefficients of a linear prediction filter, a step of generation of
an
extended excitation signal on at least one second frequency band and a step of
filtering, by a linear prediction filter, for the second frequency band, the
determination method comprising: determination of a linear prediction filter
called
additional filter, of lower order than the linear prediction filter of the
first frequency
band, the coefficients of the additional filter being obtained from the
parameters
decoded or extracted from the first frequency band; and computation of the
optimized scale factor as a function at least of the coefficients of the
additional filter.
According to another aspect of the present invention, there is provided a
device for determining an optimized scale factor to be applied to an
excitation signal
or to a filter in an audio frequency signal frequency band extension device,
the band
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9a
extension device comprising a module for decoding or extracting, in a first
frequency
band, an excitation signal and parameters of the first frequency band
comprising
coefficients of a linear prediction filter, a module for generating an
extended
excitation signal on at least one second frequency band and a module for
filtering, by
a linear prediction filter, for the second frequency band, the determination
device
comprising: a module for determining a linear prediction filter called
additional filter,
of lower order than the linear prediction filter of the first frequency band,
the
coefficients of the additional filter being obtained from the parameters
decoded or
extracted from the first frequency band; and a module for computing the
optimized
scale factor as a function at least of the coefficients of the additional
filter.
Other features and advantages of the invention will become more clearly
apparent on reading the following description, given purely as a nonlimiting
example
and with reference to the attached drawings, in which:
- figure 1 illustrates a part of a decoder of AMR-WB type implementing
frequency band extension steps of the prior art and as described
previously;
- figures 2a and 2b present the coding of the high band in the AMR-WB+
codec according to the prior art and as described previously;
- figure 3 illustrates a decoder that can interwork with the AMR-WB coding,
incorporating a band extension device used according to an embodiment
of the invention;
- figure 4 illustrates a device for determining a scale factor optimized by
a
subframe as a function of the bit rate, according to an embodiment of the
invention; and
- figures 5a and 5b illustrate the frequency responses of the filters used
for
the computation of the optimized scale factor according to an embodiment
of the invention;
- figure 6 illustrates, in flow diagram form, the main steps of a method
for
determining an optimized scale factor according to an embodiment of the
invention;
- figure 7 illustrates an embodiment in the frequency domain of a device
for
determining an optimized scale factor as part of a band extension;
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84019720
- figure 8 illustrates a hardware implementation of an optimized scale factor
determination device in a band extension according to the invention.
Figure 3 illustrates an exemplary decoder, compatible with the AMR-VVB/G.722.2
5 standard in which there is a band extension comprising a
determination of an optimized scale
factor according to an embodiment of the method of the invention, implemented
by the band
extension device illustrated by the block 309.
Unlike the AMR-WB decoding which operates with an output sampling frequency of
16 kHz, a decoder is considered here which can operate with an output signal
(synthesis) at
10 the frequency fs = 8, 16, 32 or 48 kHz. It should be noted that it
is assumed here that the
coding has been performed according to the AMR-WB algorithm with an internal
frequency of
12.8 kHz for the CELP coding in low band and at 23.85 kbit/s with a gain
coding per subframe
at the frequency of 16 kHz; even though the invention is described here at the
decoding level, it
is assumed here that the coding can also operate with an input signal at the
frequency Is
2.5 = 8, 16, 32 or 48 kHz and suitable resampling operations, beyond
the context of the
invention, are implemented in coding as a function of the value of Is. It can
be noted that,
when fs = 8 kHz, in the case of a decoding compatible with AMR-WB, it is not
necessary to
extend the 0-6.4 kHz low band, because the audio band reconstructed at the
frequency fs is
limited to 0-4000 Hz.
In figure 3, the CELP decoding (LF for low frequencies) still operates at the
internal
frequency of 12.8 kHz, as in AMR-WB, and the band extension (HF for high
frequencies) used
for the invention operates at the frequency of 16 kHz, and the LF and HF
syntheses are
combined (block 312) at the frequency Is after suitable resampling (block 306
and internal
processing in the block 311). In the variant embodiments, the combining of the
low and high
bands can be done at 16 kHz, after having resampled the low band from 12.8 to
16 kHz,
before resampling the combined signal at the frequency Is.
The decoding according to figure 3 depends on the AMR-WB mode (or bit rate)
associated with the current frame received. As an indication, and without
affecting the block
309, the decoding of the CELP part in low band comprises the following steps:
= demultiplexing of the coded parameters (block 300) in the case of a frame
correctly
received (bfi=0 where bfi is the "bad frame incficatof with a value 0 for a
frame
received and 1. for a frame lost);
= decoding of the ISF parameters with interpolation and conversion into LPC
coefficients (block 301) as described in clause 6.1 of the standard G.722.2;
= decoding of the CELP excitation (block 302), with an adaptive and fixed part
for
reconstructing the excitation (exc or u'(0) in each subframe of length 64 at
12.8
kHz:
n =0,- ,63
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by following the notations of clause 7.1.2.1 of ITU-T recommendation G.718 of
a
decoder interoperable with the AMR-WB coder/decoder, concerning the CELP
decoding, where v(n) and c(n) are respectively the code words of the adaptive
and
fixed dictionaries, and gp and R, are the associated decoded gains. This
excitation
u'(n) is used in the adaptive dictionary of the next subframe; it is then post-
processed and, as in G.718, the excitation u'(n) (also denoted exc) is
distinguished
from its modified post-processed version u(n) (also denoted exc2) which serves
as
input for the synthesis fitter, 1/ ;1(z) , in the block 303;
= synthesis filtering by 1/ 11(z) (block 303) where the decoded LPC fitter
A(z) is of
the order 16;
= narrow-band post-processing (block 304) according to clause 7.3 of G.718
if fs=8
kHz;
= de-emphasis (block 305) by the filter 1/ (1-0.68z-1) ;
= post-processing of the low frequencies (called "bass posfilter) (block
306)
attenuating the cross-harmonics noise at low frequencies as described in
clause 7.14.1.1 of G.718. This processing introduces a delay which is taken
into
account in the decoding of the high band (> 6.4 kHz);
= resampling of the internal frequency of 12.8 kHz at the output frequency
fs (block
307). A number of embodiments are possible. Without losing generality, it is
considered here, by way of example, that if fs=8 or 16 kHz, the resampling
described in clause 7.6 of G.718 is repeated here, and if fs=32 or 48 kHz,
additional
finite impulse response (FIR) filters are used;
= computation of the parameters of the "noise gate" (block 308)
preferentially
performed as described in clause 7.14.3 of G.718 to "enhance" the quality of
the
silences by level reduction.
In variants which can be implemented for the invention, the post-processing
operations
applied to the excitation can be modified (for example, the phase dispersion
can be
enhanced) or these post-processing operations can be extended (for example, a
reduction of
the cross-harmonics noise can be implemented), without affecting the nature of
the band
extension.
It can be noted that the use of blocks 306, 308, 314 is optional.
It will also be noted that the decoding of the low band described above
assumes a so-called
"active" current frame with a bit rate between 6.6 and 23.85 kbit/s. In fact,
when the DTX
mode is activated, certain frames can be coded as "inactive" and in this case
it is possible to
either transmit a silence descriptor (on 35 bits) or transmit nothing. In
particular, it will be
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recalled that the SID frame describes a number of parameters: ISF parameters
averaged over
8 frames, average energy over 8 frames, "dithering" flag for the
reconstruction of non-
stationary noise. In all cases, in the decoder, there is the same decoding
model as for an
active frame, with a reconstruction of the excitation and of an LPC filter for
the current frame,
which makes it possible to apply the band extension even to inactive frames.
The same
observation applies for the decoding of "lost frames" (or FEC, PLC) in which
the LPC model is
applied.
In the embodiment described here and with reference to figure 7, the decoder
makes
it possible to extend the decoded low band (50-6400 Hz taking into account the
50 Hz high-
2.0 .. pass filtering on the decoder, 0-6400 Hz in the general case) to an
extended band, the width
of which varies, ranging approximately from 50-6900 Hz to 50-7700 Hz depending
on the
mode implemented in the current frame. It is thus possible to refer to a first
frequency band
of 0 to 6400 Hz and to a second frequency band of 6400 to 8000 Hz. In reality,
in the
preferred embodiment, the extension of the excitation is performed in the
frequency domain
in a 5000 to 8000 Hz band, to allow a bandpass filtering of 6000 to 6900 or
7700 Hz width.
At 23.85 kbit/s, the HF gain correction information (0.8 kbit/s) transmitted
at 23.85
kbit/s is here decoded. Its use is detailed later, with reference to figure 4.
.. The .. high-
band synthesis part is produced in the block 309 representing the band
extension device used
for the invention and which is detailed in figure 7 in an embodiment.
In order to align the decoded low and high bands, a delay (block 310) is
introduced
to synchronize the outputs of the blocks 306 and 307 and the high band
synthesized at 16
kHz is resampled from 16 kHz to the frequency fs (output of block 311). The
value of the
delay T depends on how the high band signal is synthesized, and on the
frequency fs as in
the post-processing of the low frequencies. Thus, generally, the value of Tin
the block 310
will have to be adjusted according to the specific implementation.
The low and high bands are then combined (added) in the block 312 and the
synthesis obtained is post-processed by 50 Hz high-pass filtering (of IIR
type) of order 2, the
coefficients of which depend on the frequency fs (block 313) and output post-
processing with
optional application of the "noise gate" in a manner similar to G.718 (block
314).
Referring to figure 3, an embodiment of a device for determining an optimized
scale
factor to be applied to an excitation signal in a frequency band extension
process is now
described. This device is included in the band extension block 309 described
previously.
Thus, the block 400, from an excitation signal decoded in a first frequency
band
u(n), performs a band extension to obtain an extended excitation signal u
HB(n) on at least
one second frequency band.
It will be noted here that the optimized scale factor estimation according to
the
invention is independent of how the signal um(n) is obtained. One condition
concerning its
energy is, however, important. Indeed, the energy of the high band from 6000
to 8000 Hz
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must be at a level similar to the energy of the band from 4000 to 6000 Hz of
the decoded
excitation signal at the output of the block 302. Furthermore, since the low-
band signal is de-
emphasized (block 305), the de-emphasis must also be applied to the high-band
excitation
signal, either by using a specific de-emphasis filter, or by multiplying by a
constant factor
which corresponds to an average attenuation of the filter mentioned. This
condition does not
apply to the case of the 23.85 kbit/s bit rate which uses the additional
information
transmitted by the coder. In this case, the energy of the high-band excitation
signal must be
consistent with the energy of the signal corresponding to the coder, as
explained later.
The frequency band extension can, for example, be implemented in the same way
as
3.0 for the
decoder of AMR-WB type described with reference to figure 1 in the blocks 100
to
102, from a white noise.
In another embodiment, this band extension can be performed from a combination
of
a white noise and of a decoded excitation signal as illustrated and described
later for the
blocks 700 to 707 in figure 7.
3.5 Other
frequency band extension methods with conservation of the energy level
between the decoded excitation signal and the extended excitation signal as
described below,
can of course be envisaged for the block 400.
Furthermore, the band extension module can also be independent of the decoder
and
can perform a band extension for an existing audio signal stored or
transmitted to the
20 extension
module, with an analysis of the audio signal to extract an excitation and an
LPC
filter therefrom. In this case, the excitation signal at the input of the
extension module is no
longer a decoded signal but a signal extracted after analysis, like the
coefficients of the linear
prediction filter of the first frequency band used in the method for
determining the optimized
scale factor in an implementation of the invention.
25 In the
example illustrated in figure 4, the case of the bit rates < 23.85 kbit/s, for
which the determination of the optimized scale factor is limited to the block
401, is
considered first.
In this case, an optimized scale factor denoted g1182(m) is computed. In one
embodiment,
this computation is performed preferentially for each subframe and it consists
in equalizing
30 the levels of
the frequency responses of the LPC filters 1/ ;1(z) and 1/ A(z/ y) used in low
and high frequencies, as described later with reference to figure 7, with
additional
precautions to avoid the cases of overestimations which can result in an
excessive energy of
the synthesized high band and therefore generate audible artifacts.
In an alternative embodiment, it will be possible to keep the extrapolated HF
synthesis fitter
35 1/ A- (z/ y)
as implemented in the AMR-WB decoder or a decoder that can interwork with
the AMR-WB coder/decoder, for example according to the ITU-T recommendation
G.718, in
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place of the filter 1/ A(z/ 7). The compensation according to the invention is
then performed
from the filters 1/ A(z) and it Aexi (z/ y) .
The determination of the optimized scale factor is also performed by the
determination (in
401a) of a linear prediction filter called additional filter, of lower order
than the linear
prediction filter of the first frequency band 1/ A(z), the coefficients of the
additional filter
being obtained from the parameters decoded or extracted from the first
frequency band. The
optimized scale factor is then computed (in 401b) as a function at least of
these coefficients
to be applied to the extended excitation signal u (n) .
The principle of the determination of the optimized scale factor, implemented
in the
block 401, is illustrated in figures 5a and 5b with concrete examples obtained
from signals
sampled at 16 kHz; the frequency response amplitude values, denoted R, P, Q
below, of 3
filters are computed at the common frequency of 6000 Hz (vertical dotted line)
in the current
subframe, of which the index m is not recalled here in the notations of the
LPC filters
interpolated by subframe to lighten the text. The value of 6000 Hz is chosen
such that it is
close to the Nyquist frequency of the low band, that is 6400 Hz. It is
preferable not to take
this Nyquist frequency to determine the optimized scale factor. Indeed, the
energy of Lhe
decoded signal in low frequencies is typically already attenuated at 6400 Hz.
Furthermore,
the band extension described here is performed on a second frequency band,
called high
band, which ranges from 6000 to 8000 Hz. It should be noted that, in variants
of the
invention, a frequency other than 6000 Hz will be able to be chosen, with no
loss of
generality for determining the optimized scale factor. It will also be
possible to consider the
case where the two LPC filters are defined for the separate bands (as in AMR-
WB+). In this
case, R, P and Q will be computed at the separation frequency.
Figures 5a and 5b illustrate how the quantities R, P, Qare defined.
The first step consists in computing the frequency responses R and P
respectively of the
linear prediction filter of the first frequency band (low band) and of the
second frequency
band (high band) at the frequency of 6000 Hz. The following is first computed:
R= 1 =
1
, ,
1A(eiel
i=o
in which M =16 is the order of the decoded LPC filter, 1/ A(z) , and
Ocorresponds to the
frequency of 6000 Hz normalized for the sampling frequency of 12.8 kHz, that
is:
6000
= 27r
12800
Then, similarly, the following is computed:
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1 1
1=1
IA(e1 . /
¨o
in which
6000
16000
In a preferred embodiment, the quantities P and R are computed according to
the
following pseudo-code:
px = py = 0
tx = ry = 0
for 1=0 to 16
px = px Ap[i]*exp tab_p[i]
py = py Ap[i]*exp tab_p[33-8
ix = ix Aq[ii*exp_tab_q[d
iy = ry Aq[i]*exp_tab_q[33-11
end for
P = ltsgit(Px*Px+PV*PY)
R = 1/sqrt(fx*tx+fy*ty)
in which Aq[i]= a, corresponds to the coefficients of A(z) (of order 16),
Ap[0=
corresponds to the coefficient of A(z/ y), sqrt() corresponds to the square
root operation
and the tables exp_tab_p and exp_lab_ci of sire 34 (Aladin the real and
imaginary parts or
the complex exponentials associated with the frequency of 6000 Hz, with
6000"i
cos (22r i =0,- - -,16
12800
exp_tab_p[i] =
¨ sin (27r 6000 (33¨i)) i=17,===,33
12800
cos(2a- 6" i) i = 0, = = = ,16
16000
exp_tab_q[i] =
-sin (27r 6000
(33¨i)) i=17,===,33
16000
The additional prediction filter is obtained for example by suitably
truncating the polynomial
;1(z) to the order 2.
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In fact, the direct truncation to the order leads to the filter 1+ + aõ which
can pose a
problem because there is generally nothing to guarantee that this filter of
order 2 is stable. In
a preferred embodiment, the stability of the filter 1+ a1 + a, is therefore
detected and a filter
1+ ai'+ (-12 is used, the coefficients of which are drawn from 1+ 6,+ 112 as a
function of the
instability detection. More specifically, the following are initialized:
a,' = i=1,2
The stability of the filter 1+ a, + a, can be verified differently; here, a
conversion is used in
the PARCOR coefficients (or reflection coefficients) domain by computing:
k, = a,7 (1+ a,=)
3.0
The stability is verified if lk, I <1, i=1, 2. The value of k, is therefore
conditionally modified
before ensuring the stability of the filter, with the following steps:
min(0.6, k2 ) k2 >0
k
max(-0.6, k2) k2 < 0
min(0.99, >0
k
max(-0.99, k2) k,< 0
in which min(.,.) and max(.,.) respectively give the minimum and the maximum
of 2
operands.
It should be noted that the threshold values, 0.99 for k, and 0.6 for k2, will
be able to be
adjusted in variants of the invention. It will be recalled that the first
reflection coefficient, k1,
characterizes the spectral slope (or tilt) of the signal modeled to the order
1; in the invention
the value of k, is saturated at a value close to the stability limit, in order
to preserve this
slope and retain a tilt similar to that of 1/ A(z) . It will also be recalled
that the second
reflection coefficient, k2, characterizes the resonance level of the signal
modeled to the order
2; since the use of a filter of order 2 aims to eliminate the influence of
such resonances
around the frequency of 6000 Hz, the value of k2 is more strongly limited;
this limit is set at
0.6.
The coefficients of 1+ a,i-Fa2' are then obtained by:
'=(1+k2)k,
62 = k2
The frequency response of the additional filter is therefore finally computed:
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1
Q = 2
Eak e- ik0
k=0
6000
with 0 = 271- . This quantity is computed preferentially according to
the following
12800
pseudo-code:
qx = qy =
for 1=0 to 2
qx = qx As[i]*exp tab q[d;
qy = qy Aspl*exp tab q[33-11;
end for
Q = 1/sqtt(qx*qx-i-qyVy)
in which As[i]¨ .
With no loss of generality, it will be possible to compute the coefficients of
the filter of order
2 otherwise, for example by applying to the LPC filter A(z) of order 16 the
reduction
procedure of the LPC order called "STEP DOWN" described in JD. Markel and A.H.
Gray,
Linear Prediction of Speech, Springer Verlag, 1976 or by performing two
Levinson-Durbin (or
STEP-UP) algorithm iterations from the self-correlations computed on the
signal synthesized
(decoded) at 12.8 kHz and windowed.
For some signals, the quantity Q, computed from the first 3 LPC coefficients
decoded, better
takes account of the influence of the spectral slope (or tilt) in the spectrum
and avoids the
influence of "spurious" peaks or troughs close to 6000 Hz which can skew or
raise the value
of the quantity R, computed from all the LPC coefficients.
In a preferred embodiment, the optimized scale factor is deduced from the pre-
computed
quantities R, P. Q conditionally, as follows:
2s
If the tilt (computed as in AMR-WB in the block 104, by normalized self-
correlation in the
form r(1)/r(0) in which r(i) is the self-correlation) is negative (tilt < 0 as
represented in figure
5b), the computation of the scale factor is done as follows:
to avoid artifacts due to excessively abrupt variations of energy of the high
band, a
smoothing is applied to the value of R . In a preferred embodiment, an
exponential
smoothing is performed with a fixed factor in time (0.5) in the form of:
R =0.5R +0.5R põõ
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= R
in which R corresponds to the value of R in the preceding subframe and
the factor 0.5 is
optimized empirically - obviously, the factor 0.5 will be able to be changed
for another value
and other smoothing methods are also possible. It should be noted that the
smoothing makes
it possible to reduce the temporal variants and therefore avoid artifacts.
The optimized scale factor is then given by:
gmn(m)= max(min(R, Q), P) I P
In an alternative embodiment, it will be possible to replace the smoothing of
R with a
smoothing of g õB2(m) such that:
0.5 g õ2(m) + 0.5 g õ2(m -1)
If the tilt (computed as in AMR-WB in the block 104) is positive (tilt > 0 as
in figure 5a), the
computation of the scale factor is done as follows:
the quantity R is smoothed adaptively in time, with a stronger smoothing when
R is
low - as in the preceding case, this smoothing makes it possible to reduce the
temporal
variants and therefore avoids artifacts:
R.-- (1- a)Ri-aRpr, with a =1- R2
= R
Then, the optimized scale factor is given by:
g HB2(m) =min (R,P,Q)I P
In an alternative embodiment, it will be possible to replace the smoothing of
R with a
smoothing of g11B2(m) as computed above.
M = 0,...,3 , a =1- 8,,,z B(m)
where g,,,(-1) is the scale or gain factor computed for the last subframe of
the preceding
frame.
The minimum of R, P. Q is taken here in order to avoid overestimating the
scale factor.
In a variant, the above condition depending only on the tat will be able to be
extended to
take account not only of the tilt parameter but also of other parameters in
order to refine the
decision. Furthermore, the computation of gH,,,(m) will be able to be adjusted
according to
these said additional parameters.
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An example of additional parameter is the number of zero crossings (ZCR, zero
crossing rate)
which can be defined as:
N-1
Z,Cr, = ¨El sgn [s(n)]¨ sgn[s(n ¨1)] I
2 n=i
in which
{ 1 if x 0
sgn(x)
¨1 if x < 0
The parameter zcr generally gives results similar to the tilt. A good
classification criterion is
the ratio between zcrs computed for the synthesized signal s(n) and zci.
computed for the
excitation signal u(n) at 12 800 Hz. This ratio is between 0 and 1, where 0
means that the
signal has a decreasing spectrum, 1 that the spectrum is increasing (which
corresponds to
io (1- tilt)/ 2. In this case, a ratio zers I zcr. > 0.5 corresponds to the
case tilt < 0, a ratio
zcr, I zcr.< 0.5 corresponds to tilt > 0.
In a variant, it will be possible to use a function of a parameter tilt.,
where ti/thp is the tilt
computed for the synthesized signal s(n) filtered by a high-pass filter with a
cut-off
frequency for example at 4800 Hz; in this case, the response 1/ A(z/ y) from 6
to 8 kHz
3.5 (applied at 16 kHz) corresponds to the weighted response of 1/ A(z)
from 4.8 to 6.4 kHz.
Since 1/ A(z/ y) has a more flattened response, it is necessary to compensate
this change of
tilt. The scale factor function according to tilt,õ is then given in an
embodiment by:
(1 - tilthp )2 + 0.6. Q and R are therefore multiplied by min (1, (1-tilthp )2
+0.6) when
tilt >0 or by max (1, (1 - ti/thp )2 +0.6) when tilt <0.
20 The case of
the 23.85 kbit/s bit rate is now considered, for which a gain correction is
performed by the blocks 403 to 408. This gain correction could moreover be the
subject of a
separate invention. In this particular embodiment according to the invention,
the gain
correction information, denoted 2
HBcorr (in) , transmitted by the AMR-WB (compatible) coding
with a bit rate of 0.8 kbit/s, is used to improve the quality at 23.85 kbit/s.
25 It is assumed here that the AMR-WB (compatible) coding has performed a
correction gain
quantization on 4 bits as described in ITU-T clause G.722.2/5.11 or,
equivalently, in the 3GPP
clause TS 26.190/5.11.
In the AMR-WB coder, the correction gain is computed by comparing the energy
of the
original signal sampled at 16 kHz and filtered by a 6-7 kHz bandpass filter,
sHB(n) , with the
3o energy of the white noise at 16 kHz filtered by a synthesis filter 1/
A(z / y) and a 6-7 kHz
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bandpass filter (before the filtering, the energy of the noise is set to a
level similar to that of
the excitation at 12.8 kHz), s ,B2(n) . The gain is the root of the ratio of
energy of the
original signal to the energy of the noise divided by two. In one possible
embodiment, it will
be possible to change the bandpass filter for a filter with a wider band (for
example from 6 to
7.6 kHz).
go(.+0-1 ____________________
1 sns(n)2
g HBcorr (M) = 80(nm-8+ 17-1 __ , in = 0, ..., 3
S,1(fl)
To be able to apply the gain information received at 23.85 kbit/s (in the
block 407), it is
important to bring the excitation to a level similar to that expected of the
AMR-WB
3.o (compatible) coding. Thus, the block 404 performs the scaling of the
excitation signal
according to the following equation;
U HB1(n) = g HB3(m)nns(n) , n = 80m,= = = ,80(m +1)-1
in which g,,3(m) is a gain per subframe computed in the block 403 in the form:
63 _______________________
Eu(n)2
g HB 3 (M)
\5 .1u ii,(n)2
rz=O
in which the factor 5 in the denominator serves to compensate the bandwidth
difference
between the signal u(n) and the signal ullB(n) , given that, in the AMR-WB
coding, the HF
excitation is a white noise over the 0-8000 Hz band.
The index of 4 bits per subframe, denoted indexHp_gain (rn), sent at 23.85
kbit/s is
demultiplexed from the bit stream (block 405) and decoded by the block 406 as
follows:
g õõõ(m) --= 2.HP _gain(index (m))
in which HP_gain(.) is the HF gain quantization dictionary defined in the AMR-
WB coding
and recalled below:
i HP gain(1) I HP_gain(t)
0 0.110595703125000 8 0.342102050781250
1 0.142608642578125 9 0.372497558593750
2 0.170806884765625 10 0.408660888671875
3 0.197723388671875 11 0.453002929687500
4 0.226593017578125 12 0.511779785156250
5 0.255676269531250 13 0.599822998046875f
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6 0.284545898437500 14 0.741241455078125
7 0.313232421875000 15 0.998779296875000
Table 1 (gain dictionary at 23.85 kbit/s)
The block 407 performs the scaling of the excitation signal according to the
following
equation:
11 HB2(n) g mkoõ(m)unBi(n), n= 80m, = = = , 80(m +1) ¨I
Finally, the energy of the excitation is adjusted to the level of the current
subfranne with the
following conditions (block 408). The following is computed:
79 _____________
E(g(m)g ne2(m)unB(1))2
fac(m)= n=0 79
Eum2(n)2
n=0
The numerator here represents the high-band signal energy which would be
obtained in the
io mode 23.05. As explained before, for the bit rates < 23.85 kbit/s, it is
necessary to retain the
level of energy between the decoded excitation signal and the extended
excitation signal
uõ(n) , but this constraint is not necessary in the case of the 23.85 kbit/s
bit rate, since
uõ(n) is in this case scaled by the gain gõ,(m). To avoid double
multiplications, certain
multiplication operations applied to the signal in the block 400 are applied
in the block 402 by
multiplying by g(m). The value of g(m) depends on the u(n)synthesis algorithm
and
must be adjusted such that the energy level between the decoded excitation
signal in low
band and the signal g(m)um(n) is retained.
In a particular embodiment, which will be described in detail later with
reference to figure 7,
g(m) = 0.6 g õ,(m), where g111(m) is a gain which ensures, for the signal um,
the same
ratio between energy per subframe and energy per frame as for the signal u(n)
and 0.6
corresponds to the average frequency response amplitude value of the de-
emphasis filter
from 5000 to 6400 Ni.
It is assumed that, in the block 408, there is information on the tilt of the
low-band signal ¨
in a preferred embodiment, this tilt is computed as in the AMR-WB codec
according to the
blocks 103 and 104, but other methods for estimating the tilt are possible
without changing
the principle of the invention.
If fac(m)> 1 or tilt < 0, the following is assumed:
unB'(n)= unB2(n), n = 80m, = = = , 80(m +1) ¨1
Otherwise:
uõ'(n)= max fac(m)).UHB2(n) , n= 80m, = = = , 80(m +1) ¨1
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It will be noted that the optimized scale factor computation described here,
notably in the
blocks 401 and 402, is distinguished from the abovementioned equalization of
filter levels
performed in the AMR-WB+ codec by a number of aspects:
= The optimized scale factor is computed directly from the transfer
functions of the LPC
filters without involving any temporal filtering. This simplifies the method.
= The equalization is done preferentially at a frequency different from the
Nyquist
frequency (6400 Hz) associated with the low band. Indeed, the LPC modeling
implicitly represents the attenuation of the signal typically caused by the
resampling
operations and therefore the frequency response of an LPC filter may be
subject at
the Nyquist frequency to a decrease which is not at the chosen common
frequency.
= The equalization here relies on a filter of lower order (here of order 2)
in addition to
the 2 filters to be equalized. This additional filter makes it possible to
avoid the
effects of local spectral fluctuations (peaks or troughs) which may be present
at the
common frequency for the computation of the frequency response of the
prediction
3.5 filters.
For the blocks 403 to 408, the advantage of the invention is that the quality
of the signal
decoded at 23.85 kbit/s according to the invention is improved relative to a
signal decoded at
23.05 kbit/s, which is not the case in an AMR-WEI decoder. In fact, this
aspect of the
invention makes it possible to use the additional information (0.8 kbit/s)
received at
23.85 kbit/s, but in a controlled manner (block 408), to improve the quality
of the extended
excitation signal at the bit rate of 23.85.
The device for determining the optimized scale facLor as illustrated by the
blocks 401 to 408
of figure 4 implements a method for determining the optimized scale factor now
described
with reference to figure 6.
The main steps are implemented by the block 401.
Thus, an extended excitation signal uH8(n) is obtained in a frequency band
extension
method E601 which comprises a step of decoding or of extraction, in a first
frequency band
called low band, of an excitation signal and of parameters of the first
frequency band such as,
for example, the coefficients of the linear prediction filter of the first
frequency band.
A step E602 determines a linear prediction filter called additional filter, of
lower order
than that of the first frequency band. To determine this filter, the
parameters of the first
frequency band decoded or extracted are used.
In one embodiment, this step is performed by truncation of the transfer
function of
the linear prediction filter of the low band to obtain a lower filter order,
for example 2. These
coefficients can then be modified as a function of a stability criterion as
explained previously
with reference to figure 4.
From the coefficients of the additional filter thus determined, a step E603 is
implemented to compute the optimized scale factor to be applied to the
extended excitation
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signal. This optimized scale factor is, for example, computed from the
frequency response of
the additional filter at a common frequency between the low band (first
frequency band) and
the high band (second frequency band). A minimum value can be chosen between
the
frequency response of this filter and those of the low-band and high-band
filters.
This therefore avoids the overestimations of energy which could exist in the
methods of the
prior art.
This step of computation of the optimized scale factor is, for example,
described
previously with reference to figure 4 and figures 5a and 5b.
The step E604 performed by the block 402 or 409 (depending on the decoding bit
io rate) for the band extension, applies the duly computed optimized scale
factor to the
extended excitation signal so as to obtain an optimized extended extension
signal ufrern).
In a particular embodiment, the device for determining the optimized scale
factor 708
is incorporated in a band extension device now described with reference to
figure 7. This
device for determining the optimized scale factor illustrated by the block 708
implements the
method for determining the optimized scale factor described previously with
reference to
figure 6.
In this embodiment, the band extension block 400 of figure 4 comprises the
blocks
700 to 707 of figure 7 that is now described.
Thus, at the input of the band extension device, a low-band excitation signal
decoded
or estimated by analysis is received (u(n)). The band extension here uses the
excitation
decoded at 12.8 kHz (exc2 or u(n)) at the output of the block 302 of figure 3.
It will be noted that, in this embodiment, the generation of the oversampled
and
extended excitation is performed in a frequency band ranging from 5 to 8 kHz
therefore
including a second frequency band (6.4-8 kHz) above the first frequency band
(0-6.4 kHz).
Thus, the generation of an extended excitation signal is performed at least
over the
second frequency band but also over a part of the first frequency band.
Obviously, the values defining these frequency bands can be different
depending on
the decoder or the processing device in which the invention is applied.
3o For this
exemplary embodiment, this signal is transformed to obtain an excitation
signal spectrum U(k) by the time-frequency transformation module 500.
In a particular embodiment, the transform uses a DCT-IV (for "Discrete Cosine
Transform" ¨
type IV) (block 700) on the current frame of 20 ms (256 samples), without
windowing, which
amounts to directly transforming u(n) with n = 0,= = 255 according to the
following
formula:
N-1
U(k)=Eu(n)cos( ¨71-rn ¨1\\
n=0 \ 2 2,
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in which N = 256 and k =0,= = = , 255.
It should be noted here that the transformation without windowing (or,
equivalently, with an
implicit rectangular window of the length of the frame) is possible because
the processing is
performed in the excitation domain, and not the signal domain so that no
artifact (block
effects) is audible, which constitutes an important advantage of this
embodiment of the
invention.
In this embodiment, the DCT-IV transformation is implemented by FFT according
to
the so-called "Evolved DCT(EDCT)" algorithm described in the article by D.M.
Zhang, H.T. Li,
io A Low Complexity Transform - Evolved OCT, IEEE 14th International
Conference on
Computational Science and Engineering (CSE), Aug. 2011, pp. 144149, and
implemented in
the MJ-T standards G.718 Annex B and G.729.1 Annex E.
In variants of the invention, and without loss of generality, the DCT-IV
transformation
will be able to be replaced by other short-term time-frequency transformations
of the same
is length and in the excitation domain, such as an FFT (for "Fast Fourier
Transform") or a DCT-
II (Discrete Cosine Transform - type II). Alternatively, it will be possible
to replace the DCT-
IV on the frame by a transformation with overlap-addition and windowing of
length greater
than the length of the current frame, for example by using an MDCT (for
"Modified Discrete
Cosine Transform"). In this case, the delay Tin the block 310 of figure 3 will
have to be
20 adjusted (reduced) appropriately as a function of the additional delay
due to the
analysis/synthesis by this transform.
The DCT spectrum, U(k), of 256 samples covering the 0-6400 Hz band (at
12.8 kHz), is then extended (block 701) into a spectrum of 320 samples
covering the 0-
8000 Hz band (at 16 kHz) in the following form:
0
{ k = 0,=-=,199
25 UHB,(k)= U (k) k = 200,= = = ,239
U(k + start _band - 240) k =240,= = = ,319
in which it is preferentially taken that start _band =160.
The block 701 operates as module for generating an oversampled and extended
excitation signal and performs a resampling from 12.8 to 16 kHz in the
frequency domain, by
adding 1/4 of samples (k = 240, = = .,319 ) to the spectrum, the ratio between
16 and 12.8
30 being 5/4.
Furthermore, the block 701 performs an implicit high-pass filtering in the 0-
5000 Hz
band since the first 200 samples of U mi(k) are set to zero; as explained
later, this high-
pass filtering is also complemented by a part of progressive attenuation of
the spectral values
of indices k =200,= = - , 255 in the 5000-6400 Hz band; this progressive
attenuation is
35 implemented in the block 704 but could be performed separately outside
of the block 704.
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Equivalently, and in variants of the invention, the implementation of the high-
pass filtering
separated into blocks of coefficients of index k =0,== = ,199 set to zero, of
attenuated
coefficients k = 200, = = =,255 in the transformed domain, will therefore be
able to be
performed in a single step.
In this exemplary embodiment and according to the definition of U i(k) , it
will be
noted that the 5000-6000 Hz band of U ,,,i(k) (which corresponds to the
indices
k = 200, = = =,239) is copied from the 5000-6000 Hz band of U(k). This
approach makes it
possible to retain the original spectrum in this band and avoids introducing
distortions in the
5000-6000 Hz band upon the addition of the HF synthesis with the LF synthesis -
in
1.0 particular the
phase of the signal (implicitly represented in the DCT-IV domain) in this band
is
preserved.
The 6000-8000 Hz band of Um,(k) is here defined by copying the 4000-6000 Hz
band of U(k) since the value of start band is preferentially set at 160.
In a variant of the embodiment, the value of start band will be able to be
made
is adaptive
around the value of 160. The details of the adaptation of the start band value
are
not described here because they go beyond the framework of the invention
without changing
its scope.
For certain wide-band signals (sampled at 16 kHz), the high band (> 6 kHz) may
be
noisy, harmonic or comprise a mixture of noise and harmonics. Furthermore, the
level of
LO harmonicity in
the 6000-8000 Hz band is generally correlated with that of the lower frequency
bands. Thus, the noise generation block 702 performs a noise generation in the
frequency
domain, 141BN(k) for k = 240,===,319 (80 samples) corresponding to a second
frequency
band called high frequency in order to then combine this noise with the
spectrum U ,,(k) in
the block 703.
25 In a
particular embodiment, the noise (in the 6000-8000 Hz band) is generated
pseudo-randomly with a linear congruential generator on 16 bits:
0 k=0,===,239
HBN (k) {
31821UHBN(k -1)+13849 k=240,===,319
with the convention that Uõ,(239) in the current frame corresponds to the
value
HBN (319) of the preceding frame. In variants of the invention, it will be
possible to replace
30 this noise generation by other methods.
The combination block 703 can be produced in different ways. Preferentially,
an
adaptive additive mixing of the following form is considered:
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LI HB2(k) = 13U HBI(k)+ aG Imp HBõ(k) , k = 240,= = -,319
in which G ill3, is a normalization factor serving to equalize the level of
energy between the
two signals,
319
1 U11B1 (02 + e
G k=
HBN = 312940
\L uHBN(02 +E.
k=240
with e =0.01, and the coefficient a (between 0 and 1) is adjusted as a
function of
parameters estimated from the decoded low band and the coefficient fi (between
0 and 1)
depends on a.
In a preferred embodiment, the energy of the noise is computed in three bands:
2000-4000 Hz, 4000-6000 Hz and 6000-8000 Hz, with
3.0 E2, = E U a(k)
keN(80,159)
EN44 = E u12(k)
/cc N(160,239)
EN4-6 = E U(k)
keN(240,319)
in which
239
1 U2(k)
k=160 U(k) k =80,..,159
159
\1 U2 (k)
k=80
U1(k) =< U (k) k =160,...,239
239 ______________________
E u2(k)
k=160
U HB1(k) k=240,...,319
\ 319
I U11/312(k)
k=240
and N(k1,k2) is the set of the indices k for which the coefficient of index k
is classified as
being associated with the noise. This set can, for example be obtained by
detecting the local
l(r(k) k-IU '(k ¨ Oland l[r(k)1..Iti 1(k + 1)1
peaks in U '(k) that verify and by
considering
that these rays are not associated with the noise, i.e. (by applying the
negation of the
preceding condition):
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N(a, b) = ta k bill 1
I 'MI <It I Kk -1)1 or ri(k)I < '(k+1)I}
It can be noted that other methods for computing the energy of the noise are
possible, for
example by taking the median value of the spectrum on the band considered or
by applying a
smoothing to each frequency ray before computing the energy per band.
a is set such that the ratio between the energy of the noise in the 4-6 kHz
and 6-8 kHz
bands is the same as between the 2-4 kHz and 4-6 kHz bands:
P E N 6 - 8
a=
739
\U2(k)-E6_8
k=160
in which
EN4-6 = max(EN4_6, ELp= max(p, EN6_,)
EN2-4
In variants of the invention, the computation of a will be able to be replaced
by other
methods. For example, in a variant, it will be possible to extract (compute)
different
parameters (or "features") characterizing the signal in low band, including a
"tilt" parameter
similar to that computed in the AMR-WB codec, and the factor a will be
estimated as a
function of a linear regression from these different parameters by limiting
its value between 0
and 1. The linear regression will, for example, be able to be estimated in a
supervised
manner by estimating the factor a by exchanging the original high band in a
learning base.
It will be noted that the way in which a is computed does not limit the nature
of the
invention.
In a preferred embodiment, the following is taken
fi = õF__7(2
in order to preserve the energy of the extended signal after mixing.
In a variant, the factors 13 and a will be able to be adapted to take account
of the fact that
a noise injected into a given band of the signal is generally perceived as
stronger than a
harmonic signal with the same energy in the same band. Thus, it will be
possible to modify
the factors ig and a as follows:
fi -,8.f(a)
a a f (a)
3o in which f (a) is a decreasing function of a, for example f (a) = b - sr
c-r , b = 1.1,
a = 1.2, f (a) limited from 0.3 to 1. It must be noted that, after
multiplication by f (a) ,
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a2 +162 <1 so that the energy of the signal U BU
HB2 ( s ,k) = , - HB 1(k) + aG HBNU HBN (k) is
lower than the energy of U õ,(k) (the energy difference depends on a , the
more noise is
added, the more the energy is attenuated).
In other variants of the invention, it will be possible to take:
fl =1-a
which makes it possible to preserve the amplitude level (when the combined
signals are of
the same sign); however, this variant has the disadvantage of resulting in an
overall energy
(at the level of Umn(k)) which is not monotonous as a function of a.
It should therefore be noted here that the block 703 performs the equivalent
of the block 101
of figure 1 to normalize the white noise as a function of an excitation which
is, by contrast
here, in the frequency domain, already extended to the rate of 16 kHz;
furthermore, the
mixing is limited to the 6000-8000 Hz band.
In a simple variant, it is possible to consider an implementation of the block
703, in
which the spectra, U õ,(k) or Gõ,,U,B,(k), are selected (switched) adaptively,
which
amounts to allow only the values 0 or 1 for ce, this approach amounts to
classifying the type
of excitation to be generated in the 6000-8000 Hz band.
The block 704 optionally performs a double operation of application of
bandpass fitter
frequency response and of de-emphasis filtering in the frequency domain.
In a variant of the invention, the de-emphasis filtering will be able to be
performed in
the time domain, after the block 705, even before the block 700; however, in
this case, the
bandpass filtering performed in the block 704 may leave certain low-frequency
components
of very low levels which are amplified by de-emphasis, which can modify, in a
slightly
perceptible manner, the decoded low band. For this reason, it is preferred
here to perform
the de-emphasis in the frequency domain. In the preferred embodiment, the
coefficients of
index k = 0 , = = =,199 are set to zero, so the de-emphasis is limited to the
higher coefficients.
The excitation is first de-emphasized according to the following equation:
õ2.'(k) = {0 k = 0, = = = ,199
U
G deeõ,,,(k)U õ2(k) k = 200, = = = , 255
G ,,õ,p, (255)U,B2(k) k = 256, = = = ,319
in which G deemph(k) is the frequency response of the filter 1/(1-0.68z) over
a restricted
discrete frequency band. By taking into account the discrete (odd) frequencies
of the DCT-IV,
G deemph (k) is defined here as:
Gdeemph(k) = 11 . , k = 0, = = ., 255
I ejek -O.68
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in which
256-80+k+1
2
256
In the case where a transformation other than DCT-IV is used, the definition
of Ok will be
able to be adjusted (for example for even frequencies).
It should be noted that the de-emphasis is applied in two phases for k = 200,=
= = ,255
corresponding to the 5000-6400 Hz frequency band, where the response 1/ (1-
0.68z-1) is
applied as at 12.8 kHz, and for k = 256,¨,319 corresponding to the 6400-8000
Hz
frequency band, where the response is extended from 16 kHz here to a constant
value in the
6.4-8 kHz band.
It can be noted that, in the AMR-WB codec, the HF synthesis is not de-
emphasized.
In the embodiment presented here, the high frequency signal is, on the
contrary, de-
emphasized so as to bring it into a domain consistent with the low frequency
signal (0-
6.4 kHz) which leaves the block 305 of figure 3. This is important for the
estimation and the
subsequent adjustment of the energy of the HF synthesis.
In a variant of the embodiment, in order to reduce the complexity, it will be
possible
to set Griõinph(k) at a constant value independent of k, by taking for example
Gaõõ,,,h(k)= 0.6 which corresponds approximately to the average value of G
,empi,(k) for
k = 200,-- -,319 in the conditions of the embodiment described above.
In another variant of the embodiment of the extension device, the de-emphasis
will
be able to be performed in an equivalent manner in the time domain after
inverse DCT.
In addition to the de-emphasis, a bandpass filtering is applied with two
separate
parts: one, high-pass, fixed, the other, low-pass, adaptive (function of the
bit rate).
This filtering is performed in the frequency domain.
In the preferred embodiment, the low-pass filter partial response is computed
in the
frequency domain as follows:
k
Gip(k) =1-0.999
N1-1
in which No =60 at 6.6 kbit/s, 40 at 8.85 kbit/s, and 20 at the bit rates >
8.85 bit/s.
Then, a bandpass filter is applied in the form:
U k õ,() =1G0 hp(k ¨ 200)Uõ2 VC)
U ,,,,21(k) k = 0,= ==J99
k = 200, = = = , 255
k = 256, = = = ,319 ¨ N ip
Gip(k ¨320 ¨ N ,p)U õ2I(k) k = 320¨N1p, = = = ,319
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The definition of G4(k), k = 0,- = -,55, is given, for example, in table 2
below.
K gl,p(k) K gv(k) K K
0 0.001622428 14 0.114057967 28 0.403990611 42 0.776551214
1 0.004717458 15 0.128865425 29 0430149896 43 0.800503267
2 0.008410494 16 0.144662643 30 0.456722014 44 0.823611104
3 0.012747280 17 0.161445005 31 0.483628433 45 0.845788355
4 0.017772424 18 0.179202219 32 0.510787115 46 0.866951597
5 0.023528982 19 0.197918220 33 0.538112915 47 0.887020781
6 0.030058032 20 0.217571104 34 0.565518011 48 0.905919644
7 0.037398264 21 0.238133114 35 0.592912340 49 0.923576092
8 0.045585564 22 0.259570657 36 0.620204057 50 0.939922577
9 0.054652620 23 0.281844373 37 0.647300005 51 0.954896429
10 0.064628539 24 0.304909235 38 0.674106188 52 0.968440179
11 0.075538482 25 0.328714699 39 0.700528260 53 0.980501849
12 0.087403328 26 0.353204886 40 0.726472003 54 0.991035206
13 0.100239356 27 0.378318805 41 0.751843820 55 1.000000000
Table 2
It will be noted that, in variants of the invention, the values of GAP(k) will
be able to be
5 modified while keeping a progressive attenuation. Similarly, the low-
pass filtering with
variable bandwidth, Gip(k), will be able to be adjusted with values or a
frequency medium
that are different, without changing the principle of this filtering step.
It will also be noted that the bandpass filtering will be able to be adapted
by defining
a single filtering step combining the high-pass and low-pass filtering.
10 In another embodiment, the bandpass filtering will be able to be
performed in an
equivalent manner in the time domain (as in the block 112 of figure 1) with
different fitter
coefficients according to the bit rate, after an inverse DCT step. However, it
will be noted that
it is advantageous to perform this step directly in the frequency domain
because the filtering
is performed in the domain of the LPC excitation and therefore the problems of
circular
convolution and of edge effects are very limited in this domain.
It will also be noted that, in the case of the 23.85 Icbitis bit rate, the de-
emphasis of
the excitation U,,,2(k) is not performed to remain in agreement with the way
in which the
correction gain is computed in the AMR-WB coder and to avoid double
multiplications. In this
case, block 704 performs only the low-pass filtering.
The inverse transform block 705 performs an inverse DCT on 320 samples to find
the
high-frequency excitation sampled at 16 kHz. Its implementation is identical
to the block 700,
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because the DCT-IV is orthonormal, except that the length of the transform is
320 instead of
256, and the following is obtained:
NI6A I (1\7 1 \
UHBO(n) = E U HB300 COS ¨z k +¨ n + ¨
k.=0 N16k \ 2 1' 2 ii
in which N 16k = 320 and k = 0, = = =,319.
This excitation sampled at 16 kHz is then, optionally, scaled by gains defined
per subframe of
80 samples (block 707).
In a preferred embodiment, a gain gHBI(m) is first computed (block 706) per
subframe by
energy ratios of the subframes such that, in each subframe of index m=0, 1, 2
or 3 of the
3.0 current frame:
g111(m) = fle . .. )
e2(m)
in which
63
ei(m).= Eu(n+64m)2+ e
n=0
79
e2(m)= Euõo(n+80m)2+ E
n=0
319
Euueo(n)2+6.
n=0
e,(m)= e1(m)
255
Eu(n)2+ e
n=0
with g = 0.01. The gain per subframe gHBI(m) can be written in the form:
63
1 u(n+64m)2+ e
n=0
255
Eu(n)2 +e
gilai(m)= __________________
i Euõ,(n+80m)2+e
n=0
319
EuHBO (n)2 + e
I n_o
which shows that, in the signal um, the same ratio between energy per subframe
and
energy per frame as in the signal u(n) is assured.
The block 707 performs the scaling of the combined signal according to the
following
equation:
14 HB(n)
= ,, o
HB1(171)uno(n), n= 80m,= = = ,80(m +1)¨i
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It will be noted that the implementation of the block 706 differs from that of
the
block 101 of figure 1, because the energy at the current frame level is taken
into account in
addition to that of the subframe. This makes it possible to have the ratio of
the energy of
each subframe in relation to the energy of the frame. The energy ratios (or
relative energies)
are therefore compared rather than the absolute energies between low band and
high band.
Thus, this scaling step makes it possible to retain, in the high band, the
energy ratio
between the subframe and the frame in the same way as in the low band.
It will be noted here that, in the case of the 23.85 kbit/s bit rate, the
gains g HB1(m)
are computed but applied in the next step, as explained with reference to
figure 4, to avoid
the double multiplications. In this case u õ (n) -=
According to the invention, the block 708 then performs a scale factor
computation
per subframe of the signal (steps E602 to E603 of figure 6), as described
previously with
reference to figure 6 and detailed in figures 4 and 5.
Finally, the corrected excitation u õ'(n) is filtered by the filtering module
710 which
can be performed here by taking as transfer function I/ ;1(z/ y), in which y
=0.9 at 6.6
kbit/s and y= 0.6 at the other bit rates, which limits the order of the filter
to the order 16.
In a variant, this filtering will be able to be performed in the same way as
is described for the
block 111 of figure 1 of the AMR-WB decoder, but the order of the filter
changes to 20 at the
6.6 bit rate, which does not significantly change the quality of the
synthesized signal. In
another variant, it will be possible to perform the LPC synthesis filtering in
the frequency
domain, after having computed the frequency response of the filter implemented
in the block
710.
In a variant embodiment, the step of filtering by a linear prediction filter
710 for the
second frequency band is combined with the application of the optimized scale
factor, which
makes it possible to reduce the processing complexity. Thus, the steps of
filtering 1/ A(z/
and of application of the optimized scale factor g HB2 are combined in a
single step of filtering
g2 Ni(zi 7) to reduce the processing complexity.
In variant embodiments of the invention, the coding of the low band (0-6.4
kHz) will
be able to be replaced by a CELP coder other than that used in AMR-WB, such
as, for
example, the CELP coder in G.718 at 8 kbit/s. With no loss of generality,
other wide-band
coders or coders operating at frequencies above 16 kHz, in which the coding of
the low band
operates with an internal frequency at 12.8 kHz, could be used. Moreover, the
invention can
obviously be adapted to sampling frequencies other than 12.8 kHz, when a low-
frequency
coder operates with a sampling frequency lower than that of the original or
reconstructed
signal. When the low-band decoding does not use linear prediction, there is no
excitation
signal to be extended, in which case it will be possible to perform an LPC
analysis of the
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signal reconstructed in the current frame and an LPC excitation will be
computed so as to be
able to apply the invention.
Finally, in another variant of the invention, the excitation (u(n)) is
resampled, for
example by linear interpolation or cubic "spline", from 12.8 to 16 kHz before
transformation
(for example DCT-IV) of length 320. This variant has the defect of being more
complex,
because the transform (DCT-IV) of the excitation is then computed over a
greater length and
the resannpling is not performed in the transform domain.
Furthermore, in variants of the invention, all the computations necessary for
the
estimation of the gains ( GHBN g HBI(M) g 1182(M) g IffiN "') will be able to
be performed
3.0 in a logarithmic domain.
In variants of the band extension, the excitation in low band u(n) and the LPC
filter
1/ A(z) will be estimated per frame, by LPC analysis of a low-band signal for
which the band
has to be extended. The low-band excitation signal is then extracted by
analysis of the audio
signal.
In a possible embodiment of this variant, the low-band audio signal is
resampled
before the step of extracting the excitation, so that the excitation extracted
from the audio
signal (by linear prediction) is already resampled.
The band extension illustrated in figure 7 is applied in this case to a low
band which
is not decoded but analyzed.
Figure 8 represents an exemplary physical embodiment of a device for
determining
an optimized scale factor 800 according to the invention. The latter can form
an integral part
of an audio frequency signal decoder or of an equipment item receiving audio
frequency
signals, decoded or not.
This type of device comprises a processor PROC cooperating with a memory block
BM
comprising a storage and/or working memory MEM.
Such a device comprises an input module E suitable for receiving an excitation
audio signal
decoded or extracted in a first frequency band called low band (u(n) or U (k))
and the
parameters of a linear prediction synthesis filter ( A(z)). It comprises an
output module S
suitable for transmitting the synthesized and optimized high-frequency signal
(413.(n)) for
example to a filtering module like the block 710 of figure 7 or to a
resampling module like the
module 311 of figure 3.
The memory block can advantageously comprise a computer program comprising
code instructions for implementing the steps of the method for determining an
optimized
scale factor to be applied to an excitation signal or to a filter within the
meaning of the
invention, when these instructions are executed by the processor PROC, and
notably the
steps of determination (E602) of a linear prediction filter, called additional
filter, of lower
order than the linear prediction filter of the first frequency band, the
coefficients of the
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WO 2015/004373 34
PCT/FR2014/051720
additional filter being obtained from parameters decoded or extracted from the
first
frequency band, and of computation (E603) of an optimized scale factor as a
function at least
of the coefficients of the additional filter.
Typically, the description of figure 6 reprises the steps of an algorithm of
such a
computer program. The computer program can also be stored on a memory medium
that can
be read by a reader of the device or that can be downloaded into the memory
space thereof.
The memory MEM stores, generally, all the data necessary for the
implementation of
the method.
In a possible embodiment, the device thus described can also comprise
functions for
application of the optimized scale factor to the extended excitation signal,
of frequency band
extension, of low-band decoding and other processing functions described for
example in
figures 3 and 4 in addition to the optimized scale factor determination
functions according to
the invention.
CA 2917795 2019-07-02
