Note : Les descriptions sont présentées dans la langue officielle dans laquelle elles ont été soumises.
11 ~17757~ `
This invention relates to improvements in frequency
discriminators.
Center frequency tuning oiE prior art resc)nant
circuit discriminatQrs consists of basically two di$ferent
types: mechaniçally and electronically tuned center fre-
quency types. Both types use elther the amplitude or phase
ver~us requenay characteristia~ oiE a single ar dual
re~onant circulk to indicate ~he Erequency oiE khe i-nput
siynal relative to the discriminator center freguency. In
the amplitude comparison mode, a dual mode resonant circuit
is preferred with dual detectors. In the phase comparison
mode, a single resonant circuit is often suiEficient with
dual detectors being used to convert phase to amplitude.
The output of su~h prior art discriminators is very sensi-
lS tive to khe amplltude oiE the input signal, and they re~uirecritical and expensive mean~ to lessen such amplitude
dependence~
The bandwidth of the prior art discriminators is
generally fixed. Therefore, in many applications r compro-
mises must be made between the desired discriminator band-
width linearity and resolution. A~other characteristic oiE
the prior art discriminators is their difficulty in achiev-
ing a wide tuning range of the center fre~uen~y~ Th~ limi-
tations of mechanically tuned circuits are due t:~ the fact
that physical dimensions have to be changed and oiE~en
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multiple resonant circuits must be trackecl to maintain
constant discriminator bandwidth. In the microwave fre-
quency region, this limitation has been overcome to some
extent by substituting ferrimagnetic cavities that can be
electronically tuned over about an oc~:ave bandwidth. Thus,
yittrium-iron-garnet ~YIG) discriminators have been built
in the manner of Nathanson ~United States Patent No.
3,274,519) using amplitude comparison or Goodman et al,
~United States Patent No. 3,364,430), Hoover et al, (United
States Patent No. 3,562,6Sl) and Pircher ~United States
Pat~nt No. 3,622,896) u5ing phase characteri~tics of YIG
reson~tor.
In the speciEic prior ar~ approaches appl~ed to
automatic Erequency control, a single, fixed oenter fre-
quency mechanically tuned cavity has been used to stabilizethe fre~uency of high frequency generators. In one method
of approach for this application, a cyclical mechanical
modulation of the center frequency of the cavity has been
used to sense the position of the generator frequency rela-
kive to the aavity cen-tor frequency and provide a correc-
tion signal to control the gene~ator. The purpo~e oE this
approach was to use the superior mechanical stability o
the cavity to stabilize the frequency of the generator. The
cavity was tuned sinusoidally by mechanical means and the
rate of tuning was limited to slow variations inherent in
mechanical variations of the cavity, The application of the
preC~ent invention in automatic frequency control is to vary
the center frequency of the generators by tracking them to
the variable center frequency of the discriminator. Bandwidth
adjustments on the discriminator can be made to Eacilitate the
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initial capture of the generator and then to maximize the
frequency resolution.
According to the inventionr there is provided a
frequency discriminator comprising ferrimagnetic resonator
5 means adapted to receive an input signal, magnetizing means
for tuning said resonator means to a quiescent center fre-
quency, sweeping means for electronically providin~ a rela--
tiva frequency SWeQping between said resonator means and
said input signal over a predetermined frequency range
10 about the quiescent center fre~uency of said resonator
means, detector means receiving an output from said resonator
means, and mean~ for aomparing ths phase of ~aid sweeping
means to the output of said detector to provide an output
signal .
There is thus provided a frequency discriminator
whose bandwidth and center frequency can be tuned elec-
tronically. The discriminator slope will be insensitive
to input signal amplitude or changes in center frequency,
The linearity of the discriminator will be maintained for
both wide and narrow bandwidths and the center Erequency of
the discriminator can be tuned over multi-octave frequency
ranges, This discriminator will also be able to demodulate
low .level signals by u~ing high gain amplification after
crystal detection,
In various applications of the frequency dis-
criminator of this inven~ion, it can be used to demodulate
frequency modulated voice or data communications, to control
the frequency of a multiple number of o~cillators or the out-
put of a harmonic generator, to overcome the sensitivity and
inability to operate properly in the presence of multiple
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signals of heterQdyne converters used for frequency measure-
ment, and to provide a means which can be incorporated with
other ferrimagnetic components to close-loop track them to a
par~icular input signal.
More particularly, the frequency discriminator in-
cludes an electronically tunable resonant circuit, such as
a YIG rqsonator and a detector connec1ted to the resonator
output. The band or spectrum of frequencies containing the
input signal or signals of interest are linearly swept by
the resonator circuit. The relative position of the detector
output is then compared with the sweep waveform to provide
th~ disariminator output. Since the relative position o~
the detectad output does not change as a ~unction of the in-
put signal level, the discriminator slope is independent of
signal amplitude. The sweep rate of the resonator must pro-
vide at least two samples within one period of the frequency
of the highest modulation component in order to accurately
demodulate the information,
The resonator may be used in either a band-pass or
band-reject mode and several advantages accrue for each in
particular applications, In the case where a ferrimagnetic
resonator is used as the resonant element, the center fre-
quency of the discriminator can be readily changed by con-
trolling the current through an electromagnet~ The sweeping
signal can then be superimposed on the center frequency
tuning current to generate a sweep of the resonator, Prefer-
ably, however, an auxiliary air core inductor is used to
superimpose a variable magnetic field across the resonator~
By using an air core inductor, the resonator can be swept
: 30 . without magnetic hysteresis or saturation and at much faster
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rates than would be possible through the electromagnet.
Since the ferrimagnetic resonator tunes linearly with
maynetic field, a very high degree of discriminator linear-
ity is achievahle by using a linear current driving source,
The bandwidth of the discriminator can be controlled very
accurately from a few MHz to several hundred MHz by chang-
ing the magnitude of the current drive. In most of the pre-
ferred embodiments disclosed herein, -the sweeping filter
alternative is used; however, it is to be understood that
with suitable modifications within the ordinary ski.ll in
the art the ~ilter can be fixed and the requency of the
~ignal aan be linearlv swept ln like ashion creati.ng out~
puts e~uall~ suitable in several embodiments~
In order that the lnvention may be more Eully
understood, it will now be described with reference to the
accompanying dra~ings, in which:
Figure 1 is a cross-sectional view of an exemplary
YIG resonator assembly including a magnetic housing, YIG
sphere and electromagnetic and air tuning coils for use in a
fre~uency discriminator embodying the present invention;
Figure 2 is a block diayram oE a frequency dis-
criminator embodying the present invention;
Figure 3 is a graphic presentation of a cyclic
current waveform suitable for tuning the air coil oE Figure l;
Figures 4A - 4G relate to Figure 3 and are graphic
presentation of a series of waveforms illustrating typical
outputs f~om the YIG resonator of Figures 1 and 2 when the
resonator is operated in a ~and-reject mode as the air coil
tuning current tunes the resonator frequency f:rom FL to FH;
Figures 5A - 5D are a series o~ wave:Eorms illustrat-
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ing that tho position of the peaks as in Figures 4A - 4G are
independent of input signal level;
Figures 6A and 6B are block diagrams of phase com-
parison methods useful in this invention when the RF detector
output is used to generate a trigger at its peak response;
Figures 7A and 7B are block diagrams of phase com-
parison methods useful in this invention when the analog out~
put of the RF detector is processed;
Figure 8 is a bl~ck diagram of a frequency control
embodiment of the frequency discriminator according to the
present invention;
Figure 9 i~ a graphical presentation of waveforms
useEul in understanding the embodiment o~ Figure 8~
Figure 10 is a block dlagram of a heterodyne
frequency counter embodiment of the frequency discriminator
according to the present invention7
Figure 11 is a partially schematic, block diagram
of a frequency modulation receiver embodiment of the frequency
discriminator according to the present invention;
Figure 12 is a partially schematic block diagram
of a Erequency trackin~ device embodiment of the frequency
discriminator according to the present invention; and
Figure 13 is a block diagram of broadband oscil-
lator using the frequency discriminator according to the
present invention to control the low frequency oscillator
driver of a broadband harmonic generator.
Referring first to one basic preferred embodiment
of the frequency discriminator according to the present in-
vention as shown in Figures 1 and 2 r wherein Flgure 1 shows
in cross-section a YIG filter 10 which is usable in the block
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diagram of Figure 2. The YIG filter 10 includes a single
resona~or YIG sphere 12 whose center frequency is tuned by
means of an electromagnet tuning coil 14 (shown in cross-
section) as a function of the electromagnet coil 14 tuning
current, and a small air core coil 16 which surrounds the
YIG sphere 12 in the electromagnet gap 18. As explained
further hereinafter, the air core coiL 16 is driven by a
periodic waveform that displaces the resonant frequency of
the YIG filter cavity by an amount on either side of the
quiescent frequency proportional to the current in the air
core coil set by bandwidth control 19. The advantages of
an air core coil (although the same effect is possible by
scanning in a periodic wave~orm into the main tuning part of
the electromagnet) ar~: the offset is zero, i~e., i~ there
is no current in the coil 16 there is no effect on the YIG
sphere 12 resonance, the time constant is considerably
smaller because the inductance of air core coil 16 is less
and an air core coil has no magnetic time cons~ant~ con-
sequently, the resonance can be tuned faster; and an air
coil core does not exhibit magnetic hysteresis or saturation,
thereby ensuring a linearity and slope independent of center
~requency tuning.
The center frequency tuning source 20 applied to
the electromagnet tuning coil 14 on lines 22 and 24 sets the
center frequency operation for ~he YIG filter 10. The center
frequency tuning source 20 provides a current for driving the
~- electromagnet tuning coil 14. The current could be made pro-
portional to frequency so that a fixed level input $ignal
sets the center frequency of the discriminator to a fixed
frequency or, alternately, the control current could be con-
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tinually swept so that the center freguency of the dis-
criminator is continually tuned. At the same time that the
center frequency tuning sourae 20 is operating, the air
core coil 16 is being tuned by an air coil tuning source 25
with a cyclic current as shown in Figure 3 and applied to
lines 15 and 17.
The shape and generation of the driving waveforms
can be of a variety of forms, They, in fact, can be shaped
to provide a particularly deRirable ~ran~fer unction for
the discriminator. I~, as is commonly required, the desired
discriminator curve i~ linear, then the resonator tuning
~hould be linear.
~he amplitude of the tuning current ~o khe air core
coil 16 whlch ~ set by the bandwidth control 19, determines
th~ extent that the YIG re~onance is tuned of its quie9cent
(or Fo) value. Therefore, by varying the amplitude of the
drive current in 19, the width of ~he resonance sweep can be
readily adjusted. As explained further below, this i~ equiv-
alent to varying the bandwidth of the discriminator. Typical
variations of bandwidth could be -~400 ~lHZ to as low as ~ a
~ew M~z,
To this point the drive currents ~or the YIG sphere
resonator 12 have been de~cribed. The drive is made up of
two part~: (1) the bias or center frequency tuning to the
electromagnet tuning coil 14 which controls ~hP quiescent or
Fo value, (2) the cyclical delta F (~F) tuning ~rom FL to FH
the amplitude of which can be varied to var~ the bandwidth
of the YIG sphere resonator 12 and there~ore the bandwidth
o~ the discriminator~
Alternately, the bias tuning could be provided by a
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permanent magnet, a mechanical variation of the magnetic gap
or any combination of electrical and permanent magnet bias-
ing. The cyclical magnetic field can be introduced at rates
from DC up to several MHz.
As the YIG sphere resonator 12 is tuned across the
~F range with a controlled sweep waveform, the position in
the sweep where the resonator is tunecl to a particular fre-
quency must be sensed. In order to accomplish this~ the YIG
sphere resonator 12 is coupled to a transmi~sion line 21 in
such a way tha~ ~he amplitude of a signal on this line within
the ~F range is amplitude modulated by the resonant YIG
~phere resonator 12. The YIG sphere resonator can be aoupled
in~o the tran~mi3slon line in either a band-pass or a band-
re~ea~ mode. In th~ band-pass mode the RF detector 28 ls
~ coupled to the output of the transmission line 27 and as the
-~, tuning of the resonator 12 approaches the ~? signal frequency
the amplitude of the RF detector increases. In the band-
reject mode, shown in Figure l, the RF detector 28 is coupled
to the transmission line 27 and as the tuning of the resonator
12 approaches the RF signal frequency the amplitude of the
RF detector 28 decreases. The choice o coupling will be
dotermined by the partiaular applicationJ the prinaiple o
operation is the same~
. .
Figure 4 shows waveforms illustrating typical out-
puts when the YIG sphere resonator 12 is arranged in a band-
reject mode. The outputs indicated occur as the input fre-
quency is tuned from FL to FH.
In Figure 4A the signal frequency is just ~arely
below the frequency FL~ Therefore only a small portion of
the reqonance amplitude change is detected. In ]?igure 4B
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the signal frequency is at the FL frequency and the ampli-
tude response reaches a peak value. In Figure 4C the signal
frequency falls within the ~F range and therefore it is detec-
ted twice; once on the positive slope and once on the nega-
S tive slope. In Figure 4D the signal frequency is at the
center of the sweep, ~0. The resonance peak is e~ually
spaced on both the negative and positi.ve slopes from the
sweep extremes.
Figures 4E, 4F and 4G repeat the same sequence
except that they recur inverse to Figures 4A, 4B and 4C.
Therefore, if only one peak occurs during the
triangular sweep, the signal is at either FL or FH. Exactly
where is readil~ indicated ~y determininy the time in the
~weep they occur. It i~ important to note that all of the
necessary phase information is located on either the posi-
tive or negative slope. The second slope contains redundant
information, Since the YIG resonance is directly proportional
to the changing magnetic field of the air core coil caused by
the tuning current, if the current is linear the resonance
point moves linearly over the entire ~F range. Therefore,
the exaat position of the resonance gives an exact measure
of the o~f~et of the signal frequency rom the quiescent
frequency tFo) when the air core coil tuning coil current is
zero.
Referring again to the block diagram of Figure 2,
the RF detector output is amplified in video amplifier 29 and
coupled to an amplitude detector circuit 30 which processes
ths peaks or dips in t~e ~F detector 28 output amplitude.
The amplitude detector circuit 30 output is applied to a phase
comparator 32 which also receives an input from the air coil
~077S7~
tuning current source 25. The output of phase comparator
32 is the frequency discriminator output. Preferred embodi-
ments for processing the RF detector amplitude and comparing
its phase to the air coil tuning sweep to generate the con-
ventional discriminator characteristic are described herein-
after.
Figure 5A indicates that the position of the peak
i5 independent of the signal level~ As the amplitude of FS
increases, the output signal amplitude from RF detector 28
1~ also changes as shown in Figures 5B, 5C and 5D; however, the
phase or the relative timiny of the peak of the resonance to
the ~weep ~aveEorm is unchanged.
The bandwidth of the frequency discrirninator em-
bodying the present invention can be reaclily varied from a
few MHz to several hundred MHz~ This is accomplished by
varying the amplitude of current driving the air core coil
16 of Figure 1. The lower limit is set by the loaded Q of
the resonator 12: as the scan is decreased the peak of
resonance is more difficult to sense. The upper limit is
set by the maximum amount o current that can be forced
~hrough air core coil 16 until there is damage to the coil
or ~IG sphere resonator 12 due to thermal e~fects.
The frequency discriminator embodying this inven-
tion can, therefore, provide the widest possible bandwidth
2S for capturing or measuring a signal frequency and yet also
provide the narrowest bandwidth for maximum resolution and
highest tuning slope. Both wide and narrow band operation
retain the excellent linearity charac~eristics and insensi-
tivity to signal amplitude variations~
Under one principle embodimen~, operat:ion depends
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on a tunable resonant cavity which is not limi~ed to a YIG
cavity, and whether a band-reject or band-pass cavity is
used depends on the application -- whether a single pole or
multiple pole filter is used also depends on ~he application.
The electronic control of the sweep of the air
coil tuning source 25 and the capabili~y to adjust the band- ;
width of the discriminator in bandwidth control 19 of
Figure 2 make possible several preferred embodiments for
generating the characteristic output voltage versus input
frequency curve of the discriminator, Figure 6 illustrates
two methods of comparing the RF detector output to the phase
of the sweeping mean~, Both depend on generati.ng a trigger
pulse at the peak ~or fixed khreshold level) of the detected
output signal illus~rated in Fiyures 4B through 4F, These
methods are preferred for those applications in which the
discriminator bandwidth is very wide with respect to the
bandwidth of the resonator and/or accurate frequency informa-
tion is required in a single sweep,
In Figure 6A the output of the detector is coupled
through a logarithmic video amplifier 34 whose output is
proportional to the input power, When a band-reject resonator,
as illustrated in Fiyure 1, is used to yenerate the detector
output, the rejected power is independent of the absolute in-
put power and the resultiny voltage waveform from the log
~5 video amplifier 34 is normalized, This simplifies the
design of peak detector 36 which generates a standard output
triyger at the peak (or fixed threshold level) of the detected
waveform. At the same time, a sawtooth or triangular wave-
form, similar to that shown in Fiy~ 3, is used to drive the
resonator cyclically ~etween FL and F~ in a linear fashion.
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The position of the trigger with respect to the input voltage
applied to the sweeping means can then be calibrated to read
out the frequency directly relative to the center frequency
of the discriminator~
In Fig~ 6A the calibrated sweeping means consists
of a digital clock 38 driving a reversible digital counter
40 which counts sequentially in either increasing or decreas-
ing counts as controlled by a counter control 42. The
: counter then programs a standard digital to analog converter
44 to generate an output voltage linearly proportioned to
the nurnber of clock pulse~ counted~ The output of the D/A
converter 4~ is then Eed to khe alr coil drlver 46 sweeping
the resonator llnear~y across the predetermined bandwidth~
Bandwidth control 48 is a scaling control which adjusts the
current drive into the air coil to correspond to the desired
discriminator bandwidth~ The input frequency to the dis-
criminator is then determined by gating out the digital count
in counter 40 by readout gate 50 at the time of the inter-
cept trigger generated in trigger generator 52~ This pro-
vides an immediate digital indication of frequency relatlve
to discriminator center frequency~ Alternatively, the cali-
brated analog output of the bandwidth control 48 could be
gated through readout gate 54 to generate an analog measure
of relative frequency. Obviously, analog outputs could also
be provided with simpler voltage generated sweeping means
than illustrated; the prime requirement, however, is that a
linear relationship exists between the sweeping means and
the position o~ the resonator in order to obtain a linear
discriminator curve.
The digital or analog measures of input frequency
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relative to the discriminator center fre~uency represent a
sampled demodulation of the input signal a~ rates equal to
sweep rate of the particular sweeping means~ Standard pro-
cedures for processing sampled input signals can be used to
recover the input modulation.
,, Figure 6B illustrates an alternate embodiment for
comparing the phase of the RF detector output to the sweep-
ing means to generate the standard discriminator curve.
Again the trigger generating means of Fig. 6A is used to
detect the peak output of the resonator. The sweeping wave
form is generated by dividing b~ 2 the frequency of clock
generator S5 in divider 56. The output pulse Erom divider
56 is integrated in ramp generator 58 to generate a linear
triangular waveform similar to that shown in Fig. 3, The
voltage output of this waveform is scaled in bandwidth con-
trol 60 to set the desired discriminator bandwidth~ The
output of control 60 is used to drive air coil driver 62 to
generate a linear resonator sweep,
The trigger pulse from trigger generator 52 in
Figure 6A is used to control a bistable multivibrator 66
that is referenced to divider 56. This phase comparison
method is similar to that described in the prior art b~
C.E. Arnold et al. (United States Patent No~ 2,764,682)~ The
output of multivibrator 66 is integra~,ed in integrator 68 to
provide an analog voltage proportional to the position of the
trigger pulse (the detected output)relative to the driving
waveform of the resonator, The output therefore measures the
relative phase between the detected input signal and the sweep-
ing waveform. If the sweeping waveform is linear over the
predetermined bandwidth, the analog output will be a linear
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function of frequency across the entire bandwidth of the
discriminator.
In those applications where ~he bandwidth o the
,; discriminator is required to be relatively narrow with
,' S respect to the bandwidth of the resonator, or where spurious
' magnetostatic mo~es on the resonator could cause false
triggeLs in the digital processing techniques, analog pro-
ceSsing to determine relative position of the output with
respect,to the sweeping waveform is possible. Two such
approaches are shown in Fig. 7. Figure 7A is similar to
the prior art approaches associated with mechanical scanning
of the resonators except that the sweepiny waveorm is a
linear trian~ular waveform and an attempt is made to normal-
ize the logarithmic amplifier 70 at its output. 'rhe analog
phase detector comprises an analog mul~iplier 72, which
multiplies the output from the logarithmic amplifier 70 by
the output of triangular waveform generator 71 (which also
drives the bandwidth control 73 and air coil driver 75),
and an integrator 74, The relative position or phase of
the dete~ted output and t:he sweeping waveform is measured
b~ the voltage output of integrator 74. The limitation5 of
this approach are the slope dependence of the overall dis-
criminator curve on the shape and output voltage level of the
logarithmic amplifier.
,'; ~ In the present invention, one unique embodiment
of the means for comparing the phase of the resonator sweep-
ing waveform to the output of the RF detector uses a saw-
tooth waveform shown in Figure 7B to drive the resonator,
In this case the analog outputs from the detec:tor are all
in proper phase so that a narrow band filter can be used
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to extract the fundamental frequency from the waveform~ It
can readily be shown that the phase of the fundamental com-
ponent of the waveform is a linear function of the position
of the detector outpu~ relative to the sweeping wave~orm,
Figure 7B is a block diagram illustrating operation
of this uni~ue phase comparison embodiment. The output of
clock generator 76 is divided by 2 in digital divider 78.
The divider output triggers sawtooth g~enerator 80 putting
out the voltage waveform 82 shown. The sawtooth retrace is
about 10 per cent of the total period. The sawtooth voltage
is scaled in bandwidth control 84 and then applied to air
coil driver 86 to sweep the resonator in one direction only~
The RF detector 28 output is ampli~ied in logarithmic
ampli~ie~ 88 and khe fundamental component of the detecked
output is ~iltered in narrow bandwidth filter 90 tuned to
the reciprocal I/T of the sweep time T. The output o~ this
filter is applied to limiter 92 to eliminate amplitude varia-
tions in the input, and the limited signal is applied to
phase detector 94 and compared with the reference phase ~rom
divider 78. Phase detector 94 provides an output voltage
that i~ linearly proportional to the relative position of
the detector output with the sweeping sawtooth waveorm,
thereby generating the desired discriminator curve,
Some significant applications for the present
2~ discriminator include:
1. Frequency control of signal sources;
2. Frequency measurement of signal sources;
3. Demodulation of FM signals~ and
4. Tracking control for other YIG devices.
Each of these will be described along with exemplary
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preerred embodiments of the invention.
Freque~y Control of S~nal Sources
Solid state signal sources are available commercial-
ly to cover the frequency range from 500 MHz through 18,000
MHz in a single unit. These sources generally use fundamental
oscillators, which are electronically tuned over octave ranges
by either varactor or YIG devices. Each octave range oscil-
lator then requires individual calibration and control to
provide the necessary frequency stability and accuracy. ~he
net result is a system that is extremely expensive,.
The application of the wide bandwidth tunable dis-
criminator in source control is to provide a single device
which fixes the accuraay, stability, linearity and reset-
ability of frequency, reduces frequency pushincJ and pulling;
eliminates turn-on drift; and provides continuous tuning at
band cross-over frequencies.
Referring now to Figures 8 and 9, a frequency con-
trol embodiment including the wide bandwidth frequency dis-
criminator of Fig~ 2 is shown. Assume that the desired opera-
tion consists of contiguous scan of RF oscillators 98, 100
and 102, and the RF coupler 103 sample~ the oscillator output
to provide input ~or discriminator 108.
Initially, the center frequency control 104 is
tuned to the lowest frequency of oscillator 98 and a coarse
frequency tune is applied to oscillator 98 vla summing
ampliier 106. The wide bandwidth discriminator 108 is
centered at the correct Fo by tuning unit 104 (see Figure 9),
Next the wide bandwidth position (~F max~ of
~igure 9~ is switched into operation ~y bandwidth adjust con-
trol 110 and the bias voltage B~ is applied to oscillator 98
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by bias control 99, In the ~F max~ position, the oscillator
is quickly tuned near Fo by the closed loop feedback control
network 101. If necessary, the gain of the loop can then be
increased by switching in the ~F min. bandwidth on the dis-
criminator. Now as Fo is tuned by cen~er frequency control
104, the oscillator will be forced to track Fo by the feed-
back control action of the discriminator 108.
When the Fo control reaches the highest frequency
in c)scillator 98, then oscillator 98 is turned off and
oscillator 100 is turned on. The discriminator bandwidth
i5 again switched to aF max. to capture oscillator 100,
whlch is also forced to track the Fo of the discriminator.
Note that the Fo tuning is continuous and there is only a
transient as one oscillator i9 turned of while khe other
is turned on. The discriminator frequency reerence can be
stopped momentarily at this time to allow any transients to
die down before continuing the scan~
It is not necessar~ to turn the oscillator off and
on i there is sufficient isolation in the RF switch 112 to
meet the spuriouq signal requirements. However, high isola-
tion fast RF qwitches are expensive and this offers an
economical approach with significantly improved isolation
over the techniques that are commercially available,
Control of oscillator 102 is passed on from oscillator 100
in much the same way as from oscillator 98 to 100. At the
end of the tuning range of oscillator 102 the sequence can
be repeated. The sweep rate is limited by the scanning
capability of the discriminator 108.
Frequency Measurement of Signal Sources
Another useful capability of the disc:riminator is
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: . . .
~1~7757~
to augment the operation of an automatic heterodyne counter
by determining the exact frequency offset of ~he unknown in-
put frequency relative to a harmonic comb line. In this
manner, the IF bandwidth of the amplifier following the RF
converqion process can be reduced, This signiicantly avoids
two of the principle limitations of prior art heterodyne
counters: the susceptibility to interference in the presence
of multiplo signals and the difficulty in measuring low level
signals due to the wide noise bandwidths of the IF amplifiers
currently in use. It also allows for the u~e of harmonic
comb line spacing much wider than is currently possible, e.g.,
500 MHz versus 100 to 200 MHz, A wider comb line spaaing
inareAses ~he ~F comb power available rom the harmonic
generator at requencies above 12 GHz and provide~ better
i~olation between adjacent com~ line. This further improves
sensitivity and extends frequency measurement to much higher
requencies (e.g., to 40 GHz). At the same time, the
presence o an input signal and the selection of the proper
harmonic for converting it to IF can be provided by the dis-
criminator directly rom the RF input signal rather than a~ter~F proce~sing as i9 currently re~uired. A block diagram o an
embodiment of the invention for improving automatic heterodyne
counting is ~hown in Fig. 10.
The RF input signal at terminal 112 is applied
through a 3db at~enuator 113 to a band-reject type YIG filter
114 which is included in the same magnetic housing as YIG
tuned harmonic generator 116. Harmonic generator 116 is
driven from a ixed IF oscillator 118 which is phase locked
to the ba~ic digital counter 120 as is common in the prior
art. Power control }22 provides a means to turn oEf or reduce
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1C97~579
the input drive to the harmonic generator. Digital-to-analog
converter 128 in response to drlve control 124 provides the
tuning control for electromagnet 126.
Initially, the IF oscillator 118 input drive to
the harmonic generator 116 is turned off and D/A converter
128 continually step tunes YIG filter 114 over the desired
operating range ~e.g., 018 to 26 GHz) in steps coarsely equal
to the comb line spacing (equal to the frequency of IF
oscillator 118). In this mode (Mode A), YIG ilter 114
operates in conjunction with RF detector 127, video amplifier
130, phase comparison means 132 and sweeping means 134 to
form the discriminator of Figure 2. Sweeping mean~ 134
cyclically scans YIG filter 11~ over a ranye equal to or
greAter than the comb line spacincJ. When the dlsarlminator
intercepts an inpu~ signal, an output similar to Figure 3
is generated. The novel ability of this discriminator to
select the largest signal will eliminate false-locking pro-
blems.
When a signal is detected in the discriminator,
D/A drive control 124 stops the scanning of electromagnet
126. A frequency measurement with the discriminator of the
offset of the signal with respect to the nominal comb line
frequency is used to select the best comb line for hetero-
dyne conversion. The comb line number is preset in digital
counter 120.
At this point, power control 122 turns on the IF
; drive to YIG tuned harmonic generator 115. Note, however,
that electromagnet 126 has not necessarily tuned the YIG
harmonic generator exactly on to the desired comb line.
ThiS is accomplished ~y using a combination of the
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~77S~
YIG tuned harmonic generator 116 (now in a band-pass embodi-
menk), RF detector 127, video amplifier 130, phase comparison
means 132 and sweeping means 134 to form the discriminator
of Figure 2. In this mode ~Mode B), RF switch 136 is opened
to isolate the input signal from RF detector 127 and RF
coupler 138 applies the RF output from ~he harmonic generator
116 to RF detector 127. The output from pha~e compari~on
means 132 is coupled back to D/A drive control 124 through
feedback network 137. This fine control on the electro-
magnet current centers the quiescent tuning of the housingat the exact comb line ~requency.
Therefore, the harmonic comb line that: will hetero~
dyne the lnput signal frequency lnto digital counter 120 has
been selected and its output power optimized, At this time,
operation reverts back to Mode A where the input signal off-
set is now measured by the discriminator with respect to the
updated tuning of the housing (now centered on the comb line
frequency). Filter/ampliier natwork 140 is then tuned tto
the offset frequency measured in network 141 by the discrim-- !
inator. The bandwidth of network 140 can be considerably
less than the IF comb spacing thereby signi~icantly improving
the capability of the automatic heterodyne counter to measure
low level input signals and to operate in the presence o
multiple inputs. The IF oscillator 118 providing an input
drive to harmonic generator 116 is then turned on, and RF
detector 127 heterodynes the input signal and the selected
comb line in the conventional prior art manner. Periodically,
prefera~ly during the reset period of digital counter 120,
the discriminator reverts into Mode A assuring that: network
0 140 is tuned to the proper IF offset and that the YIG tuned
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16)775'79
harmonic generator 116 is tuned to the optimum comb line~
Demodulation of FM~ nals
This technique provides a discriminator that makes
possible practical low cost FM communications systems at
microwave frequencies.
Referring to Figure 11 whereln an exemplary embodi-
ment of an FM receiver having a frequency discriminator
according to the invention is shown, the FM mo~ulated input
signal is inaident upon a microwave antenna system 142. The
signal is routed directly to RF detector 144 through tran~-
mission line 146. The air coil drive 148 is set for maximum
~F to ensure capturing the signal, The offset frequency of
Fo relative to the permanent magnet 149 bia~ field is measured
in phase detector 150 and a DC current to air coil drlve 1~8
lS aenters the di~criminator (provides automatic frequency con-
trol). Next, the ~F range is optimized to match the input
signal deviation. Video amplifier 152 amplifies the signal
generated as YIG sphere resonator 154 sweeps through the RF
input signal (either band-pass or band-reject can be used)
by means of air core coil 155. Phase detector 150 basically
compares the RF detector 144 output position to the sweep
waveform creating a conventional FM discriminator curve using
methods such as depicted in Figures 6 and 7. This output is
filtered in the RC circuit 1$6 to remove the sample rate from
the discriminator and to pass the modulation rate. For ex-
ample, for voice communication, t~e sample rate should be
twice the maximum modulation rate to recreate the original
information. Audio amplifier 158 drives speaker 160.
Trackin~ Control for Other Devices
Because the wide bandwidth discriminator is so simple,
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75'7~
' :
so small and requires so little signal power, it is possible
to include it under the same magnetic pole piece as other YIG
tuned components to provide an output tha~ tracks a filter to
a given input signal,
For example, suppose it is necessary to track a
multipole YIG filter to a fundamental fre~uency of a signal
source while rejecting harmonics or spurious signals~ Cur-
rentiy, such tracking can only be done in an open l.oop which
makes it guite impractical over wide frequency ranges, parti-
cularly when narrow band filters are required, or when opera-
tion is desired over environmental extremes. Generally, in
these ca~es, it i~ not possible to monitor the output of the
filter to peak the desired response because this would ampli-
tude modulate the output. Also, because -the filter is scanning
at a ast rate, such ~uning could not be accomplished through
the main tuning coil quickly enough to establish an effective
tracking signal.
The embodiment shown in Fig.12 can be used to
. achieve such results. A small portion (>-30dbm) o~ the in-
put signal at Fo is coupled from a standard YIG filter 171
and through the band-pass/band-reject YIG filter 172 of the
wide bandwidth discriminator 174. (See Figure 2.) Using
techniques described previously, an output voltage corre-
sponding to the difference between Fo and the quiescent
operating frequency of the YIG housing is fed back into
center frequency summing amplifier 176 through feedback net-
work 178. Thiq feedback voltage is phased so that the total
tuning voltage corresponds to a quiescent tuning current
through electromagnet 180 that will center standard YIG
filter 171 at Fo~ Since the air coil tuning 182 is sampling
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~77579
at a much higher rate than the sweep rate tuning of Fo the
feedback can be very effective in tracking the electromagnet
tuning to the input signal Fo~ The only requirement is tha~
the ~IG filter 172 sphere in the discriminator 174 be under
the same coarse tuning field as the YIG filter 171 sphere
and that both spheres be aligned in frequency (basically the
anisotropic fields are on the zero temperature drift axis).
It is also possible to couple directly off a filter 171
sphere to provide the resonator for the discriminator,
Since the discriminator YIG filter 172 can be as
~mall or smaller than O.Q10 inches, such traaking i9 readily
achieved. It is also possible that the filter aan be center-
ed at a ~recluen¢y offse~ ~rom Fo by independently of~settiny
the discrlminator center fre~uency. This can be accomplished
in several ways including: a DC bias on the air core coil,
varying the reference phase of the feedback loop, varying the
spherc position or offsetting the resonance by using the
anisotropic field of the YIG~ A particularly novel way
would be to couple to one of the well known magnetostatic
tracking resonances of ferrimagnetic materials to form the
r~onator. These 0purious modes occur at a ~ixed offset from
the main mode, are in~ependent of Erequency, and the ofset
can be controlled over a considerable range by varying the
saturation magnetization of the material. Such offsets are
particularly useful in eliminating unwanted mixing products
and feedthrough in high level heterodyne mixers and tracking
local oscillators and preselectors.
In the tracking applica~ion of the wide ~andwidth
discriminator, an alternative method in certain c:ases could
0 consist of cyclical sweeping of the input signal rather than
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~LCP7757~
the resonant structure. The detected output and phase
relationships with the sweeping means are identical to those
generated by sweeping the resonant structure. A feedback
signal generated in the phase comparator could be applied
to either the signal source or the resonant device causing
the two elements to track one another.
Another tracking applicatio:n of the discriminator
is in a broadband harmonic generator where the output from
the harmonic generator is used to control the frequency of
the input source. Prior art harmonic generators use open-
loop tracking between input source and output filter, are
difficult to align and track, and have discontinuou~ output
tuning o~ ~requency~ By u~ing the wide center frequency
tuning range o~ the discriminator, the desired output ~re-
quency can be tuned continuously while succes~ively filter-
ing out the harmonics of the driving source In this manner,
an inexpensive 1 to 12.4 GHz or 2 to 18 GHz signal source
can be built with excellent linearity and frequency accuracy
using only a single fundamental oscillator. This invention
is the key element in providing closed-loop control of fre-
quency as shown in Fig 13.
A center frequ~ncy tuning control unit 184 provides
s the means for tuning the discriminator 186 over the entire
, output range. At the same time, a high power input RF source
1~8 is coarsely tuned by harmonic tuning network 190 to a
frequency equal to the output frequency divided by the
harmonic number. A sample of the output is obtained in
. - coupler 191 and an error correction signal generated in dis-
criminator 186 is fed back through network 192 and summed
with the coarse control in amplifier 194 to provide fine
-26
~'
.,
:.
775~9
frequency control of the input source 188. Thus, the out-
put frequency scan appears to be con~inueus even while the
input source is being switched to corr,espond to the proper
drive frequency for each harmonic. If the input source is
a varactor oscillator, the discontinuity in the output can
be as little as a fraction of a millisecond~
The output of step-recovery diode 196 consists of
all of the harmonics of the input source 188~ The desired
output frequency (NFo~ is filtered by YIG filter 198 to
eliminate unwanted harmonics. Since the output frequency
reerence (discriminator 186) is also a YIG device operating
at the same frequency it is easy to track them versus fre-
quen¢y. Al~ernatively, the YIG re~onator incoxporated in
discriminator 186 can be tuned within the same magnetic
tuning structure as YIG filter 198 thereby closiny the loop
around the filter as well as the input source 188.
If the YIG resonator of discriminator 186 is
placed before YIG filter 198, then a band-pass approach is
needed to select the desired harmonic in the presence of
other higher level harmonics. If the YIG resonator used
for the discriminator is placed a~ter the output filter, as
shown in Figure 13, then a band-reject approach can be used.
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