Note : Les descriptions sont présentées dans la langue officielle dans laquelle elles ont été soumises.
~ RCA 71lo24/7l/0~4A
The present invention is directed to the field of
electronic tuning systems for television receivers and is
particularly directed to the field of phase locked loop
tuning systems for television receivers.
Phase locked loops are desirably employed in many
applications because they are capable of synthesizing a
signal having a relatively accurate and stable frequency
which may readily be controlled. Typically, phase locked
loops include a source of a reference frequency signal, a
controlled oscillator, a programmable counter to divide the
frequency of the controlled oscillator output by a controll-
able factor, a phase detector to derive a signal representing
the phase and frequency relationship between the reference -
frèquency signal and the output signal of the controlled
oscillator signal and a low pass filter to derive a D.C.
control signal for the controlled oscillator from the output
signal of the phase detector. Examples oE such phase
locked loops and their applications are described in RCA
Digital Integrated Circuits, Application Note ICAN 6101
entitled, "The RCA COS/MOS Phase-Locked-Loop -- A Versatile
Building Block for Micro-Power Digital and Analog
Applications", which may, for example, be found in the RCA
1974 Databook SSD-2()3B (COS/MOS Digital Integrated Circuits).
Because of the programming, accuracy and stability
performance advantages of phase locked loops, they have
recently been suggested for use in radio and television
tuning systems. For example, phase locked loop types of
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RCA 71,024/71,024A
~5~
1 television tuning systems are described in Canadian Applica-
tion No. 265,196 filed 9 Nover,lber 1976 in -the name of J.G.N.
Henderson; United States Patent No. 4,078,212 issued 7 March
197~ in the name of R.M. Rast; and United States Patent No.
4,031,549 issued 21 June 1977 in the name of R~l. Rast et al.
Care must be taken in selecting the co~ponents of
a phase locked loop tuning system so as to insure its
compatibility with the receiver in which it is utilized.
For example, phase locked loop tuning systems generally
utilize a phase detector of the type, such as is described
for example, in the above-identified RCA Application Note,
which generates pulses at the frequency of the reference
frequency signal whose durations represent the phase and
frequency relationship between the reference frequency
signal and the local oscillator signal. Unfortunately,
because the low pass filter utilized to derive the control
signal for the controlled oscillator from the output signal
of the phase detector may not sufficiently filter the output
signal of the phase detectort pulses at the reference
frequency may undesirably amplitude modulate the control
signal. As a result, the local oscillator signal
generated in response to the ampli.tude of the control signal
may include a frequency modulated component which, when
processed by the receiver, may give rise to undesirable
interference signals in the receiver's audio or video output
signals.
For examplet frequency modulated components of
local oscillator signals generated by a phase locked loop
tuning system of a frequency modulation (FM) radio receiver
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~5~ RC~ 7I~024/71,02~A
may be demodulated by the receiver's demodulator -to produce
audible interference signals annoying to a listener. To
reduce the generation o~ such audible inter~erence signals
in FM radio receivers utilizing a phase locked loop type of
S tuning system, the reference frequency of the pha~e locked
loop may be selected to be higher than the highest frequency
in the audio frequency range. In this manner, undesirable
components at the reference frequency generated by the phase
locked loop tuning system will be outside of the audio
frequency range and therefore not be heard by the average
listener.
However, a similar technique, whereby the reference
frequency of a phaselocked loop tuning system for a
television receiver is selected to be higher than the
frequency of the highest frequency signal utilized to produce
an image to reduce the visible effects of frequency modulated
components generated by the phase locked loop tuning system
cannot be readily implemented since it would require the use
of a reference frequency considerably higher (e.g., greater
than 4.5 MHz) than that capable of being readily processed by
the digital circuitry presently available for use in phase
; locked loops.
Furthermore, although it is desirable to share
components of the phase locked loop tuning system with other
portions o a television receiver in which it is employed to
reduce the cost of the receiver, care must be taken that the
reduced cost is not at the expense of the overall performance
of the receiver. For example, it has been suggested in an
article entitled, "A Fre~uency Synthesizer for Television
Receivers", by Eric Breeze et al D published in the IEEE
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~5~ c~ 71,02~71,024A
Transactions, November 1974, and U.S. Patent 3,980,951
entitled, "Electronic Tuning Control System i-or Television",
filed in the name of Eric sreeze et al. on August 13, 1975
and issued on September 14, 1976, to couple the signal
3.58 MHz (i.e., the color subcarrier frequency in the
Unitea States) derived by a crystal oscillator of a color
demodulator of a television receiver to a phase detector
employed in a phase locked loop for tuning the receiver to
reduce the cost of the receiver by eliminating the cost of
a separate crystal oscillator from which a reference
frequency signal may be derived. However/ under such
circumstances it is advisable to divide the frequency of
the 3.58 MHz signal sufficiently to provide a ~requency
compatible with the operating frequency range of digital
circuitry employed in tha phase locked loop. The factor
by which the 3.58 MHz signal is divided should desirably
also be selected to reduce the visible effects of modulated
components of the local oscillator occurring at the reference
frequency earlier described. Although the Breeze et al.
article (on page 262) states that an additional advantage
of using the 3.58 MHz color subcarrier signal as a reference
is that any digital noise generated would be coherent with
the 15.75 KHz raster frequency and would therefore be less
objectionable when seen on the screen, for reasons later to
be explained the reierence frequencies derived by dividing
the 3.58 MHz color subcarrier frequency by means of dividers
which are capable oi~ dividing by integer number would not
be suitable for substantially cancelling interference images
generated in respon~3e to unfiltered pulse components of the
control signal of a phase locked loop tuning system occurring
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1 at the reference frequency in the manner set forth in detail
below.
In accordance with an embodlment of the present
invention, a phase
locked loop tuning system employed to generate the local
oscillator signal for a television receiver which produces
an image by scanning an image reproducing device at
predetermined scanning rates includes means for generating
a reference signal having a frequency related to the
predetermined scanning rates so that the visible effects
of an unfiltered component of the local oscillator signal
occurring at the reference frequency are substantially
cancelled as an image is formed.
The sole FIGURE shows, in block diagram form, a
television receiver including a phase locked loop type of
tuning system constructed in accordance with the present
invention.
The television receiver of the sole FIGURE includes
an antenna 12 for receiving radio frequency (RF) television
signals and an RF processing unit 14 for amplifying and
otherwise processing the received signals. The processed RF
signals are combined in a mixer 18 with local oscillator
2S signals generated by a phase locked loop tuning system 16
to form an intermediLate frequency (IF) signal. The
intermediate frequency signal is amplified, filtered and
otherwise processed in an IF processing unit 20 and a signal
processing unit 22 1:o provide chrominance, luminance,
synchronization and sound signal components. The chrominance
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RCA 71, 024/71, 024A
1 and luminance signals are coupled to appropriate electrodes
of a kinescope 24 and the sound signals are coupled to a
loudspeaker 26 by means of a signal processing unit 22
The synchronization components of the signal
provided by IF si~nal processing unit 20 are coupled to a
synchronization signal (sync) separator which extracts
horizontal and vertical synchronization pulses from the
composite signal. The horizontal synchronization pulses are
coupled to a horizontal deflection unit 30 which controls
the hori~ontal deflection of the electron beams generated
by kinescope 24. Horizontal deflection unit 30 also
couples a signal to a high voltage unit which develops a
supply voltage for kinescope 24 therefrom. Vertical
synchronization pulses are coupled to a vertical deflection
unit 32 which controls the vertical deflection of electron
beams generated by kinescope 24.
Portions of the receiver thus far described may
be formed in a manner similar to corresponding portions of
the receiver disclosed in RCA Television Service Data,
File 1975 C-lO for the CTC-74 chassis published by RC~
Corporation, Indianapolis, Indiana.
Phase locked loop tuning system 16 includes an
oscillator 34, which for example may comprise a crystal
oscillator, for generating a signal having an accurate and
stable fixed frequency fxTAL- The frequency of the fixed
frequency signal is divided by a factor R by means of a
divider 36, which may, for example, comprise a counter, to
develop a reference frequency signal having a frequency
fREF equal to R - The value of the factor R is selected
so that the reference frequency fREF is suitably low to make
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~ao~s~lg~ RCA 71,024/71,024A
1 it compatible with the operating frequency range of other
portions of phase locked loop 16. Furthermore, as will be
subsequently explained, the value of the factor R is
selected in relation to the value of the fixed frequency
fxTAL to produce a reference frequency, related in a
predetermined manner to the receiver's horizontal and vertical
electron beam scanning rates, H and V, respectively, to
reduce the visible effects of undesirable frequency modulated
components of the local oscillator signal which may be
generated by phase locked loop 16.
The reference frequency signal is directly coupled
(i.e., coupled without further frequency division) to an
input of a phase detector 38. The output signal of a
programmable divider 46 is coupled to the other input of
phase detector 38. Phase detector 38 develops a signal
representing the phase and/or fre~uency deviation between
its two input signals. A typical phase detector suitable
for use in phase locked loop 16 provides a series of pulses
at the reference frequency whose duration is related to the
phase and frequency deviation between its two input signals.
Such a phase datector is described in the aforementioned
RCA application note ICAN 6061,~
and is included in the CD4046 integrated
circuit phase locked loop available from RCA Corporation,
Somerville, New Jersey.
The output signal of phase detector 38 is coupled
to a low pass ilter 40 which integrates it to form a DC
signal the amplitude of which varies in accordance with
the phase and frequency deviations between the input signals
f phase detector 38. This varying DC signal is coupled
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~5~ RCA 71,024/71,024A
to a voltage controlled oscillator, serving as a local
oscillator 42, to control its frequency of oscillation.
The local oscillator output signal of oscillator 42 is
coupled to a mixer 18 and to divide-by-K prescaler 44.
Divide-by-K prescaler 44 comprises, for example,
a counter which divides the frequency of the local oscillator
signal by a factor K selected so that: its output signal has
a frequency which is compatible with the operating frequency
ra;nge of the remaining components of phase locked loop 16.
Specifically, the value of the factor K is selected so that
the signal coupled to programmable divider 46 has a frequency
lower than the highest frequency which programmable divider
is capable of processing.
Programmable divider 46 divides the frequency of
the output signal of prescaler 44 by a programmable factor
N dependent on the channel seleçted by a viewer by means of
a channel selector unit 48. Programmable divider 46 may,
for example, comprise a co~nter which counts N cycles of
its input signal for each cycle of its output signals. The
factor N is controlled, for example, in response to binary
coded decimal (BCD) signals provided by channel selector
unit 48. The output signal of programmable divider 46 is
coupled to phase detector 38 to complete phase locked loop
16.
In operat:ion, the control signal produced by low
pass filter 40 controls the frequency of the local oscillator
signal until the frequency and phase of the output signals
of divide-by-R divi.der 36 and divide-by-N divider 46 are
in a predetermined relationship, e.g., substantially equal.
At this point, phase locked loop 16 is said to be "locked"
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~ RCA 71,024/71,02~A
1 and the local oscillator signal will have a frequency fLO
determined by the following expression:
LO R XTAL NK REF ( 1 )
In the United States, the values, in MHz of the
local oscillator frequencies for the channels a viewer may
select include prime numbers (i.e., the lowest common divider
is 1) in a range between lOl and 931. Therefore, it is
desirable that, for each channel that may be selected by a
viewer, N be equal to the frequency, in MHz, of the
corresponding local oscillator signal. With this premise~
expression (l) can be rewritten as:
l MHZ = KfREF (2)
Theoretically, in accordance with expression (2),
the frequency of the local oscillator is as stable as the
reference frequency. Unfortunately, when a phase detector
of the type described in RCA Application Note ICAN 6061,
referenced above, is employed as phase detector 38, portions
of the error pulses, representing the phase and frequency
deviation between the output signals of divide-by-R divider
36 and divide-by-N divider ~6, occurring at the reference
frequency fREF may cause the local oscillator frequency f~O
to be frequency modulated. This is so because in practice,
low pass filter 40 cannot readily remove all traces of the
error pulses from the D.C. control signal produced by it.
As a result, the control signal applied to local oscillator
42 is amplitude modulated by error pulse components occurring
at the reference frequency fREF~ The amplitude modulated
components of the control signal tend to cause a correspond-
ing frequency modulation of the local oscillator signal.
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RCA 71,024/71,024A
l When a local oscillator signal containing frequency
modulated components is combined with a radio frequency
carrier in mixer 18, an intermediate frequency slgnal is
produced which may also contain a frequency modula-ted
component. Since conventional IE processing circuits ~hich
may be utilized as IF amplifier 20 have an asymmetric, e.g.,
decreasing, amplitude versus frequency response character-
istic with respect to the picture carrier frequency, e.g.,
45.75 MHz, IF signal componen-ts having a frequency hi~her
1 than the picture carrier are attenuated more than IF signal
components having a frequency lower than the picture carrier.
Therefore, the video signal developed by IF processing unit
20 may include an amplitude modulated component at a
frequency corresponding to the reference frequency. When
processed by signal processing unit 22, these amplitude
modulated components of the output signal IF processing
unit 20 ordinarily may produce a visible interference
pattern in the image reproduced by kinescope 24 which is
annoying to a viewer.
As earlier mentioned, to reduce the visible effects
of frequency modulated components of the local oscillator
signal occurring at the reference frequency fREF, the
reference frequency is chosen in a predetermined relationship
to the horizontal and vertical scanning rates, H and V,
respectively. Specifically, reference frequencies which
may be utilized for this purpose are defined by the
expression:
fREF m H ~ nV (3
where n is an integer and m is an odd integer. In other
3 words, this expression defines a family of frequencies all
:
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~5~ , RCA 71,02~/71,024A
1 of which are the sum o~ an odd subharmonic of -the horizontal
scanning rate and a harmonic of the vertical ~canning ra-te.
Under these conditions, amplitude modulated components of
the video signal corresponding to frequency modulated
components of the local oscillator signal occurring at the
reference frequency will tend -to be cancelled as an image
is formed.
Expression (3) will be better understood after a
brief description of some aspects of the operation oE
receiver lO to produce a color image. The image produced
by receiver lO is formed by scanning three substantially
convergent intensity modulated electron beams corresponding
to the three primary colors, i.e., red,green and blue,
in a raster pattern across the phosphor coating on the
inside face of kinescope 2~. The electron beams are
horizontally deflected along alternate horizontal lines
while they are simultaneously vertically deflected. Two
vertical transitions or fields are required to form a
complete image. During one field the odd-numbered horizontal
lines are scanned, while during the following field, the
even-numbered horizontal lines are scanned. This type of
raster scanning pattern is commonly referred to in the art
as being interlaced because the horizontal lines skipped
during one field are scanned during the next and vice versa.
In the United States to form one complete image there are a
total of 525 horizontal lines scanned at a rate H, equal to
the frequency of the horizontal synchronization pulse, e.g.,
approximately 15.734 KHz. Since there are two fields scanned
per image, the vertical scanning rate V, equal to the frequency
0 of the vertical synchronization pulses is equal to 5~5 x 2
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~ 5 ~ ~ ~ RCA 71,024/71,024A
1 or approximately 60 Hz.
~ n order to make the transmission and receptionof color television signals compatible with the operation
of monochrome ("black and white") television receivers,
clusters of signals representiny color information are
interlaced in frequency with clusters of signals representing
brightness (or luminance) informa-tion. As a result, the
signals representing color information occupy the same
frequency band as do the signals representing luminance
information. So that color signals do not produce inter-
ference patterns on the screen of a monochrome receiver the
carrier associated with the clusters of signals representing
color information has a frequency which ~s an odd harmonic
of one-half the line scanning rate (H) while the carrier
associated with the clusters of signals representing
luminance information has a frequency which is whole
multiple of the line scanning rate (H).
Because of these relationships to the horizontal
scanning rate H, which in turn is related to the scanning
rate V, brightness variations produced by a signal represent-
ing luminance information goes through a whole number of
cycles during the scanning of any given line or frame. This
means that either in the next line or in the next frame, the
brightness variations reoccur in phase and a reinforcing
2S effect is as a result produced. In the case of a signal
representing color information, however, the opposite
condition exists. Since the frequency of a signal represent-
ing color information is an odd harmonic of one-half the
line frequency H, during the scanning time of any one line,
a signal representing color information goes through a
~.0~9a~ RCA 7 1, 0 2 4 /7 1 , 0 2 4A
1 certain number of cycles plus a half-cycle. Thus, durlng
the scanning of the next line in the same frame, the signal
representing color information reoccurs out of phase by
180 degrees. It also reoccurs out o~ phase during the
scanning of the same line in the succeeding frame. As a
result of this out of phase condition, a cancellation effect
occurs, and an interference pattern due to
brightness var1ations produced on the screen of a monochrome
receiver by a signal representing color information cannot
readily be perceived by the human eye. Thus, satisfactory
monochrome reproduction can be achieved when a composite
color signal is being transmitted.
In the United States the carrier signal of the
color information, i.e.j the color subcarrier, has a
frequency of 3.5795.... MHz (approximately 3.58 MHz). The
color subcarrier~requency, i.e., 3.5795.~..MHz; is equal
to 455 times one-half of the horizontal line ra-te in the
United States,~i.e., 15,734.26573.... Hz (approximately
15,734 Hz). Therefore, interference image components
resulting from the color subcarrier occur may times during
each horizontal line. As discussed above, interference
image components resulcing from the color subcarrier are
cancelled on a line by line basis. That is, the image
interference signals resulting from the color subcarrier
on one horizontal line are cancelled by out of phase image
components resulting from the color subcarrier in the next
(in numerical order) line which occurs in the next field.
For example, color subcarrier image components in line 1 of
field 1 are cancelled by out of phase color subcarrier image
components in line 3 of field 1.
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~5~9~ RCA 71,024/71,024A
1 To understand expression (3), i-t is necessary to
appreciate tha-t the reference frequency fREF for phase locked
loop television tuning systems is limited in frequency by the
highest frequency which the components, specifically
programmable dividers such as 46, of the phase locked loop
are capable of processing. For state-of-the-art components
which may be employed in phase locked loop systems, the
reference frequency should desirably be considerably less
than 3.58 MHz. Under these conditions, the periods of
interference images related to the reference frequency
signal are so long that several horizontal
lines are scanned during the interval. In other words, while
interference images related to the color subcarrier signal
would tend to produce disconcerting patterns along the
horizontal direction of an image if not cancelled, inter-
ference images related to the reference frequency signal
would tend to produce disconcerting patterns along the
vertical direction of an image if not cancelled.
As a result, in order to minimize the visibility of
interference patterns related to a reference frequency signal
which is compatible with state-of-the-art components for
use in phase locked loop tuning systems, interference images
related to a reference frequency signal should be cancelled
on a field to fielcl basis rather than on a line to line
basis as is the case for interference images related to the
color subcarrier. This means that interference images
related to the reference frequency signal in one field should
be out of phase with interference images related to the
reference frequency signal in the nex-t field. This may be
expressed mathematically by the expression:
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1 REF jV ~ 2V (4)
where j is any integer. The term 2-V corresponds to the
necessary phase reversal to accomplish cancellation of
vertical patterns.
Recognizing that the integer j can always be
expressed as the sum or difference of -two other integers
n and p, expression (4) can be rewritten as:
fREF = (P + n)V -+ 2-V = +nV + (p + 2)V (5)
Recalling that in the United States V = 525' expression (5)
may be rewritten as:
fREF = +nV + (P + 2)525 = *nV -~ (2p +- 1)525 (6)
Letting
(2p ~ 1) = 1 ( )
525 m
where m is an integer, expression (6) can be written as
fREF = +nV + m (8)
which is the same as expression (3). Referring to expression
(7), it is seen that
(2p -+ 1) = 525 (9)
Since (2p + 1) always defines an odd integer, m must be an
integer factor of 525. Thus m must be one of the odd numbers
1, 3, 5, 7, 15, 21, 25, 35, 75, 105, 175 or 525.
The values of m, n and K which satisfy expression
(3) are defined by the expression:
lMHz = :K( m H + nV) (10)
which is obtained by combining expressions (2) and (3).
Furthermore, again recognizing that V = ~ x 2, expression
(10) may be rewritten as:
lMHz = :KH( m * 525 ) (11)
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RCA 71,024/71,024A
1 It is desirable that K be an integer number since
dividers which divide by fractional divisors are relatively
complex and therefore relatively expensive. Unfortunately,
there are no exact solutions of expression (11) for which K
is an integer number. However, there are many solutions for
which K is an integer number and whic:h provide a reference
frequency within +l Hz of an exact frequency solu-tion.
Although a ~otal cancellation of interference signals
due to a frequency modulation of the local oscillator signal
occurring at the reference frequency rate cannot be achieved
by such a non-exact solution, a significant reduction, e.g.,
between 15 and 20 decibels (dB), of the interference signal
can be expected.
One non-exact solution of expression (11) which
affords a significant reduction of the undesired interference
signal is provided by the selection of m equal to 5, n equal
to 17 and K equal to 240. With this solution, by utilizing
a 1 MHz crystal oscillator and selecting R equal to K, a
reference frequèncy of 4166.66667 Hz having a deviation of
0.8325 Hz from the corresponding exact frequency solution
defined by expression (11) is provided. This solution is a -
particularly desirable one for the following reasons.
Counters which are utilized to divide the frequency of
relatively high frequency signals, e.g., such as the local
oscillator signal for channel 82 having a frequency of 931
MHz in the United States, desirably include as large a
number as possible of binary (flip-flop) stages cascaded
without a feedback path from the output of one stage to the
input of a second stage whereby the second stage is reset
3 after a predetermined count. This is so because resetting
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RCA 71, 024/ 71, 024A
95~
1 a binary stage requires a time clelay which considerably
limits the maximum frequency o~ operation of a counter.
Selecting K as 240 permits utilizing a counter comprising
4 binary stages cascaded wi-thout a feedback path -to divide
by 16 followed by ~ binary stages with a feedback path to
divide by 15. Since the first four stages of the counter do
not utili~e a feedback path -they can readily divide the
relatively high input frequency (by 16) to provide a relative-
ly low frequency which can be further divided (by 15) by the
last four stages (the frequency of operation of which is
limited by the feedback path).
Examples of other solutions which permit the use
of at least 2 binary (flip-flop) cascaded without a feedback
path are now set forth. These solutions may be implemented
1~ to form a phase locked loop tuning system by utilizing a
1 MHz crystal oscillator and by selecting R equal to K.
Selecting m equal to 5, n equal to -5 and K equal to 344
provides a reference frequency of 2906.97674 Hz having a
0.11616 Hz deviation from the corresponding exact reference
frequency. A counter which may be utilized to divide by 344
may comprise, for example, 3 binary stages cascaded without
a eedback path to divide by 8 followed by 6 binary stages
cascaded with a feedback pa-th to divide by ~3. Selecting m
equal to 5, n equal to -1 and K equal to 324 provides a
25 reference frequency of 3086.41975 Hz having a deviation of
0.4933 Hz from the exact reference frequency defined by
expression (5). A counter which may be utilized to divide
by 324 may comprise, for example, 2 binary stages cascaded
without a feedback path to divide by 4 followed by 7 binary
stages cascaded with a feedback path to divide by 81.
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RCA 71,02~/71,024~
5~
ith respect to the above-cited Breeze et al.
article and patent where it is suggested to derive the
reference frequency of a phase locked loop tuninq system
from the 3.58 MHz color subcarrier frequency, it is noted
that there is no integer divider which may be used to
convert the color subcarrier frequency to an odd submul-tiple
of the horizontal line scanning rate H as required by
expression (3). As earlier explained, in the United States
so that color subcarrier frequency components are not visible
in an image, the color subcarrier frequency was deliberately
chosen so that it is an odd harmonic of one-half the
horizontal line scanning rates. In other words, 3.5795~....
MHz divided by 15,734.26573.... Hz equals 222.5 which is not
an integer number. Specifically, in the Breeze et al.
article, two reference frequencies are derived from the 3.58
MHz color subcarrier frequency: a 10 KHz reference
frequency for VHF channels is derived by dividing 3.58 MHz
by 358 (i.e., 179 x 2) and a 2.5 KHz reference frequency for
UHF channels is derived by dividing 3.58 MHz by 1432
(179 x 8). Applying the relationship of expression (3) to
find a null reference frequency which will substantially
cancel video components related to the reference frequency
signal close to the 10 KHz VHF reference frequency of the
Breeze et al. article, we find that a null reference
frequency is 9.98001982 KHz [see expression (3) with
m = 3 and n = 79]. Thus, the Breeze et al. reference fre-
quency for VHF channels is approximately 20Hz away from the
null reference~ frequency defined in expression (3).Sir~ilarly,
it can be shown that the 2.5 KHz UHF reference frequency of
the Breeze et al. article is approximately 12Hz away from the
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I closest null frequency defined in expression (3) [see
expression (3) with m = 7 and n = 4]. Considering that
the null reference frequencies according to the invention
occur only 59.95 Hz (i.e., the vertical scanning frequency
in the United States) apart, the reference frequencies
derived from the color subcarrier frequency set for-th in the
sreeze et al. article are rather poor choices to subs~antially
cancel video components generated in response to amplltude
modulated components of a phase locked loop control signal
o occurring at the reference frequency.
By utilizing a phase locked loop tuning system
wherein the reference frequency is substantially equal to
one of the frequencies defined by expression (3), not
only are undesired interference patterns, which may result
from a frequency modulated component of the local oscillator ;~
signal occurring at the reference frequency, reduced, but
they are reduced without additional circuitry. Indeed,
circuitry employed in the phase locked loop tuning system
may actually be simplified. For example, since the
reduction of the undesirable effects of frequency modulated
components occurring at the reference frequency will be no
longer solely dependent on the low pass filter included in
the loop, its filtering requirements may be relaxed. As
a result, a less complex and therefore less expensive
low pass filter may be utilized. Moreover, the acquisition
or pull-in time of a phase locked loop (i.e., the time
required for a phase locked loop to reach a locked condition),
wherein the reference frequency substantially equal to one
of the frequencies defined by expression (3) is provided
may be shorter than that of a conventional phase locked
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1 loop. This is so because the low pass filter employed in a
phase locked loop wherein a reference frequency substantially
- equal to one of the frequencies defined by expression (3) is
provided may utilize a low pass filter having a higher cut-
off frequency than that of a low pass filter of aconventional phase locked loop having a cut-off frequency
which is chosen sufficiently low to provide acceptable
high frequency rejection.
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