Note : Les descriptions sont présentées dans la langue officielle dans laquelle elles ont été soumises.
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The pres~nt invention relates to a power supply
apparatus. More specifically, the present invention relates to a
direct current power supply apparatus employing a direct current/
alternate current inverter, par~icularly suited or small-sized
and light-weight electrical equipment.
A sPlf-excited inver~er employing a Royer oscillator has
been proposed and put into practical use as an inverter for use
in a power supply apparatus. Since such a Royer-type inverter is
well known, it is not believed necessary ~o describe the same in
detail. Briefly described, a Royer-type inverter comprises two
power transistors which are each utilized as a switching device
such that a direct curren~ is on/off controlled or switched to
provide an alternate current voltage of a rectangular wave formO
The switching operation of the transistors is achieved by a
feedback coil coupled to the primary wlnding of a saturable
transformer. In operation, one transistox is rendered conductive
while the other transistor is rendered non-conductive for a given
time period. Then the operation state is reversed and thereafter
this sequence is repeated. In a Royer-type inverter, the excited
voltage in th~ primary winding of the saturable transformer
which serves as a common load of the respective transistors
causes a voltage to be applied between the collector and emitter
electrodes in the non-conduction stater This applied voltage
may equal approximately two times the power supply voltage.
Accordingly, transistors of a higher withstand voltag are
requlred for such an inverter.
- ~ A so-called half-bridge type inverter has been proposed
as an improvement to the ~oyer-type inverter. In a typical
example of a half-bridge-type inverter the voltage from a direct
current power supply is halved by a pair of dividing capacitors.
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Accordinyly, only half the power supply voltage is applied
between -the collector and emitter electrodes of the transistor in
a conductive state. It follows that transistors havir.g a lower
withstand voltage can be used as compared to -those used in a
Royer-type inverter.
However, even a half-bridge-type inverter still has
- several problems to be solved. Specifically, because a starting
resistor need be employed across one of two power transistors,
the biasing circuits of the two otherwise matched power
transistors becomes asymmetrical. In the result, the power
transistor across which the starting resistor is connected is
never rendered fully non~conductive but remains somewhat
conductive even during its non-conductive interval. The output
of the inverter i8 therefore asy~metrical, resulting in poor
efficiency and perhaps causing a short circuiting of the two
power transistors.
Accordingly, a principal object of the present
invention is to provide a power supply apparatus which seeks to
~ overcome the above disadvantages of the prior art.
`~ 20 According to the present invcntion, then, there is
provided a power supply apparatus comprising an inverter for
converting a direct current output from a direct current voltage
source .into an alternating current voltage, the inverter
comprising: first and second dividing capacitors connected in
ser1es, the capaci-tors being arranged to be connected in parallel
wlth the direct current voltage source ~irst and second trans-
istors having their main current-carrying paths connected in
series, the series-connected paths also being arranged to be
connected in parallel with the direct current voltage source; a
saturable transformer having a primary winding connected between
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the junction of the dividing capacitors and the junc-tion of
the main current-carrying paths of the transistors; biasing
means for each of the transistors, each biasing means comprising
a respective feedback winding magnetically coupled to the
primary winding and connected in such a manner that the voltage
induced thereacross by current flowing through the primary
winding when the respective transistor is conductive is operable
to maintain the -transistor in its conductive state until satu-
ration occurs, the arrangement being such that in use the
transistors are alternating conductive and thus cause an
alternating current to flow through the primary winding; and
the biasing means of the transistors being arranged to cause
substantially symmetrical operation of the transistors.
According to a further aspect of the present invention,
- there is also provided an electric shaver comprising: a casing
; having an openiny; a shaver cutt r assembly provided in -the
shaver casing so as to be exposed through the opening, the :
; shaver cutter assembly comprising a stationary cutter and a
movable cutter; a power supply apparatus provided within the
shaver casing, the power supply apparatus comprising an inverter
for converting a direct current output from a direct current
voltage source into an alternating current voltage, the inverter
comprising: first and second dividing capacitors connected in
seri~s, the capacitors being arranged for connection in parallel ~-
with the direct current voltage source; first and Second trans-
istors~having their main current-carrying paths connected in ;
series, *he series-connected paths also being arranged ~or
connection in parallel with the direct current voltage source;
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a saturable transformer having a primary winding connected
between -the junction of the dividiny capacitors and the junc~ion
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of the main current-carrying paths of the transistors; and
biasing means for each of the transistoxs, each biasing means
comprising a respective feedback winding magnetically coupled to
the primary winding and connected in such a manner that the
voltage induced thereacross by current ~lowing through the
primary winding when the respective transistor is conductive is
operable to maintain the transistor in its conductive state
: until ~turation occurs, the arrangement being such that in use
the transistors are alternatively conductive and -thus cause an
alternating current to flow through the primary winding; the
biasing ~eans of the transistors being arranged to cause
substantially symmetrical operation of the transistors; and
the saturable transformer further including a secondary
winding and the shaver further comprising motor means housed ~:
in the shaver casing and coupled to the secondary winding of
: the saturable transformer for driving the movable cutter of
the shaver cutter assembly.
Embodiments of the present invention will now be
: described in greater detail and will be bet-ter understood
when read in conjunction with the following drawings in
which:
FIGURE 1 is a schematic diagram of a conventional
halfbridge-type inverter;
FIGURE 2 is a schematic diagram of one embodimen-t of
the present inven-tion;
FIGURE 3 ia a schematic d.iagram of a preferred embodi-
ment of ~he prèsent invention;
. FIGURES 4A and 4B are load characteristic curves of
: the transistors employed in the Figu~e 3 emhodiment~
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FIGURE 5 is a schematic diagram of another preferred
embodiment of the present invention;
FIGURES 6A and 6B are wave ~orms of the outpu~ ~rom
the inverter shown in Figure 5,
FIGURES 7A th.rough 7J show wave forms of th~ electrical
signals at various portions in ~he Figure 5 embodiment,
FIGURE 8 is a schematic diagram of a ~urther preferred
embodiment of the present invention;
FIGURE5 9 and 10 are wave forms of the collector
current in the Figure 8 embodiment;
FIGUR~ 11 is a w~ve form of the voltage across the
time constant capacitor in the Figure 8 embodiment;
FIGURE 12 is a schematic diagram of still another
preferred embodiment of the present invention;
FIGURE 13 is a schematic diagram of still a further
prefèrred embodiment of the presen~ invention;
FIGURE 14 shows a fluctuation characteristic of a
charging curren~ with respect to the source voltage fluctuation
~in the Figure 13 embodiment;
FIGURE 15 is a schematic diagram of still a further
preferred embodiment of the present invention;
FIGURE 16 is a schematic diagram of still a ~urther
preferred embodiment of the present invention;
FIGU~E 17 is a wave ~orm of a charging voltage of the
secondary battery in the Figure 16 embodiment;
: FIGURE 18 is a schematic d1agram of still another
preferred embodiment o~ the present invention; and
FIGURE 19 ~hows a noi~e characteristic.
~ Re~erring to Figure 1, in a known half-bridge-type
in~erter, when a switch 3 is turned on, a direct current
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voltage from a direct current voltage source 2 causes a current
to flow through base resistors 7 and 11 of a pair of PNP
transistors 6 and 10. The voltage drop across the respective
base resistors 7 and 11 will rénder the pair of ~NP transistors
6 and 10 conduc~ive. A starti~g resistor 13 is connected between
the base and collector electrodes of the transistor 10. Hence,
the current flowing through the base re~istor 11 is larger than
the current flowing through the other base resistor 7 and the
transistor 10 is rendered conductive earlier than the transistor
6. When` the transistor 10 is rendered conduc~ive, the electric
charge in a dividing capacitor 9 is discharged through the
transistor 10 and a primary winding 5 wound on a saturable
magnetic core 14. On th~ other hand, a dividing capacitor 4
starts being charged by a current ~lowing thro~gh the
transistor 10 and the primary winding 5 from the direct current
voltage source 2. At that time, the current flowing from the
point a to the point b of the primary winding 5 induces a voltage
across one feedback winding 12 in the direction ~or forward
biasing the transistor 10 and al50 induces a reverse bias voltage
for the transistor 6 in the other feedback winding 8. Accor-
dingly, the transistor 10 is rendered co~ductive, while ~he
transistor 6 is rendered non-conductive, with the result that a
positive half-wave output is generated in a secondary winding 15
wound on the magnetic core 14 and coupled to the primary
winding 5.
Thusg the primary winding 5 is excited. When the
primary winding~5 is thus excited and the magnetic core 14 of the
saturable trans~ormer i~ ~aturated, the magnetomotive.force of
the magnetic core 14 disappears and a voltage is induced in the
- 30 respective feedback windings 8 and 12 in the reverse direction~
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Therefore, one transistor lO is rendered non~conductive and the
other transistor 6 is rendered conductive. Accordingly, the
electric charge in the dividing capacitor 4 is discharged through
the primary winding 5 and the transistor 6 and the dividing
capacitor 9 is charged by a current flowing through the primary
winding 5 and the transistor 6 from the direct current voltage
source 2. The above-described charging current causes the
primary winding 5 to be excited in the direction from the point
b to the point a, with the result that a negative half-wave
output is obtained in the secondary winding 15.
The output from the inverter l is obtained throu~h the
secondary winding 15 which is wound on the saturable magnetic
core 14. Winding 15 is coupled to primary winding 5O The output
is applied to a load circuit 200, which may comprise a direct
current motor, for example.
Since in such a half-bridge-type inverter as shown in
Figure l, the voltage E of the direct current voltage source 2 is
divided by two by means of a pair of dividing capacitors 4 and 9,
the voltage E/2 is applied between the collector and emitter
electrodes of the transistor in conduction. As a result,
according to a hal~-bridge-type inverter, transistors of a lower
withstand v~ltage can be employed as compared with a case of a
Royer-type inverter.
As mentioned aboYe, problems remain however.
Specifically, transistox lO need be provided with a startiny
res~stor 13 for preferentially rendering transistor lO conductive.
Because of the ~tarting resistor 130 the bias circuits for the
two transistors 6 and lO selected as;,a pair in terms of their
electrical characteristics become asymmetrical. This causes a
situation wherein the transistor 10 is not rend~red fully
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non-conductive or cut off, i.e., a somewhat conductive tendency
exists even in the non-conduction period. In order to avoid
such a situation, a reverse bias could be applied between the
base and emitter electrodes of the transistor 10, although
there remains some difficulty. Ano her problem i 5 that the
above-described conductive tendency of the transistor 10 makes
the output wave of the inverter 1 asymmetrical with respect to
the positive and negative polarities, resulting in poor
efficiency. This fact is also liable to cause short-circuiting
of transistors 6 and 10.
Referring now to Figure 2, circuit configuration of a
direct current/alternate current inverter for use according to
an en~odiment of the present invention is shown. The embodiment
shown comprises a series connection of a paix of dividing
capacitors 104 and 109 and a series connec~ion of a PNP
transistor 106 and an NPN transistor 110, both series connections
being coupled in parallel with a direct current voltage source
2 through a switch 103. A primary winding 105 wound on a
saturable ~agnetic core 114 for constituting a saturable trans-
former is interposed between the junctions a and b of the
res~ective series connections. The saturable transformer ~urther
comprises a secondary winding 115 and a pair of feedback
windings 108 and 112 wound on saturable magnetic core 114 so as
to be magnetically coupled to primary winding 105. Secondary
winding 115 is connected to a load circuit 200.
One end of each o~ feedback windings 108 and 112 is
` connected to the base electrodes o~ transistors 106 and 110
re6pectively. ~he othex ends of each of feedback windings 108
and 112 are connected to each of the emitter electrodes o~
transistors 106 and 110 respectively through each of respective
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base resistors 107 and 111 so as to constitute a respectivebase biasing circuit. A capacitor 116 is interposed between the
junction c of feedback winding 108 and base resistor 107 and the
junction d between feedback winding 112 and base resistor lllo
The capacitor 116 is shunted by a resistor 117 of a large
resistance value.
When switch 103 is closed, a current flows through base
resistors 107 and 111 and capacitor 116. The voltage drop across
respective base resistors 107 and 111 will render xespective
transistors 106 and 110 conductive. However, because of the
differin~ characteristics of the components such as transistors
106 and 110, ~eedback windings 108 and 112 and base resistors
107 and 111, one transistor is likely to ke conductive earlier
than the other. Assume that transistor 106 is likely to be
conductive earlier than transisto~ 110 in the embodiment shown~
When PNP transistor 106 becomes conduc~ive, the charge in
`~ dividing capacitor 104 is discharged through transistor 106 and
primary winding 105. At the same time, dividing capacitor 109
starts being charged by a current ~lowiny through transistor 106
and primary winding 105 from direct current voltage source 2.
At tha~ time, the current flowing from the node b of primary
windlng 105 to the node a induces a voltage in one feedback
winding 108 in the direction for forward biasing transistor 106
and a voltage in the other feedhack winding 112 for revexse
biaslng transistor 110. Accordingly, transistor 106 is rendered
conductive and transistor 110 ls rendered non-conductive, with
the result that a positive half~wave output is obtained in
second winding 115.
The primary winding 105 is thus ex~ited and the
30 ~magnetic core 114 o~ the saturable transformer is saturated.
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When magnetic core 114 is saturated, the magnetomotive force
of magnetic core 114 disappears and a reverse directional
voltage is induced in feedback windings 108 and 112. There-
fore, one transistor 106 is rendered non-conductive and the other
transistor 110 is rendered conductive. Accordingly, the electric
charge in dividing capacitor 109 is discharged through primary
winding 105 and transistor 110 and dividing capacitor 104 is
charged by a current flowing through primary winding lOS and
transistor 110 from the direct current voltage source 2. The
above-described charging current causes primary winding 105 to
be excited in the direction from node a to node b, with the
result that a negative half-wave output is obtained in secondary
winding 115.
Thereafter, the conduction state of ~he transîs~ors is
reversed each time magnetic core 114 of the saturable transformer
is saturated as described above and a positive half-wave output -
and a negative half-wave output are obtained alternately in
secondary winding 115.
One ~eature to be noted in the Figure 2 embodiment is
20 the connectlon of capacitor 116 and resistor 117. The said
connection of capacitor 116 and resistor 117 causes an ample
current to flow through the base biasing circuit at the initial
" stage of the operation, eliminating the necessity for starting
resistor 13 as illustrated in Figure 1. Additionally,
transistors 106 and 110 are implemented by compl~mentary
txansistors. The emitter electrodes of each of these transistors
are coupled to the direct current voltagç source side. As a
result, the base biasing circuits of transistors 1~6 and 110 can
be provlded on the voltage source side with respect to primary
winding 105 of the transformer, making it possible to
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symmetrically construct the above-described pair of base
biasing circui-ts. Accordingly, the output from the present
inverter is symmetrical. The above-described symmetrical circuit
configuration of the base biasing circuits facilitates the
selection of impedance values for the various impedance
components such as resistors and capacitors used in the base
biasing circuits. Further, the continuous conductivity of one of
the two power transistors described in conjunction with the prior
art can be avoided. The short-circuiting o~ transistors 106 and
110 can therefore also be avoided.
Referring to Figure 3, a further preferred emhodiment
of a power supply apparatus emp7oying the above-described
inverter will be described. It is pointed out that the
embodiment shown is particularly suited as a power supply for
electrical equipment using relatively small direct current
motors such as electric razors, motor-driven tooth brushes and
the like. Thus, the embodiment is shown comprising a storage
battery 203 and a direct current motor 205 in the load circuit
200. Both terminals of secondary winding 115 of the saturable
transformer are coupled to rectifying diodes 201 and 202
constituting a rectifying circuit. The output of the rectifying
circuit is connected to one end of storage battery 203. The
other end o~ storage battery 203 is connected to a center tap
115a of secondary winding 115 through a limiting resistor 204~
In addition, both terminals of storage battery 203 are connected
to direct current motor 205 thxough a switch contact 206.
Accordinyly, when switch contact 206 is closed, the limuting
resistor 204 i9 short-circuited and ~torage battery 203 is
charged with a full-wave rectified output obtained by
rectifying an alternate current output in a rectangular wave form
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at secondary winding 115 of inverter 100 by means of diodes 201
and 202. It is pointed out that direct current motor 205 is
also energized with the above-described full-wave rectiEied
output.
The direct current voltage source 2 of inverter 100
may comprise an alternate current voltage source 21 such as a
commercial power supply and a bridge circuit 22 for full wave
rectification of the alternate current output from alternate
current voltage source 21. Both terminals of bridge circuit 22
are connected to the input termunals of inverter 100 at the
output terminals of direct current voltage source 2.
Inverter 100 shown in Figure 3 is different from the
Figure 2 inverter in the following respects, Speci~ically, in
the Figure 3 embodiment, capacitor 116 and resistor 117 are
each implemented by series connections o~ the two capacitors
116a and 116b and two resistors 117a and 117b, swch that the
junctions of the respective series connections are each
connected to the junction a of dividing capacitors 104 and 109.
It has been observed that the connec*ion of capacitors 116a and
116b to the point a reduces a spike voltage occurring across
primary winding 105. More specifically, a spike voltage
occurring across primary winding 105 cau~es a current to flow
through capacitors 116a and 116b and feedback windings 108 and
112 to the base and collector junctions of transistors 106 and
110 respectively. This spike voltage also causes a current to
flow through capacitors 104 and 109 to the emitter and collector
junctions of transistors 106 and 110 in the reverse direction.
Accordingly, the spike voltage is absorbed in the inverter.
In the Figure 3, a capacitor 118 is connected across
primary winding 105. The said capacitor 118 also serves to
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absorb the spike voltage occurring in primary windiny 105. More
specifically, without capacitor 1~8, it is possible that the
voltage to transistors 106 and 110 will fall outside the region
of safe operation enclosed by dotted line A in Figure 4A of
the load characteristic curve. However, it has been observed
that the addition of capacitor 118 as described above serves to
confine the operation of transistors 106 and 110 within the
region of safe operation A as shown in ~igure 4B. In Figures 4A
and 4B, the abscissa indicates the collector/emltter voltage
(VcE) of the transistors, while the ordinate indicates the
collector current IC of the transistors. Since the operation
of inverter 100 in Figure 3 is the same as that of the funda-
mentàl circuit configuration shown in Figure 2, the details of
the inverter will not be repeated here.
; Figure 5 is a schematic diagram of another preferred
; embodiment of the present invention, wherein an improvement in
the inverter i9 adopted. In comparison with the Figure 3
embodiment, the Figuxe 5 embodiment comprises diodes 119 and 120
coupled between the collector and emitter electrodes of
transistors 106 and 110 respectively in the reverse direction.
As the remaining portions of the Figure 5 embodiment are
substantially the same as those illustrated in Figure 3, only
the salient features of the Flgure 5 embodiment will be
described in greater detail below. Without diodes 119 and 120,
secondary winding 115 provides a positive half-wave output o~
pulsive form as seen from Figure 6B. The no~mal wave form
should be as shown in Figure 6A. Such pulsive portions P cause
an over w ltage to be applied between the base and emitter
electrodes of transistors 106 and 110 in the reverse direction~
whereby transistors 105 and 110 create heat and the wave form
of the output from inverter 100 is distorted. Further, a low
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frequency vibratory noise emanates from the saturable transformer.
;r` Actual measurement show that the wave form of the base/emltter
voltage VBE of -transistor 110, ~or example, is as shown in
Figure 7A. Similarly, the wave forms of the base current IB,
the collector current IC and the emitter current IE f
transistor 110 are shown in Figures 7s, 7C and 7D, respectively.
Thus, a conduction period of transistor 110 is shown as W in
Figure 7A. Referring to Figure 7A, it is supposed that the
base/emitter voltage VBE should be positive in response to the
rise of the base current IB and should remain positive as shown
by the dotted line. However, in actuality, a negative valley
portion appears, represent~d by the solid line in Figure 7A. It
is presumed that the negative voltage portion cau~es the above-
described pulsive portions P.
When transistor 110 is shifted from the non-conduction
state to the conduction state, a current flows through primary
winding 105 in a direction from point a to point b~ However,
before that, a current has been flowing through primary winding
105 in a direction from point b to point a by virtue of the
conductivity of transistor 106. As a result, a counter electro-
motive force is generated across primary winding 105. The said
counter electromotive force causes a current I105 to flow ~hrough
primary winding 105 in the negative direction, i.e., in the `
direction from point b to point a, as shown in Figure 7~,
irrespective of the conductivity of transistor 110. The wave
forms of the current I116b of capacitor 116b, the current I117b
of resistor 117b and the current Illl of base resistor 111 are
shown in Figures 7E, 7F and 7G, respectively. Taking into
consideration the above-described wave forms, it is presumed that
the above-described negative current in primary winding 105 flows
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from the base electrode to the collector electrode of transistor
110 through capacitor 116b and resistor 117b of the parall 1
circuit and the feedback winding 112, whereby transistor 110 is
reverse-biased and the negativé current flows through dividing
capacitor 109 and through the emitter/collector junction o~ tran-
5istor 110. In other words,it is presumed tha the ne~ative current
causes a current to instantaneously flow through transistor 110
in the reverse direction, whereby transistor 110 instantaneously
serves as a so-called backward transistor. ~ence, the negative
voltage portion shown in Figure 7~ in base/emitter voltage VBE.
It is presumed that pulsive portions P are generated to cut off
transistor 110 which is about to become conductive due to the
counter electromotive forceO
In the embodiment shown, the counter electromotive
force occurring in primary winding 105 bypasses transistor 110
: by means of diode 120. Hence, a current is prevented from
flowin~ through the emitter/collector junction. ~s a result,
the output of the wave form as shown in Figure 6A is obtained
from secondary winding 115, wherein the pulsive portions P shown
in Figure 6B have disappeared. Thus, the wave forM of the output
from inverter 100 is not distorted and the low-frequency
vibratory noise from the transformer lS eliminated. The wave
form of the base/ mitter voltage VBE of transistor 110 in such
a situation is shown in Figure 7I and the wave form of the
current I120 of diode 120 is shown in ~igure 7J. Although in
the foregoing only one transistor 110 and diode 120 were
described, the same applies to the tran~istor 106 and the diode
119.
As described above 9 wave distortion of the output
o~ inverter 100 and transform~r vibratory noise due to spike
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voltage are eliminated by the use of diodes 119 and 120
connected in reverse direction between the emitter and collector
electrodes of transistors 106 and 110.
As will also be appreciated, if transistors 106 and 110
are selected to have reverse characteristics, then diodes 119
and 120 can be dispensed with. Specifically, since diodes 119
and 120 are connected to bypass a reverse directional current
between the collector and emitter electrodes of transistors 106
and 110, diodes 119 and 120 can be dispensed with, if transistors
106 and 110 each function satisfactorily as backward transistors.
Figure 8 shows a schematic diagram of a further
preferred embodiment of the present invention, wherein it is
contemplated that diversified current amplification factors of
transistor 106 and 110 are absorbed by connecting a pair of
RC time constant circuits 121 and 122 to the base biasing
circuits of transistors 106 and 110. The RC time constant
-` circuits 121 and 122 comprise a series connection of a
resistor 121a and a capacitor 121b and a series connection
of a resi~tor 122a and a capacitor 122h, respectively. These
resistors 121a and 122a each function as a charging/discharging
resis-tor of the corrésponding capacitors 121b and 122b, res-
pectively. More specifically, the charging/discharging time
constant of each of the circuits 121 and 122 is determined
by the resistor 121a and the capacitor 121b, and the resistor
122a and the capacitor 122b, respectively. A further
modification illustrated in Figure 8 is the noise filter
circuit 23 which includes a choke coil and a capacitor inter-
posed between an alternate current voltage source 21, such
as a commercail voltage source, and bridge circuit 22. The
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above-described time constant circuits 121 and 122 are inter-
posed in the base biasing circuits of inverter 100.
In opera-tion, when switch 103 iS turned on~ a
current flows through the pair of time constant circuits
121 and 122 and capacitors 116a and 116b. The voltage drops
in respective time constant circuits 121 and 122 cause
transistors 106 and 110 to be turned on. In such a situation,
because of the diversified characteristics of circu.it
components SUCh as transistors 106 and 110, feedback windings
108 and 112 and time constant circuits 121 and 122, either
transistor may become
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conductive before the other. Assuming thak transistor 106
becomes conductive ~irst, then substantially the same
operation as is described in conjunction with the embodiment
: of Figure 2 follows thereafter. When the other transistor 110
becomes conductive, then collector current IC flow8 as shown
in Figure ~ and the saturable magnetic core 114 of the trans-
former becomes magnetically saturated immediately before the
collector current Ic disappears. As a result, a peak current
Ic, as shown by the dotted line in Figure 9 flows. The voltage
- 10 V109 of dividing capacitor 109 abruptly decreases at such peak
point to become voltage V109,. Assuming that capacitor 116b is
not connected, the fluctuation ~V of the base voltage of
transistor 110 in such voltage transition can be expressed by
. the following equation:
~V = 122a -- (V
R122a R117b 109 109'
:~ where R122a and R117b are the resistance values of resistors
122a and 117b.
Since the abo~e-described fluctuation V is small,
: the base current of transistor 110 does not abruptly decrease
and the collector current gives rise to a peak current Ic, as
shown by the dotted line in Figure 9. By contrast, with
capacitor 116b connected, since capacitor 116b is charged to the
voltage V116b, the fluctuation ~V' o~ the base voltage of
transistor 110 when dividing capacitor 109 is discharged -through
primary winding 105 may be expressed by the followi~g e~uation:
( log V116b) - (v~og 1 ~ Vll~b)
; V109 Vlos I
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The fluctuation av~ is larger than the fluctuation ~V
described previously and can decrease the base current of
transistor 110~ As a result, the collector current IC exhibits
: certain characteristics as shown by the solid line in Figure 9,
wherein no peak characteristic is seen. The same applies to the
collector current of transistor 106. Further, the current load
and heat in respective transistors 106 and 110 can be mitigated.
According to the embodiment shown, the collector
current IC of each of transistors 106 and 110 exhibits
characteristics without a peak as shown in Figure 9. However,
the cut-off time point t is differen~ depending on the value of
the current amplification factor of the respective transistors
106 and 110. More specifically, and referring to Figure 10, the
cut-off time point t2 of the ~ransistor having a smaller
current amplification factor comes later than the cut-off time
- point tl of the transistor having a larg~r current amplification
factor. As a result, a difference ~t in the conductlon period
of the transistors of different current amplification factors
occurs. This entails asymmetry in the positive- and negative~
going directions in t~e inverter output. For example, refe.rring
to Figure 8, assuming that transistor 106 has a smaller current
ampliication factor compared with transistor 110, then the
base switching voltage or the base threshold value of transistor
: 106 ls largex than that of tran~istor 110. Therefore, the
voltage V104 o~ dividing capacitor 104 corre~ponding to
translstor 106 is higher than the voltage Vlog of the other
dividing capacitor 109. It follows that supply voltages V10~
andV1O9 will differ ~rom each other even when the time constants
of time constant circuits 121 ana 122 provided in ~he base
30 biasing circuits of transistors 106 and 110 are the sameO
- 17 ~
Accordingly~ the voltages V121b and V122b
121b and 122b exhibit the characteristics shown in Figure 11~
One transistor 106 is rendered non~conductive by virtue of the
base voltage corresponding to the v~ltage vl21b, in Figure 11
: and similarly transistor 110 is rendered non-conductive by virtue
of the base voltage corresponding to the voltage V122b, in
Figure 11. Accordingly, the conduction periods of transistors
106 and 110 become substantially the same even when their curre~t
amplification factors differ from each other~ A diversified
difference in temperature increase of transistors 106 and 110
is also reduced.
Figure 12 is a schematic diagram of still another
preferred embodiment of the present invention, wherein capacitors
116a' and 116b' are connected in parallel with the r spective
base resistors 107 and lllr In previously described embodiments,
capacitors 116a and 116b were described as being connected in ~.
parallel with resistors 117a and 117b. The modification of
~: the circuit of Figure 12 brings abouk the following advantages.
In the case where capacitors 116a and 116b are connected in
parallel with resistors 117a and 117b as shown in Figure 3,
the capacitors 116a and 116b mus~ possess an increased withstand
voltage, inasmuch as the resistance values of resistors 117a ana
117b are larger than those of base resistors 107 and 111. Such
capacitors are large and costly. By contrast, according to the
FLgure 12 embodiment, since capacitors 116a' and 116b' are
coupled in parallel with base resistors of decreassd resistance
.
; value, capacitors 116a and 116b may possess a smaller withstand
chara~teristic and may therefore be smaller-sized and less
. .
co0t1y as well. This allows for the provision of a power supply
apparatus employing an inexpensive inverter.
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ss~
Figure 13 illustrates load circuit 200 coupled to
secondary winding 115 of the saturable transformer. The direct
current voltage source 2 and inverter 100 may be of the type
already described above. Therefore, only load circuit 200
will be described in detail below.
Each terminal of secondary winding 115 of the
saturable transformer is coupled to one terminal of bat~ery
203 and direct current motor 205 through the base/collectox
junction$ of planar transistors 207 and 208, the base and emitter
electrodes of which are short-circuited in the forward direction.
The other ends of battery 203 and ~irect current motor 205 are
connected to first and second switch contacts 211 and 213.
Central tap 115a of secondary winding 115 is connec~ed to third
contact 212 and one end of secondary winding 115 is connected to
a fourth switch contact 210 through a limiting resistor 209.
A switch short-circuiting piece 206' is switched to the solid
line position when direct current motor 205 is ~o be utilized
When switch 206' i5 turned on, the output of inverter 100 is
full-wave rectified by the base/collector junctions of
transistors 207 and 208 through central tap llSa and direct
current motor 205 is energized with a half of the full~wave
rectified output from inverter l00. Battery 203 is charged
simultaneously. On the other hand, if direct current motor 205
is not to be utilized, switch short-circuitiny piece 206' is
switched to the dotted line position and battery 203 i5 charged
by the half-wave rectified output of the output voltage from
secondary winding 115,
Consideriny that the number of secondary winding turns
of secondary winding 115 is extremely small, say about 10 turns,
the circuit i9 configured such that hattery 203 is charged with
a half of the output voltage of inverter 100. Accordingly, the
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: . . . ... . .
. . ..
value of limiting resistor 204 in Figure 3 may be small. But
the fluctuation of the battery chargi~g current by virtue of any
fluctuations in the A/C voltage source is relatively large, as
shown by the line X in Figure 14 . By contras t, according to
: the embodiment shown, since battery 203 is adapted to be charged
- with the ~ull output voltage of invertex 100 appearing across
secondary winding 115, the value of limiting resistor 209 can
be selected to be larger, say several times larger, as compared
with tha~ of limi~ing resistor 204 in Figure 3. As a result,
fluctuation of the battery ~harging current with respect to the
fluctuation of the commercial power source voltag~ is reduced as
shown by line Y in Figure 14.
Generally speaking, a diode ls structured such that a
pellet is connected between two external lead wires. Therefore,
if the diode is damaged, there is a good chance that the two
external lead wires will be closed rather than opened. Accor~
dingly, if direct current motor 205 or battery 203 are
short-circuited, a large short-circuit current will flow through
secondary windlng 115, damaging the components of inverter 100~
20 To prevent this possibility, planar type transistors 207 and 208
;are employed in place of conventional diodes, wherein base/
collector junctions having a rectifying capability are utilized.
Appropriate planar type transistors are structured such that a
bonding~wire or internal lead wire of 30 microns thickness is :~.
: connected between the external lead wire and the pellet.
Accordingly, if a planar type transistor is employed, the
amplitude of the collector current lS restricted by the thinness
of the wire. It follows that should direct current motor 205 or
. : ~
: battery 203 become short~circuited, the bonding wire will melt
30 and the circuit will be opened to prevent damage to the components
of inverter 100.
Although the bas~/emitter junction o~ a transistor can
be considered as being similar to an ordinary diode, ~he voltage
VEBO, i.eO, the voltage between the base and emitter electrodes
when the base electrode is opened, is lower and the transistor
cannot be utilized as a diode when the induced voltage acxoss
secondary winding 115 is large. In order to eliminate this
inconvenience, the embodiment shown employs the base/collector
junctions of transistors 207 and 208 for the purpose of
rectification.
When load circuit 200 comprises a battery to be
char~ed, it becomes important to avoid overcharging the battery.
Figures 15 through 17 show two embodiments for that purpose.
In Fig. 15, a switching device 306 is connected between
the base and emitter electrodes of transistor 110 of inverter 100.
The battery 203 is shunted by a series connection of a resistor
302 and a constant voltage device 303 and a detector 301 for
detecting whether the battery charge level is approaching a
predetermined value. Detector 301 is structured to detect a
point where the voltage of battery 203 associated with the
battery-charge level reaches a reference voltage determined by
constant voltage element 303O Detector 301 may comprise an
operation amplifier, for example. Detector 301 is provided at
the output thereof with an active element 305 for closing the
above-described switching device 306. In the embodiment shown,
a photocoupler 304 is employed wherein switching device 306
:: ~ may;be a ~phototransistor while active device 305 may be a light-
emitting diode. i
In operation, when direct current voltage source 2
.
and inverter 100 are operative as described above in relation
~30 to Fig. 3, battery ~03 is charged ~y the outpu~ from the inverter.
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The output of the inverter is full-wave rectified by means of
diodes 201 and 202. When b~ttery 203 is charged and the charged
voltage increases to reach the reference voltage of constant
voltage element 303, then active device 305 is rendered
operative in response to the output of detector 301, thereby to
emit light. Switching device 306 is shunted by the light to
become conductive~ Accordingly, the base and emitter electrodes
of transistor 110 are short-circuited and the voltage induced
across feedback winding 112 is not applied between the base
and emitter electrodes of transistor 110 any longer. Transistor
110 is thereby rendered non-conductive. Since transistor 110
is rendered non-conductive, the charging circuit of dividing
capacitor 104 is interrupted, the dividing capacitor 104 is
no longer charged and transistor 106 becomes non-conductive~
As a result, the oscillating operation of inverter 100 is stopped
and hence the charging operation of battery 203 i9 also stopped.
Figure 16 is a schematic diagram of a further
preferred embodiment of the present invention designed to prevent
the overcharging of a battery. In comparison with the
embodiment of Figure 15, th~ embodiment of Figure 16 comprises
the following modifications7 As illustrated in Figure 15,
switching device 306 was interposed between the base and emitter
electrodes of transistor 110 so that transistor 110 of inverter
100, could be interrupted in response to the charge level of
battery 203. By contrast, according to the Figure 16 embodiment,
two active elements 305a and 30S~, ~uch as light-emitting
diodes, are coupled to the output of detector 301 In addition,
bidirectional switching device~ 3~6a and 306b, which may be
phototransistors, are sn/o~f controlled in response to active
devices 305a and 305b. Thus, each o~ switching devices 306a and
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,
5Z
306b constitutes a series connection together with capacitors
307a and 307b respectively. The series connection of capacitor
307a and switching device 306a and the series connection of
capacitor 307b and switching device 306b are each coupled between
one end of feedback windings 108 and 112 and the junction a.
In operation, when switch 103 is turned on and
inverter 100 is brought to an operating state, battery 203 is
charged by the output from inverter 100 as rectified by diodes
201 and 202. When battery 203 is charged and the battery voltage
increases to reach the reference vol~age of constant voltage
device 3Q3, active devices 305a and 305b, comprising light-
emitting diodes respond to the output of det ctor 301 by
emitting light and bidirectional switching devices 306a and 306b
(phototransistors) are shunted by the light from light emitting `
diodes 3~05a and 305b become conductive. Accordingly, because
capacitors 307a and 307b are coupled in parallel with
capacitors 116a and 116b, the composite capacitances of
capacitors 116a and 307a and the capacitors 116b and 307b become -
larger. Capacitors 116a, 307a, 116b, and 307b are charged due
to the counter-electromotive force flowing throu~h feedback
windings 108 and 112 and the base/collector junctions of
transistors 106 and 110 respectively, More specifically, and
with re~spect to capacitors 116b and 307b, in the case of a
transition from a state where transistor 106 is turned on and
tranRistor 110 is turned off to a situation where transistor 106
-~ ~ is turned off and transistor 110 is turned on, the capacitors
116b and 307b are charged due to the counter electromotive
~orce flowing throu~h feedback winding 112 and the base/collector
junction of transistor 110. Since the composite capacitance
of capacitors 116b and 307b in such a si tuation i5 larger than
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.
the capacitance of capacitor 116b only, the reverse bias period
of transistor 110, which is about to be turned on, increases.
As the peak value of the full-wave rectified output from the
output terminals of direct current voltage source 2 increases,
the base/emit~er junc~ion of transistor 110 is forward biased.
Therefore, the conduction period of transistor 110 commences
when the peak value of the full-wave rectified output becomes
high and the output from diodes 201 and ~02 is confined to the
period T shown in Figure 17, resulting in a supplemental
charging state. Accordingly, it is possible to detect whether
the battery is being charged at a predetermined rate. Further,
a reverse bias period of the transistor is provided by increasing
the capacitor capacitance of the impedance circuit of inverter
100. Thus, the on period of transistors 106 and 110 is
shortened and the charging state is switched from rapid charging
to supplemental charging~ '~he usable range of a charging
apparatus employing a half-bridge-type inverter can therefore be
expanded.
It will be appreciated that the combination of active
device 305 (305a and 305b~ and the switching device 306
(306a and 306b) may differ ~rom that described above to
comprise, for example, a combination of a relay coil and the
contacts thereof.
Generally, a power supply apparatus employing a push-
pull type inverter yields a large oscillatory output from the
inverter which is of a rectangular wave form. This type of
output is likely to cause noi~e from the inverter. Assuming
that audio equipment is placed near a small electrical device
including a charging circuit, a ~pace capacitance or coupling
may form between a battery in the load of the inverter and the
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;2
audio equipment. The noise transmitted through such a space
capacitance is liable to adversely affect the audio equipment.
Such noise comprises a common mode noise which may be defined
as noise occurring between ground and the electrical equiprnent.
Such noise is different from normal mode noise which occurs
between two lines in the equipment. Common mode noise is
difficult to remedy irrespective o~ an increased magnitude and is
more difficult to overcome than normal mo~e noise.
The embodiment of Figure 18 is intended to overcome or
mitigate the problem of common mode noise. To that end, a noise-
absorbing capacitor 308 is provided. One end of noise-absorbing
capacitor 30~ is connected to casing 203a of battery 203 or the
casing ~05a of direct current motor 205. The other end of
noise-absorbing capacitor 308 is connected to the rectifica~ion
output terminal of the full-wave rectifyi~g circuit 22 or the
output terminal of direct current voltage source 2. Capacitor
308 serves to isolate in a direct current manner the primary
secondary side of the saturable transformer whil~ coupling them
in an alternate current manner. Accordingly, noise appearing
on primary winding 105 is transferred through secondary winding
115 to casing 203a of battery 203 or to casing 205a of direct
current motor 205. The noise from casing 2~3a or casing 205a is
absorbed on the primary side of the transfoxmer by means of -~
said capacitor 308. Experimentation shows that without
capacitor 308, noise charac~eristics shown by the dotted line
in Figure 19 are ob~ained, whereas with capacitor 308, noise
characteristics as shown by the solid line are obtained. Use
of~capacitor 30d results, therefore, in a decrease of noise
levels by several deci~els.
Although the foregoing descriptions of various
... ..... ..
-- 2 5
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9~
embodiments centre mainly on the characteristic features o~
each respective embodiment, it will be appreciated that a power
supply apparatus suitable for any particular use can be provided
by a proper combination of the above-described features.
It is suggested that the present invention is
particularly suited as a power supply for a portable electric
shaver. In general, an electric shaver comprises a casing having
an opening, a shaver cutter assembly provided so as to be exposed
through the opening, and a prime mover such as an electric motor
for driving the shaver cutter assembly. Typically, the shaver
cutter assembly comprises a stationary cutter mounted in the
shaver casing so as to be exposed through the opening and a
movable cutter provided to be movable with respect to the
stationary cutter. The prime mover may comprise a direct current
motor coupled to a direct current voltage supply. According ~o
one prior art approach, the casing is formed of a space for
housing a dry cell or a rechargeable battery for providing a
direct current voltage output~ Alternatively, a prior art
electric shaver is structured to be adaptably connected to a
separate AC adapter, which is structured to convext an alternate
current voltage from a commercial power supply into a direct
current voltage suited for driving a direct current motor. Since
prior art AC adapters are bulky, it was impossible to include
both a rechargeable battery and an AC adapter within the shaver
casing without increasing the bulkiness of the shaver casing.
~b~iQu~ly~ it would be de~irable to include both a reahargeable
battery and an AC adapter within the sha~er casing. The
- present power supply is suf~iciently compact and lightweight to
afford this advantage. More speci~ically, referring to Figure 3,
` 30 for example, bat~ery 203 can be used as a rechargea~le battery for
- 26
,
5~
energizing the razor's direct current motor. The direct current
motor 205 can be used to drive ~he movable cutters of the shaver
cutter assembly. The present power supply, including in~erter
circuit 100, bridge circuit 22, rechargeable battery 203 and
direct current mo~or 205 can all be housed within a shaver
ca~ing of ordinary size.
~ ` "' '
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