RECEPTEUR TRT FM A LIMITEUR OSCILLANT AMELIORE

FM TVRO RECEIVER WITH IMPROVED OSCILLATING LIMITER

Abrégé anglais

IMPROVED FM RECEIVER
ABSTRACT
An improved FM receiver is disclosed incor-
porating improved oscillating limiter circuitry having
an electrically tunable bandpass filter which is phase
modulated by the baseband signal the phase of which
has been advanced in accordance with the formula:
<IMG>
wherein td1 equals the time delay through the limiter,
td2 equals the time delay through the feedback filter,
td3 equals the time delay through the demodulator and
fn equals the frequency of the respective baseband
components.

Revendications

We Claim:
1. A circuit for reducing noise in an FM signal
modulated by a plurality of baseband signal
components, each at a respective frequency fm,
comprising:
a limiter having an input adapted to receive
such an FM signal and an output, the limiter having a
first time delay td1;
feedback means coupling the limiter output
to the limiter input and responsive to control signals
for phase modulating signals fed back therethrough by
the control signals and for providing regenerative
feedback around the limiter, the feedback means having
a second time delay td2;
control means responsive to the limiter
output for generating the control signals, the control
means including first means having a third time delay
td3 for generating first signals corresponding to
baseband signal components of a received FM signal,
each at a respective frequency fn, and second means
advancing the phase of each of the first signals by a
respective amount ?n wherein
?n approximately equals 360° <IMG>
for generating the control signals.
-34-

-35-
2. The circuit of claim 1 wherein fn = fm.
3. The circuit of claim 1, the feedback means
including a bandpass filter having a center frequency,
the second means further for generating a DC said
control signal which is correlated to an average over
time of the frequency of the first signals, the
bandpass filter being responsive to the DC control
signal whereby to electrically offset the center
frequency by an amount approximately equal to fn - fm.

4. A circuit for reducing noise in an FM signal
modulated by at least one dominant baseband signal
component at a frequency fm, comprising:
a limiter having an input adapted to receive
such an FM signal and an output, the limiter having a
first time delay td1;
feedback means coupling the limiter output
to the limiter input and responsive to a
control signal for phase modulating signals fedback
therethrough by the control signal and for providing
regenerative feedback around the limiter, the feedback
means having a second time delay td2;
control means responsive to the limiter
output for generating the control signal, the control
means including first means for having a third time
delay td3 for generating a first signal at a frequency
fn and corresponding to a dominant baseband signal
component of a received FM signal, and second means
for advancing the phase of the first signal by an
amount ? wherein
? approximately equals 360° <IMG>
for generating the control signal.
36

-37-
5. The circuit of claim 4 wherein fn = fm.
6. The circuit of claim 4, the feedback means
including a bandpass filter having a center frequency,
the second means further for generating a DC control
signal which is correlated to an average over time of
the frequency of the first signal, the bandpass filter
being responsive to the DC control signal whereby to
electrically offset the center frequency by an amount
approximately equal to fn - fm.
7. The circuit of claim 4, the second means
further for cutting-off any said first signal having a
frequency much above fn.
8. The circuit of claim 7 wherein fn = fm.
9. The circuit of claim 7, the feedback means
including a bandpass filter having a center frequency,
the second means further for generating a DC control
signal which is correlated to an average over time of
the frequency of the first signal, the bandpass filter
being responsive to the DC control signal whereby to
electrically offset the center frequency by an amount
approximately equal to fn - fm.

-38-
10. The circuit of claim 4, wherein the FM
signal is a television signal and the dominant signal
component is at approximately 3.58 MHz.
11. The circuit of claim 10, ? approximately
equal to 45°.
12. The circuit of claim 10, the second means
further for cutting off any said first signal having a
frequency above approximately 5.4 MHz.

-39-
13. A receiver adapted to receive FM signals
modulated by a plurality of baseband signal
components, each at a respective frequency fm, and to
demodulate the FM signals to generate signals
corresponding to the baseband signal components, the
receiver comprising:
first means for receiving such FM signals;
mixer means for mixing received FM signals
with at least one local oscillator signal to generate
intermediate frequency FM signals modulated by a
plurality of baseband signal components, each at a
respective frequency fm;
a limiter having an input adapted to receive
the intermediate frequency FM signals and an output,
the limiter having a first time delay td1;
feedback means coupling the limiter output
to the limiter input and responsive to control signals
for phase modulating signals fedback therethrough by
the control signals and for providing regenerative
feedback around the limiter, the feedback means having
a second time delay td2;
control means responsive to the limiter
output for generating the control signals, the control
means including first means having a third time delay
td3 for generating first signals corresponding to
baseband signal components of a received FM signal,
each at a respective frequency fn, and second means

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advancing the phase of each of the first signals by a
respective amount ?n wherein
?n approximately equals 360° <IMG>
for generating the control signals.
14. The circuit of claim 13 wherein fn = fm.
15. The circuit of claim 13, the feedback means
including a bandpass filter having a center frequency,
said second means further for generating a DC said
control signal which is correlated to an average over
time of the frequency of the first signals, the
bandpass filter being responsive to the DC control
signal whereby to electrically offset the center
frequency by an amount approximately equal to fn - fm.

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16. A receiver adapted to receive FM signals
modulated by a plurality of baseband signal
components, each at a frequency fm, and to demodulate
the FM signals to generate signals corresponding to
the baseband signal components, one of the baseband
signal components being dominant and at a frequency
fd, the receiver comprising:
first means for receiving such FM signals:
mixer means for mixing received FM signals
with at least one local oscillator signal to generate
intermediate frequency FM signals modulated by a
plurality of baseband signal components, each at a
respective frequency fm one dominant baseband signal
component being at a frequency fd;
a limiter having an input adapted to receive
the intermediate frequency FM signal and an output,
the limiter having a first time delay td1;
feedback means coupling the limiter output
to the limiter input and responsive to a control
signal for phase modulating signals fedback
therethrough by the control signal and for providing
regenerative feedback around the limiter, the feedback
means having a second time delay td2;
control means responsive to the limiter
output for generating the control signal, the control
means including first means having a third time delay
td3 for generating a first signal at a frequency f1

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and corresponding to a dominant baseband signal
component of a received FM signal, and second means
for advancing the phase of the first signal by an
amount ? wherein
? approximately equals 360° <IMG>
for generating the control signal.
17. The circuit of claim 16 wherein f1 = fd.
18. The circuit of claim 16, the feedback means
including a bandpass filter having a center frequency,
the second means further for generating a DC control
signal which is correlated to an average over time of
the frequency of at least the first signal, the
bandpass filter being responsive to the DC control
signal whereby to electrically offset the center
frequency by an amount approximately equal to f1 - fd.

-43-
19. The circuit of claim 16, the second means
further for cutting-off any said first signal having a
frequency much above f1.
20. The circuit of claim 17 wherein f1 = fd.
21. The circuit of claim 19, the feedback filter
means including a bandpass filter having a center
frequency, the second means further for generating a
DC control signal which is correlated to an average
over time of the frequency of at least the first
signal, the bandpass filter being responsive to the DC
control signal whereby to electrically offset the
center frequency by an amount approximately equal to
f1 - fd.

22. A method of reducing noise in an FM signal modulated
by a plurality of baseband signal components, each at a
respective frequency fn, comprising:
a) amplitude limiting the FM signal through a
limiter;
b) demodulating the amplitude limited FM signal to
generate first signals corresponding to the baseband signal
components, each at a respective frequency fn;
c) feeding back the amplitude limited FM signal
through a feedback network to the limiter;
d) phase modulating the feedback amplitude limited FM
signal in the feedback network with the first signals each of
which has been phase advanced by a respective amount ?n wherein
?n approximately equals 360° times the sum of any time delays
in steps a)-c) divided by the inverse of the frequency fn of
the said first signal.
23. A method of reducing noise in an FM signal modulated
by at least one dominant signal component at a frequency fn,
comprising:
a) amplitude limiting the FM signal through a
limiter;
b) demodulating the amplitude limited FM signal to
generate a first signal corresponding to the dominant baseband
signal component at a frequency fn;
c) feeding back the amplitude limited FM signal
through a feedback network to the limiter;
44

d) phase modulating the feedback amplitude limited FM
singal in the feedback network with the first signal which has
been phase advanced by a respective amount 0 wherein 0
approximately equals 360° times the sum of any time delays in
steps a)-c) divided by the inverse of the frequency fn of the
first signal.
24. A circuit for reducing noise in an FM television
signal modulated by at least a chroma signal component at a
frequency fn, comprising:
a limiter having an input adapted to receive such an
FM television signal, and an output, the limiter having a first
time delay td1;
feedback means coupling the limiter output to the
limiter input and responsive to a control signal for phase
modulating signals fedback therethrough by the control signal
and for providing regenerative feedback around the limiter, the
feedback means having a second time delay td2;
control means responsive to the limiter output for
generating the control signal, the control means including
first means having a third time delay td3 for generating at
least a first signal at a frequency fn and corresponding to a
chroma signal component of a received FM television signal, and
second means for advancing the phase of the first signal by an
amount ? wherein
? approximately equals 360° <IMG>
for generating the control signal.

25. The circuit of claim 24, the second means further for
cutting-off any signal generated by said first means having a
frequency much above fn.
26. A method of reducing noise in an FM television signal
modulated by at least a chroma signal component at a frequency
fn, comprising:
a) amplitude limiting the FM television signal
through a limiter;
b) demodulating the amplitude limited FM television
signal to generate at least a first signal corresponding to the
chroma signal component at a frequency fn;
c) feeding back the amplitude limited FM signal
through a feedback network to the limiter;
d) phase modulating the feedback amplitude limited FM
signal in the feedback network with the first signal which has
been phase advanced by a respective amount ? wherein ?
approximately equals 360° times the sum of any time delays in
steps a)-c) divided by the inverse of the frequency fn of the
first signal.
27. The method of claim 26 further comprising:
e) cutting-off from reaching the feedback network
signals generated at a frequency much above fn by said
demodulating.
46

Description

~ ~f~ p3~
Bac~ground of the Invention
This invention relates to FM receivers, and
particularly to F~ receivers adapted to receive wea~
signals in the presence of noise such as encountered
in reception of television signals transmitted via
satellite. Specifically, the present invention
relates to receivers for satellite transmitted tele-
vision signals, i.e., television receive only (T~RO)
receiver utilizing noise threshold extension circuitry
of the oscillating limiter type.
Information signals such as television
signals having a variety of frequency components may
be frequency or angle modulated onto a carrier signal
of predetermined frequency. A typical television
signal includes among other information, a color burst
or chroma signal at a frequency of about 3.58 MHz, and
may further include audio subcarriers in a range of

--2--
frequencies between approximately 5.4 MHz and approxi-
mately 8.5 MHz. The resulting frequency modulated
(FM) signal is of a predetermined bandwidth centered
about the frequency of the carrier signal. The FM
signal may be transmitted from an earth-bound
transmitter to an orbiting satellite and subsequently
retrans~itted from the satellite to earth receivlng
stations. The earth receiving station might normally
include a reflector antenna configured to receive the
satellite signal. The reflector antenna is typically
coupled to a low noise amplifier which is further
coupled to a receiver such as a FM TVRO receiver. The
FM TVRO receiver is designed to demodulate the
television signal from the carrier signal. To that
end, the FM receiver will typically include circuitry
to heterodyne, super-heterodyne, or otherwise mix the
signal received at the antenna dish with one or more
local oscillator signals to produce an intermediate
frequency (IF) FM signal which can more easily be
operated upon by conventional receiver circuitry. The
IF signal is still an FM signal, albeit at a lower
frequency as is well understood. That is, the FM
signal transmitted by the satellite may have a certain
carrier frequency in the several gigahertz range
whereas, in television satellite communication
systems, the IF signal may typically be centered about
70 MHz with a bandwidth of approximately 30 MHz, i.e.,
about 55 MHz to 85 MHz.

~L2 ~3~
--3--
The IF signal is subse~uently coupled to a
demodulator or detector where it is demodulated to
reproduce as nearly as possible the original
modulating or baseband signal. This resultant
demodulated, or baseband, signal is provided to a
television monitor for viewing, or may be further
process as desired.
In a television satellite transmission
system, the baseband signal is, ideally, O to 8.5 MHz
and includes the video, audio subcarrier and related
information signals only. Within the baseband, the
continuous video region from below about 30 Hz to
about 4.2 MHz is of primary interest in the
demodulation of 525 line television formats, such as
NTSC, whereas for 625 line television format, such as
PAL, the continuous video region of primary interest
is from below about 25 Hz to about 5.0 MHz. The
region lying between the upper end of the video
baseband (about 4.2 MHz or about 5.0 MHz) to the
typical baseband upper limit of about 8.5 MHz is
normally used for the transmission of relatively
narrow band FM subcarriers, a common format having the
same transmission parameters as broadcast FM
transmissions. These subcarriers are normally detect-
ed by suitable narrow band detectors, and because of
the reduced bandwidth-typically associated therewith,
are more resistant to the effects of noise. Wideband
subcarriers may also be encountered.

3~;3~
In reality, due to the nature of the satel-
lite communication system involved, the FM signal
appearing at the input to the receiver of the earth
receiving station is typically extremely weak and
accompanied by a substantial amount of electrical
noise. This condition is caused, in part, by the fact
the signal transmitted by the satellite transmitter
must travel a great distance to reach the earth-bound
receiver. As a result, the strength of the
information portion of the signal received may be so
weak as to not be intelligible after demodula~ion.
Compounding the difficulty of receiving such weak
signals is the unavoidable addition of terrestrial
noise to the signal due to objects with non-zero
temperature in the view of the reflector antenna.
Objects with non-zero noise temperature are sources of
electrical noise which can be received by the
reflector antenna. In addition to the terrestrial
noise, a variety of other unavoidable electrical noise
sources are commonly encountered in typical
communication systems, as well as in satellite
communication systems. The net effect is that the
receiver must extract an extremely weak information
signal in the presence of strong noise if satisfactory
results are to be achieved. The foregoing, and the
essentially triangular spectral distribution of noise
in the baseband, result in a baseband or demodulated
signal which is not a true reproduction of the

~ 3~
original television signal but will likely also
contain a great deal of noise, i.e., spurious signals.
Such noise can degrade picture quality and/or audio
fidelity and may even preclude detection of the
information content of the signal.
As mentioned, an FM signal has a center
fre~uency which is the carrier frequency~ Ideally,
the frequency of the FM signal will thus vary about
the center or carrier frequency but the amplitude of
the signal will not vary. Hence, it can be assumed
that amplitude variations (AM) on the received signal
are noise. To eliminate such AM noise, it is common
practice to employ an amplitude limiter between the IF
stage of the receiver and the subsequent demodulator
stage. The amplitude limiter operates to limit
amplitude variations appearing on the FM signal as
appears from the IF stage thus reducing the AM noise
therein, and preventing its conversion to the baseband
output by detector imperfections.
Where the strength of the information signal
received is large compared to the noise in the signal,
an amplitude limiter alone will usually suffice to
sufficiently suppress the AM noise. However, where
the information signal strength is weak compared to
the noise, reduction of the AM noise by the amplitude
limiter will be insufficient for quality picture
reception and/or may adversely affect the weak infor-
mation signal precluding proper demodulation.

g~3~
As a measure of information or the ratio of modulation
inPormation to noise, it is typical to determine the carrier-
to-noise ratio, or CNR. In terms of CNR, about at 12 to 14
dB and higher, the amplitdue limitex is alone sufficient to
suppress AM noise. On the other hand, at CNR levels below
about 12 dB the limiter's capability is usually not adequate
to properly suppress the noise without also affecting the
information signal.
As recognized in u.s. Patent No. 3,909,725 ta saghdady~
at such low CNR, the amplitude limiter's performance can be
greatly improved by providing regenerative (i.e., in-phase)
feedback around the amplitude limiter. Regenerative feedback
results in improved reception by suppressing the noise without
degradation of the information signal. Thus, in U.S. Patent
No. 3,909,725, there is disclosed a feedback amplifier and
filter configured to provide in-phase feedback around the
limiter in the frequency band of interest. The Pilter is
typically a bandpass filter with a bandpass wide enough to
pass the entire FM signal containing the modulating signal.
Such regenerative feedback permits better reception of weaker
information signals in the presence of noise, and,
subsequently, more satisfactory demodulation for viewing
purposes, than previously possible. Hence, the lower limit
or threshold of CNR at which proper reception can occur
MLS/lcm

3~
is extended. This phenomenon or technique is
sometimes, therefore, referred to as threshold
extension.
When regenerative feedback around the
limiter is employed, the circuit will normally tend to
oscillate in the absence of an input signal. Hence, a
limiter having regenerative feedback is often referred
to as an oscillating limiter. This self-induced
oscillation has the added benefit of providing a
squelch to the receiver as described in the aforesaid
Baghdady patent.
The Baghdady oscillating limi-~er concept
appears to substantially lower the CNR threshold at
which proper demodulation can occur, however, its
boundary conditions for proper operation are exceeded
under many satellite television modulation conditions.
Subsequent developments with oscillating limiters have
attempted to further extend the boundary conditions by
providing for electrical tuning of the feedback
filter, referred to as an electrically tunable
bandpass filter. Thus, in U. S. Patents Nos.
4,035,730 and 4,101,837, the feedback filter is an
electrically tunable bandpass filter with a bandwidth
apparently narrower than the IF bandwidth and having a
center frequency nominally set at the IF center
frequency (e.g., 70 MH~). These two patents have
apparently proceeded on the assumption that the
boundary would be further extended by tuning or

3d~
"steering" the center frequency of the filter so that
it tracks or matches the frequency of the FM input
while trying to cause the tuning to ignore the noise
in the signal. The patents describe steering as
follows: the FM signal is demodulated to provide a
baseband signal (and noise); the baseband signal is
then filtered such that the high frequency or noise
components are cut off and the filtered signal coupled
to the feedback filter in an effort to cause the
center frequency of the feedback filter to
substantially match the frequency of the received IF
FM signal: specifically, the steering is to apparently
be done with at least the chroma (3.58 MHz) portion of
the baseband signal but supplied to the feedback
filter with zero phase relative to the limiter output.
The latter patent, U.S. Patent No. 4,101,837,
emphasizes this point by describing the loop delay
~between limiter output and filter control input for
the steering signal) as desirably being 360, i.e.,
substantially in-phase. If such attempts have worked
at all, they have met with only marginal success in
raising the boundary modulation conditions at which
the oscillating limiter will improve reception.
Moreover, an oscillating limiter which uses a feedback
filter having a bandwidth less than the bandwidth of
the FM signal wherein the center frequency is steered
in response to the modulation of the FM signal
apparently also interferes with proper demodulation at

~2'~ 34
high CNR. Hence, in one prior art unit, the feedback
filter is electronically decoupled
from the limiter at C~R greater than about 12 dB CNR.
Accordingly, such attempts to steer the filter may
even result in poorer quality reception rather than
improved reception.
Additionally, with ~he advent of satellite
communications, a further problem has been
encountered. Ideally, each satellite transponder
which is set to a particular channel will operate at
the same nominal or carrier frequency. That ideal is
not always achieved. Hence, the signal to be received
from one satellite may be at the correct nominal
frequency whereas the signal to be received from a
second satellite may be offset slightly in frequency
due to drift or the like. Additionally, the receiving
system may operate with some unwanted frequency offset
of its own due to changes caused by temperature
fluctuation such as in equipment mounted at the
reflector antenna.
Summary of the Invention
I have discovered that efforts to steer or
tune the feedback filter are unnecessary. Indeed, as
indicated in the Baghdady patent, the feedback filter
should be as wide as the bandwidth of the FM signal
(e.g., the IF bandwidth). Hence, steering is
generally unnecessary and may well be futile.
Further, such steerin~ apparently degrades receiver

" 3L2~i3~
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performance, at least at high CNR, requiring circuitry
to effectively remove the feedback at high CN~.
However, in order to effectively utilize the noise
reduction characteristics of the oscillating limiter,
I have discovered that by operating the electrically
tunable feedback filter as a phase modulator, further
reduction in threshold is achieved and circuitry to
eliminate or control the amount of feedback around the
limiter is eliminat~d~ Specifically, I have
determined that the so called steering signal shouLd
not have zero degrees have relative phase as taught by
the 4,035,730 and 4,101,837 patents; such relative
phasing is always non-optimal. Instead, I have
determined that if the components of the demodulated
signal are each phase-advanced in accordance with
specified criteria, and then applied to the feedback
filter, substantial improvement is obtained without
steering the filter and without the need to provide
circuitry to filter or modify the noise in the
baseband. Such advancing phase at 3.58 MHz for a
television signal, or example, is 45 and hence, the
steering signal applied to the filter is not at zero
phase relative to the limiter output.
With appropriate advanced phase to the
steering signal, the present invention provides an FM
r~ceiver with improved-reception, partiGularly when
used to receive weak information signals accompanied
by substantial electrical noise. The present

3~l
invention further provides, in an FM receiver having
an oscillating limiter circuit which is use~ul in
satellite television ground receiving stations,
enhanced picture quality and/or sound reproduction of
a television signal. The present invention also
provides an FM receiver which can satisfactorily
demodulate signals from a satellite transponder even
though it is not operating at precisely its correct
nominal frequency. Further, the present invention
provides an FM receiver which can automatically
co~pensate for undesired frequency offsets in the
receiving system. Thus, the present in~ention
represents an improvement of the Baghdady oscillating
limiter which is superior to other attempted
impro~ements.
Thus, in accordance with the present
invention and in its broadest aspect, the oscillating
limiter includes a feedback filter which phase
modulates the limiter output signal with the
demodulated signal wherein each frequency component of
the demodulated signal has been phase advanced by an
amount Pn which is approximately equal to
360 . td1 + td ~+ td~, wherein tdl equals
the time delay through the limiter, td2 equals the
time delay through the feedback filter, td3 equals the
time delay through the demodulator and fn equals the
frequency of the respective baseband components. In a

specific embodiment adapted for color television, the
major baseband component of concern is approximately
3.58 MHz. In a preferred system, the approximate time
delays are tdl = 2 nsec; td2 = 19 nsec and td3 = 13
nsec. Hence, the phase lead added, 0~, is
approximately 45 for the 3.58 MHz signal.
To compensate for offset, drift and the
like, the center frequency of the electrically tunable
bandpass filter is tunable in response to a DC
component in the detected video or baseband thereby
causing the center frequency of the filter track the
average frequency of the received signal over a
substantial period of time.
Presently, practical feedback networks are
unable to provide both the correct amplitude and phase
simultaneously at the chroma and all subcarrier
frequencies. Future circuitry may allow such
simultaneous optimization, and it should be considered
to be within the scope and spirit of my invention.
Without such proper phasing across the entire band, I
have discovered that some baseband signals are
sufficiently handled by the Baghdady oscillating
limiter without modification, yet such signals, if not
properly phase advanced when supplied to the feedback
filter, may significantly degrade performance of the
oscillating limiter. In connection with a television
signal, such problems may be presented by audio
subcarriers. However, I have found that, because the

lZ)~i3fl~
chroma signal is the dominant signal of interest, the degrading
effects of audio subcarrier power may be overcome merely by
attenuating the subcarrier signals so that they do not
contribute appreciably to the phase modulation of the
electrically tunable bandpass filter.
By the foregoing, I have provided a noise reduction
circuit for an FM receiver which extends the threshold by about
2 dB CNR and does not require additional special handling of
noise nor control over the feedback dependent in any way on the
CNR.
In its method aspect, the invention relates to a
method of reducing noise in an FM television signal modulated
by at least a chroma signal component at a frequency fn,
comprising: amplitude limiting the FM television signal
through a limiter: demodulating the amplitude limited FM
television signal to generate at least a first signal
corresponding to the chroma signal component at a frequency fn;
feeding back the amplitude limited FM signal through a feedback
network to the limiter; and phase modulating the feedback
amplitude limited FM signal in the feedback network with the
first signal which has been phase advanced by a respective
amount ~ wherein 0 approximately equals 360 times the sum of
any time delays in steps a)-c) divided by the inverse of the
frequency fn of the first signal.
Brief Description of the Drawings
These and other features and advantages of the
invention will become more readily apparent from the following
- 13 -
kh/r~

3~
detailed description taken with the accompanying drawings in
whi~h:
Fig. 1 is a block circuit diagram of a preferred
embodiment of an FM receiver according to the present
invention;
Fig. 2 is a schematic drawing of a preferred limiter,
feedback filter, demodulator and phasing circuit of Fig. l;
Fig. 3 is a graph of the amplitude and phase response
of the phasing circuit of Fig. 2;
Fig. 4 is a schematic drawing of an alternative
phasing circuit of Fig. l; and
Fig. 5 is a graph of the amplitude and phase response
of the phasing circuit of Fig. 4.
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kh/R~

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Detailed Description of the Preferred Embodiment
With reference to Fig. 1, there is shown a
block circuit diagram of a preferred embodiment of an
FM TVRO receiver 10 accordin~ to the present
invention. Receiver 10 is driven by a low noise
amplifier 12 which amplifies an FM signal received on
reflector antenna 13. Reflector antenna 13 is sized
to receive satellite transmitted television signals
which are typically in the several gigahertz range.
The output of low noise amplifier 12 is coupled to the
input 14 of FM receiver 10.
As is conventional, the FM signal received
on input 14 is mixed or heterodyned at first mixer 16
with a signal generated by a first local oscillator 18
to generate a first IF signal on output 20. The
frequency of local oscillator 18 is variable so as to
permit tuning of the receiver to the desired channel.
The first IF signal on output 20 is preferably
centered about 250 MHz. As is also conventional, the
first IF signal is further mixed or heterodyned at
second mixer 22 with the output of a second local
oscillator 24 to produce a second IF signal preferably
centered ahout 70 MHz on output 26. The second IF
signal is an FM signal the modulation of which
corresponds, ideally, to the original modulating
signals, i.e., the television signals.
Output 26 is selectively coupled to a narrow
bandwidth 4-pole IF filter 28 or a broader bandwidth

12~
-15-
4-pole IF filter 30 by switch 32. IF filters 28,30
are bandpass filters having a bandwidth of
approximately 15.75 MHz and 30 MHz, respectively. For
typical television signals in the United States, a 30
MHz bandwidth IF filter (30) is usually appropriate.
In some instances, such as with the IntelSat European
communication system, the bandwidth of the information
signal is narrower. Thus, it is more appropriate to
use the narrower bandwidth IF filter 28. In some
situations with U.S. television systems, the
terrestrial noise may be so great as to prevent
satisfactory demodulation unless some of the noise is
reduced. Hence, a very poor or no picture will
result. Use of narrower IF filter 28 will also reduce
some of that noise.
As seen in Fig. 1, receiver 10 is provided
with a switch 32 which is preferably an electronic
switch, as is well ~nown, by which output 26 may be
selectively coupled to either filter 28 or filter 30.
When switch 30 is in a first position (shown in solid
line in Fig. 1), the IF signal path includes narrow
band IF filter 28. When switch 30 is in a second
position (shown in dotted line in Fig. 1~, the IF
signal path includes wide band IF filter 30.
The outputs of IF filter 28,30 are coupled
to a variable gain circuit 34, the output of which is
coupled to amplifier/AM detector strip 36.
Amplifier/AM detector strip 36 is preferably comprised

-16-
of three capacitively coupled SL1613C integrated cir-
cuits (not shown) manufactured by Plessey Solid State,
Irvine, California, and an NPN transistor amplifier
(Q100 in Fig. 2). Amplifier/AM detector strip 36
amplifies the IF signal and supplies the amplified IF
signal on output 38 to the oscillating limiter of the
invention to be discussed. Amplifier/AM detector
strip 36 also provides an AM output 40 which drives an
AGC controller 42 to vary the gain of circuit 34 to
provide automatic gain control as is well understood.
The RF output of the last SL1613C integrated
circuit (not shown) of amplifier/limiter strip 36 is
coupled through NPN transistor amplifier Q100 thereof
to provide an FM output 38 which is coupled to the
input 44 of an amplitude limiter 46 at summing
junction 48. The output 50 of limiter 46 drives
demodulator or detector 52 to generate the baseband
signals. Limiter output 50 is also regeneratively fed
back via summing junction 48 to limiter input 44
through the series combination of electrically tunable
bandpass filter 54 and electronic switch 56. Switch
56 is preferred but may be replaced with an electri-
cally closed circuit (e.g;, a short circuit or the
like) if desired.
Demodulator 52 provides on its output 58 the
baseband or modulating~signal which, in a television
system, includes the composite video (including chroma
at about 3.58 MHz), audio subcarrier and related

3~
signals which are operated upon by the remaining FM
receiver 10 circuitry as is well understood (repre-
sented by block 66). The circuitry of the present
invention is also provided with a phasing circuit 60
which is responsive to the output 58 of demodulator
52. The output 62 of phasing circuit 60 comprises
control signals which phase modulate filter 54 on the
control input 64 thereof. Phasing circuit 60 also
includes circuitry to average the baseband signal from
detector 52 to provide a DC control signal on output
62 as well which DC contro~ signa} may ~e
advantageously utilized to vary the center frequency
of filter 54 to compensate for drift, offset, or the
like affecting the IF center frequency.
The DC component is an integrated response
to the varying frequency output from demodulator 52
and thus is proportional to the average frequency,
i.e. the center frequency, of the IF signal. Filter
54 is tunable in response to this DC component whereby
the center frequency of filter 54 corresponds to the
center frequency of the IF signal corrected for
frequency offsets as discussed above.
The feedback signal from output 50, which is
coupled to input 44 through filter 54, is phase
modulated by the control signals generated by circuit
60. As is well understood, all signals meet with some
time delay as they pass through components of an
electrical system. Limiter 46 generally presents a 2

~2~i3~
-18-
nsec (nanosecond) time delay (tdl) whereas filter 54
presents an approximate 19.8 nsec delay (td2~
(inversely proportional to bandwidth1. Similarly,
detector 52 has a time delay associated therewith of
approximately 13 nsec (td3). I have determined that
by phase modulating the feedback signal with the
baseband signal, wherein each component of the
baseband signal (fn) is advanced in phase (0n) by a
particular amount, dramatic improvement over the prior
art is obtained without added circuitry to compensate
for the affects o~ the shapi~g circuit 60. 0~ is
determined by summing the above three time delays,
dividing by the inverse of the frequency component in
question, and multiplying by 360. Thus,
0n approximately equals 360 . tdl + td2 + td3 -
l/fn
Hence, for example, with respect to the chroma signal
(about 3.58 MHz), 0 is approximately 45, which has
proved successful by experimentation.
FM receiver 10 also includes three power
supplies PSl, PS2, and PS3 to provide a positive power
supply of 12.0 volts, 6.0 volts and 5.2 volts, respec-
tively. Power supplies PSl through PS3 may be powered
by a 120 volt AC line (not shown). The reference
potential of all power supplies PSl to PS3 are tied to
the same point referred to herein as ground.
With reference to Fig. 2, there is shown a
schematic diagram of limiter 46, demodulator 52,

~t~ ~ 3,~
-19-
feedback filter 54, switch 56 and phasing circuit 60.
Limiter 46, electrically tunable bandpass filter 54,
and electronic switch 56 comprise an oscillating
limiter in accordance with the principles of the
present invention. The last stage of amplifier/AM
detector strip 38 is an NPN transistor Q100, the
collector of which is resistively coupled to summing
junction 48 through wiper arm 68 of potentiometer Rl.
Potentiometer Rl also serves to provide DC power from
power supply PS2 to the collector of transistor Q100
a~ t~ e~ectr~nic switch 5Ç, The emi~er of
transistor Q100 is coupled to ground through the
parallel combination of capacitor Cl and series
resistors R2,R3, the junction of which is bypassed to
ground by capacitor C2.
Output 38 of transistor Q100 and output 70
of switch 56 are summed at junction 48 to drive input
44 of limiter 46. Input 44 is capacitively coupled to
a first input 72 of exclusive OR gate 74 which is
powered by power supply PS3. A second input 76 of
gate 74 is coupled to ground. The output 78 of gate
74 is coupled to ground through resistor R4 and is
further coupled to a first input 80 of exclusive
ORINOR 82 also powered by power supply PS3. A second
input 84 of gate 82 is grounded.
Exclusive NOR output 86 and exclusive OR
output 88 of gate 82 provide balanced limiter
outputs which are coupled to inputs 90,92 of

rc~ 3 ~
--20--
demodulator 52, respectively. Exclusive NOR output 86
is also coupled to input 72 of gate 74 through the
series combination of resistor R5 and resistor R6.
The junction of resistors R5 and R6 is coupled to
ground by capacitor C3 and to power supply PS3 by
resistor R8. The DC feedback provided by resistors R5
and R6 services to maintain operation of gates 74 and
82 in the approximate center of their transition
region. Exclusive OR output 88 is also coupled to
electrically tunable bandpass filter 54 through a
series combination of resistor R9, capacitor C4, delay
line ~coaxial ~a~le~ ~4 a~ resistor ~lC. ~he
junction of output 88 and resistor R9 is resistively
coupled to ground by resistor Rll. Similarly, the
~unction of dely line 94 and resistor R10 is resistive
coupled to ground via resistor R12. Finally, input 96
to electrically tunable bandpass filter 54 is coupled
to ground through resistor R13.
Electrically tunable bandpass filter 54
includes the series connection of inductor Ll,
varactor diode Dl, varactor diode D2, and inductor L2.
The cathodes of varactor diodes Dl and D2 are
connected in common at node 98. Connected in parallel
between node 98 and ground is capacitor C5 and two
varactor diodes D3 and D4. The cathodes of varactor
diodes D3 and D4 are also connected in common at node
98. Varactor diodes Dl through D4 may be lSV161
matched diodes available from Matcom, Inc., Palo Alto,

C3~3~
-21-
California. Also coupled to node 98 is the output 62
of phasing circuit 60 through the series combination
of inductor L3 and resistor R14 whereby filter 54 is
responsive on a control input 64 to control signals
generated by phasing circuit 55.
The output 102 of electrically tunable
bandpass filter 54 is coupled to ground through
resistor R15 and capacitively coupled by capacitor C6
to the base of NPN transistor Q102 which is part of
switch 56. The collector of transistor Q102 is the
output 70 of switch 56. Transistor Q102 is configured
to operate ~ike an electronic switch ~s will ~e
discussed below.
The collector of transistor Q102 is biased
by power supply PS2 through potentiometer R1.
Similarly, the bases of transistors Q102 and Q104 are
biased by a voltage divider comprised of resistors R16
and R17 in series from power supply PS2. the base of
transistor Q102 is inductively coupled to the voltage
divider by inductor L4. The emitters of NPN
transistors Q102 and Q104 are connected together and
are resistively coupled to ground through series
resistor R18 and potentiometer Rl9 and switch S1.
Also, the junction of resistor R18 and potentiometer
R19 is capacitively coupled to ground through
capacitor C7 while the~.junction of voltage divider
resistors R16, R17 is further capacitively coupled to
ground through capacitor C8.

--22--
By virtue of the foregoing arrangement, when
switch Sl is in the closed position shown in Fig. 2,
transistors Q102 and Q104 are biased into their active
regions whereby signals from filter 54 can pass to
summing junction 48 via transistor Q102. However,
when switch 51 is in the open position (shown in
dotted line in Fig. 2), transistors Q102 and Q104 are
cutoff. Thus, no si~nal can pass from filter 54 to
summing junction 48 thereby electronically disabling
the oscillating limiter of the present in~ention.
As mentioned previously, limiter 46 has a
time de~ay tdl associated therewith of approximately 2
nsec. Filter 54 actually has a time delay of less
than 19 nsec but in order to provide the necessary
regenerative feedback for proper operation of the
oscillating limiter, the delay through limiter 46,
filter 54, switch 56 and delay line 94 must be modulo
2 Pi, i.e., a multiple of the inverse frequency (in
radians). In this case the IF frequency of 70 MHz
dictates that the delay be a multiple of 14.3 nsec.
In the current implementation, transistor Q102
provides a phase inversion, hence, the required phase
shift is modulo 2 Pi + Pi, i.e., an odd multiple of
7.14 nsec (approximately 21.4 nsec). This condition
is not quite satisfied by the delays inherent in
limiter 46, filter 54`and switch 56. In order to
satisfy the condition, therefore, some delay must be
added which is accomplished here with an approximate

~f~ 3~1
-23-
.8 nsec delay added by delay line 94. Hence, the
total delay from output 50 of limiter 46 to input 44
via filter or feedback path 54 is referred to as the
filter or feedback path delay of td2.
Demodulator 52 and shaping circuit 60 will
now be described. As mentioned, detector 5Z is
preferably a double-balanced phase detector driven by
the outputs from the exclusive OR/NOR logic gate 82 of
limiter 46. I~put 92 o~ demodulator S2 is directly
coupled via a first path 110 to a first input 112 of
an exclusive OR/NOR logic gate 114 which is powered
from power supply PS2. Input 90 is coupled through an
odd multiple quarter wavelength delay line 116, such
as a length of coaxial cable, to a second input 118 of
gate 114. Preferably, delay line 116 is a three
quarter wavelength delay line at 70 MHz. Input 118 of
gate 114 is further coupled to ground through the
series resistors R22 and R23. Resistor R22 terminates
coaxial cable 63. To that end, the junction of
resistors R22 and resistor R23 is coupled to ground
through capacitor C10.
Exclusive OR output 120 and Exclusive NOR
output 122 of gate 114 provide a balanced detector
output. Outputs 120 and 122 are resistively coupled
to ground through resistors R24 and R25, respectively.
Outputs 120 and 122 are further coupled to the base of
NPN transistors Q110 and Q112, respectively, through
identical T-networks each comprised of resistor R26,

~ 3
-24-
capacitor C12, and inductor L10. Further, the base of
each transistor QllO and Q112 is capacitively coupled
to the other through capacitor C13. The emitters of
transistors QllO and Q112 are resistively coupled
through the series combination of resistors R28 and
R29, the junction of which is resistively coupled to
ground through resistor R30.
Gate 114 operates in conjunction with first
path 110 and delay line 116 as a phase detector, the
outputs of which drive the above components (compris-
ing a low pass filter) to couple the video and audio
and related signals to transistors QllO and Q112. The
cutoff frequency of this low pass filter is set
sufficiently high (e.g., 18 MHz) whereby only the
undesired detector outputs components at twice the IF
frequency (hence at 140 MHz) are effected. The low
pass filter thus has no significant effect on the
baseband, which may extend to one-half the bandwidth
of the widest IF filter 30, or 15 MHz.
The collector of transistor QllO is directly
coupled to the collector and base of a first PNP
transistor Q114 and also to the bases of second and
third PNP transistors Q115 and Q116, respectively,
which function as current mirrors. Similarly, the
collector of transistor Q112 is directly coupled to
the collector and base of fourth PNP transistor Q117
and to the bases of fifth and sixth PNP transistors
Q118 and Qll9, respectively. The emitters of all six

3~
-25-
transistors Q114 through Qll9 are resistively coupled
to power supply PSl by resistors R31 through R36,
respectively.
The collectors of transistors Q116 and Qll9
provide a differential video and audio output or
baseband signal to the remaining circuitry (66) of FM
receiver 10. The collectors of transistors Q115 and
Q118 similarly provide a replica of that same
differential video and audio output signal to drive
phasing circuit 60. By provision of transistors
Q116,Qll9 on the one hand, and transistors Q115,Q118
on the other, it is possi~le to provide a aifferential
video output for the remaining circuitry 66 of FM
receiver 10, and to drive phasing circuit 60 with
identical differential video outputs while maintaining
isolation between phasing circuit 60 and the remaining
receiver circuitry 66.
With respect to the shaping circuit 60, one
of the video outputs, the collector of transistor
Q115, is directly coupled to the collector and the
base of NPN transistor Q120 (which therefore is
diode-configured) and further to the base of current
mirror NPN transistor Q121. The other video output,
the collector of transistor Q118, is directly coupled
to the collector of transistor Q121 and to the video
input 130 of filter 13~. The emitters of transistors
Q120 and Q121 are each resistively coupled to ground
by resistors R38 and R40, respectively. This

3~.k
-26-
configuration results in a single-ended video or
baseband signal on video input 130.
Filter 132 also has a DC input 134 which may
be utilized to adjust the bias on the base of PNP
transistor Q122 from output 135 of filter 132. As
will be recognized, as the bias of transistor Q122 is
varied, the center frequency of filter 54 will be
tuned in response thereto. So that the center
frequency of electrically tunable bandpass filter 54
can be manually adjusted at the factory to the nominal
70 MHz center frequency thereof for proper operation
in the ield, DC adjust circuit 136 is provided.
~ence, a~sent offset, as will be discussed, the center
frequency of electrically tunable bandpass filter 54
is approximately e~ual to the center frequency of the
IF center frequency.
DC adjust circuit 136 is a variable DC power
supply including variable voltage divider
potentiometer R42, capacitor C15, resistor R43 and NPN
transistor Q123. The emitter of transistor Q123 is
resistively coupled to ground through resistor R43,
and the base thereof capacitively coupled to ground
through capacitor C15 and to variable voltage divider
R42 through the wiper arm thereof. The collector of
transistor Q123 is directly coupled to power supply
PSl. By adjustment o the wiper arm of potentiometer
R42, a DC output is provided on the emitter of
transistor Q123. The operating point of electrically

~2/~ 3~
tuna~le bandpass filter 54 is responsive to transistor
Q122 which is biased, in large measure, by the emitter
output of transistor Q123 thus making the nominal
operating point or center frequency of filter 54
dependent upon the setting of DC adjust circuit 136.
Transistor Q122 (and hence filter 54) is
also responsive to filter 132 which is configured
according to the principles of this inv-ention. The
emitter of transistor Q122 is coupled to power supply
PSl through resistor R44, and the collector thereof is
coupled directly to ground. The emitter of transistor
Q122 is further coupled to the input 64 of
electrically tunable bandpass filter 54 for tuning
purposes as will be described.
Filter 132 preferably provides an
appropriate amount of phase lead 0n to each frequency
component fn of the baseband signal at video input 130
according to the formula 0n = 360 . t 1 td2 _ td3-
lf
Such a filter would thus require some form of
broadband phase lead network. While this concept
forms a basis for the present invention, such a
broadband phase lead network is not always achievable,
especially over as wide a band as 8 MHz, the
television bandwidth, for example. However, if almost
all the modulation "stress" on the system were
concentrated at (or close to) one frequency, it then
becomes practical to use conventional LRC phase lead

3~
-28-
networks to accomplish the desired phase lead as a
spot approximation in the band of interest.
Fortunately this is exactly the nature of a TV signal,
as transmitted by satellite. The signal is
pre-emphasized by some 13 dB between 187 KHz and 875
KHz (for NTSC; PAL and SECAM are similar), which
reduced the deviation due to horizontal scanning to
small proportions, and leaves only the chrominance
singal at approximately 3.58 MHz as it's major,
repetitive, high frequency component. While other
high frequencies do certainly exist, they are
generally the differentiated result of image edge
contours, and thus occur with far less frequency than
the chroma signal, which has continuous sidebands for
all colored image areas. Since chroma exists for long
periods, at high level, it is the major opportunity
for detector mistracking, whose visible effect is what
are knows as "sparklies". From the !'stress" analysis,
if we take the baseband output which is delayed by
about 15 nsec (td1 plus td3), and advanced it's phase
by 36 nsec, for a net advance of 21 nsec (td1 plus
td2), then applied this signal to the feedback filter
as a phase modulator, we should by appropriate vector
summation and level adjustment able to produce a
signal at the summing point with zero apparent
envelope delay, relative to the input signal.
In fact, this is exactly the situation
observed. An RL lead network (see Fig. 4 to be

3~
-29-
discussed infra) provide~ a significant improvement
over the prior Baghdady circuitry alone, or the
~chroma) modulation indices up to 3 encountered in
satellite television FM transmission. Such a circuit
would have a phase lead characteristic of
approximately 45 at about 3.58 MHz. As mentioned,
because the modulation stress is concentrated at this
frequency, for most purposes, filter 132 may be
designed to accommodate this dominant baseband
component. On the other hand, many television signals
include elaborate audio subcarrier modulation. Unless
filter 132 also provides the appropriate phase lead 0n
to each of these subcarriers, it has been found that
the composite energy due to these subcarriers may
deleteriously affect operation of the oscillating
limiter when they are supplied as modulation signals
to the feedback filter 54. The oscillating limiter
itself however, is generally sufficient to handle such
subcarriers as they generally have a modulation index
of less than one. Hence, because a filter to provide
the appropriate 0n at 3.58 MHz as well as for the
subcarriers may not be practical, and because the
oscillating limiter needs no assistance to handle
them, it has been found possible to make filter 132
such that it provides the appropriate 0n at 3.58 MHz
without regard to other baseband components and which
substantially attenuates the subcarriers. To this
end, filter 132 is preferably comprised of

3~
~ 30-
potentiometer R50, inductor L15 and capacitors C20 and
C22.
Potentiometer R50 is series coupled between
DC adjust circuit 136 and the base of transistor Q122
to provide a DC ~eedthrough. Also, the DC signal from
adjust circuit 136 may be varied over time due to the
averaging affects of filter 132 on the video signals
at input 130 to thereby vary the center frequency of
filter 54 to compensate for offset, drift and the
like. Video input 130 is coupled to the wiper arm of
potentiometer R50 to thereby permit adjustment of the
magnitude of the output of filter 132. The junction
of potentiometer R50 and base of transistor Q122 is
coupled to ground through the parallel combination of
variable inductor L15 and capacitor C20 which are
series connected to capacitor C22. Inductor L15 is
adjustable to permit fine tuning of the phase of
filter 132 to obtain the desired 0n at about 3.58 MHz
during assembly at the factor. Hence, capacitor C20
and inductor L15 generally provide the desired phase
adjust whereas capacitor C22 cooperates with
potiometer R50 to vary the DC signal from circuit 136.
As can be seen from Fig. 3, filter 132
provides approximately 45 phase lead at 3.58 MHz and
goes resonant just above that frequency so that
attenuation is provided at the subcarrier frequencies
(5.4 to 8.5) MHz to thereby cut them off.

~ .q~3~
-31-
Where subcarriers are not significant,
filter 132 may be the simpler RL circuit of ~ig. 4
(Filter 132') with the aforesaid potentiometer R50 and
capacitor C22. However, the parallel inductor L15 and
capacitor C20 branch has been replaced with inductor
L16 in series with potentiometer R50. Filter 132'
similarly provides approximately 45 phase lead at
3.58 MHz (Fig. 5) but does not significantly attenuate
subcarrier components of the baseband signal.
By virtue o~ the foregoing, filter 132 (or
filter 132') provides control signals to phase
modulate filter 54 according to a predetermined phase
relationship. Filter 132 (and 132') also provides a
DC control signal to vary the center frequency of
filter 54 to compensate for offset, drift or the like.
The DC component or control signal from
phasing circuit 60 is an integrated response to the
varying frequency output from the demodulator 52 and
thus represents a DC value reflective of the average
or center frequency over time (which corresponds to
the carrier frequency) as seen ~y the demodulator 52.
Where that average frequency is offset from what is
ideally expected, the DC voltage generated in filter
132 (actually a variation of the DC output of circuit
13~) and passed proportionately through transistor
Q122 will cause the center frequency of the
electrically tunable bandpass filter 54 to move in a
direction toward the actual center or carrier

3~
frequency of the FM signal (actually to the true
center frequency of the IF signal if it is offset from
the expected 70 MHz). Hence, frequency offsets due to
improper operation of the satellite and/or fluc-
tuations caused in the receiving circuitry such as at
the reflector antenna are compensated in a manner
analageous to automatic fine tuning as is well under-
stood. By way of example, if the expected IF center
frequency were 70 M~z, (fd) and the actual frequency
is 70.1 MHz (fl), this would imply that the center
frequency of the oscillating limiter should be tuned
or offset by 0.1 MHz ~fl-fd) Over time, phasing
circuit 60 would provide a DC control signal
sufficiently offset from that to which the unit was
nominally set as to affect the oscillating limiter
center frequency by the appropriate amount. This
adjustment, because it is derived from averaging the
demodulated signals, does not track the modulation of
the FM signal.
In operation, FM signals received on input
14 will be super-heterodyned to a 70 MHz IF signal and
then filtered through one of two IF filters. The
filtered IF signal will be amplified and the amplified
IF signal processed by the oscillating limiter and
demodulator to provide a baseband signal to the rest
of the circuitry. The feedback filter of the
oscillating limiter will function as a phase
modulator, the modulating signals being the

~Z^~34
-33-
demodulat~d video chroma signal advanced by leading
phase in a predetermined manner. The foregoing
cooperate to improve the quality of the information
signal resulting in improved picture quality.
Although not shown in the Figures, it is to
be understood that the positive power supply lines
should be bypassed to ground by several .01 microfarad
capacitors as is well known. Further, the power
supply lines should be provided with impedances
comprised of a ferrite bead or the like surrounding
each positive power supply line as is known to sup-
press electromagnetic interference.
All capacitances are in picofarads except
those marked in microfarads and those indicated to be
".01", the latter also being in microfarads. All
inductances are given in microhenries and resistances
in ohms. Finally, gates 74, 82 and 120 comprise an
MClOH107P integrated ECL circuit manufactured by
Motorola Semiconductor Products, Phoenix, Arizona.
While the invention has been described in
connection with reception of satellite transmitted
television signals, in its broader aspects, the
invention is applicable to the reception of FM signals
generally.