Note : Les descriptions sont présentées dans la langue officielle dans laquelle elles ont été soumises.
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DIGITAL MODULATION PROCESS, AND MODULATOR
IMPLEMENTING THE PROCESS
This invention concerns digital radio communication systems. It
concerns, in particular, the methods of modulation implemented in such
5 systems.
Digital modulation is generally designed to combine the following
three requirements: a high transmission rate, minimum spectrum occupancy
and a low transmission error rate under various operating conditions.
Various methods were proposed in the past with a view to
10 achieving a high transmission rate on a channel with reduced spectral
bandwidth (transmission rate exceeding 1 biVs/Hz).
The first group of methods uses multi-level frequency modulation
as a basis, together with adequate filtering of the modulating signal (e.g.
Gaussian filtering used with GMSK modulation) in order to reduce adjacent
channel interference. These methods have the advantage that they are easily
applied, and result in modulated signals of constant envelope. They
consequently permit transmitters to be fitted with power amplifiers which
operate in the saturated state. These amplifiers are readily available, cheap
and very efficient. However, in order to comply with constraints relating to
20 adjacent channel interference, the modulation index must be limited
considerably, and the modulating signal thoroughly filtered. This causes the
symbol spacing to be reduced, and this adversely affects the noise immunity
of the modulation. In other words, the sensitivity of the radio receivers is
limited .
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Another group of methods uses phase-shift keying (PSK) and, if
necessary, differential phase shift keying (DPSK) as a basis, and the resulting
signal is filtered to ensure that standards relating to adjacent channel
interference are complied with. In general, a filter satisfying the Nyquist
5 criterion is used in order to limit inter-symbol interference. These methods
generally provide satisfactory sensitivity at the expense of a large variation in
the amplitude of the radio signal. Very linear amplifiers are therefore
necessary, and they are difficult to design and set up. In addition, they are
generally inefficient, and this seriously affects the autonomy of mobile stations.
10 A non-linear amplifier can be used in conjunction with a linearising method, but
such method complicates a transmitter very considerably if there are large
envelope variations.
Other solutions have also been proposed, e.g. in U.S. Patents
5,642,384 and 5,311,552, where an appropriate choice of a constellation and
15 of a coded modulation process prevents transitions in the constellation for
which the phase change is relatively large. This permits the variation in
amplitude of a radio signal to be reduced to values compatible with the
characteristics of amplifiers which are easier to design. However, the
reduction in amplitude is achieved at the expense of a considerable reduction
20 in the symbol spacing, which is very difficult to compensate by coding gains, in
particular in the error rate range of the greatest importance to speech
communications, i.e. for bit error rates (BER) of the order of 10-2, especially
when the channel is affected by fading (Rayleigh fading).
An object of the present invention is to propose a digital
25 modulation group permitting joint optimisation of noise immunity, even in a
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channel affected by fading, adjacent channel interference, and variation in
amplitude of the radio signal.
The invention thus proposes a digital modulation process
wherein the successive symbols of a digital stream are converted into phase
increments, an accumulated phase is obtained by adding the successive
phase increments, a modulating phase is obtained by filtering the accumulated
phase, a complex signal is produced whose argument represents the
modulating phase, two phase quadrature radio waveforms are respectively
modulated on the basis of said complex signal, and a radio signal resulting
10 from a combination of the two modulated waveforms is transmitted. According
to the invention, said complex signal is digitally filtered, and digital signalsobtained from the real and imaginary components of the digitally filtered
complex signal are converted into analog form before being respectively
subjected to anti-aliasing analog filtering and then mixed with the two radio
waveforms.
Said digital signals obtained from the real and imaginary
components of the digitally fiitered complex signal typically consist of the real
and imaginary components themselves. However, if an amplifier linearising
process is used, by pre-distortion for example (see European patent
20 application No. 0 797 293), the real and imaginary components may be
subject to correction before being converted into analog form. The use of a
linearising process is not included directly in this invention. In many cases, the
invention will permit such a process to be dispensed with. In other cases, it
will permit the use of such processes to be simplified considerably (for
25 example, by not taking account of phase changes), in view of the small
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variations in the signal envelope permitted by an appropriate choice of
parameters for filtering the accumulated phase and said real and imaginary
components. The criteria for this selection will be specified further on.
The invention permits digital radio communication systems, in
5 particular professional radio communication systems, to be implemented in
accordance with applicable standards relating to adjacent channel
interference, and provides unequalled sensitivity and thus radio range, using
power amplifier components which are readily available on the market and
have a high power efficiency.
Another aspect of the invention relates to a digital modulator,
including means for converting successive symbols of a digital stream into
phase increments, a summator which accumulates the successive phase
increments to produce an accumulated phase, a phase filter receiving the
accumulated phase and producing a modulating phase, means for producing a
15 complex signal whose argument represents the modulating phase, and a
modulator for respectively modulating two phase quadrature radio waveforms
on the basis of said complex signal, and for transmitting a radio signal
resulting from a combination of the two modulated waveforms, the modulator
comprising a digital filter to which the complex signal is applied, analog-digital
20 converters respectively processing the digital signals obtained from the real
and imaginary components of the digitally filtered complex signal, anti-aliasing
analog filters receiving the output signals from the analog-digital converters,
and two mixers each receiving one of the two radio waveforms and the output
signal from one of the two anti-aliasing filters.
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Other features and advantages of the invention will become
apparent from the following description of non-limiting embodiments, with
reference to the attached drawings where:
- Figures 1 and 2 are block diagrams of a digital modulator in
accordance with the invention and an associated receiver respectively; and
- Figures 3 and 4 are graphs respectively showing the
constellation and spectrum of a digital modulator according to the invention.
The modulator shown in Figure 1 comprises a unit 10 which
converts the successive symbols aj of a digital stream into phase increments
10 ~. The successive phase increments ~ produced by the unit 10 are
accumulated by a summator 11. The unit 10 may merely consist of a register
containing the possible values of the phase increments ~ and addressed by
the current value of the symbol aj
In the embodiment of Figure 1, the symbol stream aj may
correspond either to a bit stream bj or another bit stream c; having with a lower
transmission rate, processed by a redundancy encoder 12. The bit streams b
and c; derive from digital sources such as speech encoders, data sources,
etc., generally with error correction coding applied. If the encoder 12 is used,
the modulator of Figure 1 operates in accordance with a coded modulation
20 (see G. Ungerboeck "Channel coding with multi-level/phase signals", IEEE
Trans. on Information Theory, Vol. IT-28, No. 1, January 1982, pages 55-67).
The unit 10, summator 11 and encoder 12 are timed by a clock signal CKS at
the frequency 1/Ts of the symbols aj.
The summator 11 stores a integer digital value k representing an
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accumulated phase. This accumulated phase is thus stored as whole
multiples of a sub-multiple of 7~, that is to say in the general form (k/P)~. For
each cycle of the clock CKS, the accumulated phase is incremented by a
value ~ depending on the current symbol aj. If each symbol represents m
bits, M = 2m different values of the increment may be added in each cycle.
These M values are chosen so that the set of the increments is symmetrical
with respect to the value 0 so that the spectrum is symmetrical. Values of k of
the type k = k' x K will typically be used, where K/P represents the modulation
index, and k' = -M+1, -M+3, ..., -1, 1, ..., M-3 or M-1. This choice of equally
0 distributed increments is not the only one possible. For example, k' = -7, -3, 3
or 7 could also be used if m=2. ~maX = (kmaX/P)~ designates the maximum
value of the phase increment ~.
The accumulated phase is fed to a digital filter 15, referred to as
phase filter, whose sampling frequency 1/Te, set by a clock signal CKE, is
higher than the frequency 1/TS of the symbols aj (generally a multiple of that
frequency).
The output signal from the phase filter 15 is a modulating phase
~, which a unit 16 converts into a complex signal, i.e. into two real signals, one
(I) representing the real component of the complex signal, and the other (Q)
representing the imaginary component.
That complex signal has a constant modulus, and an argument
equil to the modulating phase ~. In other words, I = cos~ and Q = sin~. The
unit 16 may merely consist of two read-only memory arrays addressed by the
output of the filter 15 at every cycle of clock CKE.
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The complex signal is filtered by a digital filter which, in the
embodiment shown, consists of two identical filters 17 which respectively filter
the components I and Q.
Two digital-analog converters 18 convert the output signals of the
5 two filters 17 into analog form. The two resulting analog signals are fed to low-
pass filters 19 in order to eliminate spectral aliasing components. Using
respective mixers 21, two quadrature radio waveforms at the carrier frequency,
deriving from a local oscillator 20, are mixed with the signals deriving from the
anti-aliasing filters 19. The two waveforms thus modulated are combined by a
10 summator 22 whose output is fed to the power amplifier 23 of the transmitter.
If the amplifier 23 were linearised by pre-distortion, it would be
necessary to correct the filtered components I and Q, between the filters 17
and converters 18, before converting them into analog form.
The receiver shown in Figure 2 includes a low-noise amplifier 30
which amplifies the signal picked up by the antenna. Its output is converted to
an intermediate frequency using a mixer 31. A band-pass filter 32 processes
the intermediate frequency signal which is then amplified further by an
amplifier 33. Two other mixers 34 provide baseband conversion by mixing
with two quadrature waveforms. The two quadrature analog components
20 deriving from the mixers 34 are fed to identical low-pass filters 35, then
converted into digital form by analog-digital converters 36. The digital
components 1' and Q' deriving from the converters 36 are fed to a channel
demodulator 37.
The demodulator 37 carries out demodulating operations
25 corresponding to the incomplete modulator consisting of the components 10,
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11, 15, 16 and (if necessary) 12 of the transmitter shown in Figure 1. Since
this incomplete modulator essentially performs continuous phase modulation
(CPM), the demodulator 37 may take the form of a conventional CPM
demodulator. It may, for example, be based on a demodulation trellis in order
5 to apply the Viterbi algorithm. The demodulator 37 delivers estimates b j or c
of the bits bj or cj fed to the modulator.
Advantageously, the demodulator 37 may include two trellis. It
uses either one of the trellis, depending on whether the encoder 12 is used at
the transmitter or not. The first trellis includes modulation states. In principle,
the number of these states is ML-1 x P, where L is the memory of the phase
filter 15 expressed in number of samples, M is the number of points on the
constellation, and P is the denominator of the modulation index. However, it is
generally possible to considerably reduce the number of states of the
demodulation trellis without adversely affecting the quality of reception
significantly. The second one of the trellis further includes the coding states of
the redundancy encoder 12, in accordance with the principle of coded
modulations. This second trellis is employed if the encoder 12 is used at the
transmitter.
In the modulator design, the values of the phase increments ~
20 are first chosen, as indicated hereabove. The filter 17 which processes the
components I and Q and determines the spectral characteristics of the
resulting signal is then constructed. The characteristics of this filter 17 must
be as close as possible to those of the receiving filter consisting of the
combination of filters 32 and 35.
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_ 9 _
A advantageous form of the digital filters 17, used to process the
components I and Q, is a filter with a finite impulse response selected to most
closely fit a time characteristic of the form:
f(t) = Sinc(at/Ts).Sinc(,~t/Ts).e~ rvTs) (1)
5 where Ts is the duration of a symbol aj, and Sinc() is the cardinal sine function.
The approximation can be made by choosing the real coefficients cc, ~ and ~.
This provides digital filters whose restriction to a finite length is as accurate as
possible by virtue of the fast decay of the Gaussian function. The secondary
lobes caused by the limitation of the digital filter length are thus minimised.
The following step consists in defining the phase filter 15. The
characteristics of this filter 15 are closely related to those of the digital filter 17.
A heuristic method is given hereunder based on the following mathematical
property: the energy of a complex function eim(t) with unitary modulus is
maximum in a filter whose spectral power template is the Fourier transform of
a function h(t) (in other words, that energy is minimum outside the filter) if it
satisfies the following equatio,n:
J h(u-t).eim(t)dt = ~,(u) eim(u)
where ~(u) is a real function.
The following algorithm is used to define the phase filter 15:
1) A power template filter is selected, i.e. a function h(t) whose
Fourier transform represents the required spectral template. A filter identical
to the one selected as the l-Q filter is typically chosen. Other choices are
obviously possible. In general, it is preferable to use a filter whose digital
implementation with a fairly short finite impulse response is possible.
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2) A function q)0 equal to 0 where t<0, equal to ~maXt/Ts where
0<t<Ts and equal to ~maX where t>Ts is used as a first approximation of the
phase change function, i.e. of the function available at the output of the phase
filter when the maximum phase increment ~maX is fed to the accumulator 11.
5 Other approximations using continuous functions equal to 0 where t<0 and
~max where t>Ts could be used.
3) A function (~n is calculated iteratively using the following
formula:
¦ h(u-t) q)n (t) dt
¦ ¦ h(u-t). ~n (t) dt¦
4) The nth approximation of the impulse response of the phase
filter, which is equal to the derivative of the function ~n~ is calculated. An
approximation of this derived function can also be made using an approximate
analytical formula in order to facilitate subsequent calculations. The analytical
formula may be as follows:
g(t) = Sinc(a't/Ts).Sinc(~'t/Ts).e~(~l~tlTs) (2)
where a', ~' and y' are real coefficients.
5) For one of these approximations (for example n = 2 or 3), the
characteristics of the modulation are evaluated with respect to the criteria of
interference power in adjacent frequency channel, variation in amplitude and
20 noise immunity. If the approximation is unsuitable, the calculations 1) to 4) are
repeated by modifying the values of the phase increments, and/or by
modifying the shape of the l-Q filter 17, and/or by modifying the shape of the
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filter referred to under 1), and/or by modifying the approximation of the phase
filter obtained using the algorithm.
The retained phase filter is then implemented as a finite impulse
response digital filter.
The phase filter 15 of the modulator of Figure 1 could be
replaced by a bank of phase filters selected in accordance with the origin of
the symbol stream aj. This could provide a phase filter 15 optimised for cases
where the redundancy encoder 12 is not used, and another one optimised for
cases where the encoder 12 is used.
In a particular embodiment of the invention, the duration Ts of a
symbol aj is 125 ,us. The number of bits per symbol is 2, the phase increments
being ~ 13, ~13 or ~. The bit rate is then 16 kbiVs. The spectral
specifications are those of the ETSI standard 300-113. The finite impulse
response of the phase filter 15 has a length of 4 symbols and the form (2) with
a' = 0.77, ,B' = 0.5 and ~' = 0. The finite impulse response of the l-Q filter 17
has a length of 8 symbols and the form (1) with a = 1.6, ~ = 0.1 and ~ = 0.12.
The values specified for the filter parameters could be replaced by values of
the same order.
The constellation corresponding to that modulation is shown in
20 Figure 3. A very small variation in amplitude will be noted, since the ratio
between the maximum instantaneous power and the average power is only 1.2
dB, whereas the ratio between the maximum and minimum instantaneous
powers is less than 2.4 dB. Because of these characteristics, the modulation
can be used with weekly linearised power amplifiers, easy to adjust and having
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an efficiency very close to that of saturated amplifiers.
The spectrum is shown in Figure 4. It can be noted that the level
of adjacent channel interference is very low and compatible with the most
demanding standards.
The noise immunity performance is excellent since, with a
channel affected by Gaussian white noise, an error rate of 1% is noted for a
signal-to-noise ratio Eb/No of 5.5 dB in the case of stationary stations, whereas
the same error rate of 1% is obtained for a signal-to-noise ratio Eb/No of 16 dB
in a dynamic case (speed of 70 km/h and carrier at 400 MHz). These error
0 rates are obtained using simple conventional demodulators (Figure 2), i.e.
trellis demodulators with a very small number of states. A trellis with only three
states can be used in the example shown.
In the same embodiment of the invention, the optional
redundancy encoder 12 permits the implementation of a coded modulation.
Here, the redundancy coding is a convolutional coding of rate 1/2, the bit rate
then being 8 kbiVs. The filter values are identical, and a trellis with only four
states may be used at the demodulator. The coding gain is of the order of 2.5
dB, and an error rate of 1% is noted with Eb/No = 3.4 dB on a channel affected
by Gaussian white noise in a stationary case.
