Note : Les descriptions sont présentées dans la langue officielle dans laquelle elles ont été soumises.
CA 02652066 2009-01-23
LINEAR ELECTRIC MOTOR CONTROLLER AND
SYSTEM FOR PROVIDING LINEAR SPEED CONTROL
This application is a divisional of Canadian Patent Application
Serial No. 2,413,865 filed on December 10, 2002
FIELD OF THE INVENTION
The invention relates to a method of, and system for, providing linear speed
control of electric direct current (DC) motors. Specifically, the present
invention is directed
to methods and systems for linear control of variable speed electric motors.
BACKGROUND OF THE INVENTION
Presently, electronic controllers are typically designed for 12 V or 24 V
electrical systems such as those used in automotive applications. Electronic
controllers in
automobiles are typically attached to dashboards, seat bottoms, rear side of
the passenger
compartment, or the like by being screwed into designated place(s).
Present-day electric motors, such as those used in heating ventilating and air
conditioning (HVAC) systems of automobiles, are controlled mainly using switch-
mode
technology, in which a fixed DC power supply is switched on and off with a
predetermined
switch frequency and pulse width modulated switch-on time as needed to control
the motor
speed. In the United States, the motor control technology has been implemented
primarily
by use of a resistive divider (e.g., blower resistor) or by the switch-mode
pulse width
modulation (PWM). A resistive divider operates by modulating the power
provided to the
electric motor by a constant or adjustable amount, resulting in a choppy or
stepwise level of
control.
PWM switch-mode works by modulating the timing of the lead and trail
edges of the power signal provided to the electric motor. PWM results in a
relatively
inaccurate control of an electric motor, and may also introduce a choppy
quality of control.
Moreover, the switch frequencies of PWM switch-mode controllers often have
impact on
system's EMC requirements.
Other known techniques involve supplying analog variable direct current
(DC) voltage for control of variable speed electric motors. These techniques
have typically
involved a low-pass filter to generate the DC voltage. One drawback of such
techniques is
that the use of a low-pass filter tends to introduce a stepwise/choppy quality
to the control
that is used for supplying the voltage to the motor due to the latency period
of control loops.
The choppiness is similar to that which was discussed above in connection with
resistive
dividers.
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Altematively, some use has been made in Europe of a type of linear motor
controller with speed set-point inputs in PWM type controller, e.g., the
linear current
threshold motor controllers shown in U.S. Patent No.5,781,385 to Permuy issued
on July
14, 1998 or the linear speed controller through motor r.p.m. feedback control
shown in U.S.
Patent No. 5,747,956 to Lamm issued May 5, 1998. A PWM set-point signal is
used
directly for driving the controller switch device through a low-pass filter.
These systems
tend to be characterized by an undesirably large latency period, i.e., the
period between
detection and correction of the desired motor speed.
A linear electric motor controller generally works by directly controlling the
motor speed by setting the voltage feeding to the electric motor. The speed of
the electric
motor has a linear relationship with the voltage supplied to the motor, hence
the term
"linear."
The widespread use of linear controllers for control of present-day variable
speed electric motors has been frustrated largely due to the large amount of
heat generated
by such controllers. The heat that is generated creates difficulty in
sufficiently cooling such
linear controllers, avoiding thermal melting and breakdown of the material
enclosing linear
controller units, and the need for placement of the controller within a
cooling air stream,
and the wide-spread use of such controllers in practical applications. For
example, some
linear controllers have required heat dissipation ratings of as high as 90-95
Watts. One
technique for addressing the heat dissipation issue has involved designs in
which the
controller is remotely located from heat-sensitive structures. This design
tends to increase
the size of the controller module.
On the other hand, PWM-type switch-mode controllers require a heat
dissipation rating of only 6 - 10 watts, which advantageously allows for the
controller to be
located adjacent to heat-sensitive components such as plastics. The housing of
contemporary electric motor controller units, however, is typically made of
standard
injection-molded polypropylene plastics, which can handle close contact with 6-
10 watts of
heat dissipation as with a PWM-type switch-mode controller, but not the
possible 90-95
watts involved with linear controllers. Thus, switch-mode or PWM-type switch-
mode
controllers tend to be highly desired for commercial production applications.
Another related aspect of electric DC motors is a locked rotor condition that
sometimes occurs and which may cause damage to the motor and the controller
due to
smoke and fire. Different techniques have been used for detecting the locked
rotor
condition. For example, one direct method is to add a rotation sensor inside
the motor,
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CA 02652066 2009-01-23
thereby detecting rotation activity. Other indirect methods are based on
changes in the
motor voltage, current and temperature due to a locked rotor condition.
It would be desirable to overcome the various problems and disadvantages of
both the heat-issues of linear control systems and the crude control of PWM-
type switch-
mode controllers in the prior art to satisfy the design requirements for
current electrical
systems and to provide smaller, more efficient controllers, in particular for
automotive
systems.
Linear Electric Motor Controller With Multiple Input Interfaces
Another drawback of known existing systems is related to fan speed
selectors/controllers. The conventional fan speed selector typically used on
an automobile
is based on a resistor card, i.e., a blower resistor is equivalent to a power
resistor pack.
Most selectors comprise multiple stages (e.g., from 3 to 8 stages) of
resistors and relays,
and include a multi-position selector switch that allows the operator to set
the speed of the
fan motor to multiple speeds, e.g., low, medium and high. Such selectors
provide only
discrete control of the fan motor 'speed, and do not allow fine-tuning of the
fan motor speed
to ensure the maximum comfort of the passengers. Furthermore, since finer
control of the
motor speed requires a greater number of stages of resistors and relays, the
cost of the
conventional selector increases_with the required degree of control of the fan
motor speed.
This is contrary to what is desired by automobile manufacturers which is to
lower unit costs
as much as possible.
In addition, the resistor card is exposed to the open air when installed
inside
the dashboard of an automobile. Consequently, it is subject to corrosion,
which negatively
impacts the life of the card. Because resistors are made mostly of ceramic
materials,
environmental heat and vibration of the automobile also negatively impact the
life of the
resistor card.
The replacement of a faulty fan speed selector can be very expensive. For
example, although the resistor card may cost only a few tens of dollars, the
labor costs to
replace the unit can run into hundreds of dollars because the interior of the
dashboard is not
easily accessible. In addition, the automobile owner is likely to experience
the
inconvenience of lost time and transportation while service is performed.
Conventional resistor card-based fan motor speed selectors also lack
configurability. Namely, a three-position selector cannot be configured to
provide four
levels of speed control. Thus, automobile manufacturers are required to
maintain an
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CA 02652066 2009-01-23 II
inventory of resistor cards for the different gradations of desired control of
motors.
Automobile manufacturers are also required to maintain an inventory of
resistor cards for
each different fan motor model to be controlled. The maintenance of multiple
inventories
of fan speed selectors is a highly undesirable cost to automobile
manufacturers.
Conventional resistor-card based fan motor speed selectors also do not
maintain a constant motor speed during operation because of changes in ambient
temperature or battery voltage.
Although automobile manufacturers have desired a configurable, low-cost
fan speed selector that provides continuous (instead of discrete) control, a
number of
technical hurdles have stood in the way of achieving these objectives.
Although fan motor
speed selectors have been utilized since before the invention of the
transistor in the early
1950s, it is only recently that compact, low-priced power switches, e.g.,
metal-oxide
semiconductor field-effect transistors (MOSFETs), powerfnl enough to control
the fan
motor of an automobile have come on the market at a cost-effective price.
However, cost-
effectively integrating the associated signal processing capabilities
compactly on a chip and
maintaining the constant speed of the fan motor using the power switches have
been
problematic.
Therefore, there is also a need for a more reliable, configurable,
continuously
controlled, compact, low-cost and long-life fan speed selector/controller that
maintains
motor speed.
Another drawback of existing blower resistor systems is that fan speed
selectors are not conveniently replaceable. Therefore, there is a need for a
drop-in
replacement fan speed selector that incorporates the characteristics above
mentioned.
Parallel FETs In Linear Mode For Reduced Thermal Impedance
Another drawback of known existing systems is related to parallel
arrangement of FETs. It is known in the power electronics industry,
particularly in
automobile applications, to use FETs in a parallel arrangement in switching
applications.
For example, parallel FET arrangements are commonly used in power supplies and
motor
drivers to achieve superior current capability at a reduced cost. Parallel
FETs are cost
effective because, for example, two 100W FETs cost less than one 200W FET.
In switching applications, it is common to connect the gates, sources, and
drains of two FETs in a parallel arrangement and achieve satisfactory current
sharing
through the two FETs when they both switch on an off at the same time. Simple
circuitry,
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CA 02652066 2009-01-23
such as a resistor connected to the gates, is all that is needed to equalize
the current shared
between the two FETs.
In switching applications operating in linear mode, however, it is desirable
to
use two FETs in a parallel arrangement in order to reduce the thermal
impedance of the
entire switch. A typical FET has a thermal impedance of approximately 0.4 to
0.5 C/W. In
a 100W application, the temperature drop on the FET alone is about 50 C. When
other
thermal impedances between the FET and the environment are taken into
consideration, i.e.,
from the FET junction through the casing to the heat sink and finally to the
ambient air, the
overall thermal impedance is often too high, which causes the temperature of
the switch to
exceed the rated operating temperature range. Using two FETs instead of one
reduces the
thermal impedance of the entire switch by half, which enables the switch to
remain in its
operating temperature range. Reduced thermal impedance also allows for the
possibility of
reducing the size of external heat dissipators, such as a heat sink, thereby
reducing the
overall size of the switch.
The use of parallel FETs in linear operation presents technical challenges.
For example, in conventional current-balancing schemes based on paralleling
FETs, there
often occurs a condition in which one of the FETs heats up more than the other
due to
differences in the FET characteristics arising from, for example, variability
in surrounding
circuitry as well as variations in the FET manufacturing process. Over the
long-term, this
uneven heating of the two FETs leads to the premature failure of one of the
FETs.
Studies have shown that reducing the thermal impedance of a switch in half
can reduce the mean-time-between-failure (MTBF) rate by as much as 20% to 30%.
Since
the typical automobile has an expected lifespan of around 7 to 10 years, the
lifespan of
automobile components such as switches is a particularly important
consideration because
the repair or replacement of an automobile component can incur high labor
costs and can
cause an inconvenience to the automobile owner.
Therefore, there is also a need for improved performance of switches
operating in linear mode.
For example, there is a need to reduce the thermal impedance of the switch,
to balance the current distribution between parallel FETs arranged in the
switch, to avoid a
thermal runaway condition in parallel FET switches, and to equalize the
temperature of
parallel FETs of a switch operating in linear mode.
In addition there is a need to increase the lifespan and the size of a switch.
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Protection Circuit Topology For Linear Power Module
Another drawback of known existing systems is related to LPM protection
circuitry. During the installation or servicing of a conventional automobile
HVAC system,
it is possible for damage to occur to the LPM from a number of sources,
including: 1) an
inadvertent reversal of polarity from the circuit's design polarity, 2)
circuit voltage
overshooting spikes that are intrinsic to all motor-driven systems, and 3)
voltage
overshooting spikes caused by a slowdown or shutdown of the electric fan
motor.
Depending on their magnitude and duration, any of these conditions are likely
to damage
the LPM in a way that it would necessitate its replacement. The replacement
cost of an
LPM can reach several hundred dollars when parts and labor costs are
considered.
Automobile owners are also likely to experience the inconvenience of lost time
and
transportation while their vehicle is being serviced. In addition, damaged
electrical
components may pose a safety concern to automobile owners and service
personnel.
Some methods for LPM protection are known in the industry. For example, U.S.
Patent No. 5,519,557, entitled, "Power supply polarity reversal protection
circuit," assigned
to Chrysler Corporation, describes a power supply polarity reversal protection
device that
includes a voltage-controlled switching device, e.g., an N-channel metal-oxide
semiconductor field-effect transistor (MOSFET), connected between the power
supply
output (e.g., a positive supply voltage) and the circuit to be powered by the
supply, i.e., the
load. A diode, preferably the parasitic diode inherent in the MOSFET between
the drain
and source, is arranged so that when the positive supply terminal is connected
to the
transistor, the diode is forward-biased and will allow a diode current to
flow. A sensing
means generates a control terminal voltage in response to the diode current
flow. This
control terminal voltage is applied to the switching means control gate and is
sufficient to
cause full conduction through the switching means. If, instead, a negative
supply voltage is
connected to the transistor, the diode will be reverse biased and no current
will flow. Since
there is no diode current, the sensing means will not generate a control
terminal voltage and
the switching means will not turn on, thus protecting the load.
However, the '557 patent does not provide protection to the power supply from
voltage excursions caused by slowdown or shutdown of the load. Therefore, what
is needed
is a linear power module with reverse bias and voltage spike protection.
In addition, there is a need for a more reliable LPM circuit with a cost-
effective solution for LPM protection. Moreover, there is a need to protect
the LPM against
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CA 02652066 2009-01-23
voltage excursions and to provide a safer automobile HVAC system with a more
reliable
means of automobile HVAC system installation and servicing.
Temp FET-Based Apparatus For Protection Against Locked Rotor In An HVAC S s~
Another drawback of known existing systems is related to locked rotor
protection. Electric motors, particularly fan motors in automobile HVAC
systems, often
include mechanisms that terminate operation of the motor in response to
thermal overload
conditions that could result in permanent damage to the motor or associated
equipment. A
thermal overload, such as an excessively high winding or rotor temperature,
may occur as a
result of a locked rotor, a faulty bearing, a high mechanical load, a supply
overvoltage, a
high ambient temperature, or some combination of these conditions.
Overcoming the problem of a locked rotor condition in the fan motor of an
HVAC system of an automobile has presented a number of challenges.
A locked rotor condition can cause an electric motor to draw many times its
rated current. Therefore, one conventional approach for handling a locked
rotor condition is
to provide overcurrent protection function, either by means of an electronic
overcurrent
protection circuit or a current fuse. When the rotor of a motor becomes locked
for some
reason, the impedance of the motor circuit decreases, which causes the current
to increase.
The increased current is then detected by the overcurrent protection function,
which cuts
power to the motor, thereby protecting the motor and the controller from
destruction.
Another conventional approach for protecting an electric motor against a
locked rotor condition is the use of position sensors, e.g., optical sensors,
rotating pulse
sensors, and proximity sensors. However, position sensors are relatively large
and increase
the costs of the motor, the control circuitry, installation, and maintenance.
Further, position
sensors are not very reliable. For example, the environment of the motor of an
HVAC
system in an automobile can be quite severe, causing optical sensors to become
dirty and
causing proximity sensors to lose their magnetic properties.
Yet another conventional approach for protecting an electric motor against a
locked rotor condition is the use of thermal cutouts. However, this approach
is undesirable
because thermal cut-outs typically include a complex arrangement of springs
and contact
elements that are mounted in a housing, which is inherently costly and does
not allow for
the direct inspection because thermal cut-outs are not usually visible through
the motor
housing.
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CA 02652066 2009-01-23
Therefore, there is a need for a low cost, high-reliability, maintenance-free
and long-lifespan locked rotor protection in an automobile HVAC system without
the use of
additional sensor components.
SUMMARY OF THE INVENTION
In accordance with the principles of the present invention, an electric motor
controller can be provided that can be both smaller and lighter in weight than
a switch-mode
controller of the prior art. This can help increase fuel economy, decrease
vehicle size, or
increase the space available within the vehicle for other uses, as well as
combinations
thereof, all of which are highly desirable achievements. The circuitry of the
controller may
be packaged into an integrated circuit to further reduce the weight and size
of the controller.
Different packaging arrangements may also be used.
The controller includes an internal feedback mechanism to minimize control
and to monitor latency, which provides for a more accurate control of the
electric motor
speed. This mechanism optionally, but preferably, incorporates a digital
conversion to
generate the motor voltage, thereby allowing for even greater accuracy as
compared to
conventional low-pass filters.
Moreover, the present invention generally allows for high precision speed
control at a resolution of less than 1% PWM duty cycle control signal,
optimized thermal
management, integrated circuits with higher energy conversion effectivity,
higher power
intensity, smaller footprint, and lighter weight, as well as the specially
configured
monitoring and protection system such as the sensorless locked rotor
protection utilizing
motor commutation pulses. In addition, the present invention provides for
strengthened
electrostatic discharge (ESD) protection (e.g., up to 8kV contacted discharge
and up to
25kV air discharge), bulk current injection (BCI) protection for high
frequency injection,
over-temperature protection with hysteresis recovery, multi-level thresholds
for over-
current limit and short circuit protection, over/under voltage protection with
hysteresis
recovery, adjustable motor soft-start control, and reversed polarity
protection for.either the
controller or DC motor power source such as a battery.
The linear motor controller can also include a suitable heat sink, capable of
conducting heat from the controller, wherein the heat sink is made of a
material and
designed so as to maximize the dissipation of heat from the controller. This
enables the
controller to be located in closer proximity to plastic components and/or a
plastic housing,
which also then allows for the controller module to be designed smaller if
desired. The
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CA 02652066 2009-01-23
controller housing and plastic components are designed to be more capable of
withstanding
the increased wattage of heat dissipated by the controller, thereby also
allowing for the
controller to be placed in closer proximity to the plastic components and/or
housing,
thereby also contributing to reduction in size of the controller module.
Additionally, the
placement of the controller is preferably optimized within a cooling airflow
so as to
facilitate heat dissipation from the controller through the heat sink.
The invention relates to an automotive electric motor linear speed controller
that includes a digital to analog converter means for converting an 8 bit
digital signal to
analog voltage for setting voltage across the electric motor, a digital state
machine means
for converting the duty cycle of a speed control input signal (e.g., in
approximately a 0,39%
precision) for output to the digital to analog converter means, and a closed
loop feedback
means for monitoring and setting the voltage across the automotive motor. It
is also
advantageous to include an over-current sense circuit for monitoring the
current across or
passing through the electric motor, an over/under voltage sense circuit for
monitoring a
supply voltage to the electric controller, or both, although neither is
strictly required.
Another embodiment of the invention relates to a circuit arrangement in a
variable speed automotive electric motor controller. The circuit arrangement
includes a
controller logic circuit for operating a controller logic finite state
machine, in which the
state machine sets the voltage supplied to an electric motor. It can also
include a closed
loop feedback for generating a signal indicating the voltage across the
electric motor, which
can then be input to the state machine for monitoring thereof.
In another embodiment, the invention includes a system incorporating at
least the above-described automotive electric motor linear speed control. In
another
embodiment, the invention includes a system for controlling the speed of an
automotive
electric motor, in which the voltage across the electric motor determines the
speed of the
electric motor. This system can include a digital to analog converter means
for converting a
digital signal to analog voltage for setting voltage across the electric
motor, a
microprocessor and associated digital memory for generating the digital
signal, where the
microprocessor is configured to instantiate and operate a digital state
machine for
converting the duty cycle of an input signal generated by an associated closed
loop
feedback means, and a closed loop feedback means for monitoring the voltage
across the
motor and generating a signal for input to the microprocessor. The invention
also relates to
an automobile including the above-described system. In a preferred embodiment,
the
system includes an operating temperature-control system.
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In yet another embodiment, the system comprises a locked rotor protecting
circuit for protecting the electric motor from locked rotor fault condition
based on a
commutating pulse noise signal obtained from said voltage across said motor.
This
embodiment also includes at least one multi-level current limiting circuit
utilizing at least
one current threshold for short circuit protection; a thermal sensing circuit
for protecting the
controller from over-heating darnage; means for protecting the controller from
damage by
reversed polarity of a power supply; and means for protecting the controller
from
electrostatic discharge and bulk current injection damage.
Alternatively, the system can include means for detecting a locked rotor in
the electric DC motor. A locked rotor signal is generated for immediate motor
disconnection, therefore avoiding overheating of the system's elements, smoke
and fire in
case of a locked rotor. Preferably, the system detects when the rotor of the
electric motor is
locked without the need of rotation sensors. The commutation pulse noise
across the motor
is used to generate the locked rotor signal, i.e., the motor rotating status
detection signal.
The signal level and pulse frequency are adjustable and resettable depending
on various
motor configurations and motor speed requirements. Consequently, the rotation
is detected
directly thus avoiding false protection triggers or missed detections. The
circuit and
methods discussed herein neither affect nor interact with the current and
voltage protections
so it is unnecessary to take special considerations to a function such as
detecting a locked
rotor or other functions.
In another embodiment, the operating temperature of the key component of
the controller is sensed by a PTC (Positive Temperature Coefficient) sensor
and produced
the thermal shut-off signal with automatic hysteresis recovery to DIN pin of
the control IC.
There are two ways of implementing the sensorless locked rotor protection
circuit. A discrete method utilizes commercial IC's and discrete components.
An integrated
method implements the configuration inside an ASIC. The integrated method is
less
expensive for mass production while the discrete method is more flexible,
universal and
immediate.
The invention also relates to a method of protecting an electric motor
comprising a linear speed control generating a speed control signal. This
method includes
the steps of detecting a locked rotor condition of the electric motor;
generating a locked
rotor signal; disconnecting the electric motor based on the locked rotor
signal; and restarting
the electric motor through turning the speed control signal on and off
Additionally, the
method can include the steps of sensing a locked rotor current; comparing the
locked rotor
CA 02652066 2009-01-23
current with a predetermined threshold; and limiting the locked rotor current
based on the
comparison. Preferably, the method includes the steps of sensing temperature
of the locked
rotor current sensitive components of the motor; and shutting down the motor
based on the
sensing of a temperature that indicates hysteresis recovery.
It is desirable to protect polarity sensitive components of the motor, and
this
can be achieved through the use of a diode. Alternatively, polarity sensitive
components,
such as one or more electrolytic capacitors, can be replaced with non-polarity
components,
such as one or more ceramic/film capacitors. Also, R-C-D network(s) can be
implemented
for electrostatio discharge and bulk current injection protection (e.g., for
bi-directional
electrostatic discharge and bulk current injection protection).
A system for protecting an electric motor comprising a linear speed control
circuit generating a speed control signal is also provided. This system
comprises means for
detecting a locked rotor condition of the electric motor; means for generating
a locked rotor
signal; means for disconnecting the electric motor based on the locked rotor
signal; and
means for restarting the electric motor through turning on and off of the
speed control
signal. Advantageously, the system further comprises means for sensing a
locked rotor
current; means for comparing the locked rotor current with a predetermined
threshold; and
means for limiting the locked rotor current based on the comparison.
Preferably, this
system includes means for sensing temperature of the locked rotor current
sensitive
components of the motor; and means for shutting down the motor based on the
temperature
with hysteresis recovery.
Further embodiments of the invention relate to a system and method of
detecting a locked rotor condition in an electric motor comprising a linear
speed control
generating a speed control signal. The method comprises the steps of:
obtaining a signal
consisting of pulses from the motor; filtering the signal thereby removing
noise and
amplifying the pulses; selecting pulses with an amplitude greater than a
predetermined
minimum amplitude; detecting whether the motor is moving based on the signal;
and
combining the signal with the speed control signal of the motor to obtain a
locked rotor
condition signal. The corresponding system comprises means for obtaining a
signal
consisting of pulses from the motor; means for filtering the signal thereby
removing noise
and amplifying the pulses; means for selecting pulses with an amplitude
greater than a
predetermined minimum amplitude; means for detecting whether the motor is
moving based
on the signal; and means for combining the signal with an ON/OFF signal of the
motor to
obtain a locked rotor condition signal.
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In one preferred embodiment, the invention relates to a linear speed control
for an automotive electric motor that includes a digital state machine for
converting the dtity
cycle of an input signal generated by an associated closed loop feedback, an
over-current
sense circuit for monitoring the current across or through said electric
motor, an over/under
voltage sense circuit for monitoring a supply voltage to the electric
controller, a digital to
analog converter for converting an 8-bit digital signal to analog voltage for
setting voltage
across said electric motor, and a closed loop feedback for monitoring the
voltage across said
motor and generating a signal for input to said digital state machine.
The controller can also be packaged inside a controller module for ease of
assembly into a final product, such as an automobile. Thus, the invention also
relates.to an
electronic controller module including an electronic controller that generates
at least about
15W of heat, an enclosure made of at least one heat-resistant material
configured and
dimensioned to substantially surround and physically protect the controller,
at least one
electrically-conductive member to provide input or output of at least one
electrical signal
through the enclosure to the controller, and a heat sink operatively
associated with the
controller to receive heat therefrom, the heat sink being configured and
dimensioned to
dissipate a sufficient amount of heat to inhibit or avoid damage to the
controller and
enclosure.
In one embodiment, the enclosure includes a lid, which comprises the heat
sink. In a preferred embodiment, the lid is made of a heat-resistant material
and the heat
sink includes a heat fin assembly mounted upon the lid that extends away
therefrom. A
portion of the heat sink can extend through the lid to a position adjacent the
controller to
facilitate heat dissipation. In one embodiment, the heat sink is made of a
material that
dissipates about 20W to 150W. The heat sink material can include any suitable
thermally
conductive material, including aluminum, copper, thermally conductive
plastic(s), or a
combination thereof. The heat-resistant material does not melt on exposure to
about 150W
and typically includes an olefinic polymer, preferably one that includes amide
units. In a
preferred embodiment, the olefinic polymer includes a polyamide-polypropylene
copolymer
and includes a filler in an amount sufficient to increase the heat-resistance
thereof.
Preferably, the filler includes talc, glass, ceramic, mica, silicate, clay,
aramid, lithopone,
silicon carbide, diatomaceous earth, carbonates, metal or an alloy or oxide
thereof,
particulate carbonaceous material, hard particulate material, or combinations
thereof. The
filler can be present in any form, preferably whiskers, fibers, strands, or
hollow or solid
microspheres.
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In a preferred embodiment, the electronic controller is a linear controller
capable of facilitating temperature control in an environment. Preferably, the
enclosure is at
least substantially rectangular. For example, the enclosure can have
dimensions of about 3
cm to 8 cm in length, about 1 cm to 4 cm in height, and about 3 cm to 6 cm in
width. As
another example, the heat sink includes at least two short fins having a
length of about 0.25
cm to 1 cm and at least two long fms having a length of about 1.5 cm to 6 cm,
each adjacent
the enclosure at one end thereof and extending away therefrom. Preferably, the
at least two
long fins include a first heat fin having a length of about 1.75 cm to 2.25 cm
and a second
heat fin having a length of about 3.5 cm to 4.5 cm.
In one embodiment, the lid and the enclosure are operatively associated via a
plurality of projections and gaps to permit the lid to securely snap into
place against the
enclosure so as to collectively completely surround the electrical controller.
In a preferred
embodiment, the module further includes an insulating member between the
controller and
the lid and in contact therewith to inhibit or avoid thermal degradation of
the controller.
Preferably, the insulating member can include at least one silicone material
that is
sufficiently flexible to at least partially conform to the controller. In
another preferred
embodiment, a thermal grout is included in the module and is disposed to
facilitate the lid
and the heat-resistant material being at least water-resistant.
In one preferred embodiment, the controller includes a single circuit board
having all controller components mounted thereon that is surrounded by the
enclosure and
the lid.
The invention also relates to a method for dissipating heat from an electronic
controller by providing an enclosure around the electronic controller which
generates at
least about 15W of heat during operation, associating a heat sink with the
controller to
receive heat therefrom, and dissipating a sufficient amount of heat to inhibit
or avoid
damage to the controller and enclosure. In one embodiment, the controlter
generates at
least about 20W to 150W during operation. In another embodiment, at least
about 90
percent of the heat generated is dissipated via the heat sink.
To further elaborate different aspects of the present invention, in some
embodiments, a linear speed control method or system is provided for an
automotive
electric motor, that may include a digital state machine, for converting the
duty cycle of a
speed control input signal to digital signal in a 0.39% precision (e.g.,
approximately a
0.39% precision), comparing said input signal digitally with closed loop
feedback of motor
13
CA 02652066 2009-01-23
voltage signal, switching the operation states, counting the timeout and
latching the control
system at fault conditions.
In accordance with an aspect of the present invention, there is provided an
electronic controller module comprising an electronic controller that
generates at least about
15 W of heat; an enclosure made of at least one heat-resistant material which
comprises an
olefinic polymer that includes amide units and that does not melt on exposure
to about 150W,
and is configured and dimensioned to substantially surround and physically
protect the
controller; at least one electrically-conductive member to provide input or
output of at least
one electrical signal through the enclosure to the controller; and a heat sink
operatively
associated with the controller to receive heat therefrom, the heat sink being
made of a
material that dissipates about 20W to 150W to inhibit or avoid damage to the
controller and
enclosure.
In accordance with another aspect of the present invention, there is provided
a
method for dissipating heat from an electronic controller which comprises
providing an
enclosure comprising an olefmic polymer including amide units around the
electronic
controller which generates at least about 15W of heat during operation;
associating a heat
sink with the controller to receive heat therefrom; and dissipating a
sufficient amount of heat
to inhibit or avoid damage to the controller and enclosure.
13a
CA 02652066 2009-01-23
BRIEF DESCRIPTION OF DRAWINGS
The purpose and advantages of the present invention will be set forth in and
apparent from the description that follows, as well as by practice of the
invention.
Additional advantages of the invention will be realized and attained by the
methods and
systems particularly pointed out in the written description and claims hereof,
as well as
from the appended drawings, wherein
Figure 1 is a block diagram of a linear controller integrated circuit in
accordance with an embodiment of the present invention;
Figure 2 is a finite state machine diagram of controller logic used in an
embodiment of the current invention;
Figure 3 is a block diagram depicting the pin-outs associated with the
controller integrated circuit of an embodiment of the current invention;
Figure 4 is a top view of the module illustrating a heat fin and module
according to an embodiment of the present invention;
Figure 5 is a perspective view of the top of the module and heat fins
according to an embodiment of the present invention;
Figure 6 is a perspective view of the bottom of the module and electrical
member(s) according to an embodiment of the present invention;
Figure 7 is a graph illustrating an example of a commutation noise during
motor operation in accordance with one embodiment of the present invention;
Figure 8 is a graph illustrating an example of a low frequency commutation
pulse during motor operation in accordance with one embodiment of the present
invention;
Figure 9 is a graph illustrating an exainple of a high frequency noise floor
commutation pulse during motor operation in accordance with one embodiment of
the
present invention;
Figure 10 is a flow diagram illustrating the sensorless locked rotor
protection
method according to an embodiment of the present invention;
Figure 11 is a flow diagram illustrating the locked rotor protection method
with multi-thresholds of motor current in accordance with another embodiment
of the
present invention;
14
CA 02652066 2009-01-23
Figure 12 is a circuit illustrating one embodiment of the locked rotor
protection circuit according to an embodiment of the present invention;
Figure 13 is a circuit element in the locked rotor protection circuit
illustrating an electrostatic discharge and bulk current injection protection
feature in
accordance with another embodiment of the present invention;
Figure 14 is a conventional fan speed selector based on a blower resistor card
according to an embodiment of the present invention;
Figure 15 is a linear electric motor controller with optional multiple hybrid
inputs conversion interface according to an embodiment of the present
invention;
Figure 16 is an optional input block of linear electric motor controller with
multiple hybrid inputs conversion interface according to another embodiment of
the present
invention;
Figure 17 is a schematic drawing of a system comprising two temperature
protected FETs operated in parallel in linear power mode having reduced
thermal
impedance according to an embodiment of the present invention;
Figure 18 is a schematic of a linear power module with reversed-bias
protection.circuit according to an embodiment of the present invention;
Figure 19 is a schematic drawing of a five-pin temperature protected FET-
based apparatus for protection against a locked rotor condition in an HVAC
system
according to an embodiment of the present invention; and
Figure 20 is a schematic drawing of a three-pin temperature protected or
fully protected FET-based apparatus for protection against a locked rotor
condition in an
HVAC system according to an embodiment of the present invention.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
In accordance with a preferred embodiment of the invention, a method and
system is provided to enable a linear electric motor controller that is
smaller, lighter in
weight, or both, compared to switch-mode controllers of the prior art. By
packaging the
linear controller of an embodiment of the current invention in an integrated
circuit,
additional weight and/or size savings can be realized. It has now been
discovered that such
linear controllers can be prepared by advantageously including an internal
feedback
mechanism to monitor the voltage and thus speed of the controlled motor, which
can
provide superior control compared to the switch mode type controller. The
inclusion of at
least one intemal feedback mechanism facilitates a decreased latency between
detection and
CA 02652066 2009-01-23
correction of deviations in speed of the controlled electric motor. This can
advantageously
minimize control latency to facilitate more accurate control of the electric
motor speed.
In one embodiment, an internal feedback that incorporates a digital
conversion to generate the motor voltage can be used, thereby allowing for
further increased
accuracy and smoothness of control over the use of low-pass filters of the
prior art.
It should be understood that the linear motor controller can be used or
adapted for a variety of suitable applications, including control of electric
motors, electric
lights, and the like. In particular, the linear motor controller of the
invention is specifically
directed to its use in controlling electric motors in automotive systems,
preferably
automobile HVAC systems.
Preferably, the linear controller of the invention is disposed on a single
circuit board having all controller components mounted thereon.
The integrated circuit block diagram of Figure 1 specified below serves to
realize the linear power module and control method according to an embodiment
of the
invention. The oscillator OSC 1 generates the 50% duty cycle internal clock
signal used by
the digital logic block 10. The clock frequency of the oscillator is about 150
kHz +/-10%.
A digital state machine means is realized in the digital logic block 10 of the
linear
integrated circuit controller. Other methods employing similar state machines
to that
described herein may be used as is well known to those of ordinary skill in
the art.
The closed loop feedback means may be embodied by operational amplifier
OPAl 16, and operational amplifier OPA3 14, in conjunction with a compe~sation
loop.
OPAl 16 is used as a differential amplifier. It divides the input voltage by a
constant value
that is determined RDIF5/RDIF6 3 and resistors RDIFI/RDIF2 4. It defines the
voltage
across the motor - load CLP 18, which is then low pass filtered. The output of
the low pass
filter is then connected via a resistor to the input of the error amplifier
OPA3 14. OPA3 14
serves as the error amplifier and gate driver. In the case of an over-current,
the gate drive is
altered to maintain a fixed load current. After a set period of time, the
output of the gate
driver is set to 0 volts.
The system transfer function CLP 18 may also be altered to allow for better
matching to optimize performance of the monitor and control of the electric
motor. The
controller has the capability to operate with a ground offset of about +/- 2
volts. Also, the
16
CA 02652066 2009-01-23
transfer function can be inverted if desired, and the slope and position of
the transfer
function can be modified within limits.
The closed loop feedback 29 circuit monitors the voltage directly across the
electric motor. The voltage is fed back into the controller integrated circuit
and compared
to the reference voltage generated by the digital to analog converter, DAC
block 12. Based
on the difference, the loop changes the gate drive voltage using a field
effect transistor
(FET) controlling the voltage across the motor so as to eliminate the
difference. This closed
feedback loop makes it possible to keep the voltage across the motor constant
under varying
loads and supply voltages. It also functions with optimal speed and
smoothness.
The digital-to-analog converter means for converting the 8-bit duty cycle
information from the digital logic block 10 into an analog signal is provided
by the DAC
block 12 and associated circuitry. This circuit converts the 8-bit digital
signal into voltage
VO 19. It is this voltage that sets the reference that determines the voltage
across the motor.
The output of this circuit is coupled with a capacitor 28 to ground which
provides low pass
filtering and prevents the reference voltage from changing too quickly. This
is important
since the duty cycle is calculated on every cycle. The output of the DAC block
12 is VO 19
(100%-duty cycle).
The ISENSE block 20 forms a part of the compensation loop of the closed
loop feedback means, and also functions as over-current and over/under-voltage
circuits for
monitoring the current and voltage supplied to the electric controller. It
fiuictions by
limiting or shutting down the output current in the motor. In order to measure
the current,
the ISENSE block 20 measures the voltage across a shunt 1 mS2 resistor (part
of ISENSE
block 20). Thus, 1 mV is equivalent to 1 Amp. Other size resistors could be
suitably
substituted, with a corresponding change in voltage and current. The ISENSE
block 20
compensates for the variation of the shunt copper resistor over a temperature
range. The
circuit is designed to provide a variable threshold of over-current. The
actual value of the
over-current threshold is determined by the speed control input signal. The
lower the
voltage across the motor, the lower the over-current threshold. The over-
current threshold
is broken down into 4 levels in the depicted embodiment. -
The over/under voltage circuit compares the system voltage to preset
references. One reference is for an over-voltage condition, the other
reference is for an
under-voltage condition. In either case, if the system voltage goes outside of
the normal
operation conditions, this circuit signals the state machine of the digital
logic block 10 to
17
CA 02652066 2009-01-23
tutn the output off. The over/under voltage circuit has a small amount of
hysteresis at each
threshold to prevent unwanted oscillations.
The DIN block 22 takes the external pulse width modulated (PWM) speed
control signal and level shifts it to a level compatible with the digital
logic block 10.
The POR block 24 resets the digital logic block 10 if the VDD1 (input
power) is below about 4.8V.
The VTOI block 25 and VP2VPH block 26 use accurate and stable voltage
references to create accurate and stable current references.
The BGAP block 27 includes a bandgap reference and a regulator. They can
provide internal temperature stable voltage references.
The HILOBAT block 23 senses the battery voltage and sends an over-
voltage or an under-voltage conditional to the digital logic block 10, should
the battery
voltage be above approximately 17 volts or below approximately 8 volts,
respectively.
In accordance with a preferred embodiment of the present invention, the
system further detects a locked rotor condition of the electric motor without
the need for
rotation sensors. The detection is based on the commutation noise over the
power feed lines
of the motor. The system, in accordance with the present invention, generates
a locked
rotor signal to immediately disconnect the motor, therefore avoiding
overheating of the
system devices, smoke and fire. Consequently, there is no need for additional
sensor in the
motor for detecting rotation activity. Preferably, the system is fully
resettable and reliable
since the system detects rotation pulses and does not use indirect variables,
such as
temperature, voltage or current, for detecting the locked rotor condition.
According to a preferred exnbodiment of the present invention, the sensorless
locked rotor protection circuit can work based on the motor commutation noise
signal. The
motor noise signal includes two parts: high frequency W) noise floor and low
frequency
(LF) commutation pulses. The noise (pulses) is typically generated by the
commutation
between the poles and the brushes in a DC motor. The pulses are combined with
a lot of
electrical noise, and its frequency is directly related to the speed and
number of poles and
brushes of the motor. A typical commutation noise during operation of the
motor is shown
in Figure 7. Th:e detailed noise features are shown in FIGS. 8 and 9.
Preferably, the
commutation pulse signal amplitude is higher than the noise floor level.
Typically, and as mentioned above, the pulse repetitive frequency of the
commutation pulses depends on the motor's configuration, such as the numbers
of poles,
18
CA 02652066 2009-01-23
brushes, coils and the supply voltages, which tend to vary proportionally to
the PWM duty-
cycle signal.
In accordance with one embodiment of the present invention, the pulse
frequency ranges from approximately 250Hz to 600Hz, while the PWM duty-cycle
ranges
from about 5% to 95%. Typically, the pulse signal level varies from several
hundred milli-
Volts at the highest speed down to less than about 100mV at the lowest speeds.
As described above, the commutation pulses in the motor circuit are used for
obtaining the rotation sensing signal, which is produced during commutating
between the
brushes and the commutator (rotor-coils head) of the motor. Consequently, the
brush DC
motor is preferred for implementation of the locked rotor protection feature.
In accordance with one embodiment of the present invention a method of
detecting a locked rotor condition in an electric motor is illustrated in
Figure 10. Figure 12
illustrates one embodiment of the circuit capable of implementing the method.
The commutation pulse noise, which is a voltage sigiial across the motor
during operation, is first sensed by a band-pass RC filter. The second order
bandpass filter
is utilized to obtain a usefal signal from the motor terminals by removing
undesired noise
and amplifying the pulses to a more useful level. Referring to Figure 12, the
steps of
filtering and amplifying are performed by three stages of operational
amplifiers.
Accordingly, the HF noise floor is filtered out and the commutation pulse is
selected with
an amplitude greater than a predetermined pulse signal level (motor rotation
sensing pulse)
and sent to the next stage through the Schmitt-Trigger. The Schmitt-Trigger
converts the
pulse signal to a stable square waveform signal, thus conditioning the signal
for the next
stage. The signal is amplified in the second stage and then sent to the multi-
vibrator.
Preferably, the multi-vibrator used is a monostable, retriggerable, resettable
multi-vibrator,
which is commercially available and is well known in the art. The multi-
vibrator modulates
the commutating pulse signal to a logic signal, which is used as the rotating
status detector
of the motor, i.e., detects whether or not the rotor is moving ("ON-OFF"
signal).
The logic signal can then be passed into a one-shot action circuit which is
used to shut down the operation of the motor if the logic signal is "OFF". The
locked rotor
sensing signal, which is the output of the one-shot action circuit, is
preferably combined
with the PWM input signal to provide a PWTv1 speed control signal which
reports the motor
19
CA 02652066 2009-01-23
locked rotor conditions. The output of the one-shot action circuit can thus
fnnction as an
enable signal during controller's general operation.
Consequently, after the locked rotor protecting action has been taken, e.g.,
shutting down the motor, the LPM module can be restarted when the PWM speed
control
signal is tu.rned off and then switched on again by the HVAC system.
According to a preferred embodiment of the present invention, the locked
rotor protecting circuit can be integrated into an existing IC chip with
addition of two or
three pins for the adjustment of band-pass filter and the ampli$er gain.
In accordance with another embodiment of the present invention, an
alternative circuit topology may be applied to the.locked rotor protection
feature utilizing
multi-thresholds of the motor current. A precision current shunt resistor or a
current
transformer may be applied to this topology. Figure 11 illustrates one
embodiment of such
a method. The motor current, including the locked rotor over-current, is
sensed with a
current shunt resistor or a current transformer. The sense signal is sent to a
comparator and
compared with a predetennined current threshold or with condition dependent
current
thresholds. Consequently, the output of the comparator is sent to the motor
driving control
circuit to limit the current or shutdown the LPM controller in accordance with
the
application requirements. The applicable number of the current thresholds
could be 2N (N =
0,1, 2, ..., n), depending on the operating conditions. Preferably, an
embodiment of this
method is integrated into the control chip.
Additionally, and in accordance with another embodiment of the present
invention, a method of, and system for, protecting the electric motor based on
a locked rotor
fault condition can be provided through thermal shutdown with hysteresis
recovery. When
a locked rotor fault condition occurs, the linear electric motor controller
and its motor could
be damaged by a high locked rotor current producing overheat. As illustrated
in Figure 12,
a circuit with a positive temperature coefficient thermistor, (e.g., a "PTC
thermal sensor" or
"PTC temperature sensor"), also known as a thermal sensor, is implemented in
the locked
rotor protection circuit and identified as RTI in Figure 12. PTCs, for
example, are
commercially available from Therm.ometrics of Edison, NJ. Preferably, the
thermal sensor
is coupled to the locked rotor current sensitive position(s), such as the
areas closest to the
switch component or I/O current traces. Preferably, the reaction time (warm-up
time) of the
selected sensor in the locked rotor protection circuit is short. The locked
rotor protection
CA 02652066 2009-01-23
circuit can be integrated within the existing IC chip with one additional pin
for the PTC
temperature sensor.
In accordance with another embodiment of the present invention, the locked
rotor protecting circuit provides for a strengthened electrostatic discharge
protection and
bulk current injection protection for high frequency high energy injection.
In the application environment of a linear electric motor controller, the
sensitivity classification of electrostatic discharge for packaging and
handling ESD model
(C=150pF, R=2kS2) is shown in a table below.
Type of Indicated Minimum Number of Minimum delay
Discharge Voltage Level Discharges between Discharges
kV
Direct +4 3
Discharge/
Contact 6 or
Discharge
8 10
5 Seconds
4
Air Discharge 8 3
or
10
10 Referring to the table above, the ESD protection is strengthened up to 8kV
contact discharge and up to 25kV air discharge.
The immunity of the linear electric motor controller to radiated
electromagnetic fields with bulk current injection can be classified per ISO
11452-4 and
SAE J1113-4 standards as shown in a table below.
21
CA 02652066 2009-01-23
Frequency Range BCI Level Injection Probe Threshold
Positions
MHz mA
mm
1 ... 400 50 120 5 No deviation in
linear controller's
100 450 5 function
750 f 5 (Function normally)
Optionally, a specially designed R-C-D network can be implemented in
accordance with another embodiment of the present invention for the above
mentioned
ESDBCI protection in the application environment. Figure 13 illustrates such a
circuit.
Consequently, the network allows the linear motor controller to withstand ESD
up to 8KV
contact discharge and up to 25KV air discharge as well as BCI up to 100niA in
all
injection probe positions without function deviation.
Preferably, the network is implemented as external circuitry outside the
integrated IC chip AA539 in accordance with the invention.
Additionally, and in accordance with yet another embodiment, the present
invention provides for protection of the controller from damage by reversed
polarity of a
power supply, such as a battery. The general criteria of engineering standard
for the
reversed battery polarity of the invention are about I to 2 minute(s)
operating without any
damage to the controller. In accordance with the preferred embodiment of the
invention,
the control circuit, including all of the polarity sensitive components is
protected with a
diode.
Altematively, and in accordance with another embodiment of the invention
the non-polarity components, such as ceramic/film capacitors, can replace the
polarity
sensitive components, such as electrolytic/tantalum capacitors. Preferably,
the switch
MOSFET is selected to be able to carry the reversed current with its body-
diode for more
than about 2 minutes when reversed battery polarity occurs.
Figure 2 illustratively shows a finite state machine diagram for digital logic
block 10. The finite state machine performs the function of converting the
duty cycle of the
input signal and determining the state of at least one linear power module.
The duty cycle
can be determined by comparirig the value of one counter to another. The first
counter
22
CA 02652066 2009-01-23 It
typically keeps track of the time that the input signal is in a low condition.
The second
counter typically keeps track of the period of the input signal. Usually, the
first counter is
divided by the second counter once every cycle, and the result is an 8-bit
value that can be
passed on to the digital to analog converter. Since the counters are greater
than 8-bits (16-
bits), the linear motor controller can determine the duty cycle over a wide
range of input
frequencies.
The controller also optionally, but preferably includes guard bands at each
end of the duty cycle scale. After the duty cycle is calculated, the result
can be compared to
see if it falls into either of the optional guard bands. If so, the state
machine can command
the output to be off if desired. The purpose of the guard bands, when present,
is typically to
cause the controller to enter a fail-safe mode when the input is shorted to a
battery or to
ground. This can facilitate protection of the sensitive and/or expensive
electronic circuitry
and prevent it from being damaged.
The state machine is also responsible for the control of the output when an
over current situation is detected. When an over current situation is detected
for longer than
a predetermined time, the state machine can turn the output off for
approximately one
second, and then turn the output back on to the level that the input line is
signaling. If the
fault is still present, the controller can repeat the same re-try procedure.
This can be
arranged to continue until either the fault is removed or the unit is turned
off by the operator
to conduct diagnostics or repairs.
When the input signal indicates that the electric motor should be off, the
state machine can put the controller into a sleep mode until a valid input
signal is applied.
The state machine of a preferred embodiment has six states and has signal
lines and logic which cause state changes. The precise states, signal lines,
and logic
describing the state transitions may vary in a given embodiment, as is
understood in the art.
The description of these items for a preferred embodiment is not intended to
limit the
invention, but rather to be exemplary in nature.
The state machine is typically instantiated and controlled by the controller
integrated circuit, as the digital logic block 10 of Figure 1. As is well
known in the art, the
use of such an integrated circuit can also be accomplished by other means,
including but not
limited to the use of a general-purpose microprocessor and associated memory
that are
properly configured.
The signal lines used by the state machine include registers that contain
either binary or other data. A binary signal line has a value of either a'0'
or a'1.' A non-
23
CA 02652066 2009-01-23
binary signal may contain any value that can be stored in the register. These
signals have
both a default value, which is used on power up, and a reset value, which is
used upon a
reset. The default and reset values are not always the same. In a preferred
embodiment, the
various signals and registers used to control the state machine and their
power-on default
and reset values are given in the following table.
Table 1 - State Machine Signals and Defaults
SIGNAL DEFAULT RESET
FaultCount '0'
FaultReset '0'
Lockout '0' '0'
Run '0' '0'
SOAFaultReset '0' '1'
Sleep '0' '1'
State Current State
TimeOutStart '0' '0'
dacrst '0' '0'
"State" is a register value and is updated to contain the current state of the
state machine, so
no reset value is needed. FaultCount and FaultReset are used to track and
signal a reset
condition, so they have no given reset value.
The controller typically operates on any suitable duty cycle, such as from
about 15 to 150 Hz. The applicable frequency of the speed control duty-cycle
signal is
limited by the rise/fall time of the signal pulse, the latency period of the
state machine and
the process/propagation time of the signal. Preferably, the duty cycle
frequency can be
about 35 Hz with a precision of about 0.4%. Generally, the resolution of the
duty-cycle
signal will be reduced when the signal frequency is out of range. The higher
the frequency
of duty cycle, the lower the resolution of control precision. The InRange
signal is set to 1'
if the duty cycle is from about 5% and 95%. The InRange signal is set to '0'
if the duty
cycle falls outside this range.
By default, the system powers on in SLEEP state 30, waiting for a valid
PWM input. Whenever the controller receives a power on reset or an InRange
='0'
condition, it returns back to the SLEEP state 30. Also, if the InRange line is
'0' and the
SOAFaultReset line is '0,' the state is set to the SLEEP state 30.
24
CA 02652066 2009-01-23
The SLEEP state 30 remains valid and current while the SOAFaultReset line
is '0,' the Sleep line is '0,' and the cacrst line is ' 1.' If the InRange
line is set to ' 1' while in
the SLEEP state 30, the controller logic shifts to the RUN state 31. RUN state
31 is the,
normal mode of operation, and the controller logic will remain in this state
so long as the
Run interrupt signal line (run int) is '0' and the dacrst interrupt signal
line (dacrst_int) is'1,'
so long as another state change is not caused by other signal lines. It is in
the RUN state 31
that the state machine allows the controller circuits to control the output.
Should a battery problem occur, the Hilobat analog block the sets either the
LowBat or HighBat line to '1'. If this occurs while in the RUN state 31, it
will cause the
controller logic to change to the BATTERY state 32. The HighBat and LowBat
signal lines
are latched. This helps to prevent false readings from transient events.
While in the BATTERY state 32, if both the LowBat and HighBat lines are
shift again to'0;' the controller logic will change back to the RUN state 31.
The LowBat
signal line is set when a low battery condition is detected. An overcharged
battery causes
the HighBat signal line to be set. When a battery overcharge or undercharge
condition is
corrected, the relevant signal lines are reset, and the state is changed to
RUN state 31. A
potential oscillation can be prevented by the hysteresis in the Hilobat block.
Detection of an over ourrent state causes the OverCurrent signal line to be
set
to '1' and changes the current state to OVERCIJRRENT state 34. The OverCurrent
signal
line is latched to prevent erroneous detection of the over-current condition.
If the state
machine is in OVERCURRENT state 34, and Fault 5 signal line is '1' (indicating
that the
over-current condition has occurred a predetermined maximum number of times),
and the
Lockout signal line is '1' (indicating that lockout is enabled); the state
machine will change
to the LOCKOUT state 35. While in the LOCKOUT state 35, the controller can be
arranged so it will not operate unless the power is cycled or a reset is
sigaaled.
While in the OVERCURRENT state 34, after a specified period of time, a
timeout will be signaled via the TimeOutStart interrupt line. This causes the
state machine
to shift to the TIMEOUT state 33 until the timeout is complete, at which time
the
TimeOutDone signal line is set to'1' and the state machine returns the RUN
state 31. The
TIIVIEOUTCOUNTER block is used to create a delay of approximately 1 second
after an
over current problem is detected.
Thus, the digital state machine, over-current sense circuit, under/over
voltage
sense circuit, digital to analog converter, and closed loop feedback circuit,
when optionally
but preferably all used in combination, include functional blocks of a custom
integrated
CA 02652066 2009-01-23
circuit that performs the functions of the linear power controller module of
the current
invention. Of these, the functionality of the state machine, digital to analog
converter, and
closed loop feedback circuit are preferable embodiments of the current
invention.
Figure 3 depicts the physical and logical arrangement of the pin-outs and
5. associated analog circuitry of the controller integrated circuit of an
embodiment of the
current invention. The actual signals, pin-outs and physical packaging of a
controller
integrated circuit may, of course, be modified by methods well known to those
of ordinary
skill in the art. The signals and their associated pin numbers of the
controller integrated
circuit of the embodiment herein described are given for exemplary purposes as
follows.
Table 2- Pin assignments of the Motor Controller Integrated Circuit
SIGNAL PIN BRIEF DESCRIPTION
M2 1 Input to the -ve input of OPA3. Connected to CLP
as part of the closed loop feedback.
M1 2 Connected to the output of OPA3 and is
capacitively connected to M2 to provide stability
and prevent oscillations.
CIDLY 3 Connected to a.capacitor. Capacitor value
determines the time delay before the circuit can
detect an overcurrent state. Provides a time delay to
prevent false triggering of the overcurrent detection
circuitry,
ROSC 4 Connected to Ground through a resistor which sets
the oscillator bias which determines the oscillator
frequency.
CLP 5 The output of OPAl 18 defines the voltage across the motor
and is low pass filtered.
RBIAS- 6 Connected to GROUND through a resistor. the
resistor value determines the BIAS current of the
ASIC to ensure proper operation.
CLIM 7 Connected to GROUND through a capacitor. The
value of the capacitor determines the "slew" rate
regarding the change of the output voltage across
the motor.
26
CA 02652066 2009-01-23
TEST 8 A special pin used by the manufacturer to
test the integrated circuit. Left "Grounded" or
unconnected in the circuit in normal use.
DI' T 9 Input pin that the speed control duty cycle signal is applied to.
DRV 10 Output that supplies the appropriate voltage to the
gate of the FET for a given voltage across the motor.
DGND 11 The GROUND connection for the digital circuitry
on the ASIC.
DVDD 12 Connected to the 5 volt bus for the digital circuitry
on the ASIC and provides a connection for a bypass
capacitor to ensure proper operation.
DINIlN'V 13 Used to determine the polarity of the input signal.
If connected to DVDD the polarity is positive, if
connected to GROUND the polarity is negative.
MOTORN 15 Connected to the negative side of the motor. It is
the input that is used to feedback the voltage across
the motor to ensure the proper value of the voltage
across the motor.
AGND 16 The GROUND connection for the analog circuitry
inside the ASIC.
ISENSEN 17 Connected to the negative side of the shunt and
used to monitor the current flowing through the
load. .
ISENSEP 18 Connected to the positive side of the shunt and used
with ISENSEN to monitor the current flowing
through the load.
VCC 19 Connected to the positive side of the voltage system
through a resistor. This is the positive voltage
connection of the ASIC.
VDD1 20 Connected to the 5 Volt bus of the analog circuitry
on the ASIC. Also provides a connection to a
bypass capacitor to ensure proper operation.
Pin 14 is not connected.
27
CA 02652066 2009-01-23 It
The linear electronic controller of the invention is typically packaged in a
module to protect the circuitry during transport, assembly into a final end
product, and use
thereof. The module includes a casing to protect the circuitry and facilitate
transport,
assembly, and use, and includes a suitable heat sink to dissipate sufficient
heat to inhibit or
avoid degradation of the efficiency of the controller and the module itself.
Preferably, the
module casing, also referred to herein as the enclosure, includes materials
designed to
withstand the high temperatures generated by the linear controller. It is
additionally
advantageous to optimize the placement of the controller in an embodiment so
as to
facilitate heat dissipation. This is facilitated by placement of the
controller in an aar stream,
such as an HVAC system in which it can be employed.
The present invention can provide one or more of the following benefits.
The module of the invention including the linear electronic controller can be
smaller than
conventional automotive controller modules, which can advantageously permit
placement
of the module in a wider variety of locations. For example, HVAC electronic
controllers
can be disposed within the airflow in an increased number of locations between
the intake
vent(s) and the vent(s) to the passenger compartment.
The module of the invention is an enclosure that at least substantially
surrounds the electronic controller, and in one preferred embod'unent it is at
least
substantially rectangular. Preferably, the enclosure completely surrounds the
controller
except for any electrically conductive members or holes of less than 2 mm for
additional
cooling that exist therein. In one preferred embodiment, the enclosure
includes a lid. In
one more preferred erribodiment, the lid is made of a heat-resistant material
and the heat
sink comprises a heat fin assembly mounted upon the lid that extends away
therefrom. In
another more preferred embodiment, the lid is the heat sinlc.
The lid and remainder of the enclosure are operatively associated to surround
the controller circuitry. In one embodiment, this operative association
includes a plurality
of projections and recesses on the lid and remainder of the enclosure so they
can snap
together. The modules of the invention can typically be at least about 25%,
preferably at
least about 33% smaller than conventional electronic controller modules. In
one preferred
embodiment, the modules can be at least about 50% smaller than conventional
modules.
Conventional controller modules are typically at least 9 cm long x 4,75 cm
across x 2 cm in
height, not including any appendages for mounting or any heat fins attached
thereto. Size
can refer to either the footprint area or the volume, or both. A rectangular
module typically
has a length of about 3 cm to 12 cm, preferably about 4 cm to 8 cm. In one
preferred
28
CA 02652066 2009-01-23
embodiment, the length is about 4.5 cm to 7 cm, while in another preferred
embodiment it
is about 5 cm to 6.5 cm. The width of such a rectangular module can, in one
embodiment,
be from about 3 cm to 7 cm, preferably from about 3.5 cm to 5 cm, while the
height can be
from about 1 cm to 4 cm, preferably from about 1.5 cm to 2.5 cm. For example,
the module
can be 6 cm long x 4.75 cm across x 2 cm high, which is a 33% size reduction
in footprint
area compared to the conventional size module described above.
The modules are.preferably sized sufficiently to increase the placement
options thereof, particularly when used in airflow for HVAC control. For
example, a
conventional 9 cm long module has a certain number of places in which it can
be disposed
in the airflow of an HVAC system it is designed to control, but a 6 cm long
module with
otherwise identical size characteristics will have a greater number of
suitable placement
locations. Preferably, the modules of the invention are snapped into place in
the desired
system being electronically controlled, but they can alternatively or
additionally use screws,
clasps, brackets, nails, or any other suitable fastener to be secured. One
desired location for
HVAC electronic controllers is in the airflow being controlled. In particular,
the modules
can be disposed witlun 10 cm of the fan blades or motor used to move the
airflow. The
controller module can also have one or more tabs, or mounting areas,
integrally formed
therewith to use in facilitating mounting of the module for the desired end-
use. These tabs
can extend away from the module in any direction, but in one preferred
embodiment, they
extend away in the same direction as the length of the module.
The module of the invention is typically lighter weight than conventional
modules due to improved electronic design, use of circuit boards, less
material in the
smaller modules, and the like. This can reduce repetitive motion injuries in
assembly
personnel. More importantly, this can reduce the overall weight of the
automotive vehicle,
thereby providing improved fuel efficiency. Such modules can have their weight
reduced
by at least about 10 weight percent, preferably by at least about 20 weight
percent according
to the invention. The modules of the invention produce less noise and heat
than
conventional controller packages, as well as providing increased durability.
The improved
design includes various features such as use of an integrated chip instead of
various off-the-
shelf components or even the footprint or volume size reduction of the module,
which
reduces weight both in the smaller circuit board and the smaller module
casing.
Advantageously, the electronic controllers and modules of the invention can
operate in various types of automotive electrical systems, including
conventional systems
and even 42 V systems. The controllers of the invention can be used in any
suitable
29
CA 02652066 2009-01-23
automotive application, including HVAC, motor control, lighting, or the like,
or
combinations thereof. In one preferred embodiment, the electronic controllers
are blowout-
resistant or blowout-proof, such that the electrical system has minimized or
avoids damage
when the car is jump-started improperly.
The module casing of the invention is typically an enclosure formed of a
material, which in one preferred embodiment is non-conductive, that is
sufficiently heat-
resistant to inhibit or prevent melting thereof under normal operating
conditions of the
electronic controller therein. At least one electrically-conductive member is
typically
present to provide input or output of at least one electrical signal through
the enclosure to
the controller. The member can include a plurality of terminals, wire
connections, or the
like, or combinations thereof. Preferably, the enclosure is formed of a heat-
resistant plastic
that will not melt or lose structural integrity when the controller is
operated in an 85 C
atmosphere while dissipating at least about 15 W to 150 W. In one preferred
embodiment,
the module material is capable of withstanding operation at extremely low
temperatures on
the order of -40 C, as well. Testing showed that the modules of the invention
survived at
least 1000 cycles of normal operation at these temperature extremes. The heat-
resistant
plastic preferably includes an olefinic polymer, particularly one including
amide units.
Preferably, a polyamide-polypropylene copolymer blend is included in forming
the
enclosure. In a preferred embodiment, the polyamide includes nylon units or is
entirely
nylon. An exemplary module material includes a nylon-polypropylene blend.
More preferably, the enclosure of the module includes a mineral filler, such
as talc, glass, mica, precipitated hydrated silica or other silicates, such as
calcium silicates;
clay; ceramic; aramid; lithopone; silicon carbide; diatomaceous earth;
carbonates such as
calcium carbonate and magnesium carbonate; metals such as titanium, tungsten,
aluminum,
bismuth, nickel, molybdenum, iron, copper, boron, cobalt, beryllium, zinc, and
tin; metal
alloys such as steel, brass, bronze, boron carbide, and tungsten carbide,
metal oxides such as
zinc oxide, iron oxide, aluminum oxide, titanium oxide, magnesium oxide, and
zirconium
oxide; particulate carbonaceous materials such as graphite, carbon black, and
natural
bitumen; fly ash; or a hard particulate material as noted below, or the like,
or combinations
thereof. Each filler may be included as noted above, or in the form of
whiskers, fibers,
strands, or hollow or solid microspheres, or the like, or in a combination
thereof.
Preferably, the filler includes talc, mica, or glass in some foxm. In one
embodiment, fibers
are preferred.
CA 02652066 2009-01-23
"Hard particulate materials," as defined herein, for optional use in the
material of the invention include, but are not limited to: Actinolite;
Aegirine; Akermanite;
Almandine; Analcite; Anatase; Andalusite; Andesine; Andradite; Anorthite;
Anorthoclase;
Anthophyllite; Apatite; Arsenopyrite; Augelite; Augite; Axinite; Baddeleyite;
Benitoite;
Bertrandite; Beryl; Beryllonite; Bixbyite; Boracite; Braunite; Bravoite;
Breithauptite;
Brookite; Cancrinite; Cassiterite; Celsian; Chloritoid; Chondrodite; Chromite;
Chrysoberyl;
Clinozoisite; Cobaltite; Columbite; Cordierite; Cordundum; Cristobalite;
Cummingtonite;
Danburite; Datolite; Derbylite; Diamond; Diaspore; Diopside; Dioptase;
Enstatite; Epidote;
Euclasite; Eudialite; Euxenite; Fayalite; Fergussonite; Forsterite;
Franklinite; Gahnite;
Gehienite; Geikielite; Glaucophane; Goethite; Grossularite; Hambergite;
Hausmannite;
Haiiyne; Hendenbergite; Helvite; Hematite; Hemimorphite; Hercynite; Herderite;
Hornblende; Humite; Hydrogrossularite; Ilmenite; Jadeite; Kaliophyllite;
Kyanite;
Lawsonite; Lazulite; Lazurite; Lepidocrocite; Leucite; Loellingite;
Manganosite; Marcasite;
Marialite; Meionite; Melilite; Mesolite; Microcline; Microlite; Monticellite;
Nepheline;
Niccolite; Nosean; Oligoclase; Olivine; Opal; Orthoclase; Orthopyroxene;
Periclase;
Pekovskite; Petalite; Phenakite; Piemontite; Pigeonite; Pollucite; Prehnite;
Pseudobrookite;
Psilomelane; Pumpellyite; Pyrite; Pyrochlore; Pyrolusite; Pyrope; Quartz;
Rammelsbergite;
Rhodonite; Rutile; Samarsldte; Sapphirine; Scapolite; Silica; Sodalite;
Sperrylite;
Spessartite; Sphene; Spinel; Spodumene; Staurolite; Stibiotantalite;
Tantalite; Tapiolite;
Thomsonite; Thorianite (R); Topaz; Tourmaline; Tremolite; Tridymite;
Ullmannite;
Uraninite (R); Uvarovite; Vesuvianite; Wagernite; Willemite; Zircon; and
Zoisite; and
combinations thereof, as named in the table "Physical Constants of Minerals"
from the CRC
HANDBOOK OF CHEMISTRY & PHYSICS, 52ND EDITION 1971-1972 (P. 193-197)
THE CHEMICAL RUBBER CO., CLEVELAND, OHIO.
The module preferably includes a nylon-polypropylene blend that is glass-
filled, preferably with glass fibers. An exemplary glass-filled nylon-
polypropylene blend
used to form the module casing includes the GAPEX class of materials,
commercially
available from Ferro Corp. of Cleveland, Ohio.
The module includes a heat sink, which can be a heat fin portion attached to
the module casing, a metal lid forming a portion or all of at least one side
of the module
casing, or both. Other suitable heat sinks may be envisioned and used. The
heat sink
typically includes a heat fin assembly mounted upon the lid that extends away
therefrom.
In one embodiment, a portion of the heat sink extends through the lid into the
enclosure and
adjacent the controller. When a heat fin is used, any suitable shape for
dissipating the heat
31
CA 02652066 2009-01-23
produced by the electronic controller of the invention can be used. The heat
fin can be cast
or extruded, for example, but in one embodiment it is preferably extruded. For
example, an
extruded heat fin having two protrusions spaced about 0.75 cm to 1.25 cm apart
with one
being about 3.5 to 4.5 cm long and the other being about 1.5 cm to 2.5 long
can be used.
Alternatively, a heat fin with an alternating pattern of 3 small fins and 2
long fins as noted
above can be used. The small fins can have a height of about 0.25 cm to 0.75
cm. Without
being bound by theory, it is believed that the extrusion of the fin provides
an improved
grain structure that provides for more desired thermal performance in
connection with the
invention.
Any sufficiently conductive material can be used for the heat sink or heat fin
assembly so long as it can dissipate at least about 15 W to 150 W, preferably
about 20 W to
100 W. Typically, the heat fin assembly on the module of the invention should
be able to
dissipate the about 85 W to 95 W that the electronic controllers disclosed
herein typically
produce, and any suitable material can be used. Preferred fin materials
include, for
example, aluminum, copper, thermally conductive plastics, and any combinations
thereof.
When a lid forming at least a portion of one or more sides of the module
casing is used, it is
preferred that a sufficiently thermally conductive lid be used that forms an
entire side of the
module casing. It is also preferred that the lid snaps or otherwise connects
directly to the
remaining module casing, which can be made of heat-resistant plastic. Again,
aluminum,
copper, and thermally conductive plastic(s), or combinations thereof, are
among the
preferred thermally conductive materials for the lid. Protective coatings can
be applied if
needed to inhibit or avoid oxidation, such as rust, or other forms of
degradation typical in
the environment in which the controller will be used.
The heat sink preferably does not directly contact the electronic controller
circuitry, particularly the transistor component(s), since electrical contact
therebetween
would likely damage some or all of the circuitry. A gap therefore typically
exists between
the circuitry and the heat sink. It is preferred to use an electrically
insulating and thermally
conducting material between at least the transistor component of the circuitry
and the heat
sink. Preferably, the electrically insulating material is in contact with a
portion of both the
heat sink and the controller. While any suitable electrically insulating
material can be used,
including silicone grease, mica wafers, SIL-PAD glass-epoxy materials, and the
like, it is
preferred that the electrically insulating material includes one or more dry
silicone-based
materials, i.e., non-greases. Preferably, at least one filler material is
included in the
silicone-based materials, and any of the other fillers disclosed herein may be
used for such
32
CA 02652066 2009-01-23
purpose. In particular; the SARCON class of silicones commercially available
from
FUJIPOLY of Carteret, NJ are preferred. Exemplary SARCONs include GRI, GRM,
and
GRN, or combinations thereof. The insulating material can be used in sheet
form, and is
preferably pre-cut in the shape of small squares that can be pulled off a
backing and placed
in the module to thermally conduct the desired circuitry and heat sink.
Preferably, the
insulating material is soft and pliable so that it at least partially conforms
to the shape of the
adjacent circuitry and heat sink, particularly when the modular casing is
closed to trap the
insulating material therebetween. The electrically insulating, but thermally
conductive,
material typically has a thiclrness of about 0.25 mm to 5 mm, preferably of
about 0.25 mm
to 2 mm. In one preferred embodiment, the thickness is from about 0.5 to 1 mm.
It is critical that the module be at least water-resistant, and preferably
waterproof, to avoid short circuits or other electrical problems or dangers
when moisture is
present in the airflow around the module or on nearby surfaces. For example,
water from
the air may condense on a nearby surface and drip onto the module. Thus, a
water-resistant
or waterproof seal is desired because any water that enters the module casing
may
compromise the electrical circuitry. When the heat sink is a heat-fin rather
than a metal lid,
a sealant is optionally, but preferably, used between the fin and heat-
resistant plastic lid.
The sealant preferably includes a thermal grout capable of providing
sufficient sealing to
inhibit or prevent the flow of water into the module casing. More preferably,
the thermal
grout includes one or more adhesives, such as a silicon adhesive. An exemplary
silicon
adhesive is a room temperature (RT) vulcanizing silicon adhesive (RTV).
In another embodiment, a mechanical fastener is optionally but preferably
used to hold the lid tightly against the rest of the module, which has the
desirable effect of
pressing the flexible insulating material between the heat sink and the
circuit board to
prevent lateral slippage of the insulating material. For example, such
mechanical fasteners
include one or more clips, clamps, screws, bolts, nails, interlocking
portions, or the like, or
a combination thereof. Alternatively, one or more adhesives can be used in
place of or in
addition to one or more mechanical fasteners. In one preferred einbodiment,
the lid is held
to the rest of the module casing with one or more screws.
In one embodiment, a plurality of terminals can be rammed into the module
casing to provide electrical connections between the circuit board containing
the electronic
controller board inside the module and the plug or other wiring outside the
module casing.
Terminals so rammed into the module are optionally, but preferably, pasted to
the circuit
board with an electrically conductive paste. This can advantageously inhibit
or avoid
33
CA 02652066 2009-01-23
plastic waste, bent pins or terminals, and even the need for subsequent
soldering of the
terminals to the circuit board. In an exemplary embodiment, there are about 2
to 6
terminals and they extend through the module on a side opposite the heat sink.
Preferably,
the termin.als and heat sink are each disposed facing in the same direction as
the height of
the module, which is typically the smallest dimension of the module when
considering the
only heat-resistant enclosure.
The invention also relates to methods of dissipating heat from electronic
controllers, particularly the controller of the present invention. This is
accomplished by
providing an enclosure that is at least partially non-metallic around the
electronic controller,
which generates at least about 15 W of heat during operation and associating a
heat sink
with the controller to receive heat therefrom. The heat sink dissipates a
sufficient amount
of heat via the heat conducting heat sink to inhibit or avoid damage to the
controller and the
enclosure, particularly the non-metallic portions thereof. In one embodiment,
at least about
90 percent of the heat generated is dissipated via the heat sink.
Figure 4 shows an embodiment of a heat sink in the form of heat fin 5
attached to lid 7 of module 1 according to the invention. In this embodiment,
heat fin 5 is
metallic and lid 7 is a heat-resistant material that can be the same or
different as enclosure
10 of the module 1 adjacent the lid 7.
Figure 5 shows heat fin 5 attached to lid 7 of module 1 according to the
invention. In this perspective, lid 7 is releasably secured to the remainder
of enclosure 10
by a plurality of projections and recesses 12, 15. Also depicted in this
embodiment are two
projections 17, 20 integrally formed with the module that can be used to
releasably or non-
releasably fasten module 1 securely to a desired location, e.g., in an
automobile system
having a motor.
Figure 6 shows the bottom of module I containing the electronic controller
(not shown) in one embodiment. Two projections 17, 20 are included for
fastening module
1 to a desired location. The perspective view shows Iid 7 attached to the
remainder of
enclosure 10 via the plurality of projections and recesses 12, 15. The bottom
of module 1
also preferably contains a protective area 22 designed to protect an
electrically conductive
member 25, 26, 27 that pennits information in the form of electronic signals
to be
transmitted between the electronic controller inside module 1 and external
motors, sensors,
power sources, and the like. The electrically conductive member depicted
includes three
terminals. In another embodiment (not shown), the electrically conductive
member can
include a pair of wires that are connected to the controller inside the
enclosure 7, 10 and
34
CA 02652066 2009-01-23
pass therethrough for connection to external electrical or other components as
desired.
Such wires can be in the form of a plug that can be readily connected to
another component
as desired to facilitate assembly of the module I when being installed in an
end-use
application.
In accordance with another embodiment of the present invention, a linear
electric motor controller having multiple input interfaces is provided.
Figure 14 shows a system 100 that is representative of a typical conventional
fan speed selector based on a blower resistor card that is used in an
automobile.
System 100 comprises a series of resistors 110, 120, and 130; a.series of
relays 115, 125, and 135; a motor 140; and a selector switch 150, further
including a series
of switches 152; 154, and 156, arranged as shown in Figure 14. The resistance
of resistor
110 is greater than the resistance of resistor 120, and the resistance of
resistor 120 is greater
than that of resistor 130. Switches 152, 154, and 156 correspond to the
positions of fan
speed selector switch 150, which is typically installed in the dashboard of
the automobile.
When relay 115 is activated by the closing of switch 156, resistor 110 is
placed in serial connection with motor 140; in this case, since resistor 110
has the greatest
resistance of the three resistors 110, 120, and 130, motor 140 operates at low
speed.
Likewise, when relay 125 is activated by the closing of switch 154, resistor
120 is placed in
serial connection with motor 140; in this case, since resistor 120 has a
resistance between
that of resistors 110 and 120, motor 140 operates at medium speed. Likewise,
when relay
135 is activated by the closing of switch 152, resistor 130 is placed in
serial connection with
motor 140; in this case, since resistor 130 has the lowest resistance of the
three resistors
110, 120, and 130, motor 140 operates at high speed. The three different
speeds at which
motor 140 operates depend on the resistance values of resistors 110, 120 and
130.
Furthermore, selector switch 150 may also be constituted with a fourth switch
and a fourth
relay that connects power source VBet to motor 140 directly without any
resistor in series, in
which case motor 140 operates at its highest speed with fully rated power
supply.
In addition, some arrangements of conventional fan speed selectors allow for
the activation of more than one relay at a time, which makes possible the
selection of motor
speeds between the low, medium and high speeds. Such arrangements allow for
finer
control of the speed of the motor while maintaining compactness of the system
by
minimizing the number of relays and resistors. However, the fineness of
control is still
discrete rather than continuous. The arrangement of three stages of resistors,
relays, and
switch terminals shown in Figure 14 is depicted to represent a typical
conventional fan
CA 02652066 2009-01-23
speed selector system; however, conventional systems are not limited to three
stages, and
may contain fewer or greater than three stages. The choice of the number of
stages is
determined by the desired degree of control of motor 140, and by the allocated
budget for
the system. The cost of the system increases with the number of stages.
System 100 has a disadvantage in that it is not configurable. For example, a
three-stage system cannot be re-configured to provide the functionality of a
six-stage
system. Instead, a different system must be provided, which requires
automobile
manufacturers to maintain an inventory of three-stage systems, four-stage
systems, and so
on. Furthermore, since resistors 110, 120, and 130 are fixed resistors,
automobile
manufacturers must also maintain an inventory of systems for different fan
motor models,
which is very costly.
Figure 15 shows a block diagram of a linear electric motor controller system
200 of the present invention. System 200 comprises a controller 210 further
having an
application-specific integrated circuit (ASIC) 220 and a power MOSFET
1(~etallic Oxide
Semiconductor Field Effect Transistor) 230. Controller 210 is connected to a
power source
VBu and a motor 250, for example, the motor of a HVAC system in an automobile,
arranged
as shown in Figure 15. ASIC 220 is connected to a four-stage selector switch
260, further
having switches 262, 264, 266, and 268 that correspond to positions A, B, C
and D on
selector switch 260. ASIC 220 is a linear motor controller integrated circuit
as described in
U.S. Patent Application No. 10/017,232, filed December 13, 2001, by Carter
Group
Canada. As described in the '232 patent application, ASIC 220 provides linear
control of
the motor speed through the use of a voltage feedback loop across the motor
connections.
Thereby, even if the battery voltage varies due to environmental factors such
as changes in
ambient temperature or the operating condition changes of the motor load, the
speed of the
motor is maintained at a constant. Power MOSFET 230 works as a variable
electronic
resistor as well as an electronic switch in implementation.
Controller 210 emulates the function of a resistor card by controlling the
voltage across motor 250 by means of power MOSFET 230 connected in series with
motor
250.
In operation, when the operator selects position A of selector switch 260 by
closing switch 262, ASIC 220 detects the grounding of the corresponding pin,
and
accordingly controls power MOSFET 230 to produce a predetermined first voltage
that is
supplied to motor 250. Thereby, the desired voltage is supplied to motor 250
so as to
36
CA 02652066 2009-01-23
generate a first desired fan rotation speed. Thus, this operates as an
equivalent to the first
stage of a resistor card in a conventional fan speed selector.
Likewise, when the operator selects position B of selector switch 260 by
closing switch 264, ASIC 220 detects the grounding of the corresponding pin,
and
accordingly controls power MOSFET 230 to produce a predetermined second
voltage that
is supplied to motor 250. Thereby, the desired voltage is supplied to motor
250 so as to
generate a second desired fan rotation speed. Thus, this operates as an
equivalent to the
second stage. of a resistor card in a conventional fan speed selector.
The same logic applies to the selection of positions C and D of selector
switch 260. Furthermore, selector switch 260 shown in Figure 15 is not limited
to four
stages, and may have a fewer or greater number of stages.
ASIC 220 is further coupled to a setting input signal port for generating
predetermined motor selection speeds. The input signal port may comprise a
pulse-width
modulation (PWM) motor speed setting signal li.ne and/or a DC voltage speed
setting signal
line to enable either duty cycle pulse-width modulation control or DC analog
voltage
continuously variable control of motor 250 with predetermined input option,
respectively.
Figure 16 shows a schematic of an optional input block 300 of ASIC 220.
Input block 300 comprises a PWM/digital conversion block 310, a
discrete/digital
conversion block 320, a DC/digital conversion block 330, an adder 340, a
Digital/Analog
converter 345, and a slope controller 350.
The function of adder 340 is to add the 8-bit digital signals applied to its
inputs.
Slope control block 350 controls the rate of change of the voltage appearing
at VControl=
PWM/digital block 310 converts the input PWM control signal to an 8-bit
digital signal. The output signal from PWM/digital block 310 is active when
the duty cycle
of the input signal is between 5% and 95%.
Discrete/digital block 320 processes the signals from the foiur positions (A,
B, C, and D) of selector switch 260, which correspond to switches 262, 264,
266, and 268.
For example, when switch position A is selected by closing switch 262,
discrete/digital
block 320 produces a first digital output signal. Likewise, when switch
position B, C, or D
is selected, a second, third, or fourth digital output signal, respectively,
is produced.
Discrete/digital block 320 is active only when one of switches 262, 264, 266,
or 268 is
closed, thus grounding one of the corresponding inputs A, B, C, or D to input
block 300.
37
CA 02652066 2009-01-23
Grounding of switches 262, 264, 266, and 268 (A, B, C, D) corresponds to
appropriate DC voltage values, e.g., 5 V, 3.75 V, 2.5 V, and 1.5 V,
respectively. Further,
discrete/digital block 320 converts each such voltage to an 8-bit digital
value in binary
system. For example, if 5 (Volt) is set as the highest value of the 8-bit
digital description, it
will be converted to the value of 11111111 in binary system, corresponding to
the value of
255 in decimal system. Therefore, 5V is converted to 1111111 1B, 3.75V to
11000000B,
2.5V to 10000000B, and 1.5V to 01001101B.
Further, although Figure 16 depicts discrete/digital block 320 with four
inputs, the present invention is not limited thereto, and discrete/digital
block 320 may have
fewer or greater than four inputs. Further, in a case where the number of
positions on the
selector switch is less than the number of inputs on discrete/digital block
320, some number
of the inputs of discrete/digital block 320 may remain unconnected. In such a
case, the
lowest and highest ranking positions of the inputs of discrete digital block
320 are
preferably used for the highest and lowest positions of the selector switch,
and the
remaining selector switch positions are preferably assigned as evenly as
possible across the
remaining inputs of discrete/digital block 320.
Discrete/digital block 320 enables system 200 of the present invention to
serve as a drop-in replacement for conventional resistor card-based fan speed
selectors in
automobiles. The design of the circuitry of discrete/digital block 320 is well
known in the
art, and library modules are commercially available to incorporate such a
block in an ASIC.
DC/digital block 330 has two functions: (1) to process the input signal from
a control potentiometer, and (2) to generate an output voltage offset for
transfer function.
DC/digital block 330 converts the DC voltage to an 8-bit digital value, e.g.,
5V is converted
to 1111111 1B, similar to the conversion of block 320.
When DC/digital block 330 is used to process the input signal from a control
potentiometer (not shown), the resistive value of an external potentiometer is
measured, and
the block 330 produces an output digital signal that is proportional (or
inversely
proportional) to the measured resistance. Thereby, motor 250 can be
continuously
controlled rather than discretely controlled. Further, DC/digital block 330
produces a
digital output only when the UDc is greater than 0.5 V. .
When DC/digital block 330 is used to generate a transfer function, it outputs
an offset digital signal based on the input signal, and that offset digital
signal is combined,
by adder 340, with the output from PWMJdigital block 310 or the output from
38
CA 02652066 2009-01-23
discrete/digital block 320. The input signal to DC/digital block 330 is
determined by the
specifications of the motor 250 to be controlled.
According to the above description, only one of PWM/digital block 310,
discrete/digital block 320, and DC/digital block 330 operates at any one time.
However,
when DC/digital block 330 operates in offset mode, it operates in combination
with either
PWM/digital block 310 or discrete/digital block.
PWM/digital block 310 converts a duty cycle of 100% (high DC value) to an
8-bit digital value of 11111111B, and a duty cycle of 50% to 10000000B.
The output of adder 340 is a digital signal that is converted to an analog
signal by D/A converter 345, which is then processed by slope controller 3 50
so that Vcontroi
is appropriate for the particular motor 250 to be controlled.
Thereby, linear controller 210 is provided with the optional multiple inputs
conversion interface.
Therefore, a reliable, configurable, compact, and low-cost linear electric
motor controller for a vehicular HVAC system is provided. The controller
features optional
multiple input interfaces, specifically an interface for pulse width
modulation control; an
interface for discrete, stepwise control; and an interface for continuous
variable control. The
controller is implemented on an application-specific integrated circuit, and
voltage to the
electric motor is varied by a power MOSFET.
In accordance with another embodiment of the present invention, an
arrangement of parallel FETs operating in linear mode is provided for reducing
thermal
impedance in a switch that controls the motor in the HVAC system of a vehicle
such as an
automobile.
Figure 17 shows a schematic of system 700 of parallel temperature protected
FETs in linear mode having reduced thermal impedance. System 700 comprises the
first
temperature protected FET 710, the second temperature protected FET 720, DC
electric
motor 730, and application-specific integrated circuit (ASIC) 740 that are
arranged as
shown.
Temperature protected FET 710 further includes FET 714, and a pair of
build-in temperature sensor diodes 716 and 718. Temp FET 710 is a single
integrated 5-pin
component having diodes 716 and 718 disposed directly over the junction of FET
714, such
that the temperature of diodes 716 and 718 is always essentially the same as
that of FET
714. Furthermore, diodes 716 and 718 are temperature-sensitive diodes, i.e.,
the resistance
39
CA 02652066 2009-01-23
of the diodes decreases as their temperature increases. For example,
temperature protected
FET 710 is typically rated 50 A at 60 V. An example of a commercially
available
temperature protected FET 710 is the IRLBD59N04E, manufactured by
International
Rectifier (El Segundo, CA).
Similarly, temperature protected FET 720 comprises FET 724, and diodes
726, 728 and has the same features and functions as FET 710.
Motor 730 is a direct current electric blower motor of a vehicular HVAC
system.
ASIC 740 further includes FET A drive block 750, constant current source
block 755, FET B drive block 760, constant current source block 765,
temperature
balancing block 770, summing block 780, and voltage control loop block 790.
The function of temperature balancing block 770 is to compare voltages VTl
with Vn. Voltage VTI from diodes 716 and 718 is proportional to the
temperature of FET
714. Likewise, voltage Vn from diodes 726 and 728 is proportional to the
temperature of
FET 724.
The fimction of constant current source blocks 755 and 765 is to supply
diodes 716 and 718 and diodes 726 and 728, respectively, with a constant
current of, for
example, 250 A, so as to sense a temperature related voltage across the
diodes on a
referable basis.
The function of voltage control loop block 790 is to compare voltage across
motor 730 with Vo~ftl, which is supplied by input block 300 as shown in Figure
16 and
described above. If voltage across motor 730 is less than Vcoõt,ot, then
voltage control loop
block 790 provides a reference voltage to FET drive blocks 750 and 760.
The function of FET drive blocks 750 and 760 is to drive FETs 714 and 724,
respectively.
To ensure that both FETs operate at the same temperature in order to ensure
an approximately equal lifespan, the function of temperature balancing block
770 is to
compare voltages VTl and Vn, which are proportional to the tempera.tures of
temp FETs
710 and 720, respectively. If VT2 is greater than VTI, then temperature
balancing block 770
outputs an offset signal. Temperature balancing block 770 preferably comprises
a rail-to-
rail operational amplifier.
In turn, summing block 780 offsets the drive voltage from FET drive block
760 to drive FET 724 based on the difference between VTl and V-n. Thus, the
drive voltage
CA 02652066 2009-01-23
to FET 724 is raised or lowered in order to keep the temperature of FET 724
the same as
that of FET 714.
Thereby, the present invention ensures an equal lifespan of parallel FETs
operating in linear mode for controlling the motor in a vehicular HVAC system,
prevents
thermal runaway, and reduces the thermal impedance of the switch by virtue of
using two
FETs instead of one.
In an alternative embodiment, more than two temperature protected FETs are
arranged in parallel, each additional temperature protected FET having a
corresponding
FET drive block, constant current source block, and temperature balancing
block in a
similar arrangement as shown for temperature protected FET 720. In this
embodiment, teFap
FET 720 and additional temperature protected FETs would have their drive
voltages offset
so that their temperatures track the temperature of temperature protected FET
710. The
implementation of additional temperature protected FETs would provide the
advantages of
greater current capability with less/lower thermal impedance.
In yet another alternative embodiment, one of the two parallel FETs is a
temperature protected FET and the other is a standard FET, and the objective
is to equalize
the current flowing through them and thereby achieve temperature protection
for the entire
switch. In this embodiment, the standard FET is slightly oversized, and the
temperature
protected FET is used to handle the power as well as to sense the temperature
of the switch.
Compared with the conventional approach of using two standard FETs, this
embodiment
reduces overall thermal impedance of the switch and also provides thermal
protection of the
switch at the same time. Since a temperature protected FET is twice the price
of a standard
FET, the present approach is also cost effective. However, this embodiment
does not deal
with the problem of thermal runaway.
Therefore, an arrangement of parallel temperature protected FETs operating
in linear mode is provided for reduced thermal impedance in a switch for
controlling the
motor in the HVAC system of a vehicle such as an automobile. This arrangement
has the
benefit of reducing thermal impedance while ensuring a balanced current
distribution
flowing between the two FETs. Furthermore, since the temperature of each of
the parallel
temperature FETs is monitored, it is possible to fully load each FET with
current in this
arrangement. Further, since each temperature FET includes temperature-
sensitive diodes
proximate to the FET junction, the temperature of the junction can be measured
more
accurately than with an external sensor, which provides for better thermal
protection.
41
CA 02652066 2009-01-23
In accordance with another embodiment of the present invention, a
protection circuit topology for a linear power module is provided.
Figure 18 shows a schematic of LPM system 800 in accordance with one
embodiment of the present invention. System 800 comprises application-specific
integrated
circuit (ASIC) 810 and power MOSFET 820 with its parasitic body diode 825.
System 800
also comprises zener diode 830, diode 840, resistor 850, diode 860, and diode
870. In
reality, diode 870 is an integrated component of ASIC 810 but is illustrated
as a separate
element for the purposes of clarity. System 800 also includes a power source
Vbat and
electric motor 880, for example the motor of a HVAC system in an automobile,
arranged as
shown in Figure 18. ASIC 810 is a linear motor controller integrated circuit
as described in
U.S. Patent Application No. 10/017,232, filed December 13, 2001, by Carter
Group
Canada. As described in the '232 patent, ASIC 810 provides linear control of
the motor
speed through the use of a voltage feedback loop across the motor connections.
Power
MOSFET 820 is capable of handling, for example, 60 A and 50 V.
When the polarity of Vbat is reversed from its design polarity, diode 860 and
diode 870 prevent wrong current flow to ASIC 810 by obstructing and
restricting current
flow to ASIC 810, thereby protecting ASIC 810 from damage. Simultaneously,
parasitic
body diode 825 conducts the opposite current while motor 880 operates in the
direction
opposite of its design direction. Based on the nominal Vbat of 9 V to 16 V,
the reverse
polarity condition produces a motor current of 5-6 amps and a parasitic body
diode 825
voltage of less than one volt. This serves to protect power MOSFET 820 from
damage by
limiting the power dissipation to well under the power MOSFET 820 power limit
of 100
watts. Because of this low power dissipation level, power MOSFET 820 is able
to sustain
the reverse polarity condition for an unlimited time without damage.
Parasitic body diode 825 is not an actual diode but indicates the diode-like
behavior of power MOSFET 820 when system 800 is in reverse polarity mode.
Under the
condition of design polarity, parasitic body diode 825 is arranged such that
it is forward-
biased and allows current to flow through power MOSFET 820. Because parasitic
body
diode 825 is not an actual diode, its performance characteristics are not
explicitly specified,
though its typical performance characteristics are approximately 0.8 volts and
15-20 amps
when operating with a 60-70 amp device (approximately one-third of the nominal
current of
system 800). The behavior of parasitic body diode 825 is evident only when
system 800
operates in a reverse polarity state.
42
CA 02652066 2009-01-23
When current is reduced to motor 880 in cases of motor slowdown and
shutdown, a large voltage spike (up to 200 volts) may result from the natural
inductance of
the windings in motor 880. Smaller voltage overshooting spikes within the
circuit may also
occur due simply to the inductive nature of the motor-driven system. A voltage
spike in
system 800 results in current traveling through zener diode 830, diode 840,
and resistor 850,
thus switching on power MOSFET 820. Resistor 850 limits the current through
zener diode
830 and diode 840, and zener diode 830 acts as a voltage Clamper. Because
power
MOSFET 820 is switched on, and the breakdown voltage of zener diode 830 is
less than the
breakdown voltage of power MOSFET 820, power MOSFET 820 is able to absorb the
energy from motor 880 without damage.
Therefore, a reliable and cost-effective linear power module protection
circuit for a vehicular HVAC system is provided. The circuit utilizes a state-
of-the=art
ASIC controller module, a power MOSFET, and several diode networks to
eliminate
potential damage to the circuit during a reverse polarity or voltage
overshooting spike
conditions. The circuit also provides over-voltage protection to the power
MOSFET during
voltage excursions caused by any transient processes of electric motor during
slowdown
and shutdown. The circuit is implemented on an application-specific integrated
circuit, and
power is provided to the electric motor through switching of the power MOSFET.
In accordance with another embodiment of the present invention, a
temperature protected FET-based apparatus is provided for protection against a
locked rotor
condition in an HVAC system.
Figure 19 shows first embodiment of a system 900 of the present invention
based on a 5-pin temperature protected FET. System 900 comprises temperature
protected
FET 910, DC electric motor 920, and application-specific integrated circuit
(ASIC) 930,
arranged as shown. Temperature protected FET 910 further comprises a FET 914,
diode
916, and diode 918. ASIC 930 further comprises temperature comparator block
940, current
source block 950, timer block 960, capacitor 980, and FET drive block 970.
Temperature protected FET 910 is a single integrated 5-pin component
having diodes 916 and 918 disposed directly over the junction of FET 914, such
that the
temperature of diodes 916 and 918 is always essentially the same as that of
FET 914.
Furthennore, diodes 916 and 918 are temperature-sensitive diodes, i.e., the
resistance of the
diodes decreases as their temperature increases. Temperature protected FET 910
is typically
rated for 50 A at 60 V. An example of a commercially available temperature
protected FET
is the IRLBD59NO4E, manufactured by Intemational Rectifier (El Segundo, CA).
43
CA 02652066 2009-01-23
The function of FET drive block 970 is to drive FET 914 in response to
inputs from other control logic either internal or external to ASIC 930.
The function of temperature comparator block 940 is to compare voltages VT
with VTemp. Voltage VT from diodes 916 and 918 is proportional to the
temperature of FET
914. Voltage VT,,,,Ip is a reference voltage generated by a divider circuit
(not shown)
generally external to ASIC 930, and is set to a value corresponding to the
maximum
permissible temperature of temperature protected FET 910. The actual value of
VTe~
depends upon the particular characteristics of FET 910 and the parameters of
diodes 916
and 918. In one embodiment of the present invention, VTw is set to a range of
approximately 0.4 to 0.5 V.
Diodes 916 and 918 in temperature protected FET 910 are supplied with a
constant current source Ic (e.g., 250 A) by constant current source block 950
to sense the
temperature related voltage across the diodes. In the event that motor 920
enters a locked
rotor condition, the failure of the fan will stop the supply of cool air to
FET 910, causing an
over-temperature condition in FET 910, which occurs regardless of whether an
overcurrent
condition exists. Since diodes 916 and 918 in FET 910 are supplied with a
constant current
source Ic, the voltage VT across diodes 916 and 918 decreases as the
temperature of diodes
916 and 918 rises and their resistance decreases. This drop in voltage
triggers temperature
comparator block 940 when the voltage VT falls below VTVMpõ which in turn
triggers timer
block 960.
The function of timer block 960 is to output a shutdown signal when VTimeoõt
is triggered. The length of time that the shutdown signal is output is set by
capacitor 980,
generally extemal to the ASIC 930, which is connected to the timer 960. The
timer 960 can
typically be set in a range from 1 to 20 seconds, but is not limited thereto.
For purposes of
explaining the present invention, the timer is assumed to be set to a value of
10 seconds.
When timer block 960 is triggered by the triggering of temperature
comparator block 940, timer block 960 outputs a shutdown signal to FET drive
block 970
for 10 seconds, which turns off FET 914 for 10 seconds by setting VD low,
which in turn
shuts off motor 920 for 10 seconds. After 10 seconds have passed, timei block
960 again
examines VTin,eO1t from temperature comparator block 940. If VTimeout is still
high because it
is being triggered by temperature comparator block 940 due to the continuing
over-
temperature condition of temperature protected FET 910, timer block 960
continues to
output a shutdown signal to FET drive block 970, keeping FET 914 and motor 920
off for
another 10 seconds. If VT;meoõt has gone low, timer block 960 ceases
outputting the
44
CA 02652066 2009-01-23
shutdown signal to FET drive block 970, which then drives FET 914 to operate
motor 920.
If temperature protected FET 910 overheats again, the above process is
repeated.
Occasionally, the repetitive process of shutting down and starting up motor
920 may have
the beneficial effect of enabling motor 920 to extricate itself from its
locked rotor condition,
for example by ejecting a foreign object that may have become jammed in the
rotor.
In one embodiment of the present invention, timer block 960 includes a
function that counts the number of attempts made to stop and start motor 920
once VT;.out
has gone high and, after a specified number of attempts, permanently outputs a
shutdown
signal to FET drive block 970 and permanently turns off FET 914 and motor 920.
An advantage of system 900 based on a 5-pin FET is that it is amenable to
application in a parallel FET arrangement of system 700 as shown in Figure 17,
which
allows current sharing.
Figure 20 shows an alternative embodiment of a system 1000 of the present
invention based on a 3-pin temperatt:re protected FET. System 1000 comprises
temperature
protected FET 1010, electric motor 1020, ASIC 1030, resistor 1050, capacitor
1070, and
zener diode 1080, which are arranged as shown. Temperature protected FET 1010
further
includes FET 1014 and thyristor 1016. ASIC 1030-further comprises FET drive
block 1040
and timer block 1060.
In operation, thyristor 1016 automatically shuts down FET 1014 when an
over-temperature condition arises by shorking.the gate of FET 1014 to the
source of FET
1014, thereby stopping operation of motor 1020. FET 1014 remains shut down
until the
temperature falls below the cutoff temperature, at which point thyristor 1016
stops shorting
the gate to the source, thereby starting operation of FET 1014 and motor 1020
again.
Typically, the cutoff temperature is about 170 C. Resistor 1050 limits the
current through
the thyristor, thereby protecting the thyristor 1016.
However, there are some 3-pin temperature protected FETs on the market
that are configured so that, once an over-temperature condition arises, FET
1014 remains
off permanently until a power reset. System 1000 overcomes this disadvantage
as follows.
When the temperature of temperature protected FET 1010 rises above the cutoff
temperature and thyristor 1016 shuts down FET 1014 and motor 1020, the
shutting down of
motor 1020 induces a voltage spike at the drain of FET 1014, which causes
zener diode
1080 to conduct. In turn, timer block 1060 triggers a shutdown signal to FET
drive block
1040 for 10 seconds, which turns the drive signal VD to FET 1010 off for 10
seconds.
CA 02652066 2009-01-23
Driving VD to zero resets thyristor 1016 so that FET 1014 and motor 1020 can
be started up
again in 10 seconds.
Zener diode 1080 is specified so that it does not break down under normal
operation of motor 1020. A typical breakdown voltage for zener diode 1080 is
25 V.
One of the advantages of the present embodiment is that 3-pin temperature
protected FETs are less-expensive than 5-pin temperature protected FETs.
Therefore, a temperature protected FET-based apparatus for protection
against a locked rotor condition in an HVAC system is provided. The apparatus
comprises
a te'mperature protected FET, an electric motor, and an ASIC having a
temperature
comparator block, a current source block, a timer block, a capacitor, and a
FET drive block.
When an over-temperature condition arises in the temperature protected FET,
the
temperature comparator block triggers the timer block to send a shutdown
signal for 10
seconds to the FET drive block, after which the timer block ends the shutdown
signal if the
over-temperature condition has ended.
The term "about," as used herein, should generally be understood to refer to
both numbers in a range of numerals. Moreover, all numerical ranges herein
should be
understood to include each whole integer within the range.
Although preferred embodiments of the invention have been illustrated in the
accompanying drawings and described in the foregoing Detailed Description, it
will be
understood that the invention is not limited to the embodiments disclosed but
is capable of
numerous rearrangements and modifications of parts and elements without
departing from
the spirit and scope of the invention or the equivalent(s) of various features
described
herein. It will be understood that the chemical and/or mechanical details of
every design
may be slightly different or modified by one of ordinary skill in the art
without departing
from the teachings of the present invention. Thus, the present invention is
not iintended to
be limited to the embodiments described and illustrated herein, but is to be
accorded the
broadest scope consistent with the teachings of the invention set forth
herein.
46
